Radio Handbook 16 1962

1.1161ffliale:n0

This book is revised and brought up

to date (at irregular intervals) os

necessitated by technical progress.

111E

11111110

II 1111ß001i

Sixteenth Edition

WILLIAM I. ORR, W6SAI

Editor, 16th Edition

The Standard of the Field -

for advanced amateurs

practical radiomen

practical engineers

practical technicians

Published and distributed to the electronics trade by

EDITORS and ENGINEERS, Ltd. Summerland , California

Dealers: Electronic distributors, order from us. Bookstores, libraria. newsdealers order from Baker li

Taylor, Hillside, N.J. Export (exc. Canada). order from N.M. Snyder Co., 440 Park Ave. So., N.Y. 16.

www.americanradiohistory.com

THE RADIO HANDBOOK

SIXTEENTH EDITION

Copyright, 1962, by

Editors and Engineers, Ltd.

Summerland, California, U.S.A.

Copyright under Pan -American Convention

All Translation Rights Reserved

Printed in U.S.A.

The "Radio Handbook" is also available on special order in Spanish and

Italian editions; French, German, and Flemish -Dutch editions are in

preparation or planned.

Outside North America, if more convenient, write: (Spanish) Marcombo, S.A., Av.

Jose Antonio, 584, Barcelona, Spain; (Italian) Edizione C.E.L.I., Via Gandino 1,

Bologna, Italy; (French, German, Flemish- Dutch) P. H. Brans, Ltd., 28 Prins Leopold

St., Borgerhout, Antwerp, Belgium.

Other Outstanding Books from the Same Publisher

(See Announcements at Back of Book)

THE RADIOTELEPHONE LICENSE MANUAL

THE SURPLUS RADIO CONVERSION MANUALS

THE SURPLUS HANDBOOK

THE WORLD'S RADIO TUBES ( RADIO TUBE VADE MECUM)

TILE WORLD'S EQUIVALENT TUBES ( EQUIVALENT TUBE VADE MECUM)

THE WORLD'S TELEVISION TUBES (TELEVISION TUBE VADE MECUM)

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THE RADIO HANDBOOK

16th Edition

Table of Contents

Chapter One. INTRODUCTION TO RADIO 11

1 -1 Amateur Radio 11

1 -2 Station and Operator Li 12

1 -3 The Amateur Bands 12

1 -4 Starting Your Study 14

Chapter Two. DIRECT CURRENT CIRCUITS 21

2 -1 The Atom 21

2 -2 Fundamental Electrical Units and Relationships 22

2 -3 Electrostatics - Capacitors 30

2 -4 Magnetism and Electromagnetism 35

2 -5 RC and RL Transients 38

Chapter Three. ALTERNATING CURRENT CIRCUITS 41

3 -1 Alternating Current 41

3 -2 Resonant Circuits 53

3 -3 Nonsinusoidal Waves and Transients 58

3 -4 Transformers 61

3 -5 Electric Filters 63

Chapter Four. VACUUM TUBE PRINCIPLES 67

4 -1 Thermionic Emission 67

4 -2 The Diode 71

4 -3 The Triode 72

4 -4 Tetrode or Screen Grid Tubes 77

4 -5 Mixer and Converter Tubes 79

4 -6 Electron Tubes at Very High Frequencies 80

4 -7 Special Microwave Electron Tubes 81

4 -8 The Cathode -Ray Tube 84

4 -9 Gas Tubes 87

4 -10 Miscellaneous Tube Types 88

Chapter Five. TRANSISTORS AND SEMI -CONDUCTORS 90

5 -1 Atomic Structure of Germanium and Silicon 90

5 -2 Mechanism of Conduction 90

5 -3 The Transistor 92

5 -4 Transistor Characteristics 94

5 -5 Transistor Circuitry 96

5 -6 Transistor Circuits 103

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Chapter Six. VACUUM TUBE AMPLIFIERS 106

6 -1 Vacuum Tube Parameters 106

6 -2 Classes and Types of Vacuum -Tube Amplifiers 107

6 -3 Biasing Methods 108

6 -4 Distortion in Amplifiers 109

6 -5 Resistance- Capacitance Coupled Audio- Frequency Amplifiers 109

6 -6 Video -Frequency Amplifiers 113

6 -7 Other Interstage Coupling Methods 113

6 -8 Phase Inverters 115

6 -9 D -C Amplifiers 117

6 -10 Single -ended Triode Amplifiers 118

6 -11 Single -ended Pentode Amplifiers 120

6 -12 Push -Pull Audio Amplifiers 121

6 -13 Class B Audio Frequency Power Amplifiers 123

6 -14 Cathode- Follower Power Amplifiers 127

6 -15 Feedback Amplifiers 129

6 -16 Vacuum -Tube Voltmeters 130

Chapter Seven. HIGH FIDELITY TECHNIQUES 134

7 -1 The Nature of Sound 134

7 -2 The Phonograph 136

7 -3 The High Fidelity Amplifier 138

7 -4 Amplifier Construction 142

7 -5 The "Baby Hi Fi" 143

7 -6 A Transformerless 25 Watt Music Amplifier 146

Chapter Eight. RADIO FREQUENCY VACUUM TUBE AMPLIFIERS 151

Tuned RF Vacuum Tube Amplifiers 151

8 -1 Grid Circuit Considerations 151

8 -2 Plate- Circuit Considerations 153

Radio- Frequency Power Amplifiers 154

8 -3 Class C. R -F Power Amplifiers 154

8 -4 Class B Radio Frequency Power Amplifiers 159

8 -5 Special R -F Power Amplifier Circuits 162

8 -6 Class ABI Radio Frequency Power Amplifiers 166

Chapter Nine. THE OSCILLOSCOPE 170

9 -1 A Typical Cathode -Ray Oscilloscope 170

9 -2 Display of Waveforms 175

9 -3 Lissajous Figures 176

9 -4 Monitoring Transmitter Performance with the Oscilloscope 179

9 -5 Receiver I -F Alignment with an Oscilloscope 180

9 -6 Single Sideband Applications 182

Chapter Ten. SPECIAL VACUUM TUBE CIRCUITS 185

10 -1 Limiting Circuits 185

10 -2 Clamping Circuits 187

10 -3 Multivibrators 188

10 -4 The Blocking Oscillator 190

10 -5 Counting Circuits 190

10 -6 Resistance - Capacity Oscillators 191

10 -7 Feedback 192

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Chapter Eleven. ELECTRONIC COMPUTERS 194

11 -1 Digital Computers 195

11 -2 Binary Notation 195

11 -3 Analog Computers 197

11 -4 The Operational Amplifier 199

11 -5 Solving Analog Problems 200

11 -6 Non -linear Functions 202

11 -7 Digital Circuitry 204

Chapter Twelve. RADIO RECEIVER FUNDAMENTALS 205

12 -1 Detection or Demodulation 205

12 -2 Superregenerative Receivers 207

12 -3 Superheterodyne Receivers 208

12 -4 Mixer Noise and Images 210

12 -5 R -F Stages 211

12 -6 Signal- Frequency Tuned Circuits 214

12 -7 I -F Tuned Circuits 216

12 -8 Detector, Audio, and Control Circuits 223

12 -9 Noise Suppression 225

12 -10 Special Considerations in U -H -F Receiver Design 229

12 -11 Receiver Adjustment 233

12 -12 Receiving Accessories 234

Chapter Thirteen. GENERATION OF RADIO FREQUENCY ENERGY 237

13 -1 Self -Controlled Oscillators 237

13 -2 Quartz Crystal Oscillators 242

13 -3 Crystal Oscillator Circuits 245

13 -4 Radio Frequency Amplifiers 249

13 -5 Neutralization of R.F. Amplifiers 250

13 -6 Neutralizing Procedure 253

13 -7 Grounded Grid Amplifiers 256

13 -8 Frequency Multipliers 256

13 -9 Tank Circuit Capacitances 259

13 -10 L and Pi Matching Networks 263

13 -11 Grid Bias 265

13 -12 Protective Circuits for Tetrode Transmitting Tubes 267

13 -13 Interstage Coupling 268

13 -14 Radio- Frequency Chokes 270

13 -15 Parallel and Push -Pull Tube Circuits 271

Chapter Fourteen. R -F FEEDBACK 272

14 -1 R -F Feedback Circuits 272

14 -2 Feedback and Neutralization of a Two -Stage R -F Amplifier 275

14 -3 Neutralization Procedure in Feedback -Type Amplifiers 277

Chapter Fifteen. AMPLITUDE MODULATION 280

15 -1 Sidebands .. .. .. 280

15 -2 Mechanics of Modulation 281

15 -3 Systems of Amplitude Modulation 283

15 -4 Input Modulation Systems 290

15 -5 Cathode Modulation 295

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15 -6 The Doherty and the Terman- Woodyard Modulated Amplifiers 296

15 -7 Speech Clipping 298

15 -8 The Bias -Shift Heising Modulator 305

Chapter Sixteen. FREQUENCY MODULATION AND RADIOTELETYPE

TRANSMISSION 308

16 -1 Frequency Modulation 308

16 -2 Direct FM Circuits 311

16 -3 Phase Modulation 315

16 -4 Reception of FM Signals 317

16 -5 Radio Teletype 322

Chapter Seventeen. SIDEBAND TRANSMISSION 323

17 -1 Commercial Applications of SSB 323

17 -2 Derivation of Single -Sideband Signals 324

17 -3 Carrier Elimination Circuits 328

17 -4 Generation of Single -Sideband Signals 330

17 -5 Single Sideband Frequency Conversion Systems 336

17 -6 Distortion Products Due to Nonlinearity of R -F Amplifiers 340

17 -7 Sideband Exciters 342

17 -8 Reception of Single Sideband Signals 347

17 -9 Double Sideband Transmission 349

17 -10 The Beam Deflection Modulator 350

Chapter Eighteen. TRANSMITTER DESIGN 352

18 -1 Resistors 352

18 -2 Capacitors 354

18 -3 Wire and Inductors 356

18 -4 Grounds 358

18 -5 Holes, Leads and Shafts 358

18 -6 Parasitic Resonances 360

18 -7 Parasitic Oscillation in R -F Amplifiers 361

18 -8 Elimination of V -H -F Parasitic Oscillations 362

18 -9 Checking for Parasitic Oscillations 364

Chapter Nineteen. TELEVISION AND BROADCAST INTERFERENCE 367

19 -1 Types of Television Interference 367

19 -2 Harmonic Radiation 369

19 -3 Low -Pass Filters 372

19 -4 Broadcast Interference 375

19 -5 HI -FI Interference 382

Chapter Twenty. TRANSMITTER KEYING AND CONTROL 383

20 -1 Power Systems 383

20 -2 Transmitter Control Methods 387

20 -3 Safety Precautions 389

20 -4 Transmitter Keying 391

20 -5 Cathode Keying 393

20 -6 Grid Circuit Keying 394

20 -7 Screen Grid Keying 395

20 -8 Differential Keying Circuits 396

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Chapter Twenty -One. RADIATION, PROPAGATION AND TRANSMISSION

LINES .._ 399

21 -1 Radiation from an Antenna 399

21 -2 General Characteristics of Antennas .- 400

21 -3 Radiation Resistance and Feed -Point Impedance 403

21 -4 Antenna Directivity 406

21 -5 Bandwidth 409

21 -6 Propagation of Radio Waves 409

21 -7 Ground -Wave Communication 410

21 -8 Ionospheric Propagation -_ 412

21 -9 Transmission Lines 416

21 -10 Non -Resonant Transmission Lines 417

21 -11 Tuned or Resonant Lines 420

21 -12 Line Discontinuities 421

Chapter Twenty -Two. ANTENNAS AND ANTENNA MATCHING 422

22 -1 End -Fed Half -Wave Horizontal Antennas __ 422

22 -2 Center -Fed Half -Wave Horizontal Antennas 423

22 -3 The Half -Wave Vertical Antenna 426

22 -4 The Ground Plane Antenna 427

22 -5 The Marconi Antenna 428

22 -6 Space- Conserving Antennas 430

22 -7 Multi -Band Antennas 432

22 -8 Matching Non -Resonant Lines to the Antenna 438

22 -9 Antenna Construction 444

22 -10 Coupling to the Antenna System 447

22 -11 Antenna Couplers 450

22 -12 A Single -Wire Antenna Tuner 452

Chapter Twenty- Three. HIGH FREQUENCY ANTENNA ARRAYS 455

23 -1 Directive Antennas 455

23 -2 Long Wire Radiators 457

23 -3 The V Antenna 458

23 -4 The Rhombic Antenna 460

23 -5 Stacked -Dipole Arrays 461

23 -6 Broadside Arrays 464

23 -7 End -Fire Directivity 469

23 -8 Combination End -Fire and Broadside Arrays 471

Chapter Twenty -Four. V -H -F AND U -H -F ANTENNAS 473

24 -1 Antenna Requirements 473

24 -2 Simple Horizontally- Polarized Antennas 475

24 -3 Simple Vertical -Polarized Antennas 476

24 -4 The Discone Antenna _..... 477

24 -5 Helical Beam Antennas 479

24 -6 The Corner -Reflector and Horn -Type Antennas 481

24 -7 VHF Horizontal Rhombic Antenna 482

24 -8 Multi- Element V -H -F Beam Antennas 484

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Chapter Twenty -Five. ROTARY BEAMS 490

25 -1 Unidirectional Parasitic End -Fire Arrays (Yogi Type) 490

25 -2 The Two Element Beam .___ _ 490

25 -3 The Three -Element Array 492

25 -4 Feed Systems for Parasitic (Yogi) Arrays 494

25 -5 Unidirectional Driven Arrays 500

25 -6 Bi- Directional Rotatable Arrays 501

25 -7 Construction of Rotatable Arrays 502

25 -8 Tuning the Array 505

25 -9 Antenna Rotation Systems 509

25 -10 Indication of Direction 510

25 -11 "Three- Band" Beams 510

Chapter Twenty -Six. MOBILE EQUIPMENT DESIGN AND INSTALLATION 511

26 -1 Mobile Reception 511

26 -2 Mobile Transmitters 517

26 -3 Antennas for Mobile Work 518

26 -4 Construction and Installation of Mobile Equipment 520

26 -5 Vehicular Noise Suppression 523

Chapter Twenty- Seven. RECEIVERS AND TRANSCEIVERS 526

27 -1 Circuitry and Components 529

27 -2 A Simple Transistorized Portable B -C Receiver 529

27 -3 An Inexpensive Bandpass- Filter Receiver 530

27 -4 A Compact Transceiver for 10 and 15 Meters 539

27 -5 "Siamese" Converter for Six and Two Meters 547

27 -6 A Deluxe Mobile Transceiver 555

27 -7 A Deluxe Receiver for the DX Operator 564

Chapter Twenty- Eight. LOW POWER TRANSMITTERS AND EXCITERS .... 577

28 -1 A Transistorized 50 Mc. Transmitter and Power Supply 578

28 -2 A Deluxe 200 -Watt Tabletop Transmitter 581

28 -3 Strip -Line Amplifiers for VHF Circuits 595

28 -4 A "9T0" Electronic Key 597

Chapter Twenty -Nine. HIGH FREQUENCY POWER AMPLIFIERS 602

29 -1 Power Amplifier Design 602

29 -2 Push -Pull Triode Amplifiers 604

29 -3 Push -Pull Tetrode Amplifiers 606

29 -4 Tetrode Pi- Network Amplifiers 609

29 -5 Grounded -Grid Amplifier Design 612

29 -6 A 350 Watt P.E.P. Grounded -Grid Amplifier 617

29 -7 The "Tri-Bander" Linear Amplifier for 20 -15 -10 622

29 -8 An 813 Grounded -Grid Linear Amplifier 627

29 -9 The KW -2. An Economy Grounded -Grid Linear Amplifier 634

29 -10 A Pi- Network Amplifier for C -W, A -M, or SSB 643

29 -11 Kilowatt Amplifier for Linear or Class C Operation 649

29 -12 A 2- Kilowatt P.E.P. All -Band Amplifier 654

29 -13 A 3 -1000Z Linear Amplifier 661

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Chapter Thirty. SPEECH AND AMPLITUDE MODULATION EQUIPMENT 669

30 -1 Modulation 669

30 -2 Design of Speech Amplifiers and Modulators 672

30 -3 General Purpose Triode Class B Modulator 673

30 -4 A 10 -Watt Amplifier- Driver 677

30 -5 A 15 -Watt Clipper- Amplifier 678

30 -6 A 200 -Watt 811 -A De -Luxe Modulator 679

30 -7 Zero Bias Tetrode Modulators 683

Chapter Thirty -One. POWER SUPPLIES 684

31 -1 Power Supply Requirements 684

31 -2 Rectification Circuits 689

31 -3 Standard Power Supply Circuits 690

31 -4 Selenium and Silicon Rectifiers 695

31 -5 100 Watt Mobile Power Supply 697

31 -6 Transistorized Power Supplies 703

31 -7 Two Transistorized Mobile Supplies 706

31 -8 Power Supply Components 707

31 -9 Special Power Supplies 709

31 -10 Power Supply Design 713

31 -11 300 Volt, 50 Ma. Power Supply . . -_ 716

31 -12 1500 Volt, 425 Milliampere Power Supply 717

31 -13 A Dual Voltage Transmitter Supply _ 718

31 -14 A Kilowatt Power Supply 718

Chapter Thirty -Two. WORKSHOP PRACTICE 720

32 -1 Tools 720

32 -2 The Material 723

32 -3 TVI -Proof Enclosures 724

32 -4 Enclosure Openings 725

32 -5 Summation of the Problem 725

32 -6 Construction Practice 726

32 -7 Shop Layout 729

Chapter Thirty- Three. ELECTRONIC TEST EQUIPMENT 731

33 -1 Voltage, Current and Power 731

33 -2 Measurement of Circuit Constants _ 737

33 -3 Measurements with a Bridge 738

33 -4 Frequency Measurements 739

33 -5 Antenna and Transmission Line Measurements 740

33 -6 A Simple Coaxial Reflectometer 742

33 -7 Measurements on Balanced Transmission Lines 744

33 -8 A "Balanced" SWR Bridge 745

33 -9 The Antennascope 747

33 -10 A Silicon Crystal Noise Generator 749

33 -11 A Monitor Scope for AM and SSB 750

Chapter Thirty -Four. RADIO MATHEMATICS AND CALCULATIONS 752

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FOREWORD TO THE SIXTEENTH EDITION

Over two decades ago the historic first edition of the RADIO HANDBOOK

was published as a unique, independent, communications manual written

especially for the advanced radio amateur and electronic engineer. Since that early

issue, great pains have been taken to keep each succeeding edition of the RADIO

HANDBOOK abreast of the rapidly expanding field of electronics.

So quickly has the electron invaded our everyday affairs that it is now no

longer possible to segregate one particular branch of electronics and define it as

radio communications; rather, the transfer of intelligence by electrical means

encompasses more than the vacuum tube, the antenna, and the tuning capacitor.

Included in this new, advanced Sixteenth Edition of the RADIO HANDBOOK

are fresh chapters covering electronic computers, r.f. feedback amplifiers, and high

fidelity techniques, plus greatly expanded chapters dealing with semi- conductors

and special vacuum tube circuits. The other chapters of this Handbook have been

thoroughly revised and brought up to date, touching briefly on those aspects in

the industrial and military electronic fields that are of immediate interest to the

electronic engineer and the radio amateur. The construction chapters have been

completely re- edited. All new equipments described therein are of modern

design, free of TV! producing problems and various unwanted parasitic

oscillations.

The writing and preparation of this Handbook would have been impossible

without the lavish help that was tended the editor by fellow amateurs and sym-

pathetic electronic organizations. Their friendly assistance and helpful suggestions

were freely given in the true amateur spirit to help make the 16th edition of the

RADIO HANDBOOK an outstanding success.

The editor and publisher wish to thank these individuals and companies whose

unselfish support made the compilation and publication of this book an inter-

esting and inspired task. -WILLIAM I. ORR, W6SAI, 3A2AF, Editor

Thomas Consalvi, W3EOZ,

Barker & Williamson, Inc.

Claude E. Doner, W3FAL,

Radio Corporation of

America

John A. Evans, W9HRH,

Potter & Brumfield Co.

Wayne Green, W2NSD,

73 Magazine

Jo Jennings, W6EI,

Jennings Radio Mfg. Co.

E. A. Neal, W4ITC,

General Electric Co.

Harold Vance, K2FF,

Radio Corporation of

America

Blackhawk Engineering Co.

H. E. Blaksley, K7ASK

Byron Hunter, W6VML

Clifford Johnson, WOURQ

Herbert Johnson, W7GRA

Thomas Lamb, K8ERV

James G. Lee, W6VAT

Hugh MacDonald, W6CDT

Otto Miller, K6ENX

Robert Moore, W7JNC

B. A. Ontiveros, W6FFF

(drafting)

A. L. Patrick, W9EHW

Raymond Rinaudo, W6KEV

Robert Sutherland, W6UOV

W. H. Sayer, Jr., WA6BAN

Mel Whiteman, W6BZ

www.americanradiohistory.com

CHAPTER ONE

Introduction to Radio

The field of radio is a division of the much

larger field of electronics. Radio itself is such

a broad study that it is still further broken

down into a number of smaller fields of which

only shortwave or high- frequency radio is cov-

ered in this book. Specifically the field of com-

munication on frequencies from 1.8 to 450 meg-

acycles is taken as the subject matter for this

work. The largest group of persons interested in

the subject of high -frequency communication is

the more than 350,000 radio amateurs located

in nearly all countries of the world. Strictly

speaking, a radio amateur is anyone interested

in radio non -commercially, but the term is ordi-

narily applied only to those hobbyists possess-

ing transmitting equipment and a license from

the government.

It was for the radio amateur, and particu-

larly for the serious and more advanced ama-

teur, that most of the equipment described in

this book was developed. However, in each

equipment group, simple items also are shown

for the student or beginner. The design prin-

ciples behind the equipment for high- frequency

radio communication are of course the same

whether the equipment is to be used for com-

mercial, military, or amateur purposes, the

principal differences lying in construction

practices, and in the tolerances and safety

factors placed upon components.

With the increasing complexity of high -fre-

quency communication, resulting primarily from

increased utilization of the available spec-

trum, it becomes necessary to delve more deep-

ly into the basic principles underlying radio

communication, both from the standpoint of

equipment design and operation and from the

standpoint of signal propagation. Hence, it will

be found that this edition of the RADIO HAND-

BOOK has been devoted in greater proportion

11

to the teaching of the principles of equipment

design and signal propagation. It is in response

to requests from schools and agencies of the

Department of Defense, in addition to persist-

ent requests from the amateur radio fraternity,

that coverage of these principles has been ex-

panded.

1 -1 Amateur Radio

Amateur radio is a fascinating hobby with

many phases. So strong is the fascination of-

fered by this hobby that many executives, en-

gineers, and military and commercial operators

enjoy amateur radio as an avocation even

though they are also engaged in the radio field

commercially. It captures and holds the inter-

est of many people in all walks of life, and in

all countries of the world where amateur acti-

vities are permitted by law.

Amateurs have rendered much public ser-

vice through furnishing communications to and

from the outside world in cases where disaster

has isolated an area by severing all wire com-

munications. Amateurs have a proud record of

heroism and service in such occasion. Many

expeditions to remote places have been kept

in touch with home by communication with ama-

teur stations on the high frequencies. The ama-

teur's fine record of performance with the

"wireless" equipment of World War I has been

surpassed by his outstanding service in World

War II.

By the time peace came in the Pacific in

the summer of 1945, many thousand amateur

operators were serving in the allied armed

forces. They had supplied the army, navy,

marines, coast guard, merchant marine, civil

service, war plants, and civilian defense or-

ganizations with trained personnel for radio,

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12 Introduction to Radio THE RADIO

radar, wire, and visual communications and

for teaching. Even now, at the time of this

writing, amateurs are being called back into

the expanded defense forces, are returning to

defense plants where their skills are critically

needed, and are being organized into communi-

cation units as an adjunct to civil defense

groups.

1 -2 Station and Operator Licenses

Every radio transmitting station in the

United States no matter how low its power

must have a license from the federal govern-

ment before being operated; some classes of

stations must have a permit from the govern-

ment even before being constructed. And every

operator of a transmitting station must have

an operator's license before operating a trans-

mitter. There are no exceptions. Similar laws

apply in practically every major country.

"Classes of Amateur There are at present six

Operator Li classes of amateur oper-

ator licenses which have

been authorized by the Federal Communica-

tions Commission. These classes differ in

many respects, so each will b e discussed

briefly. (a) Amateur Extra Class. This class of li-

cense is available to any U. S. citizen who at

any time has held for a period of two years or

more a valid amateur license, issued by the

FCC, excluding licenses of the Novice and

Technician Classes. The examination for the

license includes a code test at 20 words per

minute, the usual tests covering basic amateur

practice and general amateur regulations, and

an additional test on advanced amateur prac-

tice. All amateur privileges are accorded the

holders of this operator's license.

(b) General Class. This class of amateur

license is equivalent to the old Amateur Class

B license, and accords to the holders all ama-

teur privileges except those which may be set

aside for holders of the Amateur Extra Class

license. This class of amateur operator's li-

cense is available to any U. S. citizen. The

examination for the license includes a code

test at 13 words per minute, and the usual ex-

aminations covering basic amateur practice

and general amateur regulations.

(c) Conditional Class. This class of ama-

teur license and the privileges accorded by it

are equivalent to the General Class license.

However, the license can be issued only to

those whose residence is more than 125 miles

airline from the nearest location at which FCC

examinations are held at intervals of not more

than three months for the General Class ama-

teur operator license, or to those who for any

of several specified reasons are unable to ap-

pear for examination.

(d) Technician Class. This is a new class

of license which is available to any citizen of

the United States. The examination is the same

as that for the General Class license, except

that the code test is at a speed of 5 words per

minute. The holder of a Technician class li-

cense is accorded all authorized amateur privi-

leges in the amateur frequency bands above

220 megacycles, and in the 50-Mc. band.

(e) Novice (.lass. this is a new class of

license which is available to any U. S. citizen

who has not previously held an amateur li-

cense of any class issued by any agency of

the U. S. government, military or civilian. The

examination consists of a code test at a speed

of 5 words per minute, plus an examination on

the rules and regulations essential to begin-

ner's operation, including sufficient elemen-

tary radio theory for the understanding of those

rules. The Novice Class of license affords

severely restricted privileges, is valid for only

a period of one year (as contrasted to all other

classes of amateur licenses which run for a

term of five years), and is not renewable.

All Novice and Technician class examina-

tions are given by volunteer examiners, as reg-

ular examinations for these two classes are

not given in FCC offices. Amateur radio clubs

in the larger cities have established examin

ing committees to assist would -be amateurs

of the area in obtaining their Novice and Tech-

nician licenses.

1 -3 The Amateur Bands

Certain small segments of the radio frequen-

cy spectrum between 1500 kc. and 10,000 .fc.

are reserved for operation of amateur radio

stations. These segments are in general agree-

ment throughout the world, although certain

parts of different amateur bands may be used

for other purposes in various geographic re-

gions. In particular, the 40 -meter amateur band

is used legally (and illegally) for short wave

broadcasting by many countries in Europe,

Africa and Asia. Parts of the 80 -meter band

are used for short distance marine work in Eu-

rope, and for broadcasting in South America.

The amateur bands available to American ra-

dio amateurs aree

160 Meters The 160 -meter band is di-

(1800 Kc. -2000 Kc.) vided into 25- kilocycle

segments on a regional

basis, with day and night power limitations,

and is available for amateur use provided no

interference is caused to the Loran (Long

Range Navigation) stations operating in this

band. This band is least affected by the 11-

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HANDBOOK Amateur Bands 13

year solar sunspot cycle. The Maximum Us-

able Frequency (MUF) even during the years

of decreased sunspot activity does not usually

drop below 4 Mc., therefore this band is not

subject to the violent fluctuations found on

the higher frequency bands. DX contacts on

on this band are limited by the ionospheric

absorption of radio signals, which is quite

high. During winter nighttime hours the ab-

sorption is often of a low enough value to per-

mit trans -oceanic contacts on this band. On

rare occasions, contacts up to 10,000 miles

have been made. As a usual rule, however,

160 -meter amateur operation is confined to

ground -wave contacts or single -skip contacts

of 1000 miles or less. Popular before World

War II, the 160 -meter band is now only sparse-

ly occupied since many areas of the country

are blanketed by the megawatt pulses of the

Loran chains.

80 Meters The 80 -meter band is the

(3500 Kc. -4000 Kc.) most popular amateur

band in the continental

United States for local "rag- chewing" and

traffic nets. During the years of minimum sun-

spot activity the ionospheric absorption on

this band may be quite low, and long distance

DX contacts are possible during the winter

night hours. Daytime operation, in general, is

limited to contacts of 500 miles or less. Dur-

ing the summer months, local static and high

ionospheric absorption limit long distance con-

tacts on this band. As the sunspot cycle ad-

vances and the MUF rises, increased iono-

spheric absorption will tend to degrade the

long distance possibilities of this band. At

the peak of the sunspot cycle, the 80 -meter

band becomes useful only for short -haul com-

munication.

40 Meters The 40 -meter band is high

(7000 Kc. -7300 Kc) enough in frequency to be

severely affected by the

11 -year sunspot cycle. During years of mini-

mum solar activity, the MUF may drop below

7 Mc., and the band will become very erratic,

with signals dropping completely out during

the night hours. Ionospheric absorption of sig-

nals is not as large a problem on this band as

it is on 80 and 160 meters. As the MUF grad-

ually rises, the skip- distance will increase on

40 meters, especially during the winter months.

At the peak of the solar cycle, the daylight

skip distance on 40 meters will be quite long,

and stations within a distance of 500 miles or

so of each other will not be able to hold com-

munication. DX operation on the 40 -meter band

is considerably hampered by broadcasting sta-

tions, propaganda stations, and jamming trans-

mitters. In Europe and Asia the band is in a

chaotic state, and amateur operation in this re-

gion is severely hampered.

20 Meters At the present time,

the 20 -meter band is

by far the most popular

band for long distance contacts. High enough

in frequency to be almost obliterated at the

bottom of the solar cycle, the band neverthe-

less provides good DX contacts during years

of minimal sunspot activity. At the present

time, the band is open to almost all parts of

the world at some time during the year. Dur-

ing the summer months, the band is active un-

til the late evening hours, but during the win-

ter months the band is only good for a few

hours during daylight. Extreme DX contacts

are usually erratic, but the 20 -meter band is

the only band available for DX operation the

year around during the bottom of the DX cycle.

As the sunspot count increases and the MUF

rises, the 20 -meter band will become open for

longer hours during the winter. The maximum

skip distance increases, and DX contacts are

possible over paths other than the Great Circle

route. Signals can be heard the "long paths,"

180 degrees opposite to the Great Circle path.

During daylight hours, absorption may become

apparent on the 20 -meter band, and all signals

except very short skip may disappear. On the

other hand, the band will be open for world-

wide DX contacts all night long. The 20 -meter

band is very susceptible to "fade- outs"

caused by solar disturbances, and all except

local signals may completely disappear for

periods of a few hours to a day or so.

(14,000 Kc.-14,350 Kc.)

15 Meters This is a relatively

(21,000 Kc.- 21,450 Kc.) new band for radio

amateurs since it has

only been available for amateur operation

since 1952. Not too much is known about the

characteristics of this band, since it has not

been occupied for a full cycle of solar activi-

ty. However, it is reasonable to assume that

it will have characteristics similar to both the

20 and 10 -meter amateur bands. It should have

a longer skip distance than 20 meters for a

given time, and sporadic -E (short -skip) should

be apparent during the winter months. During

a period of low sunspot activity, the MUF will

rarely rise as high as 15 meters, so this band

will be "dead" for a large part of the year.

During the next few years, 15 -meter activity

should pick up rapidly, and the band should

support extremely long DX contacts. Activity

on the 15 -meter band is limited in some areas,

www.americanradiohistory.com

14 I n t r o d u c t i o n t o R a d i o T H E R A D I O

since the older model TV receivers have a

21 Mc. i -f channel, which falls directly in the

15 -meter band. The interference problems

brought about by such an unwise choice of

intermediate frequency often restrict operation

on this band by amateur stations unfortunate

enough to be situated near such an obsolete

receiver.

10- Meters During the peak of the

(28.000 Kc.- 29,700 Kc.) sunspot cycle, the 10-

meter band is without

doubt the most popular

amateur band. The combination of long skip

and low ionospheric absorption make reliable

DX contacts with low powered equipment pos-

sible. The great width of the band (1700 kc.)

provides room for a large number of amateurs.

The long skip(1500 miles or so) prevents near-

by amateurs from hearing each other, thus

dropping the interference level. During the win-

ter months, sporadic -E (short skip) signals

up to 1200 miles or so will be heard. The 10-

meter band is poorest in the summer months,

even during a sunspot maximum. Extremely

long daylight skip is common on this band, and

and in years of high MUF the 10 -meter band

will support intercontinental DX contacts dur-

ing daylight hours.

The second harmonic of stations operating

in the 10 -meter band falls directly into tele-

vision channel 2, and the higher harmonics of

10 -meter transmitters fall into the higher TV

channels. This harmonic problem seriously

curtailed amateur 10 -meter operation during

the late 40's. However, with the new circuit

techniques and TVI precautionary measures

stressed in this Handbook, 10 -meter operation

should cause little or no interference to near-

by television receivers of modern design.

Six Meters At the peak of the sunspot

(50 Mc. -54 Mc.) cycle, the MUF occasional-

ly rises high enough to per-

mit DX contacts up to 10,000 miles or so on

6 meters. Activity on this band during such a

period is often quite high. Interest in this band

wanes during a period of lesser solar activity,

as contacts, as a rule, are restricted to short -

skip work. The proximity of the 6 -meter band

to television channel 2 often causes interfer-

ence problems to amateurs located in areas

where channel 2 is active. As the sunspot cy-

cle increases, activity on the 6 -meter band will

increase.

The V -HF Bands The v -h -f bands are

(Two Meters and "Up ") the least affected by

the vagaries of the

sunspot cycle and the Heaviside layer. Their

predominant use is for reliable communication

over distances of 150 miles or less. These

bands are sparsely occupied in the rural sec-

tions of the United States, but are quite heavi-

ly congested in the urban areas of high popu-

lation.

In recent years it has been found that v -h -f

signals are propagated by other means than by

line -of -sight transmission. "Scatter signals,"

Aurora reflection, and air -mass boundary bend-

ing are responsible for v -h -f communication up

to 1200 miles or so. Weather conditions will

often affect long distance communication on

the 2 -meter band, and all the v -h -f bands are

particularly sensitive to this condition.

The other v -h -f bands have had insufficient

occupancy to provide a clear picture of their

characteristics. In general, they behave much

as does the 2 -meter band, with the weather

effects becoming more pronounced on the high-

er frequency bands.

1 -4 Starting Your Study

When you start to prepare yourself for the

amateur examination you will find that the cir-

cuit diagrams, tube characteristic curves, and

formulas appear confusing and difficult of un-

derstanding. But after a few study sessions

one becomes sufficiently familiar with the

notation of the diagrams and the basic con-

cepts of theory and operation so that the ac-

quisition of further knowledge becomes easier

and even fascinating.

As it takes a considerable time to become

proficient in sending and receiving code, it is

a good idea to intersperse technical study ses-

sions with periods of code practice. Many

short code practice sessions benefit one more

than a small number of longer sessions. Alter-

nating between one study and the other keeps

the student from getting "stale" since each

type of study serves as a sort of respite from

the other.

When you have practiced the code long

enough you will be able to follow the gist of

the slower sending stations. Many stations

send very slowly when working other stations

at great distances. Stations repeat their calls

many times when calling other stations before

contact is established, and one need not have

achieved much code proficiency to make out

their calls and thus determine their location.

The Code The applicant for any class of ama-

teur operator license must be able

to send and receive the Continental Code

(sometimes called the International Morse

Code). The speed required for the sending and

receiving test may be either 5, 13, or 20 words

per minute, depending upon the class of li-

cense, assuming an average of five characters

to the word in each case. The sending and re-

www.americanradiohistory.com

HANDBOOK Learning the Code 15

A = N MD )

6 O Mo =El 2

C MI P ME Ma 3

D . Q IEM 4 GM

E R NEI 5

F S 6 =..

G MO T 7 fm. gm, .

H U EMI 8 OM 4=1 1M

I V 9 ME, MED 4=1 IMP

,J W =El 41 0

K -. X MIMEE 0 MEANS ZERO. AND IS WRITTEN IN THIS

WAY TO DISTINGUISH IT FROM THE LETTER 'O''

L Y MIMED MI IT OFTEN IS TRANSMITTED INSTEAD AS ONE

LONG DASH (EQUIVALENT TO 5 DOTS)

M MED MD Z MI

PERIOD (.)

COMMA (,)

INTERROGATION (7)

QUOTATION MARK (")

COLON ( )

SEMICOLON (I)

PARENTHESIS ( I

WAIT SIGN (AS) _ .

DOUBLE DASH (BREAK) MMD

ERROR (ERASE SIGN)

FRACTION BAR( /) mo

END OF MESSAGE (AR) IMID

END OF TRANSMISSION (SK) e

INTERNAT. DISTRESS SIG. (SOS) moo

Figure 1

The Continental (or International Morse) Code is used for substantially all non -automatic rodio

communication. DO NOT memorize from the printed page; code is a language of SOUND, and

must not be learned visually; learn by listening as explained in the text.

ceiving tests run for five minutes, and one

minute of errorless transmission or reception

must be accomplished within the five -minute

interval.

If the code test is failed, the applicant must

wait at least one month before he may again

appear for another test. Approximately 30% of

amateur applicants fail to pass the test. It

should be expected that nervousness and ex-

citement will at least to some degree tempo-

rarily lower the applicant's code ability. The

best prevention against this is to master the

code at a little greater than the required speed

under ordinary conditions. Then if you slow

down a little due to nervousness during a test

the result will not prove fatal.

Memorizing There is no shortcut to code pro -

the Code ficiency. To memorize the al-

phabet entails but a few eve-

nings of diligent application, but considerable

time is required to build up speed. The exact

time required depends upon the individual's

ability and the regularity of practice.

While the speed of learning will naturally

vary greatly with different individuals, about

70 hours of practice (no practice period to be

over 30 minutes) will usually suffice to bring

a speed of about 13 w.p.m.; 16 w.p.m. requires

about 120 hours; 20 w.p.m., 175 hours.

Since code reading requires that individual

letters be recognized instantly, any memoriz-

ing scheme which depends upon orderly se-

quence, such as learning all "dab" letters

and all "dit" letters in separate groups, is to

be discouraged. Before beginning with a code

practice set it is necessary to memorize the

whole alphabet perfectly. A good plan is to

study only two or three letters a day and to

drill with those letters until they become part

of your consciousness. Mentally translate each

day's letters into their sound equivalent

wherever they are seen, on signs, in papers,

indoors and outdoors. Tackle two additional

letters in the code chart each day, at the same

time reviewing the characters already learned.

Avoid memorizing by routine. Be able to

sound out any letter immediately without so

much as hesitating to think about the letters

preceding or following the one in question.

Know C, for example, apart from the sequence

ABC. Skip about among all the characters

learned, and before very long sufficient letters

will have been acquired to enable you to spell

out simple words to yourself in "dit dabs."

This is interesting exercise, and for that rea-

son it is good to memorize all the vowels first

and the most common consonants next.

Actual code practice should start only when

the entire alphabet, the numerals, period, corn-

www.americanradiohistory.com

16 Introduction to Radio THE RADIO

Figure 2

These code characters are used in languages

other than English. They may occasionally

be encountered so it is well to know them.

ma, and question mark have been memorized

so thoroughly that any one can be sounded

without the slightest hesitation. Do not bother

with other punctuation or miscellaneous sig-

nals until later.

Sound - Each letter and figure must be

Not Sight memorized by its sound rather

than its appearance. Code is a

system of sound communication, the same as

is the spoken word. The letter A, for example,

is one short and one long sound in combina-

tion sounding like dit dab, and it must be re-

membered as such, and not as "dot dash."

Practice Time, patience, and regularity are

required to learn the code properly.

Do not expect to accomplish it within a few

days. Don't practice too long at one stretch; it

does more harm than good. Thirty minutes at

a time should be the limit.

Lack of regularity in practice is the most

common cause of lack of progress. Irregular

practice is very little better than no practice

at all. Write down what you have heard; then

forget it; do not look back. If your mind dwells

even for an instant on a signal about which

you have doubt, you will miss the next few

characters while your attention is diverted.

While various automatic code machines,

phonograph records, etc., will give you prac-

tice, by far the best practice is to obtain a

study companion who is also interested in

learning the code. When you have both memo-

rized the alphabet you can start sending to

each other. Practice with a key and oscillator

or key and buzzer generally proves superior

to all automatic equipment. Two such sets

operated between two rooms are fine -or be-

tween your house and his will be just that

much better. Avoid talking to your partner

while practicing. If you must ask him a ques-

tion, do it in code. It makes more interesting

practice than confining yourself to random

practice material.

hen two co- learners have memorized the

code and are ready to start sending to each

other for practice, it is a good idea to enlist

the aid of an experienced operator for the first

practice session or two so that they will get

an idea of how properly formed characters

sound.

During the first practice period the speed

should be such that substantially solid copy

can be made without strain. Never mind if this

is only two or three words per minute. In the

next period the speed should be increased

slightly to a point where nearly all of the

characters can be caught only through con-

scious effort. When the student becomes pro-

ficient at this new speed, another slight in-

crease may be made, progressing in this man-

ner until a speed of about 16 words per minute

is attained if the object is to pass the amateur

13 -word per minute code test. The margin of

3 w.p.m. is recommended to overcome a possi-

ble excitement factor at examination time.

Then when you take the test you don't have to

worry about the "jitters" or an "off day."

Speed should not be increased to a new

level until the student finally makes solid

copy with ease for at least a five -minute

period at the old level. How frequently in-

creases of speed can be made depends upon

individual ability and the amount of practice.

Each increase is apt to prove disconcerting,

but remember "you are never learning when

you are comfortable."

A number of amateurs are sending code

practice on the air on schedule once or twice

each week; excellent practice can be obtained

after you have bought or constructed your re-

ceiver by taking advantage of these sessions.

If you live in a medium -size or large city,

the chances are that there is an amateur radio

club in your vicinity which offers free code

practice lessons periodically.

Skill When you listen to someone speaking

you do not consciously think how his

words are spelled. This is also true when you

read. In code you must train your ears to read

code just as your eyes were trained in school

to read printed matter. With enough practice

you acquire skill, and from skill, speed. In

other words, it becomes a habit, something

which can be done without conscious effort.

Conscious effort is fatal to speed; we can't

think rapidly enough; a speed of 25 words a

minute, which is a common one in commercial

operations, means 125 characters per minute

or more than two per second, which leaves

no time for conscious thinking.

www.americanradiohistory.com

HANDBOOK Learning the Code 17

Perfect Formation When transmitting on the

of Characters code practice set to your

partner, concentrate on the

quality of your sending, not on your speed.

Your partner will appreciate it and he could

not copy you if you speeded up anyhow.

If you want to get a reputation as having an

excellent "fist" on the air, just remember that

speed alone won't do the trick. Proper execu-

tion of your letters and spacing will make

much more of an impression. Fortunately, as

you get so that you can send evenly and accu-

rately, your sending speed will automatically

increase. Remember to try to see how evenly

you can send, and how fast you can receive.

Concentrate on making signals properly with

your key. Perfect formation of characters is

paramount to everything else. Make every sig-

nal right no matter if you have to practice it

hundreds or thousands of times. Never allow

yourself to vary the slightest from perfect for-

mation once you have learned it.

If possible, get a good operator to listen to

your sending for a short time, asking him to

criticize even the slightest imperfections.

Timing It is of the utmost importance to

maintain uniform spacing in charac-

ters and combinations of characters. Lack of

uniformity at this point probably causes be-

ginners more trouble than any other single fac-

tor. Every dot, every dash, and every space

must be correctly timed. In other words, ac-

curate timing is absolutely essential to intel-

ligibility, and timing of the spaces between

the dots and dashes is just as important as

the lengths of the dots and dashes themselves.

The characters are timed with the dot as a

"yardstick." A standard dash is three times

as long as a dot. The spacing between parts

of the same letter is equal to one dot; the

space between letters is equal to three dots,

and that between words equal to five dots.

The rule for spacing between letters and

words is not strictly observed when sending

slower than about 10 words per minute for the

benefit of someone learning the code and de-

siring receiving practice. When sending at,

say, 5 w.p.m., the individual letters should be

made the same as if the sending rate were

about 10 w.p.m., except that the spacing be-

tween letters and words is greatly exaggerated.

The reason for this is obvious. The letter L,

for instance, will then sound exactly the same

at 10 w.p.m. as at 5 w.p.m., and when the

speed is increased above 5 w.p.m. the student

will not have to become familiar with what

may seem to him like a new sound, although

it is in reality only a faster combination of

dots and dashes. At the greater speed he will

merely have to learn the identification of the

same sound without taking as long to do so.

0oó000boá

ins

B

Or-:C,t,.

taMS C

tmo

A

tit tt> riti IMP

O N E

Figure 3

Diagram illustrating relative lengths of

dashes and spaces referred to the duration

of o dot. A dash is exactly equal in duration

to three dots; spaces between parts of a

letter equal one dot; those between letters,

three dots; space between words, five dots.

Note that a slight increase between two parts

of a letter will make it sound like two

letters.

Be particularly careful of letters like B.

Many beginners seem to have a tendency to

leave a longer space after the dash than that

which they place between succeeding dots,

thus making it sound like TS. Similarly, make

sure that you do not leave a longer space after

the first dot in the letter C than you do be-

tween other parts of the same letter; otherwise

it will sound like NN.

Sending vs.

Receiving Once you have memorized the

code thoroughly you should con-

centrate on increasing your re-

ceiving speed. True, if you have to practice

with another newcomer who is learning the

code with you, you will both have to do some

sending. But don't attempt to practice sending

just for the sake of increasing your sending

speed.

When transmitting on the code practice set

to your partner so that he can get receiving

practice, concentrate on the quality of your

sending, not on your speed.

Because it is comparatively easy to learn

to send rapidly, especially when no particular

care is given to the quality of sending, many

operators who have just received their licenses

get on the air and send mediocre or worse code

at 20 w.p.m. when they can barely receive

good code at 13. Most oldtimers remember their

own period of initiation and are only too glad

to be patient and considerate if you tell them

that you are a newcomer. But the surest way

to incur their scorn is to try to impress them

with your "lightning speed," and then to re-

quest them to send more slowly when they

come back at you at the same speed.

Stress your copying ability; never stress

your sending ability. It should be obvious that

that if you try to send faster than you can re-

ceive, your ear will not recognize any mis-

takes which your hand may make.

www.americanradiohistory.com

1 8 I n t r o d u c t i o n t o R a d i o T H E R A D I O

Figure 4

PROPER POSITION OF THE FINGERS FOR

OPERATING A TELEGRAPH KEY

The fingers hold the knob and act os a cush-

ion. The hand rests lightly on the key. The

muscles of the forearm provide the power,

the wrist acting as the fulcrum. The power

should not come from the fingers, but rather

from the forearm muscles.

Using the Key Figure 4 shows the proper posi-

tion of the hand, fingers and

wrist when manipulating a telegraph or radio

key. The forearm should rest naturally on the

desk. It is preferable that the key be placed

far enough back from the edge of the table

(about 18 inches) that the elbow can rest on

the table. Otherwise, pressure of the table

edge on the arm will tend to hinder the circu-

lation of the blood and weaken the ulnar nerve

at a point where it is close to the surface,

which in turn will tend to increase fatigue

considerably.

The knob of the key is grasped lightly with

the thumb along the edge; the index and third

fingers rest on the top towards the front or far

edge. The hand moves with a free up and down

motion, the wrist acting as a fulcrum. The

power must come entirely from the arm mus-

cles. The third and index fingers will bend

slightly during the sending but not because of

deliberate effort to manipulate the finger mus-

cles. Keep your finger muscles just tight

enough to act as a cushion for the arm motion

and let the slight movement of the fingers take

care of itself. The key's spring is adjusted to

the individual wrist and should be neither too

stiff nor too loose. Use a moderately stiff ten-

sion at first and gradually lighten it as you

become more proficient. The separation be-

tween the contacts must be the proper amount

for the desired speed, being somewhat under

1/16 inch for slow speeds and slightly closer

together (about 1/32 inch) for faster speeds.

Avoid extremes in either direction.

Do not allow the muscles of arm, wrist, or

fingers to become tense. Send with a full, free

arm movement. Avoid like the plague any fin-

ger motion other than the slight cushioning

effect mentioned above.

Stick to the regular hand key for learning

code. No other key is satisfactory for this pur-

pose. Not until you have thoroughly mastered

both sending and receiving at the maximum

speed in which you are interested should you

tackle any form of automatic or semi -automatic

key such as the Vibroplex ( "bug ") or an elec-

tronic key.

Difficulties Should you experience difficulty

in increasing your code speed

after you have once memorized the characters,

there is no reason to become discouraged. It

is more difficult for some people to learn code

than for others, but there is no justification

for the contention sometimes made that "some

people just can't learn the code." It is not a

matter of intelligence; so don't feel ashamed

if you seem to experience a little more than

the usual difficulty in learning code. Your re-

action time may be a little slower or your co-

ordination not so good. If this is the case,

remember you can still learn the code. You

may never learn to send and receive at 40

w.p.m., but you can learn sufficient speed for

all non -commercial purposes and even for most

commercial purposes if you have patience,

and refuse to be discouraged by the fact that

others seem to pick it up more rapidly.

When the sending operator is sending just

a bit too fast for you (the best speed for prac-

tice), you will occasionally miss a signal or a

small group of them. When you do, leave a

blank space; do not spend time futilely trying

to recall it; dismiss it, and center attention

on the next letter; otherwise you'll miss more.

Do not ask the sender any questions until the

transmission is finished.

To prevent guessing and get equal practice

on the less common letters, depart occasional-

ly from plain language material and use a jum-

ble of letters in which the usually less com-

monly used letters predominate.

As mentioned before, many students put a

greater space after the dash in the letter B

than between other parts of the same letter so

it sounds like TS. C, F, Q, V, X, Y and Z

often give similar trouble. Make a list of words

or arbitrary combinations in which these let-

ters predominate and practice them, both send-

ing and receiving until they no longer give you

trouble. Stop everything e l s e and stick at

them. So long as they give you trouble you are

not ready for anything else.

Follow the same procedure with letters

which you may tend to confuse such as F and

L, which are often confused by beginners.

www.americanradiohistory.com

HANDBOOK Learning the Code 19

Figure 5

THE SIMPLEST CODE PRACTICE

SET CONSISTS OF A KEY AND A

BUZZER

The buzzer is adjusted to give a

steady, high -pitched whine. If de-

sired, the phones may be omitted,

in which case the buzzer should be

mounted firmly on a sounding board.

Crystal, magnetic, or dynamic ear-

phones may be used. Additional

sets of phones should be connected

in parallel, not in series.

1.5 TO S VOLTS

= OF BATTERY

KEY

INEXPENSIVE 500

OHM POTENTIOMETER

VOLUME CONTROL

PHONES.

1 TO 4

PAIR

THESE PARTS REQUIRED

ONLY IF HEADPHONE

OPERATION IS DESIRED

Keep at it until you always get them right

without having to stop even an instant to think

about it.

If you do not instantly recognize the sound

of any character, you have not learned it; go

back and practice your alphabet further. You

should never have to omit writing down every

signal you hear except when the transmission

is too fast for you.

Write down what you hear, not what you

think it should be. It is surprising how often

the word which you guess will be wrong.

Copying Behind All good operators copy sev-

eral words behind, that is,

while one word is being received, they are

writing down or typing, say, the fourth or fifth

previous word. At first this is very difficult,

but after sufficient practice it will be found

actually to be easier than copying close up.

It also results in more accurate copy and en-

ables the receiving operator to capitalize and

CH-722 COLLECTOR

2= BASE

3= EMITTER

REO 00T

10K KEY

0.5 W

2000 n

PHONES

Figure 6

SIMPLE TRANSISTOR CODE

PRACTICE OSCILLATOR

An inexpensive Raytheon CK -722 transistor

requires only a single 11,2 -volt flashlight

battery for power. The inductance of the ear-

phone windings forms part of the oscillatory

circuit. The pitch of the note may be changed

by varying the value of the two capacitors

connected across the earphones.

punctuate copy as he goes along. It is not rec-

ommended that the beginner attempt to do this

until he can send and receive accurately and

with ease at a speed of at least 12 words a

minute.

It requires a considerable amount of train-

ing to dissociate the action of the subcon-

scious mind from the direction of the conscious

mind. It may help some in obtaining this train-

ing to write down two columns of short words.

Spell the first word in the first column out loud

while writing down the first word in the second

column. At first this will be a bit awkward,

but you will rapidly gain facility with practice.

Do the same with all the words, and then re-

verse columns.

Next try speaking aloud the words in the one

column while writing those in the other column;

then reverse columns.

After the foregoing can be done easily, try

sending with your key the words in one col-

umn while spelling those in the other. It won't

be easy at first, but it is well worth keeping

after if you intend to develop any real code

proficiency. Do not attempt to catch up. There

is a natural tendency to close up the gap, and

you must train yourself to overcome this.

Next have your code companion send you a

word either from a list or from straight text;

do not write it down yet. Now have him send

the next word; after receiving this second

word, write down the first word. After receiv-

ing the third word, write the second word; and

so on. Never mind how slowly you must go,

even if it is only two or three words per minute.

Stay behind.

It will probably take quite a number of prac-

tice sessions before you can do this with any

facility. After it is relatively easy, then try

staying two words behind; keep this up until

it is easy. Then try three words, four words,

and five words. The more you practice keep-

ing received material in mind, the easier it

will be to stay behind. It will be found easier

at first to copy material with which one is

fairly familiar, then gradually switch to less

familiar material.

www.americanradiohistory.com

20 Introduction to R adio

Automatic Code The two practice sets which

Machines are described in this chapter

are of most value when you

have someone with whom to practice. Automa-

tic code machines are not recommended to any-

one who can possibly obtain a companion with

whom to practice, someone who is also inter-

ested in learning the code. If you are unable

to enlist a code partner and have to practice

by yourself, the best way to g e t receiving

practice is by the use of a tape machine (auto-

matic code sending machine) with several

practice tapes. Or you can use a set of phono-

graph code practice records. The records are

of use only if you have a phonograph whose

turntable speed is readily adjustable. The tape

machine can be rented by the month for a rea-

sonable fee.

Once you can copy about 10 w.p.m. you can

also get receiving practice by listening to slow

sending stations on your receiver. Many ama-

teur stations send slowly particularly when

working far distant stations. When receiving

conditions are particularly poor many commer-

cial stations also send slowly, sometimes re-

peating every word. Until you can copy around

10 w.p.m. your receiver isn't much use, and

either another operator or a machine or records

are necessary for getting receiving practice

after you have once memorized the code.

Code Practice

Sets If you don't feel too foolish

doing it, you can secure a

measure of code practice with

the help of a partner by sending "dit -dah"

messages to each other while riding to work,

eating lunch, etc. It is better, however, to use

a buzzer or code practice oscillator in con-

junction with a regular telegraph key.

As a good key may be considered an invest-

ment it is wise to make a well -made key your

first purchase. Regardless of what type code

practice set you use, you will need a key, and

later on you will need one to key your trans-

mitter. If you get a good key to begin with,

you won't have to buy another one later.

The key should be rugged and have fairly

heavy contacts. Not only will the key stand

up better, but such a key will contribute to

the "heavy" type of sending so desirable for

radio work. Morse (telegraph) operators use

a "light" style of sending and can send some-

what faster when using this light touch. But,

in radio work static and interference are often

present, and a slightly heavier dot is desir-

able. If you use a husky key, you will find

yourself automatically sending in this manner.

To generate a tone simulating a code signal

as heard on a receiver, either a mechanical

buzzer or an audio oscillator may be used. Fig-

ure 5 shows a simple code -practice set using

a buzzer which may be used directly simply

by mounting the buzzer on a sounding board,

or the buzzer may be used to feed from one to

four pairs of conventional high -impedance

phones.

An example of the audio -oscillator type of

code -practice set is illustrated in figures 6

and 7. An inexpensive Raytheon CK -722 trans-

istor is used in place of the more expensive,

power consuming vacuum tube. A single "pen -

lite" 1i-volt cell powers the unit. The coils

of the earphones form the inductive portion

of the resonant circuit. 'Phones having an

impedance of 2000 ohms or higher should be

used. Surplus type R -14 earphones also work

well with this circuit.

Figure 7

The circuit of Figure 6 is used in this

miniature transistorized code Practice

oscillator. Components are mounted in a

small plastic case. The transistor is

attached to a three terminal phenolic

mounting strip. Sub- miniature jacks are

used for the key and phones connections.

A hearing aid earphone may also be used,

as shown. The phone is stored in the

plastic case when not in use.

www.americanradiohistory.com

CHAPTER TWO

Direct Current Circuits

All naturally occurring matter (excluding

artifically produced radioactive substances) is

made up of 92 fundamental constituents called

elements. These elements can exist either in

the free state such as iron, oxygen, carbon,

copper, tungsten, and aluminum, or in chemi-

cal unions commonly called compounds. The

smallest unit which still retains all the origi-

nal characteristics of an element is the atom.

Combinations of atoms, or subdivisions of

compounds, result in another fundamental

unit, the molecule. The molecule is the small-

est unit of any compound. All reactive ele-

ments when in the gaseous state also exist

in the molecular form, made up of two or more

atoms. The nonreactive gaseous elements

helium, neon, argon, krypton, xenon, and

radon are the only gaseous elements that ever

exist in a stable monatomic state at ordinary

temperatures.

2 -1 The Atom

An atom is an extremely small unit of

matter - there are literally billions of them

making up so small a piece of material as a

speck of dust. To understand the basic theory

of electricity and hence of radio, we must go

further and divide the atom into its main

components, a positively charged nucleus and

a cloud of negatively charged particles that

surround the nucleus. These particles, swirling

around the nucleus in elliptical orbits at an

incredible rate of speed, are called orbital

electrons.

It is upon the behavior of these electrons

when freed from the atom, that depends the

study of electricity and radio, as well as

allied sciences. Actually it is possible to sub-

divide the nucleus of the atom into a dozen or

21

so different particles, but this further sub-

division can be left to quantum mechanics and

atomic physics. As far as the study of elec-

tronics is concerned it is only necessary for

the reader to think of the normal atom as being

composed of a nucleus having a net positive

charge that is exactly neutralized by the one

or more orbital electrons surrounding it.

The atoms of different elements differ in

respect to the charge on the positive nucleus

and in the number of electrons revolving

around this charge. They range all the way

from hydrogen, having a net charge of one

on the nucleus and one orbital electron, to

uranium with a net charge of 92 on the nucleus

and 92 orbital electrons. The number of orbital

electrons is called the atomic number of the

element.

Action of the From the above it must not be

Electrons thought that the electrons re-

volve in a haphazard manner

around the nucleus. Rather, the electrons in

an element having a large atomic number are

grouped into rings having a definite number of

electrons. The only atoms in which these rings

are completely filled are those of the inert

gases mentioned before; all other elements

have one or more uncompleted rings of elec-

trons. If the uncompleted ring is nearly empty,

the element is metallic in character, being

most metallic when there is only one electron

in the outer ring. If the incomplete ring lacks

only one or two electrons, the element is

usually non- metallic. Elements with a ring

about half completed will exhibit both non-

metallic and metallic characteristics; carbon,

silicon, germanium, and arsenic are examples.

Such elements are called semi- conductors.

In metallic elements these outer ring elec-

trons are rather loosely held. Consequently,

www.americanradiohistory.com

22 Direct Current Circuits THE RADIO

there is a continuous helter -skelter movement

of these electrons and a continual shifting

from one atom to another. The electrons which

move about in a substance are called free

electrons, and it is the ability of these elec-

trons to drift from atom to atom which makes

possible the electric current.

Conductors and If the free electrons are nu-

Insulators merous and loosely held, the

element is a good conductor.

On the other hand, if there are few free elec-

trons, as is the case when the electrons in an

outer ring are tightly held, the element is a

poor conductor. If there are virtually no free

electrons, the element is a good insulator.

2 -2 Fundamental Electrical

Units and Relationships

Electromotive Force: The free electrons in

Potential Difference a conductor move con-

stantly about and change

their position in a haphazard manner. To

produce a drift of electrons or electric current

along a wire it is necessary that there be a

difference in "pressure" or potential between

the two ends of the wire. This potential dif-

ference can be produced by connecting a

source of electrical potential to the ends of

the wire.

As will be explained later, there is an ex-

cess of electrons at the negative terminal of

a battery and a deficiency of electrons at the

positive terminal, due to chemical action.

When the battery is connected to the wire, the

deficient atoms at the positive terminal attract

free electrons from the wire in order for the

positive terminal to become neutral. The

attracting of electrons continues through the

wire, and finally the excess electrons at

the negative terminal of the battery are at-

tracted by the positively charged atoms at the

end of the wire. Other sources of electrical

potential (in addition to a battery) are: an

electrical generator (dynamo), a thermocouple,

an electrostatic generator (static machine), a

photoelectric cell, and a crystal or piezo-

electric generator.

Thus it is seen that a potential difference

is the result of a difference in the number of

electrons between the two (or more) points in

question. The force or pressure due to a

potential difference is termed the electro-

motive force, usually abbreviated e. m. f. or

E.M.F. It is expressed in units called volts.

It should be noted that for there to be a

potential difference between two bodies or

points it is not necessary that one have a

positive charge and the other a negative

charge. If two bodies each have a negative

charge, but one more negative than the other,

the one with the lesser negative charge will

act as though it were positively charged with

respect to the other body. It is the algebraic

potential difference that determines the force

with which electrons are attracted or repulsed,

the potential of the earth being taken as the

zero reference point.

The Electric The flow of electrons along a

Current conductor due to the application

of an electromotive force con-

stitutes an electric current. This drift is in

addition to the irregular movements of the

electrons. However, it must not be thought

that each free electron travels from one end

of the circuit to the other. On the contrary,

each free electron travels only a short distance

before colliding with an atom; this collision

generally knocking off one or more electrons

from the atom, which in turn move a short

distance and collide with other atoms, knock-

ing off other electrons. Thus, in the general

drift of electrons along a wire carrying an

electric current, each electron travels only a

short distance and the excess of electrons at

one end and the deficiency at the other are

balanced by the source of the e.m.f. When this

source is removed the state of normalcy re-

turns; there is still the rapid interchange of

free electrons between atoms, but there is no

general trend or "net movement" in either

one direction or the other.

Ampere and There are two units of measure -

Coulomb ment associated with current,

and they are often confused.

The rate of flou of electricity is stated in

amperes. The unit of quantity is the coulomb.

A coulomb is equal to 6.28 x 10" electrons,

and when this quantity of electrons flows by

a given point in every second, a current of

one ampere is said to be flowing. An ampere

is equal to one coulomb per second; a coulomb

is, conversely, equal to one ampere- second.

Thus we see that coulomb indicates amount,

and ampere indicates rate of flow of electric

current.

Older textbooks speak of current flow as

being from the positive terminal of the e.m.f.

source through the conductor to the negative

terminal. Nevertheless, it has long been an

established fact that the current flow in a

metallic conductor is the electronic flow from

the negative terminal of the source of voltage

through the conductor to the positive terminal.

The only exceptions to the electronic direction

of flow occur in gaseous and electrolytic con-

ductors where the flow of positive ions toward

the cathode or negative electrode constitutes

a positive flow in the opposite direction to the

electronic flow. (An ion is an atom, molecule,

www.americanradiohistory.com

HANDBOOK Resistance 23

or particle which either lacks one or more

electrons, or else has an excess of one or

more electrons.)

In radio work the terms "electron flow" and

"current" are becoming accepted as being

synonymous, but the older terminology is still

accepted in the electrical (industrial) field.

Because of the confusion this sometimes

causes, it is often safer to refer to the direc-

tion of electron flow rather than to the direc-

tion of the "current." Since electron flow

consists actually of a passage of negative

charges, current flow and algebraic electron

flow do pass in the same direction.

Resistance The flow of current in a material

depends upon the ease with

which electrons can be detached from the

atoms of the material and upon its molecular

structure. In other words, the easier it is to

detach electrons from the atoms the more free

electrons there will be to contribute to the

flow of current, and the fewer collisions that

occur between free electrons and atoms the

greater will be the total electron flow.

The opposition to a steady electron flow

is called the resistance of a material, and is

one of its physical properties.

The unit of resistance is the ohm. Every

substance has a specific resistance, usually

expressed as ohms per mil -foot, which is deter-

mined by the material' s molecular structure

and temperature. A mil -foot is a piece of

material one circular mil in area and one foot

long. Another measure of resistivity frequently

used is expressed in the units microhms per

centimeter cube. The resistance of a uniform

length of a given substance is directly pro-

portional to its length and specific resistance,

and inversely proportional to its cross- section-

al area. A wire with a certain resistance for a

given length will have twice as much resist-

ance if the length of the wire is doubled. For

a given length, doubling the cross -sectional

area of the wire will halve the resistance,

while doubling the diameter will reduce the

resistance to one fourth. This is true since

the cross -sectional area of a wire varies as

the square of the diameter. The relationship

between the resistance and the linear dimen-

sions of a conductor may be expressed by the

following equation: rl

R =- A

Where

R = resistance in ohms

r = resistivity in Ohms per mil -foot

l = length of conductor in feet

A = cross - sectional area in circular mils

TABLE OF RESISTIVITY

Material

' esist vny in

Ohms per

Circular

Mil -Foot

Temp. Coeff. of

resistance per =C

at 20° C.

Aluminum 17 0.0049

Bross 45 0.003 to 0.007

Cadmium 46 0.0038

Chromium 16 0.00

Copper 10.4 0.0039

Iron 59 0.006

Silver 9.8 0.004

Zinc 36 0.0035

Nichrome 650 0.0002

Const 295 0.00001

Manganin 290 0.00001

Monet 255 0.0019

FIGURE 1

The resistance also depends upon tempera-

ture, increasing with increases in temperature

for most substances (including most metals),

due to increased electron acceleration and

hence a greater number of impacts between

electrons and atoms. However, in the case of

some substances such as carbon and glass the

temperature coefficient is negative and the

resistance decreases as the temperature in-

creases. This is also true of electrolytes. The

temperature may be raised by the external ap-

plication of heat, or by the flow of the current

itself. In the latter case, the temperature is

raised by the heat generated when the electrons

and atoms collide.

Conductors and In the molecular structure of

Insulators many materials such as glass,

porcelain, and mica all elec-

trons are tightly held within their orbits and

there are comparatively few free electrons.

This type of substance will conduct an elec-

tric current only with great difficulty and is

known as an insulator. An insulator is said to

have a high electrical resistance.

On the other hand, materials that have a

large number of free electrons are known as

conductors. Alost metals, those elements which

have only one or two electrons in their outer

ring, are good conductors. Silver, copper, and

aluminum, in that order, are the best of the

common metals used as conductors and are

said to have the greatest conductivity, or low-

est resistance to the flow of an electric

current.

Fundamental These units are the volt,

Electrical Units the ampere, and the ohm.

They were mentioned in the

preceding paragraphs, but were not completely

defined in terms of fixed, known quantities.

The fundamental unit of current, or rate of

flow of electricity is the ampere. A current of

one ampere will deposit silver from a speci-

fied solution of silver nitrate at a rate of

1.118 milligrams per second.

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24 Direct Current Circuits THE RADIO

1111111111111111

lu 1

Figure 2

TYPICAL RESISTORS

Shown above are various types of resistors used in electronic circuits. The larger units are

power resistors. On the left is a variable power resistor. Three precision -type resistors ore

shown in the tenter with two small composition resistors beneath them. At the right is o

composition -type potentiometer, used for audio circuitry.

The international standard for the ohm is

the resistance offered by a uniform column of

mercury at 0°C., 14.4521 grams in mass, of

constant cross - sectional area and 106.300

centimeters in length. The expression megohm

(1,000,000 ohms) is also sometimes used

when speaking of very large values of resist-

ance.

A volt is the e.m.f. that will produce a cur-

rent of one ampere through a resistance of

one ohm. The standard of electromotive force

is the Weston cell which at 20 °C. has a

potential of 1.0183 volts across its terminals.

This cell is used only for reference purposes

in a bridge circuit, since only an infinitesimal

RESISTANCE

vor

CONDUCTORS

BATTERY

Ri B2

-vw-v-

E

_-

Figure 3

SIMPLE SERIES CIRCUITS

At (A) the battery is in series with a single

resistor. At (B) the battery is in series with

two resistors, the resistors themselves being

in series. The arrows indicate the direction of

electron flow.

amount of current may be drawn from it with-

out disturbing its characteristics.

Ohm's Law The relationship between the

electromotive force (voltage),

the flow of current (amperes), and the resist-

ance which impedes the flow of current (ohms),

is very clearly expressed in a simple but

highly valuable law known as Ohm's laun.

This law states that the current in amperes is

equal to the voltage in volts divided by the

resistance in ohms. Expressed as an equation:

E

I =- R

If the voltage (E) and resistance (R) are

known, the current (I) can be readily found.

If the voltage and current are known, and the

resistance is unknown, the resistance (R) is

E

equal to - . When the voltage is the un-

known quantity, it can be found by multiply-

ing I x R. These three equations are all secured

from the original by simple transposition.

The expressions are here repeated for quick

reference:

E

I =- R

E

R=- I E = IR

www.americanradiohistory.com

HANDBOOK Resistive Circuits 25

Figure 4

SIMPLE PARALLEL

CIRCUIT

The two resistors RI and R2 are said to be in

parallel since the flow of current is offered

two parallel paths. An electron leaving point

A will pass either through R1 or R2, but not

through both, to reach the positive terminal

of the battery. If a large number of lectrons

are considered, the greater number will pass

through whichever of the two resistors has

the lower resistance.

where I is the current in amperes,

R is the resistance in ohms,

E is the electromotive force in volts.

Application of All electrical circuits fall in-

Ohm's Law to one of three classes: series

circuits, parallel circuits, and

series -parallel circuits. A series circuit is

one in which the current flows in a single

continuous path and is of the same value at

every point in the circuit (figure 3). In a par-

allel circuit there are two or more current

paths between two points in the circuit, as

shown in figure 4. Here the current divides at

A, part going through R, and part through R2i

and combines at B to return to the battery.

Figure 5 shows a series -parallel circuit. There

are two paths between points A and B as in

the parallel circuit, and in addition there are

two resistances in series in each branch of

the parallel combination. Two other examples

of series -parallel arrangements appear in fig-

ure 6. The way in which the current splits to

flow through the parallel branches is shown by

the arrows.

In every circuit, each of the parts has some

resistance: the batteries or generator, the con-

necting conductors, and the apparatus itself.

Thus, if each part has some resistance, no

matter how little, and a current is flowing

through it, there will be a voltage drop across

it. In other words, there will be a potential

difference between the two ends of the circuit

element in question. This drop in voltage is

equal to the product of the current and the

resistance, hence it is called the IR drop.

The source of voltage has an internal re-

sistance, and when connected into a circuit

so that current flows, there will be an IR drop

in the source just as in every other part of the

circuit. Thus, if the terminal voltage of the

source could be measured in a way that would

cause no current to flow, it would be found

to be more than the voltage measured when a

current flows by the amount of the IR drop

Figure 5

SERIES-PARALLEL

CIRCUIT

In this type of circuit the resistors are ar-

ranged in series groups, and these serlesed

groups ore then placed in parallel.

in the source. The voltage measured with no

current flowing is termed the no load voltage;

that measured with current flowing is the load

voltage. It is apparent that a voltage source

having a low internal resistance is most de-

sirable.

Resistances The current flowing in a series

in Series circuit is equal to the voltage

impressed divided by the total

resistance across which the voltage is im-

pressed. Since the same current flows through

every part of the circuit, it is merely nec-

essary to add all the individual resistances to

obtain the total resistance. Expressed as a

formula:

Riotai =RI +R2 +R, +... +RN .

Of course, if the resistances happened to be

all the same value, the total resistance would

be the resistance of one multiplied by the

number of resistors in the circuit.

Resistances Consider two resistors, one of

in Parallel 100 ohms and one of 10 ohms,

connected in parallel as in fig-

ure 4, with a voltage of 10 volts applied

across each resistor, so the current through

each can be easily calculated.

E = 10 volts

R = 100 ohms

E = 10 volts

R 10 ohms

E

I= -- R

10

I, = = 0.1 ampere

100

10

I2 = -= 1.0 ampere

10

Total current = I, + 12 = 1.1 ampere

Until it divides at A, the entire current of

1.1 amperes is flowing through the conductor

from the battery to A, and again from B through

the conductor to the battery. Since this is more

current than flows through the smaller resistor

it is evident that the resistance of the parallel

combination must be less than 10 ohms, the

resistance of the smaller resistor. We can find

this value by applying Ohm's law.

www.americanradiohistory.com

26 Direct Current Circuits THE RADIO

E

R=- I

E = 10 volts 10

I = 1.1 amperes R 1.1 = 9.09 ohms

The resistance of the parallel combination is

9.09 ohms.

Mathematically, we can derive a simple

formula for finding the effective resistance of

two resistors connected in parallel.

This formula is:

R, x R,

R - R, +R,

where R is the unknown resistance,

R, is the resistance of the first resistor,

R2 is the resistance of the second re-

sistor.

If the effective value required is known,

and it is desired to connect one unknown re-

sistor in parallel with one of known value,

the following transposition of the above for-

mula will simplify the problem of obtaining

the unknown value:

R2 - R, x R

R, -R

where R is the effective value required,

R, is the known resistor,

R2 is the value of the unknown resist-

ance necessary to give R when

in parallel with R,.

The resultant value of placing a number of

unlike resistors in parallel is equal to the re-

ciprocal of the sum of the reciprocals of the

various resistors. This can be expressed as:

R= 1

1 1 1 1

- +- + -+... -

R, R, R, R.

The effective value of placing any number

of unlike resistors in parallel can be deter-

mined from the above formula. However, it

is commonly used only when there are three

or more resistors under consideration, since

the simplified formula given before is more

convenient when only two resistors are being

used. From the above, it also follows that when

two or more resistors of the same value are

placed in parallel, the effective resistance of

the paralleled resistors is equal to the value

of one of the resistors divided by the number

of resistors in parallel.

The effective value of resistance of two or

A

Figure 6

OTHER COMMON SERIES -PARALLEL

CIRCUITS

more resistors connected in parallel is always

less than the value of the lowest resistance in

the combination. It is well to bear this simple

rule in mind, as it will assist greatly in ap-

proximating the value of paralleled resistors.

Resistors in To find the total resistance of

Series Parallel several resistors connected in

series -parallel, it is usually

easiest to apply either the formula for series

resistors or the parallel resistor formula first,

in order to reduce the original arrangement to

a simpler one. For instance, in figure 5 the

series resistors should be added in each

branch, then there will be but two resistors in

parallel to be calculated. Similarly in figure 7,

although here there will be three parallel re-

sistors after adding the series resistors in

each branch. In figure 6B the paralleled re-

sistors should be reduced to the equivalent

series value, and then the series resistance

values can be added.

Resistances in series -parallel can be solved

by combining the series and parallel formulas

into one similar to the following (refer to

figure 7):

R- 1

1 1 1

+--

R,+ R, R, + R, Rs +R6+R,

Voltage Dividers A voltage divider is ex-

actly what its name im-

plies: a resistor or a series of resistors con-

nected across a source of voltage from which

various lesser values of voltage may be ob-

tained by connection to various points along

the resistor.

A voltage divider serves a most useful pur-

pose in a radio receiver, transmitter or ampli-

fier, because it offers a simple means of

obtaining plate, screen, and bias voltages of

different values from a common power supply

www.americanradiohistory.com

HANDBOOK Voltage Divider 27

Figure 7

ANOTHER TYPE OF

SERIES -PARALLEL CIRCUIT

i__

source. It may also be used to obtain very low

voltages of the order of .01 to .001 volt with

a high degree of accuracy, even though a

means of measuring such voltages is lacking.

The procedure for making these measurements

can best be given in the following example.

Assume that an accurately calibrated volt-

meter reading from 0 to 150 volts is available,

and that the source of voltage is exactly 100

volts. This 100 volts is then impressed through

a resistance of exactly 1,000 ohms. It will,

then, be found that the voltage along various

points on the resistor, with respect to the

grounded end, is exactly proportional to the

resistance at that point. From Ohm's law, the

current would be 0.1 ampere; this current re-

mains unchanged since the original value of

resistance (1,000 ohms) and the voltage source

(100 volts) are unchanged. Thus, at a 500 -

ohm point on the resistor (half its entire re-

sistance), the voltage will likewise be halved

or reduced to 50 volts.

The equation (E = I x R) gives the proof:

E = 500 x 0.1 = 50. At the point of 250 ohms

on the resistor, the voltage will be one -fourth

the total value, or 25 volts (E = 250 x 0.1 = 25).

Continuing with this process, a point can be

found where the resistance measures exactly

1 ohm and where the voltage equals 0.1 volt.

It is, therefore, obvious that if the original

source of voltage and the resistance can be

measured, it is a simple matter to predeter-

mine the voltage at any point along the resist-

or, provided that the current remains constant,

and provided that no current is taken from the

tap -on point unless this current is taken into

consideration.

Voltage Divider Proper design of a voltage

Calculations divider for any type of radio

equipment is a relatively

simple matter. The first consideration is the

amount of "bleeder current" to be drawn.

In addition, it is also necessary that the de-

sired voltage and the exact current at each tap

on the voltage divider be known.

Figure 8 illustrates the flow of current in a

simple voltage divider and load circuit. The

light arrows indicate the flow of bleeder cur-

rent, while the heavy arrows indicate the flow

of the load current. The design of a combined

BLEEDER CURRENT

FLOWS BETWEEN

POINTS A AND B EATERNAL

LOAD

Figure 8

SIMPLE VOLTAGE DIVIDER

CIRCUIT

The arrows indicate the manner in which the

current flow divides between the voltage divider

itslf and th externo! load circuit.

bleeder resistor and voltage divider, such as

is commonly used in radio equipment, is illus-

trated in the following example:

A power supply delivers 300 volts and is

conservatively rated to supply all needed cur-

rent for the receiver and still allow a bleeder

current of 10 milliamperes. The following volt-

ages are wanted: 75 volts at 2 milliamperes

for the detector tube, 100 volts at 5 milli-

amperes for the screens of the tubes, and

250 volts at 20 milliamperes for the plates of

the tubes. The required voltage drop across R,

is 75 volts, across R, 25 volts, across R, 150

volts, and across R, it is 50 volts. These

values are shown in the diagram of figure 9.

The respective current values are also indi-

cated. Apply Ohm's law:

E

R, 75

= 01 = 7,500 ohms.

-

E 25 2,083

R, I ohms.

012

E 150

R, = -_ -= 8,823 ohms.

I .017

E 50

R, =- = .037 = 1,351 ohms.

RTotal = 7,500 + 2,083 + 8,823 +

1,351 = 19,757 ohms.

A 20,000 -ohm resistor with three sliding taps

will be of the approximately correct size, and

would ordinarily be used because of the diffi-

culty in securing four separate resistors of the

exact odd values indicated, and because no

adjustment would be possible to compensate

for any slight error in estimating the probable

currents through the various taps.

When the sliders on the resistor once are

set to the proper point, as in the above ex-

www.americanradiohistory.com

28 Direct Current Circuits THE RADIO

300 VOLTS

10 + 2 +5 +20 MA.

50 VOLTS DROP

10 +2 +5 MA

150 VOLTS DROP

10 +2 MA.

25 VOLTS DROP

/ 0V 0MA

BLEEDE75 R CURRENT10 .A.J

VOLTS D,ROP l

POWER SUPPLY - LOA D - - - - -

Figure 9

MORE COMPLEX VOLTAGE DIVIDER

The method for computing the values of the

resistors is discussed in the accompanying text.

ample, the voltages will remain constant at

the values shown as long as the current re-

mains a constant value.

Disadvantages of One of the serious disadvan-

Voltage Dividers rages of the voltage divider

becomes evident when the

the current drawn fromone of the taps changes.

It is obvious that the voltage drops are inter-

dependent and, in turn, the individual drops

are in proportion to the current which flows

through the respective sections of the divider

resistor. The only remedy lies in providing a

heavy steady bleeder current in order to make

the individual currents so small a part of the

total current that any change in current will

result in only a slightchange in voltage. This

can seldom be realized in practice because of

the excessive values of bleeder current which

would be required.

Kirchhoff's Laws Ohm's law is all that is

necessary to calculate the

values in simple circuits, such as the pre-

ceding examples; but in more complex prob-

lems, involving several loops or more than

one voltage in the same closed circuit, the

use of Kirchhoff's laws will greatly simplify

the calculations. These laws are merely rules

for applying Ohm's law.

Kirchhoff's first law is concerned with net

current to a point in a circuit and states that:

At any point in a circuit the current

flowing toward the point is equal to

the current flowing away from the

point.

Stated in another way: if currents flowing to

the point are considered positive, and those

flowing from the point are considered nega-

A

-2 AMPS 1M

(--

RI -1

L1, -2-AMPS

R2

1

AMPi -

Figure 10

ILLUSTRATING KIRCHHOFF'S

FIRST LAW

The current flowing toward point "A" is qual

to the current flowing away from point "A."

tive, the sum of all currents flowing toward

and away from the point - taking signs into

account - is equal to zero. Such a sum is

known as an algebraic sum; such that the law

can be stated thus: The algebraic sum of all

currents entering and leaving a point is zero.

Figure 10 illustrates this first law. Since the

effective resistance of the network of resistors

is 5 ohms, it can be seen that 4 amperes flow

toward point A, and 2 amperes flow away

through the two 5 -ohm resistors in series. The

remaining 2 amperes flow away through the 10-

ohm resistor. Thus, there are 4 amperes flowing

to point A and 4 amperes flowing away from

the point. If R is the effective resistance of

the network (5 ohms), R, = 10 ohms, R, = 5

ohms, R, = 5 ohms, and E = 20 volts, we can

set up the following equation:

E E E =0

R R, R2 +R,

20 20 20

5 10 5 +5

4 -2 -2 =0

Kirchhoff's second law is concerned with

net voltage drop around a closed loop in a

circuit and states that:

In any closed path or loop in a circuit

the sum of the IR drops must equal

the sum of the applied e.m. f.'s.

The second law also may be conveniently

stated in terms of an algebraic sum as: The

algebraic sum of all voltage drops around a

closed path or loop in a circuit is zero. The

applied e.m.f.'s (voltages) are considered

positive, while IR drops taken in the direction

of current flow (including the internal drop

of the sources of voltage) are considered

negative.

Figure 11 shows an example of the applica-

tion of Kirchhoff's laws to a comparatively

simple circuit consisting of three resistors and

www.americanradiohistory.com

HANDBOOK Kirchoff's Laws 29

1. SET VOLTAGE DROPS AROUND EACH LOOP EQUAL TO ZERO.

1121DHMS)+2(t -12)+3 =0 (FIRST LOOP)

-6+2 (12-11) +312 °0 (SECOND LOOP)

2. SIMPLIFY

211+211-212+3.0

411 +3

2 - 1 2 211+6 - I z

5

21a- 2It+31z -6 =0

512- 211 -6 =0

3. EQUATE

411 +3 - 2It +6

2 5

4 SIMPLIFY

2011+15= 411 +12

11 ß-t6 AMPERE

5 RE- SUBSTITUTE

3 2 t

Iz- 2 - 2 I á AMPERE

Figure 11

ILLUSTRATING KIRCHHOFF'S

SECOND LAW

The voltage drop around any closed loop In a

network Is qual to zero.

two batteries. First assume an arbitrary direc-

tion of current flow in each closed loop of the

circuit, drawing an arrow to indicate the as-

sumed direction of current flow. Then equate

the sum of all IR drops plus battery drops

around each loop to zero. You will need one

equation for each unknown to be determined.

Then solve the equations for the unknown cur-

rents in the general manner indicated in figure

11. If the answer comes out positive the di-

rection of current flow you originally assumed

was correct. If the answer comes out negative,

the current flow is in the opposite direction to

the arrow which was drawn originally. This is

illustrated in the example of figure 11 where

the direction of flow of I, is opposite to the

direction assumed in the sketch.

Power in In order to cause electrons

Resistive Circuits to flow through a conductor,

constituting a current flow,

it is necessary to apply an electromotive force

(voltage) across the circuit. Less power is

expended in creating a small current flow

through a given resistance than in creating

a large one; so it is necessary to have a unit

of power as a reference.

The unit of electrical power is the watt,

which is the rate of energy consumption when

an e.m.f. of 1 volt forces a current of 1 ampere

through a circuit. The power in a resistive

circuit is equal to the product of the volt-

age applied across, and the current flowing

in, a given circuit. Hence: P (watts) = E

(volts) x I (amperes).

Since it is often convenient to express

power in terms of the resistance of the circuit

and the current flowing through it, a substi-

tution of IR for E (E = IR) in the above formula

gives: P = IR x I or P = 12R. In terms of volt-

age and resistance, P = E' /R. Here, I = E/R

and when this is substituted for I the original

formula becomes P = E x E /R, or P = E' /R.

To repeat these three expressions:

P = EI, P = I2R, and P = E2 /R,

where P is the power in watts,

E is the electromotive force in volts,

and

I is the current in amperes.

To apply the above equations to a typical

problem: The voltage drop across a cathode

resistor in a power amplifier stage is 50 volts;

the plate current flowing through the resistor

is 150 milliamperes. The number of watts the

resistor will be required to dissipate is found

from the formula: P = El, or 50 x .150 = 7.5

watts (.150 amperes is equal to 150 milli-

amperes). From the foregoing it is seen that

a 7.5 -watt resistor will safely carry the re-

quired current, yet a 10- or 20 -watt resistor

would ordinarily be used to provide a safety

factor.

In another problem, the conditions being

similar to those above, but with the resistance

(R = 333`/2 ohms), and current being the known

factors, the solution is obtained as follows:

P = I2R = .0225 x 333.33 = 7.5. If only the volt-

age and resistance are known, P = E2 /R =

2500/333.33 = 7.5 watts. It is seen that all

three equations give the same results; the

selection of the particular equation depends

only upon the known factors.

Power, Energy It is important to remember

and Work that power (expressed in watts,

horsepower, etc.), represents

the rate of energy consumption or the rate of

doing work. But when we pay our electric bill

Figure 12

MATCHING OF

RESISTANCES

RL

I

To deliver the greatest amount of power to the

load, the load resistance RL should be equal to

the Internal reslstonce of the battery RI.

www.americanradiohistory.com

30 Direct Current Circuits THE RADIO

nErg sm.

Figure 13

TYPICAL CAPACITORS

The two large units ore high value filter capaci-

tors. Shown beneath these ore various types of

by -pass capacitors for r -f and audio application.

to the power company we have purchased a

specific amount of energy or work expressed

in the common units of kilowatt- hours. Thus

rate of energy consumption (watts or kilowatts)

multiplied by time (seconds, minutes or hours)

gives us total energy or work. Other units of

energy are the watt- second, BTU, calorie, erg,

and joule.

Heating Effect Heat is generated when a

source of voltage causes a

current to flow through a resistor (or, for that

matter, through any conductor). As explained

earlier, this is due to the fact that heat is

given off when free electrons collide with the

atoms of the material. More heat is generated

in high resistance materials than in those of

low resistance, since the free electrons must

strike the atoms harder to knock off other

electrons. As the heating effect is a function

of the current flowing and the resistance of

the circuit, the power expended in heat is

given by the second formula: P = I'R.

2 -3 Electrostatics - Capacitors

Electrical energy can be stored in an elec-

trostatic field. A device capable of storing

energy in such a field is called capacitor

(in earlier usage the term condenser was

frequently used but the IRE standards call for

the use of capacitor instead of condenser) and

is said to have a certain capacitance. The

energy stored in an electrostatic field is ex-

pressed in joules (watt seconds) and is equal

to CE' /2, where C is the capacitance in farads

(a unit of capacitance to be discussed) and E

is the potential in volts. The charge is equal

to CE, the charge being expressed in coulombs.

Capacitance and Two metallic plates sep-

Capacitors arated from each other by

a thin layer of insulating

material (called a dielectric, in this case),

becomes a capacitor. When a source of d -c

potential is momentarily applied across these

plates, they may be said to become charged.

If the same two plates are then joined to-

gether momentarily by means of a switch, the

capacitor will discharge.

When the potential was first applied, elec-

trons immediately flowed from one plate to the

other through the battery or such source of

d -c potential as was applied to the capacitor

plates. However, the circuit from plate to

plate in the capacitor was incomplete (the two

plates being separated by an insulator) and

thus the electron flow ceased, meanwhile es-

tablishing a shortage of electrons on one plate

and a surplus of electrons on the other.

Remember that when a deficiency of elec-

trons exists at one end of a conductor, there

is always a tendency for the electrons to move

about in such a manner as to re- establish a

state of balance. In the case of the capacitor

herein discussed, the surplus quantity of elec-

trons on one of the capacitor plates cannot

move to the other plate because the circuit

has been broken; that is, the battery or d -c po-

tential was removed. This leaves the capaci-

tor in a charged condition; the capacitor plate

with the electron deficiency is positively

charged, the other plate being negative.

In this condition, a considerable stress

exists in the insulating material (dielectric)

which separates the two capacitor plates, due

to the mutual attraction of two unlike poten-

tials on the plates. This stress is known as

electrostatic energy, as contrasted with elec-

tromagnetic energy in the case of an inductor.

This charge can also be called potential

energy because it is capable of performing

work when the charge is released through an

external circuit. The charge is proportional to

the voltage but the energy is proportional to

the voltage squared, as shown in the following

analogy.

The charge represents a definite amount of

electricity, or a given number of electrons.

The potential energy possessed by these

electrons depends not only upon their number,

but also upon their potential or voltage.

Compare the electrons to water, and two

capacitors to standpipes, a 1 fifd. capacitor to

www.americanradiohistory.com

HANDBOOK Capacitance 31

SHORTAGE

OF ELECTRONS

A- EILECT

ELD ROSTATIC

- SURPLUS

OF ELECTRONS

CHARGING CURRENT

Figure 14

SIMPLE CAPACITOR

Illustrating the imaginary lines of force repre

Renting the paths along which the repelling force

of the electrons would act on o free electron

located between the two capacitor plates.

a standpipe having a cross section of 1 square

inch and a 2 pfd. capacitor to a standpipe hav-

ing a cross section of 2 square inches. The

charge will represent a given volume of water,

as the "charge" simply indicates a certain

number of electrons. Suppose the water is

equal to 5 gallons.

Now the potential energy, or capacity for

doing work, of the 5 gallons of water will be

twice as great when confined to the 1 sq. in.

standpipe as when confined to the 2 sq. in.

standpipe. Yet the volume of water, or "charge"

is the same in either case.

Likewise a 1 pfd. capacitor charged to 1000

volts possesses twice as much potential

energy as does a 2 pfd. capacitor charged to

500 volts, though the charge (expressed in

coulombs: Q = CE) is the same in either case.

The Unit of Capac- If the external circuit of

the two capacitor plates is

completed by joining the

terminals together with a piece of wire, the

electrons will rush immediately from one plate

to the other through the external circuit and

establish a state of equilibrium. This latter

phenomenon explains the discharge of a capac-

itor. The amount of stored energy in a charged

capacitor is dependent upon the charging po-

tential, as well as a factor which takes into

account the size of the plates, dielectric

thickness, nature of the dielectric, and the

number of plates. This factor, which is de-

termined by the foregoing, is called the capaci-

tanceof a capacitor and is expressed in farads.

The farad is such a large unit of capaci-

tance that it is rarely used in radio calcula-

tions, and the following more practical units

have, therefore, been chosen.

itance: The Farad

1 micro farad = 1 /1,000,000 of a farad, or

.000001 farad, or 10-6 farads.

1 micro- microfarad = 1 /1,000,000 of a micro -

farad, or .000001 microfarad, or 10'6 mi-

cro farads.

1 micro-microlarad = one - millionth of one -

millionth of a farad, or 10'E' farads.

If the capacitance is to be expressed in

microfarads in the equation given for energy

storage, the factor C would then have to be

divided by 1,000,000, thus:

Stored energy in joules - CxE'

2 x 1,000,000

This storage of energy in a capacitor is one

of its very important properties, particularly

in those capacitors which are used in power

supply filter circuits.

Dielectric Although any substance which has

Materials the characteristics of a good in-

sulator may be used as a dielec-

tric material, commercially manufactured ca-

pacitors make use of dielectric materials

which have been selected because their char-

acteristics are particularly suited to the job at

hand. Air is a very good dielectric material,

but an air - spaced capacitor does not have a

high capacitance since the dielectric constant

of air is only slightly greater than one. A

group of other commonly used dielectric mate -

ials is listed in figure 15.

Certain materials, such as bakelite, lucite,

and other plastics dissipate considerable

energy when used as capacitor dielectrics.

1

MATERIAL DIELECTRIC

CONSTANT

1O MC.

POWER

FACTOR

1O MC.

SOFTENING

POINT

FAHRENHEIT

ANILINE- FORMALDEHYDE 3 4 0.004 260

RESIN

BARIUM TITANATE 1200 1.0 -

CASTOR OIL .67

CELLULOSE ACETATE 3.7 0.04 IRO

GLASS.WINDOW 6 -6 POOR 2000

GLASS, PYREX 4.5 0.02

XEL -F FLUOROTHENE U.S 0.6 -

METHYL - METHACRYLATE -

LUCITE 2.6 0.007 160

MICA 5.4 0.0003

MYCALEX, MYKROY 7.0 0.002 650

PHENOL -FORMALDEHYDE, 5.0 0.015 270

LOW-LOSS YELLOW

PHENOL -FORMALDEHYDE 5.5 0.03 330

BLACK BAKELITE

PORCELAIN 7.0 0.005 _2600

POLYETHYLENE 2 25 0.0003 220

POLYSTYRENE 2.55 0.0002 175'

QUARTZ FUSED 4.2 0 0002 2600

RUBBER, HARD-EBONITE 2.6 0.007 150

STEATITE 6.1 0.003 2700'

SULFUR 3.6 0.003 236

TEFLON 2.1 0.02 -

TITANIUM DIOXIDE 100 -175 0.0006 2700

TRANSFORMER OIL 2.2 0.003

UREA -FORMALDEHYDE 5.0 0.05 260

VINYL RESINS 4.0 0.02 200

WOOD. MAPLE . POOR

FIGURE 15

www.americanradiohistory.com

34 Direct Current Circuits THE RADIO

C 1 1 1

1 1 1 1 1 1

- +- - +- - +-

C, C, C, C, C, C,

Capacitors in A -C When a capacitor is con -

and D -C Circuits nected into a direct -cur-

rent circuit, it will block

the d.c., or stop the flow of current. Beyond

the initial movement of electrons during the

period when the capacitor is being charged,

there will be no flow of current because the

circuit is effectively broken by the dielectric

of the capacitor.

Strictly speaking, a very small current may

actually flow because the dielectric of the

capacitor may not be a perfect insulator. This

minute current flow is the leakage current

previously referred to and is dependent upon

the internal d -c resistance of the capacitor.

This leakage current is usually quite notice-

able in most types of electrolytic capacitors.

When an alternating current is applied to

a capacitor, the capacitor will charge and dis-

charge a certain number of times per second

in accordance with the frequency of the alter-

nating voltage. The electron flow in the charge

and discharge of a capacitor when an a -c

potential is applied constitutes an alternating

current, in effect. It is for this reason that a

capacitor will pass an alternating current yet

offer practically infinite opposition to a direct

current. These two properties are repeatedly

in evidence in a radio circuit.

Voltage Rating Any good paper dielectric

of Capacitors filter capacitor has such a

in Series high internal resistance (in-

dicating a good dielectric)

that the exact resistance will vary consider-

ably from capacitor to capacitor even though

they are made by the same manufacturer and

are of the same rating. Thus, when 1000 volts

d.c. is connected across two 1-pfd. 500 -volt

capacitors in series, the chances are that the

voltage will divide unevenly and one capacitor

will receive more than 500 volts and the other

less than 500 volts.

Voltage Equalizing By connecting a half -

Resistors megohm 1 -watt carbon re-

sistor across each capac-

itor, the voltage will be equalized because the

resistors act as a voltage divider, and the

internal resistances of the capacitors are so

much higher (many megohms) that they have

but little effect in disturbing the voltage di-

vider balance.

Carbon resistors of the inexpensive type

are not particularly accurate (not being de-

signed for precision service); therefore it is

EQUAL

CAPACITANCE EQUAL

RESISTANCE

Figure 18

SHOWING THE USE OF VOLTAGE EQUAL-

IZING RESISTORS ACROSS CAPACITORS

CONNECTED IN SERIES

advisable to check several on an accurate

ohmmeter to find two that are as close as

possible in resistance. The exact resistance

is unimportant, just so it is the same for the

two resistors used.

Capacitors in

Series on A.C. When two capacitors are con-

nected in series, alternating

voltage pays no heed to the

relatively high internal resistance of each

capacitor, but divides across the capacitors

in inverse proportion to the capacitance. Be-

cause, in addition to the d.c. across a capac-

itor in a filter or audio amplifier circuit there

is usually an a -c or a -f voltage component, it

is inadvisable to series -connect capacitors

of unequal capacitance even if dividers are

provided to keep the d.c. within the ratings of

the individual capacitors.

For instance, if a 500 -volt 1 -µfd. capacitor

is used in series with a 4-pfd. 500 -volt capac-

itor across a 250 -volt a -c supply, the 1 -µfd.

capacitor will have 200 volts a.c. across it

and the 4-pfd. capacitor only 50 volts. An

equalizing divider to do any good in this case

would have to be of very low resistance be-

cause of the comparatively low impedance of

the capacitors to a.c. Such a divider would

draw excessive current and be impracticable.

The safest rule to follow is to use only

capacitors of the same capacitance and volt-

age rating and to install matched high resist-

ance proportioning resistors across the various

capacitors to equalize the d -c voltage drop

across each capacitor. This holds regardless

of how many capacitors are series -connected.

Electrolytic Electrolytic capacitors use a very

Capacitors thin film of oxide as the dielec-

tric, and are polarized; that is,

they have a positive and a negative terminal

which must be properly connected in a circuit;

otherwise, the oxide will break down and the

capacitor will overheat. The unit then will no

longer be of service. When electrolytic capac-

itors are connected in series, the positive ter-

minal is always connected to the positive lead

of the power supply; the negative terminal of

www.americanradiohistory.com

HANDBOOK M agnetism 35

the capacitor connects to the positive terminal

of the next capacitor in the series combination.

The method of connection for electrolytic ca-

pacitors in series is shown in figure 18. Elec-

trolytic capacitors have very low cost per

microfarad of capacity, but also have a large

power factor and high leakage; both dependent

upon applied voltage, temperature and the age

of the capacitor. The modern electrolytic ca-

pacitor uses a dry paste electrolyte embedded

in a gauze or paper dielectric. Aluminium foil

and the dielectric are wrapped in a circular

bundle and are mounted in a cardboard or metal

box. Etched electrodes may be employed to

increase the effective anode area, and the

total capacity of the unit.

The capacity of an electrolytic capacitor is

affected by the applied voltage, the usage of

the capacitor, and the temperature and humidity

of the environment. The capacity usually drops

with the aging of the unit. The leakage current

and power factor increase with age. At high

frequencies the power factor becomes so poor

that the electrolytic capacitor acts as a series

resistance rather than as a capacity.

2 -4 Magnetism

and Electromagnetism

The common bar or horseshoe magnet is

familiar to most people. The magnetic field

which surrounds it causes the magnet to at-

tract other magnetic materials, such as iron

nails or tacks. Exactly the same kind of mag-

netic field is set up around any conductor

carrying a current, but the field exists only

while the current is flowing.

Magnetic Fields Before a potential, or volt-

age, is applied to a con-

ductor there is no external field, because there

is no general movement of the electrons in

one direction. However, the electrons do pro-

gressively move along the conductor when an

e.m.f. is applied, the direction of motion de-

pending upon the polarity of the e.m.f. Since

each electron has an electric field about it, the

flow of electrons causes these fields to build

up into a resultant external field which acts in

a plane at right angles to the direction in

which the current is flowing. This field is

known as the magnetic field.

The magnetic field around a current -carrying

conductor is illustrated in figure 19. The

direction of this magnetic field depends en-

tirely upon the direction of electron drift or

current flow in the conductor. When the flow

is toward the observer, the field about the

conductor is clockwise; when the flow is away

from the observer, the field is counter- clock-

wise. This is easily remembered if the left

hand is clenched, with the thumb outstretched

ELECTRON DRIFT

.."-SWITCH

Figure 19

LEFT -HAND RULE

Showing the direction of the magnetic lines of

force produced around a conductor carrying an

electric current.

and pointing in the direction of electron flow.

The fingers then indicate the direction of the

magnetic field around the conductor.

Each electron adds its field to the total ex-

ternal magnetic field, so that the greater the

number of electrons moving along the con-

ductor, the stronger will be the resulting field.

One of the fundamental laws of magnetism

is that like poles repel one another and unlike

poles attract one another. This is true of cur-

rent- carrying conductors as well as of perman-

ent magnets. Thus, if two conductors are placed

side by side and the current in each is flowing

in the same direction, the magnetic fields will

also be in the same direction and will combine

to form a larger and stronger field. If the cur-

rent flow in adjacent conductors is in opposite

directions, the magnetic fields oppose each

other and tend to cancel.

The magnetic field around a conductor may

be considerably increased in strength by wind-

ing the wire into a coil. The field around each

wire then combines with those of the adjacent

turns to form a total field through the coil

which is concentrated along the axis of the

coil and behaves externally in a way similar

to the field of a bar magnet.

If the left hand is held so that the thumb

is outstretched and parallel to the axis of a

coil, with the fingers curled to indicate the

direction of electron flow around the turns of

the coil, the thumb then points in the direc-

tion of the north pole of the magnetic field.

The Magnetic In the magnetic circuit, the

Circuit units which correspond to cur-

rent, voltage, and resistance

in the electrical circuit are flux, magneto -

motive force, and reluctance.

Flux, Flux As a current is made up of a drift

Density of electrons, so is a magnetic

field made up of lines of force, and

the total number of lines of force in a given

magnetic circuit is termed the flux. The flux

depends upon the material, cross section, and

length of the magnetic circuit, and it varies

directly as the current flowing in the circuit.

www.americanradiohistory.com

36 Direct Current Circuits THE RADIO

The unit of flux is the maxwell, and the sym-

bol is the Greek letter cp (phi).

Flux density is the number of lines of force

per unit area. It is expressed in gauss if the

unit of area is the square centimeter (1 gauss

= 1 line of force per square centimeter), or

in lines per square inch. The symbol for flux

density is B if it is expressed in gausses, or

B if expressed in lines per square Inch.

magnetomotive The force which produces a

Force flux in a magnetic circuit

is called magnetomotive force.

It is abbreviated m.m.f. and is designated by

the letter F. The unit of magnetomotive force

is the gilbert, which is equivalent to 1.26 x NI,

where N is the number of turns and I is the

current flowing in the circuit in amperes.

The m.m.f. necessary to produce a given

flux density is stated in gilberts per centi-

meter (oersteds) (H), or in ampere -turns per

inch (H).

Reluctance Magnetic reluctance corresponds

to electrical resistance, and is

the property of a material that opposes the

creation of a magnetic flux in the material.

It is expressed in rels, and the symbol is the

letter R. A material has a reluctance of 1 rel

when an m.m.f. of 1 ampere -turn (NI) generates

a flux of 1 line of force in it. Combinations

of reluctances are treated the same as re-

sistances in finding the total effective reluc-

tance. The specific reluctance of any sub-

stance is its reluctance per unit volume.

Except for iron and its alloys, most common

materials have a specific reluctance very

nearly the same as that of a vacuum, which,

for all practical purposes, may be considered

the same as the specific reluctance of air.

Ohm's Law for The relations between flux,

Magnetic Circuits magnetomotive force, and

reluctance are exactly the

same as the relations between current, volt-

age, and resistance in the electrical circuit.

These can be stated as follows:

F

R

F

R F=chR

where ç = flux, F = m.m.f., and R = reluctance.

Permeability Permeability expresses the ease

with which a magnetic field may

be set up in a material as compared with the

effort required in the case of air. Iron, for ex-

ample, has a permeability of around 2000

times that of air, which means that a given

amount of magnetizing effect produced in an

iron core by a current flowing through a coil

of wire will produce 2000 times the flux density

that the same magnetizing effect would pro-

duce in air. It may be expressed by the ratio

B/H or B /H. In other words,

B

R H or B

H

where p is the premeability, B is the flux

density in gausses, B is the flux density in

lines per square inch, H is the m.m.f. in

gilberts per centimeter (oersteds), and H is

the m.m.f. in ampere -turns per inch. These

relations may also be stated as follows:

B B

H=- or H=-, and B=Hit or B= Hµ

fi f

It can be seen from the foregoing that per-

meability is inversely proportional to the

specific reluctance of a material.

Saturation Permeability is similar to electric

conductivity. There is, however,

one important difference: the permeability of

magnetic materials is not independent of the

magnetic current (flux) flowing through it,

although electrical conductivity is substan-

tially independent of the electric current in a

wire. When the flux density of a magnetic

conductor has been increased to the saturation

point, a further increase in the magnetizing

force will not produce a corresponding in-

crease in flux density.

Calculations To simplify magnetic circuit

calculations, a magnetization

curve may be drawn for a given unit of ma-

terial. Such a curve is termed a B -H curve, and

may be determined by experiment. When the

current in an iron core coil is first applied,

the relation between the winding current and

the core flux is shown at A -B in figure 20. If

the current is then reduced to zero, reversed,

brought back again to zero and reversed to the

MAGNETIZING FORCE

H -

Figure 20

TYPICAL HYSTERESIS LOOP

(B -H CURVE = A -B)

Showing relationship between the current in the

winding of on iron core inductor and the core

flux. A direct current flowing through th Induc-

tance brings the magnetic state of the core to

some point on the hysteresis loop, such as C.

www.americanradiohistory.com

HANDBOOK Inductance 37

original direction, the flux passes through a

typical hysteresis loop as shown.

Residual Magnetism; The magnetism remaining

Retentivity in a material after the

magnetizing force is re-

moved is called residual magnetism. Reten-

tivity is the property which causes a magnetic

material to have residual magnetism after

having been magnetized.

Hysteresis; Hysteresis is the character -

Coercive Force istic of a magnetic system

which causes a loss of power

due to the fact that a negative magnetizing

force must be applied to reduce the residual

magnetism to zero. This negative force is

termed coercive /orce. By "negative" mag-

netizing force is meant one which is of the

opposite polarity with respect to the original

magnetizing force. Hysteresis loss is apparent

in transformers and chokes by the heating of

the core.

Inductance If the switch shown in figure 19

is opened and closed, a pulsating

direct current will be produced. When it is

first closed, the current does not instanta-

neously rise to its maximum value, but builds

up to it. While it is building up, the magnetic

field is expanding around the conductor. Of

course, this happens in a small fraction of a

second. If the switch is then opened. the cur-

rent stops and the magnetic field contracts

quickly. This expanding and contracting field

will induce a current in any other conductor

that is part of a continuous circuit which it

cuts. Such a field can be obtained in the way

just mentioned by means of a vibrator inter -

ruptor, or by applying a.c. to the circuit in

place of the battery. Varying the resistance of

the circuit will also produce the same effect.

This inducing of a current in a conductor due

to a varying current in another conductor not

in acutal contact is called electromagnetic in-

duction.

Self -inductance If an alternating current flows

through a coil the varying

magnetic field around each turn cuts itself and

the adjacent turn and induces a voltage in the

coil of opposite polarity to the applied e.m.f.

The amount of induced voltage depends upon

the number of turns in the coil, the current

flowing in the coil, and the number of lines

of force threading the coil. The voltage so

induced is known as a counter -e.m. f. or back -

e.m.f., and the effect is termed self -induction.

When the applied voltage is building up, the

counter- e.m.f. opposes the rise; when the ap-

plied voltage is decreasing, the counter- e.m.f.

is of the same polarity and tends to maintain

the current. Thus, it can be seen that self -

induction tends to prevent any change in the

current in the circuit.

The storage of energy in a magnetic field

is expressed in joules and is equal to (LI3) /2.

(A joule is equal to 1 watt- second. L is de-

fined immediately following.)

The Unit of Inductance is usually denoted by

Inductance; the letter L, and is expressed in

The Henry henrys. A coil has an inductance

of 1 henry when a voltage of 1

volt is induced by a current change of 1 am-

pere per second. The henry, while commonly

used in audio frequency circuits, is too large

for reference to inductance coils, such as

those used in radio frequency circuits; milli-

henry or microhenry is more commonly used,

in the following manner:

1 henry = 1,000 millihenrys, or 10' milli -

henrys.

1 millihenry = 1 /1,000 of a henry, .001 henry,

or 10' henry.

1 microhenry = 1 /1,000,000 of a henry, or

.000001 henry, or 10-e henry.

1 microhenry =1/1,000 of a millihenry, .001

or 10-' millibenrys.

1,000 microbenrys = 1 millihenry.

Mutual Inductance When one coil is near an-

other, a varying current in

one will produce a varying magnetic field

which cuts the turns of the other coil, inducing

a current in it. This induced current is also

varying, and will therefore induce another cur-

rent in the first coil. This reaction between

two coupled circuits is called mutual induction,

and can be calculated and expressed in henrys.

The symbol for mutual inductance is M. Two

circuits thus joined are said to be inductively

coupled.

The magnitude of the mutual inductance de-

pends upon the shape and size of the two cir-

cuits, their positions and distances apart, and

the premeability of the medium. The extent to

u

i I i., I I 2 I

Figure 21

MUTUAL INDUCTANCE

The quantity M represents the mutual inductance

between the two coils L1 and L,.

www.americanradiohistory.com

38 Direct Current Circuits THE RADIO

i-- L ---¡ INDUCTANCE OF

SINGLE- LAYER

SOLENOID COILS

R2 N2

L 9R +10 L MICRONENRIES

WHERE R = RADIUS OF COIL TO CENTER OF WIRE

L = LENGTH OF COIL

N = NUMBER OF TURNS

Figure 22

FORMULA FOR

CALCULATING INDUCTANCE

Through the usa of the equation and the sketch

shown above th inductance of single -layer

solenoid coils can be calculated with an ac-

curacy of about on. per cent for tho types of

coils normally used in the h -f and v -h -f range.

which two inductors are coupled is expressed

by a relation known as coefficient of coupling.

This is the ratio of the mutual inductance ac-

tually present to the maximum possible value.

The formula for mutual inductance is L

L, + L, + 2M when the coils are poled so that

their fields add. When they are poled so that

their fields buck, then L = L, + L, - 2M

(figure 21).

Inductors in Inductors in parallel are corn -

Parallel bined exactly as are resistors in

parallel, provided that they are

far enough apart so that the mutual inductance

is entirely negligible.

Inductors in Inductors in series are additive,

Series just as are resistors in series,

again provided that no mutual

inductance exists. In this case, the total in-

ductance L is:

L = L, + L2 + etc.

Where mutual inductance does exist:

L =L, +L,+ 2M,

where M is the mutual inductance.

This latter expression assumes that the

coils are connected in such a way that all flux

linkages are in the same direction, i.e., ad-

ditive. If this is not the case and the mutual

linkages subtract from the self -linkages, the

following formula holds:

L =L, +L,- 2M,

where M is the mutual inductance.

Core Material Ordinary magnetic cores can-

not be used for radio frequen-

cies because the eddy current and hysteresis

losses in the core material becomes enormous

as the frequency is increased. The principal

use for conventional magnetic cores is in the

audio -frequency range below approximately

15,000 cycles, whereas at very low frequencies

(50 to 60 cycles) their use is mandatory if

an appreciable value of inductance is desired.

An air core inductor of only 1 henry in-

ductance would be quite large in size, yet

values as high as 500 henrys are commonly

available in small iron core chokes. The in-

ductance of a coil with a magnetic core will

vary with the amount of current (both a -c and

d -c) which passes through the coil. For this

reason, iron core chokes that are used in power

supplies have a certain inductance rating at a

predetermined value of d -c.

The premeability of air does not change

with flux density; so the inductance of iron

core coils often is made less dependent upon

flux density by making part of the magnetic

path air, instead of utilizing a closed loop of

iron. This incorporation of an air gap is nec-

essary in many applications of iron core coils,

particularly where the coil carries a consider-

able d -c component. Because the permeability

of air is so much lower than that of iron, the

air gap need comprise only a small fraction of

the magnetic circuit in order to provide a sub-

stantial proportion of the total reluctance.

Iron Cored Inductors Iron -core inductors may

at Radio Frequencies be used at radio frequen-

cies if the iron is in a

very finely divided form, as in the case of the

powdered iron cores used in some types of r -f

coils and i -f transformers. These cores are

made of extremely small particles of iron. The

particles are treated with an insulating mater-

ial so that each particle will be insulated from

the others, and the treated powder is molded

with a binder into cores. Eddy current losses

are greatly reduced, with the result that these

special iron cores are entirely practical in cir-

cuits which operate up to 100 Mc. in frequency.

2 -5 R C and R L Transients

A voltage divider may be constructed as

shown in figure 23. Kirchhoff's and Ohm's

Laws hold for such a divider. This circuit is

known as an RC circuit.

Time Constant - When switch S in figure 23 is

RC and RL placed in position 1, a volt -

Circuits meter across capacitor C will

indicate the manner in which

the capacitor will become charged through the

resistor R from battery B. If relatively large

values are used for R and C, and if a v -t volt-

meter which draws negligible current is used

www.americanradiohistory.com

HANDBOOK Time Constant 39

t001

r

: 60

L < 60

W V40

< ti 20

Wo

rt OTIME t. IN TERMS OF TIME CONSTANT PC'

44 100

<` <

FaA 60

óóZ

030 60

Hi

Iug 40.

4.9 20¡C

óZW -

ózó 0 - -- 1-

á <7 0 2 3

7ló1 TIME t, IN TERMS OF TIME CONSTANT RC

ñ51

Figure 23

TIME CONSTANT OF AN R -C CIRCUIT

Shown at (A) is the circuit upon which is based

the curves of (B) and (C). (8) shows the rate at

which capacitor C will charge from the instant

at which switch S is placed in position 1. (C)

shows the discharge curve of capacitor C from

the instant at which switch S is placed in

position 3.

to measure the voltage e, the rate of charge of

the capacitor may actually be plotted with the

aid of a stop watch.

Voltage Gradient It will be found that the volt-

age e will begin to rise

rapidly from zero the instant the switch is

closed. Then, as the capacitor begins to

charge, the rate of change of voltage across

the capacitor will be found to decrease, the

charging taking place more and more slowly

as the capacitor voltage e approaches the bat-

tery voltage E. Actually, it will be found that

in any given interval a constant percentage of

the remaining difference between e and E

will be delivered to the capacitor as an in-

crease in voltage. A voltage which changes in

this manner is said to increase logarithmically,

or is said to follow an exponential curve.

Time Constant A mathematical analysis of

the charging of a capacitor in

this manner would show that the relationship

between the battery voltage E and the voltage

across the capacitor e could be expressed in

the following manner:

e = E (1 _ f -t /Rc)

where e,E,R, and C have the values discussed

above. f = 2.716 (the base of Naperian or

natural logarithms), and t represents the time

which has elapsed since the closing of the

switch. With t expressed in seconds, R and C

Figure 24

TYPICAL INDUCTANCES

The large inductance is a 1000 -watt transmitting coil. To the right and left of this coil are small r -f

chokes. S I varieties of low power capability coils are shown below, along with various types of r -f

chokes intended for high- frequency operation.

www.americanradiohistory.com

40 Direct Current Circuits

i4cc

R (INCLUDING D.C. RESISTANCE

OF INDUCTOR L)

TIME t, IN TERMS OF TIME CONSTANT }

Figure 25

TIME CONSTANT OF AN R -L CIRCUIT

Nota that the time constant for the Increase In

current through an R-L. circuit Is identical to

the rate of Increase in voltage across the

capacitor In on R -C circuit.

may be expressed in farads and ohms, or R

and C may be expressed in microfarads and

megohms. The product RC is called the time

constant of the circuit, and is expressed in

seconds. As an example, if R is one megohm

and C is one microfarad, the time constant

RC will be equal to the product of the two,

or one second.

When the elapsed time t is equal to the

time constant of the RC network under con-

sideration, the exponent of E becomes -1.

Now e' is equal to 1 /e, or 1/2.716, which

is 0.368. The quantity (1 - 0.368) then is equal

to 0.632. Expressed as percentage, the above

means that the voltage across the capaci-

tor will have increased to 63.2 per cent of

the battery voltage in an interval equal to the

time constant or RC product of the circuit.

Then, during the next period equal to the time

constant of the RC combination, the voltage

across the capacitor will have risen to 63.2

per cent of the remaining difference in voltage,

or 86.5 per cent of the applied voltage E.

RL Circuit In the case of a series combination

of a resistor and an inductor, as

shown in figure 25, the current through the

combination follows a very similar law to that

given above for the voltage appearing across

the capacitor in an RC series circuit. The

equation for the current through the combina-

tion is:

E

i=- (1-E-tR/L)

where i represents the current at any instant

through the series circuit, E represents the

applied voltage, and R represents the total

resistance of the resistor and the d -c resist-

ance of the inductor in series. Thus the time

constant of the RL circuit is L /R, with R ex-

pressed in ohms and L expressed in henrys.

Voltage Decoy When the switch in figure 23 is

moved to position 3 after the

capacitor has been charged, the capacitor volt-

age will drop in the manner shown in figure

23 -C. In this case the voltage across the ca-

pacitor will decrease to 36.8 per cent of the

initial voltage (will make 63.2 per cent of the

total drop) in a period of time equal to the

time constant of the RC circuit.

TYPICAL IRON -CORE INDUCTANCES

At the right is an upright mounting filter choke intended for use in low powered trans-

mitters and audio equipment. At the center is o hermetically sealed inductance for use

under poor environmental conditions. To the left is an inexpensive receiving -type choke,

with a small iron -core r -f choke directly in front of it.

www.americanradiohistory.com

CHAPTER THREE

Alternating Current Circuits

The previous chapter has been devoted to

a discussion of circuits and circuit elements

upon which is impressed a current consisting

of a flow of electrons in one direction. This

type of unidirectional current flow is called

direct current, abbreviated d. c. Equally as im-

portant in radio and communications work,

and power practice, is a type of current flow

whose direction of electron flow reverses

periodically. The reversal of flow may take

place at a low rate, in the case of power sys-

tems, or it may take place millions of times

per second in the case of communications

frequencies. This type of current flow is

called alternating current, abbreviated a. c.

3 -1 Alternating Current

Frequency of on An alternating current is

one whose amplitude of

current flow periodically

rises from zero to a maximum in one direction,

decreases to zero, changes its direction,

rises to maximum in the opposite direction,

and decreases to zero again. This complete

process, starting from zero, passing through

two maximums in opposite directions, and re-

turning to zero again, is called a cycle. The

number of times per second that a current

passes through the complete cycle is called

the frequency of the current. One and one

quarter cycles of an alternating current wave

are illustrated diagrammatically in figure 1.

Alternating Current

41

Frequency Spectrum At present the usable fre-

quency range for alternat-

ing electrical currents extends over the enor-

mous frequency range from about 15 cycles per

second to perhaps 30,000,000,000 cycles per

second. It is obviously cumbersome to use a

frequency designation in c.p.s. for enormously

high frequencies, so three common units which

are multiples of one cycle per second have

been established.

z 4-

Y.1

¢ K

U a

I- z w a

CC J U

DIRECT CURRENT

t CYCLE

-i CYCLE -01

ALTERNATING CURRENT

TIME-41.

TIME -

Figure 1

ALTERNATING CURRENT

AND DIRECT CURRENT

Graphical comparison between unidrectionai

(direct) current and alternating current as plotted

against time.

www.americanradiohistory.com

42 Alternating Current Circuits THE RADIO

These units are:

(1) the kilocycle (abbr., kc.), 1000 c.p.s.

(2) the Megacycle (abbr., Mc.), 1,000,000

c.p.s. or 1000 kc.

(3) the kilo -Megacycle (abbr., kN1c.),

1,000,000,000 c.p.s. or 1000 Mc.

With easily handled units such as these we

can classify the entire usable frequency range

into frequency bands.

The frequencies falling between about 15

and 20,000 c.p.s. are called audio frequencies,

abbreviated a.f., since these frequencies are

audible to the human ear when converted from

electrical to acoustical signals by a loud-

speaker or headphone. Frequencies in the

vicinity of 60 c.p.s. also are called power fre-

quencies, since they are commonly used to

distribute electrical power to the consumer.

The frequencies falling between 10,000

c.p.s. (10 kc.) and 30,000,000.000 c.p.s. (30

kMc.) are commonly called radio frequencies,

abbreviated r. J., since they are commonly used

in radio communication and allied arts. The

radio- frequency spectrum is often arbitrarily

classified into seven frequency bands, each

one of which is ten times as high in frequency

as the one just below it in the spectrum (ex-

cept for the v -1 -f band at the bottom end of

the spectrum). The present spectrum, with

classifications, is given below.

Frequency Classification Abbrev.

10 to 30 kc. Very -low frequencies v.l.f

30 to 300 kc. Low frequencies l.f.

300 to 3000 kc. Medium frequencies m.f.

3 to 30 Mc. High frequencies h. f.

30 to 300 Mc. Very -high frequencies v.h.f.

300 to 3000 Mc. Ultra -high frequencies u.h.f.

3 to 30 kMc. Super -high frequencies s.h.f.

30 to 300 kMc. Extremely -high

frequencies e.h.f.

Generation of Faraday discovered that if

Alternating Current a conductor which forms

part of a closed circuit is

moved through a magnetic field so as to cut

across the lines of force, a current will flow

in the conductor. He also discovered that, if a

conductor in a second closed circuit is brought

near the first conductor and the current in the

first one is varied, a current will flow in the

second conductor. This effect is known as

induction, and the currents so generated are

induced currents. In the latter case it is the

lines of force which are moving and cutting

the second conductor, due to the varying cur-

rent strength in the first conductor.

A current is induced in a conductor if there

is a relative motion between the conductor

and a magnetic field, its direction of flow de-

pending upon the direction of the relative

Figure 2

THE ALTERNATOR

Semi -schematic representation of the simplest

form of the alternator.

motion between the conductor and the field,

and its strength depends upon the intensity of

the field, the rate of cutting lines of force, and

the number of turns in the conductor.

Alternators A machine that generates an alter-

nating current is called an alter-

nator or a -c generator. Such a machine in its

basic form is shown in figure 2. It consists of

two permanent magnets, M. the opposite poles

of which face each other and are machined so

that they have a common radius. Between

these two poles, north (N) and south (S),

a substantially constant magnetic field exists.

If a conductor in the form of C is suspended

so that it can be freely rotated between the

two poles, and if the opposite ends of con-

ductor C are brought to collector rings, there

will be a flow of alternating current when con-

ductor C is rotated. This current will flow out

through the collector rings R and brushes B

to the external circuit, X -Y.

The field intensity between the two pole

pieces is substantially constant over the entire

area of the pole face However, when the

conductor is moving parallel to the lines of

force at the top or bottom of the pole faces,

no lines are being cut. As the conductor moves

on across the pole face it cuts more and more

lines of force for each unit distance of travel,

until it is cutting the maximum number of

lines when opposite the center of the pole.

Therefore, zero current is induced in the con-

ductor at the instant it is midway between

the two poles, and maximum current is in-

duced when it is opposite the center of the

pole face. After the conductor has rotated

through 180° it can be seen that its position

with respect to the pole pieces will be exactly

opposite to that when it started. Hence, the

second 180° of rotation will produce an alter-

nation of current in the opposite direction to

that of the first alternation.

The current does not increase directly as

the angle of rotation, but rather as the sine

of the angle; hence, such a current has the

mathematical form of a sine wave. Although

www.americanradiohistory.com

HANDBOOK The Sine /lave 43

LINES OF FORCE

E

A B C O E

LINES OF FORCE

(UNIFORM DENSITY

Figure 3

OUTPUT OF THE ALTERNATOR

Graph showing sine -wave output current of the

alternator of figure 2.

most electrical machinery does not produce a

strictly pure sine curve, the departures are

usually so slight that the assumption can be

regarded as fact for most practical purposes.

All that has been said in the foregoing para-

graphs concerning alternating current also is

applicable to alternating voltage.

The rotating arrow to the left in figure 3

represents a conductor rotating in a constant

magnetic field of uniform density. The arrow

also can be taken as a vector representing the

strength of the magnetic field. This means that

the length of the arrow is determined by the

strength of the field (number of lines of force),

which is constant. Now if the arrow is rotating

at a constant rate (that is, with constant

angular velocity), then the voltage developed

across the conductor will be proportional to

the rate at which it is cutting lines of force,

which rate is proportional to the vertical

distance between the tip of the arrow and the

horizontal base line.

If EO is taken as unity or a voltage of 1,

then the voltage (vertical distance from tip of

arrow to the horizontal base line) at point C

for instance may be determined simply by

referring to a table of sines and looking up the

sine of the angle which the arrow makes with

the horizontal.

When the arrow has traveled from A to point

E, it has traveled 90 degrees or one quarter

cycle. The other three quadrants are not shown

because their complementary or mirror relation-

ship to the first quadrant is obvious.

It is important to note that time units are

represented by degrees or quadrants. The fact

that AB, BC, CD, and DE are equal chords

(forming equal quadrants) simply means that

the arrow (conductor or vector) is traveling

at a constant speed, because these points on

the radius represent the passage of equal

units of time.

The whole picture can be represented in

another way, and its derivation from the fore-

going is shown in figure 3. The time base is

represented by a straight line rather than by

t CYCLE

le2 CYCLE HM - --+ CYCLE- w

90

60 t20

30 ISO

te0

a 2

t CYCLE' +-

3 CYCLE'

3

2143 330

240 300

2t0

WHERE F = FREQUENCY IN CYCLES

-- TIMES

Figure 4

THE SINE WAVE

illustrating one cycle of o sine wave. One

complete cycle of alternation is broken up

into 360 degrees. Then one -half cycle is 180

degrees, one -quarter cycle is 90 degrees, and

so on down to the smallest division of the

wave. A cosine wave has a shape identical to

a sine wave but is shifted 90 degrees in phase

- In other words the wove begfna at full am

plilude, the 90- degree point comes at zero am-

plitude, the 180 -degree point comes at full

amplitude in the opposite direction of current

How, etc.

angular rotation. Points A, B, C, etc., rep-

resent the same units of time as before. When

the voltage corresponding to each point is

projected to the corresponding time unit, the

familiar sine curve is the result.

The frequency of the generated voltage is

proportional to the speed of rotation of the

alternator, and to the number of magnetic poles

in the field. Alternators may be built to produce

radio frequencies up to 30 kilocycles, and

some such machines are still used for low

frequency communication purposes. By means

of multiple windings, three -phase output may

be obtained from large industrial alternators.

Radian Notation From figure 1 we see that the

value of an a -c wave varies

continuously. It is often of importance to know

the amplitude of the wave in terms of the

total amplitude at any instant or at any time

within the cycle. To be able to establish the

instant in question we must be able to divide

the cycle into parts. We could divide the cycle

into eighths, hundredths, or any other ratio that

suited our fancy. However, it is much more

convenient mathematically to divide the cycle

either into electrical degrees (360° represent

one cycle) or into radians. A radian is an arc

of a circle equal to the radius of the circle;

hence there are 2n radians per cycle-or per

circle (since there are n diameters per circum-

ference, there are 2rr radii).

Both radian notation and electrical degree

www.americanradiohistory.com

44 Alternating Current Circuits THE RADIO

notation are used in discussions of alternating

current circuits. However, trigonometric tables

are much more readily available in terms of

degrees than radians, so the following simple

conversions are useful.

2n radians = 1 cycle = 3600

n radians ='/2 cycle = 180°

n - radians ='4 cycle = 90°

2

n

- radians ='4 cycle = 60°

3

n

- 4 radians =' /R cycle = 45°

1

1 radian = -cycle = 57.3°

2n

When the conductor in the simple alter-

nator of figure 2 has made one complete revo-

lution it has generated one cycle and has ro-

tated through 2n radians. The expression 2nf

then represents the number of radians in one

cycle multiplied by the number of cycles per

second (the frequency) of the alternating

voltage or current. The expression then repre-

sents the number of radians per second through

which the conductor has rotated. Hence 27rf

represents the angular velocity of the rotating

conductor, or of the rotating vector which

represents any alternating current or voltage,

expressed in radians per second.

In technical literature the expression 2nf

is often replaced by al, the lower -case Greek

letter omega. Velocity multiplied by time

gives the distance travelled. so 2nft (or an)

represents the angular distance through which

the rotating conductor or the rotating vector

has travelled since the reference time t = 0.

In the case of a sine wave the reference time

t = 0 represents that instant when the voltage

or the current, whichever is under discussion,

also is equal to zero.

Instantaneous Value The instantaneous volt -

of Voltage or age or current is propor-

Current tional to the sine of the

angle through which the

rotating vector has travelled since reference

time t = 0. Hence, when the peak value of the

a -c wave amplitude (either voltage or cur-

rent amplitude) is known, and the angle through

which the rotating vector has travelled is

established, the amplitude of the wave at

this instant can be determined through use

of the following expression:

e = Erna: sin left,

B

A

o

WHERE

e (THETA). PHASE ANGLE .277F T

A = RADIANS OR 90'

B'/r RADIANS OR IRO

T RADIANS OR 270

D. 2 A RADIANS OR 350

t RADIAN a 57.324 DEGREES

Figure 5

ILLUSTRATING RADIAN NOTATION

The radian is a unit of phase angle, equal to

57.324 degrees. It is commonly used in mathe-

matical relationships involving phase angles

since such relationships are simplified when

radian notation is used

where e = the instantaneous voltage

E = maximum crest value of voltage,

f = frequency in cycles per second, and

t = period of time which has elapsed

since t = 0 expressed as a fraction

of one second.

The instantaneous current can be found from

the same expression by substituting i for e

and Imax for Emaz.

It is often easier to visualize the process of

determining the instantaneous amplitude by

ignoring the frequency and considering only

one cycle of the a -c wave. In this case, for a

sine wave, the expression becomes:

e=Ema: sin 9

where O represents the angle through which

the vector has rotated since time (and ampli-

tude) were zero. As examples:

when 0 = 30°

sin O = 0.5

so e = 0.5 Emu

when O = 60°

sin O = 0.866

so e = 0.866 Erna:

when O = 90°

sin 0 = 1.0

so e = Emax

when O = 1 radian

sin O = 0.8415

so e = 0.8415 Etna:

www.americanradiohistory.com

H A N D B O O K A-C Relationships 45

Effective Value The instantaneous value

of an of an alternating current

Alternating Current or voltage varies continu-

ously throughout the cycle.

So some value of an a -c wave must be chosen

to establish a relationship between the effec-

tiveness of an a -c and a d -c voltage or cur -

rent/ The heating value of an alternating

current has been chosen to establish the refer-

ence between the effective values of a.c. and

d.c. Thus an alternating current will have an

effective value of 1 ampere when it produces

the same heat in a resistor as does 1 ampere

of direct current.

The effective value is derived by taking the

instantaneous values of current over a cycle of

alternating current, squaring these values.

taking an average of the squares, and then

taking the square root of the average. By this

procedure, the effective value becomes known

as the root mean square or r.m.s. value. This

is the value that is read on a -c voltmeters and

a -c ammeters. The r.m.s. value is 70.7 (for

sine waves only) per cent of the peak or maxi-

mum instantaneous value and is expressed as

follows:

Eetf. or Er.m.s. = 0.707 x Erna: or

left. or Ir.m.s. = 0.707 x !ma:.

The following relations are extremely useful

in radio and power work:

Er m. s. = 0.707 x &max, and

Ems = 1.414 x Er.m.s.

Rectified Alternating If an alternating current

Current or Pulsat- is passed through a recti-

ing Direct Current fier, it emerges in the

form of a current of

varying amplitude which flows in one direc-

tion only. Such a current is known as rectified

a. c. or pulsating d. c. A typical wave form of a

pulsating direct current as would be obtained

from the output of a full -wave rectifier is

shown in figure 6.

Measuring instruments designed for d -c

operation will not read the peak for instantan-

eous maximum value of the pulsating d -c out-

put from the rectifier; they will read only the

average value. This can be explained by as-

suming that it could be possible to cut off

some of the peaks of the waves, using the cut-

off portions to fill in the spaces that are open,

thereby obtaining an average d -c value. A

milliammeter and voltmeter connected to the

adjoining circuit, or across the output of the

rectifier, will read this average value. It is re-

lated to peak value by the following expres-

sion:

Eavg = 0.636 x Fina:

Figure 6

FULL -WAVE RECTIFIED

SINE WAVE

Waveform obtained at the output of a fullwave

rectifier being fed with a sine wave and having

100 per cent rectification efficiency. Each

pulse has the same shape os one -half cycle of

a sine wave. This type of current is known as

pulsating direct current.

It is thus seen that

per cent of the peak

Relationship Between

Peak, R.M.S. or

Effective, and

Average Values

the average value is 63.6

value.

To summarize the three

most significant values

of an a -c sine wave: the

peak value is equal to

1.41 times the r.m.s. or

effective, and the r.m.s. value is equal to

0.707 times the peak value; the average value

of a full -wave rectified a -c wave is 0.636

times the peak value, and the average value

of a rectified wave is equal to 0.9 times the

r.m.s. value.

R.M.S. = 0.707 x Peak

Average = 0.636 x Peak

Average = 0.9 x R.M.S.

R.M.S. = 1.11 x Average

Peak = 1.414 x R.M.S.

Peak = 1.57 x Average

Applying Ohm's Law Ohm's law applies

equally to direct or al-

ternating current, pro-

vided the circuits under consideration are

purely resistive, that is, circuits which have

neither inductance (coils) nor capacitance

(capacitors). Problems which involve tube

filaments, drop resistors, electric lamps,

heaters or similar resistive devices can be

solved from Ohm's law, regardless of whether

the current is direct or alternating. When a

capacitor or coil is made a part of the circuit,

a property common to either, called reactance,

must be taken into consideration. Ohm's law

still applies to a -c circuits containing react-

ance, but additional considerations are in-

volved; these will be discussed in a later

paragraph.

to Alternating Current

www.americanradiohistory.com

46 Alternating Current Circuits THE RADIO

E

TIME

CURRENT LAGGING VOLTAGE BY 90°

(CIRCUIT CONTAINING PURE INDUCTANCE ONLY)

Figure 7

LAGGING PHASE ANGLE

Showing the manner in which the current lags

the voltage in an a -c circuit containing pure

inductance only. The lag is equal to one -quarter

cycle or 90 degrees.

TIME

CURRENT LEADING VOLTAGE BY 90°

(CIRCUIT CONTAINING PURE CAPACITANCE ONLY)

Figure 8

LEADING PHASE ANGLE

Showing the manner in which the current leads

the voltage in on o -c circuit containing pure

capacitance only. The lead is equal to one -

quarter cycle or 90 degrees.

Inductive As was stated in Chapter Two,

Reactance when a changing current flows

through an inductor a back- or

counter -electromotive force is developed,

opposing any change in the initial current.

This property of an inductor causes it to offer

opposition or impedance to a change in cur-

rent. The measure of impedance offered by an

inductor to an alternating current of a given

frequency is known as its inductive reactance.

This is expressed as XL.

XL = 2rrf L,

where XL = inductive reactance expressed in

ohms.

n= 3.1416 (2n= 6.283),

f = frequency in cycles,

L = inductance in henrys.

Inductive Reactance It is very often neces-

at Rodio Frequencies cary to compute induc-

tive reactance at radio

frequencies. The same formula may be used,

but to make it less cumbersome the inductance

is expressed in millihenrys and the frequency

in kilocycles. For higher frequencies and

smaller values of inductance, frequency is

expressed in megacycles and inductance in

microhenrys. The basic equation need not be

changed, since the multiplying factors for

inductance and frequency appear in numerator

and denominator, and hence are cancelled out.

However, it is not possible in the same equa-

tion to express L in millihenrys and f in cycles

without conversion factors.

Capacitive It has been explained that induc-

Reactance tive reactance is the measure of

the ability of an inductor to offer

impedance to the flow of an alternating current.

Capacitors have a similar property although

in this case the opposition is to any change in

the voltage across the capacitor. This property

is called capacitive reactance and is ex-

pressed as follows:

1

Xc 2nfC

where Xc = capacitive reactance in ohms,

n = 3.1416

f = frequency in cycles,

C = capacitance in farads.

Capacitive Re-

actance at

Radio Frequencies

into smaller units

countered in radio

be written:

Here again, as in the case

of inductive reactance, the

units of capacitance and

frequency can be converted

for practical problems en-

work. The equation may

1,000,000

Xc 2nfC

where f = frequency in megacycles,

C w capacitance in micro- microfarads.

In the audio range it is often convenient to

express frequency (f) in cycles and capac-

itance (C) in micro /arads, in which event the

same formula applies.

Phase When an alternating current flows

through a purely resistive circuit, it

will be found that the current will go through

maximum and minimum in perfect step with

the voltage. In this case the current is said to

be in step or in phase with the voltage. For

this reason, Ohm's law will apply equally well

for a. c. or d. c. where pure resistances are con-

cerned, provided that the same values of the

www.americanradiohistory.com

HANDBOOK Reactance 47

wave (either peak or r.m.s.) for both voltage

and current are used in the calculations.

However, in calculations involving alternat-

ing currents the voltage and current are not

necessarily in phase. The current through the

circuit may lag behind the voltage, in which

case the current is said to have lagging phase.

Lagging phase is caused by inductive react-

ance. If the current reaches its maximum value

ahead of the voltage (figure 8) the current is

said to have a leading phase. A leading phase

angle is caused by capacitive reactance.

In an electrical circuit containing reactance

only, the current will either lead or lag the

voltage by 90 °. If the circuit contains induc-

tive reactance only, the current will lag the

voltage by 90 °. If only capacitive reactance is

in the circuit, the current will lead the voltage

by 90 °.

Reactances Inductive and capacitive re-

in Combination actance have exactly opposite

effects on the phase relation

between current and voltage in a circuit.

Hence when they are used in combination

their effects tend to neutralize. The combined

effect of a capacitive and an inductive react-

ance is often called the net reactance of a

circuit. The net reactance (X) is found by sub-

tracting the capacitive reactance from the in-

ductive reactance, X = XL Xc.

The result of such a combination of pure

reactances may be either positive, in which

case the positive reactance is greater so that

the net reactance is inductive, or it may be

negative in which case the capacitive react-

ance is greater so that the net reactance is

capacitive. The net reactance may also be

zero in which case the circuit is said to be

resonant. The condition of resonance will be

discussed in a later section. Note that induc-

tive reactance is always taken as being posi-

tive while capacitive reactance is always

taken as being negative.

Impedance; Circuits Pure reactances intro -

Containing Reactance duce a phase angle of

and Resistance 90° between voltage and

current; pure resistance

introduces no phase shift between voltage and

current. Hence we cannot add a reactance and

a resistance directly. When a reactance and a

resistance are used in combination the re-

sulting phase angle of current flow with re-

spect to the impressed voltage lies somewhere

between plus or minus 90° and 0° depending

upon the relative magnitudes of the reactance

and the resistance.

The term impedance is a general term which

can be applied to any electrical entity which

impedes the flow of current. Hence the term

may be used to designate a resistance, a pure

Y-AniS

Figure 9

Operation on the vector (+A) by the quantity ( -1)

vector to rotate through 180 degrees.

reactance, or a complex combination of both

reactance and resistance. The designation for

impedance is Z. An impedance must be de-

fined in such a manner that both its magnitude

and its phase angle are established. The

designation may be accomplished in either of

two ways-one of which is convertible into

the other by simple mathematical operations.

The "J" Operator The first method of des-

ignating an impedance is

actually to specify both the resistive and the

reactive component in the form R + jX. In this

form R represents the resistive component in

ohms and X represents the reactive component.

The "j" merely means that the X component

is reactive and thus cannot be added directly

to the R component. Plus jX means that the

reactance is positive or inductive, while if

minus jX were given it would mean that the

reactive component was negative or capacitive.

In figure 9 we have a vector ( +A) lying along

the positive X -axis of the usual X -Y coordi-

nate system. If this vector is multiplied by the

quantity ( -1), it becomes ( A) and its position

now lies along the X -axis in the negative

direction. The operator ( -1) has caused the

vector to rotate through an angle of 180 de-

grees. Since ( -1) is equal to (V -1 x V77-1), the

same result may be obtained by operating on

the vector with the operator (VIT x V-1).

However if the vector is operated on but once

by the operator (V-1), it is caused to rotate

only 90 degrees (figure 10). Thus the operator

( V- 11)rotates a vector by 90 degrees. For con-

venience, this operator is called the j operator.

In like fashion, the operator ( j) rotates the

vector of figure 9 through an angle of 270

degrees, so that the resulting vector ( jA)

falls on the ( Y) axis of the coordinate system.

www.americanradiohistory.com

48 Alternating Current Circuits THE RADIO

Y-AXIS

( +A) X (AI ) ROTATES

VECTOR THROUGH 90

+jA k b

A X AXIS

Figure 10

Operation on the vector ( +A) by the quantity (j)

causes vector to rotate through 90 degrees.

Polar Notation The second method of repre-

senting an impedance is to

specify its absolute magnitude and the phase

angle of current with respect to voltage, in

the form Z L O. Figure 11 shows graphically

the relationship between the two common ways

of representing an impedance.

The construction of figure 11 is called an

impedance diagram. Through the use of such

a diagram we can add graphically a resistance

and a reactance to obtain a value for the re-

sulting impedance in the scalar form. With

zero at the origin, resistances are plotted to

the right, positive values of reactance (induc-

tive) in the upward direction, and negative

values of reactance (capacitive) in the down-

ward direction.

Note that the resistance and reactance are

drawn as the two sides of a right triangle,

with the hypotenuse representing the resulting

impedance. Hence it is possible to deter-

mine mathematically the value of a resultant

impedance through the familiar right -triangle

relationship-the square of the hypotenuse is

equal to the sum of the squares of the other

two sides:

Z2 =R' +X2

or IZI = ß/R2 + X

Note also that the angle O included between

R and Z can be determined from any of the

following trigonometric relationships:

X

sin o Z

R

cos O = Z

X

tan O = - R

One common problem is that of determining

the scalar magnitude of the impedance, IZI,

X i o RESISTANCE -R

- R. OHMS

L 3e.e5

4

Z. 4+J3

IZI' 5 t0/-' 0.73

IZI= 5 ae.e5

Figure 11

THE IMPEDANCE TRIANGLE

Showing the graphical construction of a triangle

for obtaining the net (scalar) impedance re-

sulting from the connection of o resistance and

a reactance in series. Shown also alongside is

the alternative mathematical procedure for

obtaining the values associated with the triangle.

and the phase angle 0, when resistance and

reactance are known; hence, of converting

from the Z = R + jX to the IZI LO form. In this

case we use two of the expressions just given:

IZI = V/R2+X2

X X

tan O = -, (or O = tan' - )

R R

The inverse problem, that of converting

from the IZI LO to the R + jX form is done

with the following relationships, both of which

are obtainable by simple division from the

trigonometric expressions just given for de-

termining the angle 0:

R =IZI cos0

jX =IZIj sin O

By simple addition these two expressions may

be combined to give the relationship between

the two most common methods of indicating

an impedance:

R + jX =IZI (cos B + j sin 0)

In the case of impedance, resistance, or re-

actance, the unit of measurement is the ohm;

hence, the ohm may be thought of as a unit of

opposition to current flow, without reference

to the relative phase angle between the ap-

plied voltage and the current which flows.

Further, since both capacitive and inductive

reactance are functions of frequency, imped-

ance will vary with frequency. Figure 12

shows the manner in which IZI will vary with

frequency in an RL series circuit and in an

RC series circuit.

Series RLC Circuits In a series circuit contain-

ing R, L, and C, the im-

www.americanradiohistory.com

HANDBOOK Impedance 49

pedance is determined as discussed before ex-

cept that the reactive component in the ex-

pressions becomes: (The net reactance-the

difference between XL and Xc.) Hence (XL

Xc) may be substituted for X in the equations.

Thus:

IZI = VR' +(XL Xc)'

O = tan '(XL (XL Xc )

R

A series RLC circuit thus may present an

impedance which is capacitively reactive if

the net reactance is capacitive, inductively

reactive if the net reactance is inductive, or

resistive if the capacitive and inductive re-

actances are equal.

Addition of The addition of complex

Complex Quantities quantities (for example,

impedances in series) is

quite simple if the quantities are in the rect-

angular form. If they are in the polar form

they only can be added graphically, unless

they are converted to the rectangular form by

the relationships previously given. As an ex-

ample of the addition of complex quantities

in the rectangular form, the equation for the

addition of impedances is:

( R, +jX,) +(R, +jX,)= (R, + Rs) +j(X,+X,)

For example if we wish to add the imped-

ances (10 + j50) and (20 j30) we obtain:

(10 + j50) + (20 j30)

= (10 + 20) + j(50 + ( -30) )

= 30 + j(50 -30)

= 30 + j20

Multiplication and It is often necessary in

Division of solving certain types of

Complex Quantities circuits to multiply or di-

vide two complex quan-

tities. It is a much simplier mathematical

operation to multiply or divide complex quan-

tities if they are expressed in the polar form.

Hence if they are given in the rectangular

form they should be converted to the polar

form before multiplication or division is begun.

Then the multiplication is accomplished by

multiplying the IZ1 terms together and adding

algebraically the L e terms, as:

(IZ,1Le,)(I4,I Le,) =IZ,1 Iz,I (L0,+ L0,)

For example, suppose that the two impedances

1201 L43 and 1321 L -23° are to be multi-

plied. Then:

( 1201 L43°) (1321 L -23 °) = 120.321

(L43° + L -23 °)

= 640 L 20°

o

Figure 12

IMPEDANCE AGAINST FREQUENCY

FOR R L AND R -C CIRCUITS

The impedance of an R -C circuit approaches

infinity os the frequency approaches zero (d.c.),

while the impedance of o series R -L circuit

approaches infinity as the frequency approaches

infinity. The impedance of an R -C circuit ap-

proaches the impedance of the series resistor

os the frequency approaches infinity, while the

impedance of o series R -L circuit approaches

the impedance of the resistor as the frequency

approaches zero.

Division is accomplished by dividing the

denominator into the numerator, and sub-

tracting the angle of the denominator from

that of the numerator, as:

IZ,I Le, IZ,I

IZ,IL0,- 1z21(Le, tee,)

For example, suppose that an impedance of

1501 L 67° is to be divided by an impedance

of 1101 L45°. Then:

1501 L67° 1501

1101 L45° 1101 = 151 (L 22 °)

Ohm's Law for The simple form of Ohm's

Complex Quantities Law used for d -c circuits

may be stated in a more

general form for application to a -c circuits

involving either complex quantities or simple

resistive elements. The form is:

E

z

in which, in the general case, I, E, and Z are

complex (vector) quantities. In the simple

case where the impedance is a pure resistance

with an a -c voltage applied, the equation

simplifies to the familiar I = E /R. In any case

the applied voltage may be expressed either

as peak, r.m.s., or average; the resulting

www.americanradiohistory.com

50 Alternating Current Circuits THE RADIO

200 n. Since the applied voltage will be the reference

for the currents and voltages within the cir-

cuit, we may define it as having a zero phase

oo angle: E = 100 LO °. Then:

100 LO °

_ -0 .354 L0 °- ( -45 °)

282 L -45°

= 0.354 L45° amperes.

Figure 13

SERIES R -L -C CIRCUIT

current always will be in the same type of

units as used to define the voltage.

In the more general case vector algebra

must be used to solve the equation. And,

since either division or multiplication is in-

volved, the complex quantities should be ex-

pressed in the polar form. As an example,

take the case of the series circuit shown in

figure 13 with 100 volts applied. The imped-

ance of the series circuit can best be obtained

first in the rectangular form, as:

200 + j(100

;(l00-.300) = 200 -j200

Now, to obtain the current we must convert

this impedance to the polar form.

IZI = V200'+(-200)'

= N/40,000 + 40,000

= 80,000

= 282 f2

X -200

O= tan' -=tan' = tan' -1

R 200

= -45 °.

Therefore Z = 282 L -45°

Note that in a series circuit the resulting im-

pedance takes the sign of the largest react-

ance in the series combination.

Where a slide -rule is being used to make

the computations, the impedance may be found

without any addition or subtraction operations

by finding the angle O first, and then using

the trigonometric equation below for obtain-

ing the impedance. Thus:

X -200

O =tan'- =tan' -tan' -1

R 200

= -45°

R

Then IZI = cos O

200

IZI 0.707 - 28212

cos -45° = 0.707

This same current must flow through all three

elements of the circuit, since they are in

series and the current through one must al-

ready have passed through the other two.

Hence the voltage drop across the resistor

(whose phase angle of course is 0 °) is:

E =1R

E = (0.354 L 45 °) (200 L0°)

= 70.8 L 45° volts

The voltage drop across the inductive react-

ance is:

E = I XL

E = (0.354 L45 °) (100 L 90 °)

= 35.4 L 135° volts

Similarly, the voltage drop across the capac-

itive reactance is:

E = I Xc

E = (0.354 1_45°) (300 /- - -90 °)

= 106.2 L -45°

Note that the voltage drop across the capac-

itive reactance is greater than the supply

voltage. This condition often occurs in a

series RLC circuit, and is explained by the

fact that the drop across the capacitive react-

ance is cancelled to a lesser or greater ex-

tent by the drop across the inductive reactance.

It is often desirable in a problem such as

the above to check the validity of the answer

by adding vectorially the voltage drops across

the components of the series circuit to make

sure that they add up to the supply voltage -

or to use the terminology of Kirchhoff's Second

Law, to make sure that the voltage drops

across all elements of the circuit, including

the source taken as negative, is equal to zero.

In the general case of the addition of a

number of voltage vectors in series it is best

to resolve the voltages into their in -phase

and out -of -phase components with respect to

the supply voltage. Then these components

may be added directly. Hence:

ER = 70.8L45°

= 70.8 ( cos 45° + j sin 45 °)

= 70.8 (0.707 + j0.707)

= 50 + j50

www.americanradiohistory.com

HANDBOOK Vector Algebra 51

VOLTAGE DROP ACROSS

X1.= 35.4 i laa

±leo

ao

DROP ACROSS RESISTOR .

,o e 45

f0

1

NET DROP ACROSS XL + XL .70.111/-45.

°'- LI NE VOLTAGE =100

DROP ACROSS XC =1Oe.2 -43

Figure 14

Graphical construction of the voltage drops

associated with the serles R -L -C circuit of

figure 13.

EL = 35.4L135°

= 35.4 ( cos 135° + j sin 135 °)

= 35.4 ( -0.707 + j0.707)

= -25 + j25

Ec= 106.2L45°

= 106.2 ( cos -45 °+ j sin -45 °)

= 106.2 (6.707 -j0.707)

= 75 -j75

ER + EL +EC =(50+ j50) + (-25 + j25)

+(75 -j75)

= (50 -25+ 75) + j(50 + 25-75)

= 100 +j0

= 100 LO °, which is equal to the

supply voltage.

Checking by

Construction on the

Complex Plane

It is frequently desirable

to check computations in-

volving complex quantities

by constructing vectors

representing the quantities on the complex

plane. Figure 14 shows such a construction

for the quantities of the problem just com-

pleted. Note that the answer to the problem

may be checked by constructing a parallel-

ogram with the voltage drop across the re-

sistor as one side and the net voltage drop

across the capacitor plus the inductor (these

may be added algrebraically as they are 180°

out of phase) as the adjacent side. The vector

sum of these two voltages, which is repre-

sented by the diagonal of the parallelogram,

is equal to the supply voltage of 100 volts at

zero phase angle.

Resistance and Re- In a series circuit, such

actonce in Parallel as just discussed, the cur-

rent through all the ele-

:_e.i T - ,.n

PARALLEL CIRCUIT EQUIVALENT SERIES CIRCUIT

Figure 15

THE EQUIVALENT SERIES CIRCUIT

Showing a parallel R -C circuit and the equiv-

alent series R -C circuit which represents the

same net impedance os the parallel circuit.

ments which go to make up the series circuit

is the same. But the voltage drops across

each of the components are, in general, dif-

ferent from one another. Conversely, in a

parallel RLC or RX circuit the voltage is,

obviously, the same across each of the ele-

ments. But the currents through each of the

elements are usually different.

There are many ways of solving a problem

involving paralleled resistance and reactance;

several of these ways will be described. In

general, it may be said that the impedance of

a number of elements in parallel is solved

using the same relations as are used for

solving resistors in parallel, except that com-

plex quantities are employed. The basic re-

lation is:

1 1 1 1

-+ -+ -+

Zrot Z, Z2 Z,

or when only two impedances are involved:

Z, Z2

Z`o` Z + Z

t :

As an example, using the two- impedance

relation, take the simple case, illustrated in

figure 15, of a resistance of 6 ohms in paral-

lel with a capacitive reactance of 4 ohms. To

simplify the first step in the computation it is

best to put the impedances in the polar form

for the numerator, since multiplication is in-

volved, and in the rectangular form for the

addition in the denominator.

(6 L0°) (4 -90°)

Zrot - 6-j4

24 L-90°

6 -j4

Then the denominator is changed to the polar

form for the division operation:

-4

O = tañ ' 6 = tan-' - 0.667 = - 33.7°

www.americanradiohistory.com

52 Alternating Current Circuits THE RADIO

IZI =

Then:

6 6 7.21 ohms

cos 33.7° 0.832

6 j4 = 7.21 L-33.7°

24 L -90°

Ztat = 7.21 L 33.7 °= 3.33 L -56.3°

= 3.33 ( cos 56.3° + j sin 56.3°)

= 3.33 [ 0.5548 + j (- 0.832)1

= 1.85 j 2.77

Equivalent Series Through the series of op-

Circuit erations in the previous

paragraph we have convert-

ed a circuit composed of two impedances in

parallel into an equivalent series circuit com-

posed of impedances in series. An equivalent

series circuit is one which, as far as the ter-

minals are concerned, acts identically to the

original parallel circuit; the current through

the circuit and the power dissipation of the

resistive elements are the same for a given

voltage at the specified frequency.

We can check the equivalent series circuit

of figure 15 with respect to the original cir-

cuit by assuming that one volt a.c. (at the

frequency where the capacitive reactance in

the parallel circuit is 4 ohms) is applied to

the terminals of both.

In the parallel circuit the current through

the resistor will be 2/6 ampere (0.166a.) while

the current through the capacitor will be j

ampere (+ j 0.25 a.). The total current will be

the sum of these two currents, or 0.166 +

j 0.25 a. Adding these vectorially we obtain:

III = x/0.1662 + 0.252 = x/0.09 = 0.3 a.

The dissipation in the resistor will be 12/6 =

0.166 watts.

In the case of the equivalent series circuit

the current will be:

E 1

Ill 3.33 =0.3a.

And the dissipation in the resistor will be:

W =I2R =0.32x1.85

= 0.9 x 1.85

= 0.166 watts

So we see that the equivalent series circuit

checks exactly with the original parallel cir-

cuit.

Parallel RLC In solving a more complicated

Circuits circuit made up of more than

two impedances in parallel we

EzEi Rz

R1+Rz

E. I ,o

1 Ez

° +o

Ez=E1 xCz

xo +502

Cz

Ez-E G+Cz

E2-E1 Lz

LI+Lz

OA O

Figure 16

SIMPLE A -C VOLTAGE DIVIDERS

may elect to use either of two methods of

solution. These methods are called the admit-

tance method and the assumed - voltage method.

However, the two methods are equivalent

since both use the sum-of-reciprocals equation:

1 1 1 1

- +- +-

Ztot Z1 Z2 Zs

In the admittance method we use the relation

Y = 1 /Z, where Y = G + jB; Y is called the

admittance, defined above, G is the conduct-

ance or R /Z' and B is the susceptance or

X/Z2. Then Ytot = 1 /Ztot = Y1 + Y2 + Y,

In the assumed- voltage method we multiply

both sides of the equation above by E, the

assumed voltage, and add the currents, as:

E E E E

-+ -+- ... = 1zt +Iz2 +1zt ..

Ztot Zt Z, Z3

Then the impedance of the parallel com-

bination may be determined from the relation:

Ztot = Eí IZ tot

AC Voltage Voltage dividers for use with

Dividers alternating current are quite simi-

lar to d-c voltage dividers. How-

ever, since capacitors and inductors oppose

the flow of a -c current as well as resistors,

voltage dividers for alternating voltages may

take any of the configurations shown in fig-

ure 16.

Since the impedances within each divider

are of the same type, the output voltage is in

phase with the input voltage. By using com-

binations of different types of impedances, the

phase angle of the output may be shifted in

relation to the input phase angle at the same

time the amplitude is reduced. Several di-

viders of this type are shown in figure 17.

Note that the ratio of output voltage to input

voltage is equal to the ratio of the output

impedance to the total divider impedance.

This relationship is true only if negligible

current is drawn by a load on the output ter-

minals.

www.americanradiohistory.com

HANDBOOK Resonant Circuits 53

E2Ei Xc

R2+XC2

E2E, XL

XL-Xc

E2 E XL

R2 +XL2

©

Ez E, Xc

R2+2

DO Ea Ei X

R2.2

Es - Ei XL-XCXC

- R2+I (L-XC12

Figure 17

COMPLEX A -C VOLTAGE DIVIDERS

3 -2 Resonant Circuits

A series circuit such as shown in figure 18

is said to be in resonance when the applied

frequency is such that the capacitive react-

ance is exactly balanced by the inductive re-

actance. At this frequency the two reactances

will cancel in their effects, and the impedance

of the circuit will be at a minimum so that

maximum current will flow. In fact, as shown

in figure 19 the net impedance of a series

circuit at resonance is equal to the resistance

which remains in the circuit after the react-

ances have been cancelled.

R t Frequency Some resistance is always

present in a circuit be-

cause it is possessed in some degree by both

the inductor and the capacitor. If the fre-

quency of the alternator E is varied from

nearly zero to some high frequency, there will

be one particular frequency at which the in-

ductive reactance and capacitive reactance

will be equal. This is known as the resonant

frequency, and in a series circuit it is the

frequency at which the circuit current will be

a maximum. Such series resonant circuits are

chiefly used when it is desirable to allow a

certain frequency to pass through the circuit

(low impedance to this frequency), while at

the same time the circuit is made to offer

considerable opposition to currents of other

frequencies.

Figure 18

SERIES RESONANT CIRCUIT

If the values of inductance and capacitance

both are fixed, there will be only one resonant

frequency.

If both the inductance and capacitance are

made variable, the circuit may then be changed

or tuned, so that a number of combinations

of inductance and capacitance can resonate at

the same frequency. This can be more easily

understood when one considers that inductive

reactance and capacitive reactance travel in

opposite directions as the frequency is changed.

For example, if the frequency were to remain

constant and the values of inductance and

capacitance were then changed, the following

combinations would have equal reactance:

Frequency is constant at 60 cycles.

L is expressed in henrys.

C is expressed in microfarads (.000001

farad.)

L XL C Xc

.265 100 26.5 100

2.65 1,000 2.65 1,000

26.5 10,000 .265 10.000

265.00 100,000 .0265 100,000

2,650.00 1,000,000 .00265 1,000,000

Frequency

of Resonance From the formula for reson-

ance, 2rrfL = 1 /2nfC. the res-

onant frequency is determined:

f= 1

2rr N/ LC

where f = frequency in cycles,

L = inductance in henrys,

C = capacitance in farads.

It is more convenient to express L and C

in smaller units, especially in making radio -

frequency calculations; f can also be ex-

pressed in megacycles or kilocycles. A very

useful group of such formulas is:

25,330 25,330 25,330

f2= orL= orC=

LC f2C f2L

where f = frequency in megacycles,

L = inductance in microhenrys,

C = capacitance in micromicrofarads.

www.americanradiohistory.com

54 Alternating Current Circuits THE RADIO

Figure 19

IMPEDANCE OF A

SERIES -RESONANT CIRCUIT

Showing the variation in reactance of the sepa-

rate elements and In the net impedance of o

series resonant circuit (such as figure 18) with

changing frequency. The vertical line is drawn

at the point of resonance (XL - Xc = 0) in the

series circuit.

Impedance of Series The impedance across

Resonant Circuits the terminals of a series

resonant circuit (figure

18) is:

Z = /r3 + (XL Xc)2,

where Z = impedance in ohms,

r = resistance in ohms,

Xc = capacitive reactance in ohms,

XL = inductive reactance in ohms.

From this equation, it can be seen that the

impedance is equal to the vector sum of the

circuit resistance and the difference between

the two reactances. Since at the resonant fre-

quency XL equals Xc. the difference between

them (figure 19) is zero, so that at resonance

the impedance is simply equal to the resist-

ance of the circuit; therefore, because the

resistance of most normal radio- frequency

circuits is of a very low order, the impedance

is also low.

At frequencies higher and lower than the

resonant frequency, the difference between

the reactances will be a definite quantity and

will add with the resistance to make the im-

pedance higher and higher as the circuit is

tuned off the resonant frequency.

If Xc should be greater than XL, then the

term (XL Xc) will give a negative number.

However, when the difference is squared the

product is always positive. This means that

the smaller reactance is subtracted from the

larger, regardless of whether it be capacitive

or inductive, and the difference squared.

FREQUENCY

Figure 20

RESONANCE CURVE

Showing the increase in impedance at reson-

ance for o parallel- resonant circuit, and simi-

larly, the increase in current at resonance for

a series- rsonant circuit. The sharpness of

resonance is determined by the Q of the circuit,

as illustrated by a comparison between A,

B, and C.

Current and Voltage

in Series Resonant

Circuits

Ohm's law.

Formulas for calculating

currents and voltages in

a series resonant circuit

are similar to those of

E

I =- Z E = IZ

The complete equations:

E

I

V' r= + (XL Xc)2

E = 1 + (XL Xc)'

Inspection of the above formulas will show

the following to apply to series resonant cir-

cuits: When the impedance is low, the current

will be high; conversely, when the impedance

is high, the current will be low.

Since it is known that the impedance will

be very low at the resonant frequency, it fol-

lows that the current will be a maximum at

this point. If a graph is plotted of the current

against the frequency either side of resonance,

the resultant curve becomes what is known as

a resonance curve. Such a curve is shown in

figure 20, the frequency being plotted against

current in the series resonant circuit.

Several factors will have an effect on the

shape of this resonance curve, of which re-

www.americanradiohistory.com

HANDBOOK Circuit Q 55

sistance and L -to -C ratio are the important

considerations. The curves B and C in figure

20 show the effect of adding increasing values

of resistance to the circuit. It will be seen

that the peaks become less and less prominent

as the resistance is increased; thus, it can be

said that the selectivity of the circuit is

thereby decreased. Selectivity in this case

can be defined as the ability of a circuit to

discriminate against frequencies adjacent to

the resonant frequency.

Voltage Across Coil Because the a.c. or r -f

and Capacitor in voltage across a coil and

Series Circuit capacitor is proportional

to the reactance (for a

given current), the actual voltages across the

coil and across the capacitor may be many

times greater than the terminal voltage of the

circuit. At resonance, the voltage across the

coil (or the capacitor) is Q times the applied

voltage. Since the Q (or merit factor) of a

series circuit can be in the neighborhood of

100 or more, the voltage across the capacitor,

for example, may be high enough to cause

flashover, even though the applied voltage is

of a value considerably below that at which

the capacitor is rated.

Circuit Q -Sharp- An extremely important

ness of Resonance property of a capacitor or

an inductor is its factor -

of- merit, more generally called its Q. It is this

factor, Q, which primarily determines the

sharpness of resonance of a tuned circuit.

This factor can be expressed as the ratio of

the reactance to the resistance, as follows:

2rrfL

Q- R

where R = total resistance.

Skin Effect The actual resistance in a wire

or an inductor can be far greater

than the d -c value when the coil is used in a

radio -frequency circuit; this is because the

current does not travel through the entire

cross -section of the conductor, but has a tend-

ency to travel closer and closer to the surface

of the wire as the frequency is increased. This

is known as the skin effect.

The actual current -carrying portion of the

wire is decreased, as a result of the skin

effect, so that the ratio of a -c to d -c resist-

ance of the wire, called the resistance ratio,

is increased. The resistance ratio of wires to

be used at frequencies below about 500 kc.

may be materially reduced through the use of

Utz wire. Litz wire, of the type commonly used

to wind the coils of 455 -kc. i -f transformers,

may consist of 3 to 10 strands of insulated

wire, about No. 40 in size, with the individual

strands connected together only at the ends of

the coils.

Variation of Q

with Frequency Examination of the equation

for determining Q might give

rise to the thought that even

though the resistance of an inductor increases

with frequency, the inductive reactance does

likewise, so that the Q might be a constant.

Actually, however, it works out in practice

that the Q of an inductor will reach a relative-

ly broad maximum at some particular frequency.

Hence, coils normally are designed in such a

manner that the peak in their curve of Q with

frequency will occur at the normal operating

frequency of the coil in the circuit for which

it is designed.

The Q of a capacitor ordinarily is much

higher than that of the best coil. Therefore,

it usually is the merit of the coil that limits

the overall Q of the circuit.

At audio frequencies the core losses in an

iron -core inductor greatly reduce the Q from

the value that would be obtained simply by

dividing the reactance by the resistance. Ob-

viously the core losses also represent circuit

resistance, just as though the loss occurred

in the wire itself.

Parallel In radio circuits, parallel reson-

Resonance ance (more correctly termed anti -

resonance) is more frequently

encountered than series resonance; in fact, it

is the basic foundation of receiver and trans-

mitter circuit operation. A circuit is shown in

figure 21.

The "Tank" In this circuit, as contrasted with

Circuit a circuit for series resonance, L

(inductance) and C (capacitance)

are connected in parallel, yet the combination

can be considered to be in series with the

remainder of the circuit. This combination

of L and C, in conjunction with R, the re-

sistance which is principally included in L, is

sometimes called a tank circuit because it ef-

fectively functions as a storage tank when in-

corporated in vacuum tube circuits.

Contrasted with series resonance, there are

two kinds of current which must be considered

in a parallel resonant circuit: (1) the line cur-

rent, as read on the indicating meter M (2)

the circulating current which flows within the

parallel L -C -R portion of the circuit. See

figure 21.

At the resonant frequency, the line current

(as read on the meter M,) will drop to a very

low value although the circulating current in

the L -C circuit may be quite large. It is inter-

esting to note that the parallel resonant cir-

cuit acts in a distinctly opposite manner to

that of a series resonant circuit, in which the

www.americanradiohistory.com

56 A!ternating Current Circuits THE RADIO

Figure 21

PARALLEL- RESONANT CIRCUIT

The inductance L and capacitance C comprise

the reactive elements of the parallel -resonant

(anti -resonant) tank circuit, and the resistance

R indicates the sum of the r -f resistance of the

coil and capacitor, plus the resistance coupled

into the circuit from the external load. In most

coses the tuning capacitor has much lower r -f

resistance than the coil and can therefore be

ignored in comparison with the coil resistance

and the coupled -in resistance. The instrument

M1 indicates the "line current" which keeps

the circuit in a state of oscillation -this cur-

rent is the some as the fundamental component

of the plate current of a Class C amplifier which

might be feeding the tank circuit. The in-

strument M2 indicates the "tank current" which

is equal to the line current multiplied by the

operating Q of the tank circuit.

current is at a maximum and the impedance is

minimum at resonance. It is for this reason

that in a parallel resonant circuit the principal

consideration is one of impedance rather than

current. It is also significant that the imped-

ance curve for parallel circuits is very nearly

identical to that of the current curve for series

resonance. The impedance at resonance is

expressed as:

Z- R

(2trf L)2

where Z = impedance in ohms,

L = inductance in henrys,

f = frequency in cycles,

R = resistance in ohms.

Or, impedance can be expressed as a func-

tion of Q as:

Z= 2nfLQ,

showing that the impedance of a circuit is

directly proportional to its effective Q at

resonance.

The curves illustrated in figure 20 can be

applied to parallel resonance. Reference to the

curve will show that the effect of adding re-

sistance to the circuit will result in both a

broadening out and lowering of the peak of the

curve. Since the voltage of the circuit is

directly proportional to the impedance, and

since it is this voltage that is applied to the

grid of the vacuum tube in a detector or am-

plifier circuit, the impedance curve must have

a sharp peak in order for the circuit to be

selective. If the curve is broad- topped in

shape, both the desired signal and the inter-

fering signals at close proximity to resonance

will give nearly equal voltages on the grid of

the tube, and the circuit will then be non-

selective; i.e., it will tune broadly.

Effect of L/C Ratio In order that the highest

in Parallel Circuits possible voltage can be

developed across a paral-

lel resonant circuit, the impedance of this

circuit must be very high. The impedance will

be greater with conventional coils of limited

Q when the ratio of inductance-to-capacitance

is great, that is, when L is large as compared

with C. When the resistance of the circuit is

very low. XL will equal XC at maximum im-

pedance. There are innumerable ratios of L

and C that will have equal reactance, at a

given resonant frequency, exactly as in the

case in a series resonant circuit.

In practice, where a certain value of in-

ductance is tuned by a variable capacitance

over a fairly wide range in frequency, the

L/C ratio will be small at the lowest fre-

quency end and large at the high -frequency end.

The circuit, therefore, will have unequal gain

and selectivity at the two ends of the band of

frequencies which is being tuned. Increasing

the Q of the circuit (lowering the resistance)

will obviously increase both the selectivity

and gain.

Circulating Tank The Q of a circuit has

Current at Resonance a definite bearing on

the circulating tank

current at resonance. This tank current is

very nearly the value of the line current multi-

plied by the effective circuit Q. For example:

an r -f line current of 0.050 amperes, with a

circuit Q of 100, will give a circulating tank

current of approximately 5 amperes. From this

it can be seen that both the inductor and the

connecting wires in a circuit with a high Q

must be of very low resistance, particularly in

the case of high power transmitters, if heat

losses are to be held to a minimum.

Because the voltage across the tank at

resonance is determined by the Q, it is pos-

sible to develop very high peak voltages

across a high Q tank with but little line cur-

rent.

Effect of Coupling

on Impedance

output circuit, the

Q of the parallel

coupling becomes

(tighter) coupling

If a parallel resonant cir-

cuit is coupled to another

circuit, such as an antenna

impedance and the effective

circuit is decreased as the

closer. The effect of closer

is the same as though an

www.americanradiohistory.com

HANDBOOK Circuit Impedance 57

Z

LOOSE COUPLING

HIGH 0 4E011.14 COUPLING

MEDI U4 0

t

O vE N COUPLING

LOP

Figure 22

EFFECT OF COUPLING ON CIRCUIT IMPEDANCE AND Q

actual resistance were added in series with

the parallel tank circuit. The resistance thus

coupled into the tank circuit can be con-

sidered as being reflected from the output or

load circuit to the driver circuit.

The behavior of coupled circuits depends

largely upon the amount of coupling, as shown

in figure 22. The coupled current in the sec-

ondary circuit is small, varying with frequency,

being maximum at the resonant frequency of

the circuit. As the coupling is increased

between the two circuits, the secondary res-

onance curve becomes broader and the reso-

nant amplitude increases, until the reflected

resistance is equal to the primary resistance.

This point is called the critical coupling

point. With greater coupling, the secondary

resonance curve becomes broader and develops

double resonance humps, which become more

pronounced and farther apart in frequency as

the coupling between the two circuits is

increased.

Tank Circuit When the plate circuit of a

Flywheel Effect Class B or Class C operated

tube is connected to a par-

allel resonant circuit tuned to the same fre-

quency as the exciting voltage for the ampli-

fier, the plate current serves to maintain this

L/C circuit in a state of oscillation.

The plate current is supplied in short pulses

which do not begin to resemble a sine wave,

even though the grid may be excited by a sine -

wave voltage. These spurts of plate current

are converted into a sine wave in the plate

tank circuit by virtue of the "Q" or "flywheel

effect" of the tank.

If a tank did not have some resistance

losses, it would, when given a "kick" with a

single pulse, continue to oscillate indefinitely.

With a moderate amount of resistance or "fric-

tion" in the circuit the tank will still have

inertia, and continue to oscillate with de-

creasing amplitude for a time after being given

a "kick." With such a circuit, almost pure

sine -wave voltage will be developed across

the tank circuit even though power is supplied

to the tank in short pulses or spurts, so long

as the spurts are evenly spaced with respect

to time and have a frequency that is the same

as the resonant frequency of the tank.

Another way to visualize the action of the

tank is to recall that a resonant tank with

moderate Q will discriminate strongly against

harmonics of the resonant frequency. The dis-

torted plate current pulse in a Class C ampli-

fier contains not only the fundamental fre-

quency (that of the grid excitation voltage)

but also higher harmonics. As the tank offers

low impedance to the harmonics and high im-

pedance to the fundamental (being resonant to

the latter), only the fundamental - a sine -

wave voltage - appears across the tank circuit

in substantial magnitude.

Loaded and Confusion sometimes exists as

Unloaded 0 to the relationship between the

unloaded and the loaded Q of the

tank circuit in the plate of an r -f power ampli-

fier. In the normal case the loaded Q of the

tank circuit is determined by such factors as

the operating conditions of the amplifier, band-

width of the signal to be emitted, permissible

level of harmonic radiation, and such factors.

The normal value of loaded Q for an r -f ampli-

fier used for communications service is from

perhaps 6 to 20. The unloaded Q of the tank

circuit determines the efficiency of the output

circuit and is determined by the losses in the

tank coil, its leads and plugs and jacks if any,

and by the losses in the tank capacitor which

ordinarily are very low. The unloaded Q of a

good quality large diameter tank coil in the

high - frequency range may be as high as 500

www.americanradiohistory.com

58 Alternating Current Circuits THE RADIO

to 800, and values greater than 300 are quite

common.

Tank Circuit Since the unloaded Q of a tank

Efficiency circuit is determined by the

minimum losses in the tank,

while the loaded Q is determined by useful

loading of the tank circuit from the external

load in addition to the internal losses in the

tank circuit, the relationship between the two

Q values determines the operating efficiency

of the tank circuit. Expressed in the form of

an equation, the loaded efficiency of a tank

circuit is:

Tank efficiency = 1 _Qt x 100

Qu

where Qu = unloaded Q of the tank circuit

Qi = loaded Q of the tank circuit

As an example, if the unloaded Q of the

tank circuit for a class C r -f power amplifier

is 400, and the external load is coupled to the

tank circuit by an amount such that the loaded

Q is 20, the tank circuit efficiency will be:

eff. = (1 - 20/400) x 100, or (1 - 0.05) x 100,

or 95 per cent. Hence 5 per cent of the power

output of the Class C amplifier will be lost

as heat in the tank circuit and the remaining

95 per cent will be delivered to the load.

3 -3 Nonsinusoidal Waves

and Transients

Pure sine waves, discussed previously, are

basic wave shapes. Waves of many different

and complex shape are used in electronics,

particularly square waves, saw -tooth waves,

and peaked waves.

Wave Composition Any periodic wave (one that

repeats itself in definite

time intervals) is composed of sine waves of

different frequencies and amplitudes, added

together. The sine wave which has the same

frequency as the complex, periodic wave is

called the fundamental. The frequencies higher

than the fundamental are called harmonics,

and are always a whole number of times higher

than the fundamental. For example, the fre-

quency twice as high as the fundamental is

called the second harmonic.

The Square Wave Figure 23 compares a square

wave with a sine wave (A)

of the same frequency. If another sine wave

(B) of smaller amplitude, but three times the

frequency of (A), called the third harmonic, is

added to (A), the resultant wave (C) more

nearly approaches the desired square wave.

-FUNDAMENTAL SINE WAVE(A

- FUNDAMENTAL PLUS

3RD HARMONIC(C)

-SQUARE WAVE

3RD HARMONIC (B)

Figure 23

COMPOSITE WAVE-FUNDAMENTAL

PLUS THIRD HARMONIC

FUNDAMENTAL PLUS 3RD HARMONIC

FUNDAMENTAL PLUS 3RD AND

5TH HARMONICS(E)

VI% HARMONIC (D)

Figure 24

THIRD HARMONIC WAVE PLUS

FIFTH HARMONIC

FUNDAMENTAL PLUS 3RD. 5TH,

AND 7TH HARMONICS (G)

FUNDAMENTAL PLUS 3RD AND

STM HARMONICS

SQUARE WAVE

7TM HARMONIC (F )

Figure 25

RESULTANT WAVE, COMPOSED OF

FUNDAMENTAL, THIRD, FIFTH,

AND SEVENTH HARMONICS

This resultant curve (figure 24) is added to

a fifth harmonic curve (D), and the sides of

the resulting curve (E)are steeper than before.

This new curve is shown in figure 25 after a

7th harmonic component has been added to it,

making the sides of the composite wave even

steeper. Addition of more higher odd harmonics

will bring the resultant wave nearer and nearer

to the desired square wave shape. The square

wave will be achieved if an infinite number of

odd harmonics are added to the original sine

wave.

www.americanradiohistory.com

HANDBOOK Nonsinusoidal Waves 59

FUND PLUS 210 HARM.

FUNDAMENTAL

2ND HARM.

FUND, PLUS AND AND

HARMONICS

f YNO. PLUS END MARY.

3RD MARYON IC

Z

FUND. PLUS 2ND, 3RD,

RM ONI C4

&IL" H XA

A M S 2CS AN

CATM

SA

FUND.PLUS 2ND ]RD, 4TH,

iikkatIPUZTD"' NAR MENICS

PLUS 2ND 3RD, AND

TM NARMOrMCS

]TM HARMONIC

-4111111

FVND. PLU12Np3RD ATM\

STM AND STM NARMISN ICS

FUND. PLUS 2ND, 3RD ATM.

\AND STM MARMONIOS

T TH HARMONIC

FUND PLUS 2040, 3R0,4T14,

STM. TH, AND

DL

?TM HARMS.

S:w ÁNDU TMNthreCCaiTa

7TH HARMONIC

TOOTH WADE

FUND. PLUS 2ND, 3RD4 ATM, STM, ATM,

AND 7TH HARMONICS

Figure 26

COMPOSITION OF A SAWTOOTH WAVE

The Sawtooth Wave In the same fashion, a

sawtooth wave is made up

of different sine waves (figure 26). The ad-

dition of all harmonics, odd and even, produces

the sawtooth wave form.

The Pecked Wave Figure 27 shows the com-

position of a peaked wave.

Note how the addition of each sucessive har-

monic makes the peak of the resultant higher

and the sides steeper.

Other Waveforms The three preceeding ex-

amples show how a complex

periodic wave is composed of a fundamental

wave and different harmonics. The shape of

the resultant wave depends upon the harmonics

that are added, their relative amplitudes, and

relative phase relationships. In general, the

steeper the sides of the waveform, the more

harmonics it contains.

AC Transient Circuits If an a -c voltage is sub-

stituted for the d -c in-

put voltage in the RC Transient circuits dis-

cussed in Chapter 2, the same principles may

be applied in the analysi* of the transient

behavior. An RC coupling circuit is designed

to have a long time constant with respect to

the lowest frequency it must pass. Such a

circuit is shown in figure 28. If a nonsinus-

oidal voltage is to be passed unchanged

through the coupling circuit, the time constant

FUNDAMENTAL PLUS

3RD HARMONIC

FUNDAMENTAL

3RD HARMONIC

FUNDAMENTAL PLUS 3RD

AND STD HARMONICS

,FUNDAMENTAL PLUS 3RD HARM.

5TH HARMONIC

/

FUNDAMENTAL PLUS 3RD, STN,

AND 7TH HARMONICS

FUNDAMENTAL PLUS 3RD

ANO STH HARMONIC

7TH HARMONIC

/

Figure 27

COMPOSITION OF A PEAKED WAVE

must be long with respect to the period of the

lowest frequency contained in the voltage

wave.

RC Differentiator An RC voltage divider that

and Integrator is designed to distort the in-

put waveform is known as a

differentiator or integrator, depending upon

the locations of the output taps. The output

from a differentiator is taken across the re-

sistance, while the output from an integrator

is taken across the capacitor. Such circuits

will change the shape of any complex a -c

waveform that is impressed upon them. This

distortion is a function of the value of the

time constant of the circuit as compared to

the period of the waveform. Neither a dif-

ferentiator nor an integrator can change the

www.americanradiohistory.com

60 Alternating Current Circuits THE RADIO

100 v.

1000 C.P 5

Cr 0.1 .Uf

R o.M OUTPUT

VOLTAGE

R 1lC 50000 USECONDS

PERIOD OP e' 1000 USECONDS

Figure 28

R -C COUPLING CIRCUIT WITH

LONG TIME CONSTANT

E'100v.

(PEAR )

10001

100 v

et

INTEGRATOR

OUTPUT

eR -100v

DIP IATOR

OUTPUT

+30v

AO PR o

,-1- 1 I I

I I 1

T

o T

Figure 29

R -C DIFFERENTIATOR AND

INTEGRATOR ACTION ON

A SINE WAVE

shape of a pure sine wave, they will merely

shift the phase of the wave (figure 29). The

differentiator output is a sine wave leading

the input wave, and the integrator output is a

sine wave which lags the input wave. The sum

of the two outputs at any instant equals the

instantaneous input voltage.

Square Wave Input If a square wave voltage is

impressed on the circuit of

figure 30, a square wave voltage output may

be obtained across the integrating capacitor

if the time constant of the circuit allows the

capacitor to become fully charged. In this

particular case, the capacitor never fully

charges, and as a result the output of the

integrator has a smaller amplitude than the

input. The differentiator output has a maximum

value greater than the input amplitude, since

the voltage left on the capacitor from the

previous half wave will add to the input volt-

age. Such a circuit, when used as a differen-

tiator, is often called a peaker. Peaks of

twice the input amplitude may be produced.

e. ,00v

(PEAR)

1000 C.P.S.

eo

C'o.1 V

1T

cc. INTEGRATOR OUTPUT

R'10R JJreR'DIPPERENTIATOROUTPUT

+100V

óf iE [Ñ1TÓR"N

100 V

+25 V.

OUTPUT Of

ATOR (ec)

Figure 30

RC DIFFERENTIATOR AND

INTEGRATOR ACTION ON

A SQUARE WAVE

Sawtooth Wave Input If a back -to -back saw -

tooth voltage is applied

to an RC circuit having a time constant one -

sixth the period of the input voltage, the re-

sult is shown in figure 31. The capacitor

voltage will closely follow the input voltage,

if the time constant is short, and the integra-

tor output closely resembles the input. The

amplitude is slightly reduced and there is a

slight phase lag. Since the voltage across the

capacitor is increasing at a constant rate, the

charging and discharging current is constant.

The output voltage of the differentiator, there-

fore, is constant during each half of the saw -

tooth input.

Miscellaneous Various voltage waveforms

Inputs other than those represented

here may be applied to short

RC circuits for the purpose of producing

across the resistor an output voltage with an

amplitude proportional to the rate of change of

the input signal. The shorter the RC time con-

stant is made with respect to the period of the

input wave, the more nearly the voltage across

www.americanradiohistory.com

HANDBOOK Transformers 61

e=100 v.

(PEAR)

1000 C.P.S.

INTEGRATOR

OUTPUT (ec)

1 DIFFERENTIATOR

JOUTPUT (e0)

e0

- +100

OUTPUT WAVEFORM

OF GENERATOR

-100

-ppt{ IO

uJ ÌA

ERÉITTÓRp I(eR)

OUTPUT OF

INTEGRATOR (eE)

Figure 31

R -C DIFFERENTIATOR AND

INTEGRATOR ACTION ON

A SAWTOOTH WAVE

the capacitor conforms to the input voltage.

Thus, the differentiator output becomes of

particular importance in very short RC cir-

cuits. Differentiator outputs for various types

of input waves are shown in figure 32.

Square Wave Test The application of a square

for Audio Equipment wave input signal to audio

equipment, and the ob-

servation of the reproduced output signal on

an oscilloscope will provide a quick and ac-

curate check of the overall operation of audio

equipment. Low -frequency and high- frequency

response, as well as transient response can be

examined easily. If the amplifier is deficient

in low- frequency response, the flat top of the

square wave will be canted, as in figure 33.

If the high- frequency response is inferior, the

rise time of the output wave will be retarded

(figure 34). An amplifier with a limited high -

and low- frequency response will turn the

square wave into the approximation of a saw -

tooth wave (figure 35).

3-4 Transformers

When two coils are placed in such inductive

relation to each other that the lines of force

from one cut across the turns of the other

inducing a current, the combination can be

called a transformer. The name is derived from

the fact that energy is transformed from one

winding to another. The inductance in which

Figure 32

Dlfferentlator outputs of short r -c circuits for

various input voltage waveshapes. The output

voltage is proportional to the rote of change

of the input voltage.

the original flux is produced is called the

primary; the inductance which receives the

induced current is called the secondary. In a

radio receiver power transformer, for example,

the coil through which the 110 -volt a.c. passes

is the primary, and the coil from which a higher

or lower voltage than the a -c line potential is

obtained is the secondary.

Transformers can have either air or mag-

netic cores, depending upon the frequencies at

which they are to be operated. The reader

should thoroughly impress upon his mind the

fact that current can be transferred from one

circuit to another only if the primary current

is changing or alternating. From this it can be

seen that a power transformer cannot possibly

function as such when the primary is supplied

with non -pulsating d.c.

A power transformer usually has a magnetic

core which consists of laminations of iron,

built up into a square or rectangular form,

with a center opening or window. The second-

ary windings may be several in number, each

perhaps delivering a different voltage. The

secondary voltages will be proportional to the

turns ratio and the primary voltage.

www.americanradiohistory.com

62 Alternating Current Circuits THE RADIO

Figure 33

Amplifier deficient In low frequency response will distort square wave applied to the input circuit, as

shown. A 60 -cycle square wave may he used.

A: Drop in gain at low frequencies

B: Lending phase shift at low frequencies

C: Logging phase shift at low frequencies

D: Accentuated low frequency gain

Types of Transformers are used in alter -

Transformers oaring- current circuits to trans-

fer power at one voltage and im-

pedance to another circuit at another voltage

and impedance. There are three main classifi-

cations of transformers: those made for use

in power -frequency circuits, those made for

audio -frequency applications, and those made

for radio frequencies.

The Transformation In a perfect transformer all

Ratio the magnetic flux lines

produced by the primary

winding link every turn of the secondary wind-

ing. For such a transformer, the ratio of the

primary and secondary voltages is exactly

the same as the ratio of the number of turns

in the two windings:

Np Ep

Ns Es

where Np = number of turns in the primary

winding

Ns = number of turns in the secondary

winding

Ep = voltage across the primary winding

Es = voltage across the secondary

winding

In practice, the transformation ratio of a

transformer is somewhat less than the turns

ratio, since unity coupling does not exist

between the primary and secondary windings.

Ampere Turns (NI) The current that flows in

the secondary winding as a

result of the induced voltage must produce a

flux which exactly equals the primary flux.

The magnetizing force of a coil is expressed

as the product of the number of turns in the

coil times the current flowing in it:

NpxIp =NsXIs, or Np = Is

Ns IP

where Ip = primary current

Is = secondary current

It can be seen from this expression that

when the voltage is stepped up, the current

is stepped down, and vice -versa.

Leakage Reactance Since unity coupling does

not exist in a practical

Figure 34

Output waveshape of amplifier having deficiency

in high- frequency response. Tested with 10 -kc.

square wave.

Figure 35

Output waveshape of amplifier having limited

low -frequency and high- frequency response.

Tested with 1 -kc. square wave.

www.americanradiohistory.com

HANDBOOK Electric Filters 63

ZL

Figure 36

IMPEDANCE -MATCHING TRANSFORMER

The reflected impedance Zp varies directly In

proportion to the secondary load IL, and

directly In proportion to the square of the

primary -to- secondary turns ratio.

transformer, part of the flux passing from the

primary circuit to the secondary circuit fol-

lows a magnetic circuit acted upon by the

primary only. The same is true of the second-

ary flux. These leakage fluxes cause leakage

reactance in the transformer, and tend to

cause the transformer to have poor voltage

regulation. To reduce such leakage reactance,

the primary and secondary windings should

be in close proximity to each other. The more

expensive transformers have interleaved wind-

ings to reduce inherent leakage reactance.

Impedance In the ideal transformer, the

Transformation impedance of the secondary

load is reflected back into the

primary winding in the following relationship:

Zp = N'Zs , or N = N/Zp/Zs

where Zp = reflected primary impedance

N = turns ratio of transformer

Zs = impedance of secondary load

Thus any specific load connected to the

secondary terminals of the transformer will

be transformed to a different specific value

appearing across the primary terminals of the

transformer. By the proper choice of turns

ratio, any reasonable value of secondary load

impedance may be "reflected" into the pri-

mary winding of the transformer to produce the

desired transformer primary impedance. The

phase angle of the primary "reflected" im-

pedance will be the same as the phase angle

of the load impedance. A capacitive second-

ary load will be presented to the transformer

source as a capacity, a resistive load will

present a resistive "reflection" to the primary

source. Thus the primary source "sees" a

transformer load entirely dependent upon the

secondary load impedance and the turns ratio

of the transformer (figure 36).

The Auto The type of transformer in figure

Transformer 37, when wound with heavy wire

over an iron core, is a common

device in primary power circuits for the pur-

pose of increasing or decreasing the line volt-

1

STEP -UP

STEP -OOW N

INPUT

VOLTAGE OUTPUT

VOLTAGE

Figure 37

THE AUTO -TRANSFORMER

Schematic diagram of an auto- transformer

showing the method of connecting it to the line

and to the load. When only a small amount of

step up or step down Is required, the auto -

transformer may be much smaller physically

thon would be a transformer with o separate

secondary winding. Continuously variable

auto -transformers (Variar and Powerstat) are

widely used commercially.

age. In effect, it is merely a continuous wind-

ing with taps taken at various points along

the winding, the input voltage being applied

to the bottom and also to one tap on the wind-

ing. If the output is taken from this same

tap, the voltage ratio will be 1 -to -1; i.e., the

input voltage will be the same as the output

voltage. On the other hand, if the output tap

is moved down toward the common terminal,

there will be a step -down in the turns ratio

with a consequent step -down in voltage. The

initial setting of the middle input tap is chosen

so that the number of turns will have suffi-

cient reactance to keep the no -load primary

current at a reasonably low value.

3 -5 Electric Filters

There are many applications where it is

desirable to pass a d -c component without

passing a superimposed a -c component, or to

ELEMENTARY FILTER SECTIONS

L- SECTIONS

Pi - NETWOR

T- NET WONIt

rs

T

Figure 38

Complex filters may be mode up from these basic

filter sections.

www.americanradiohistory.com

64 A l t e r n a t i n g C u r r e n t C i r c u i t s T H E R A D I O

LOW -PASS SHUNT -DERIVE FILTER

(SERIES -ARM RESONATED

2

2CI

C2

2C1

O

<

z

f2 fQ

FREQUENCY

R. LOAD RESISTANCE

L M LE

Ci

1 4 M x C K

C2= MCK

HIGH-PASS SERIES -DERI ED FILTER

(5J.UNT -ARM RESONATE

CI

LK= M= ,/I -()2 Cx= 777_tF-

f2 = CUT -OFF FREQUENCY. fk =FREQUENCY OF

NIGH ATTENUATION

fq

FREQUENCY

R. LOAD RESISTANCE

CI'

C2- 14M> -x Cs

L2- M -

LK = 7- M-I /ßa`2 CS 477fIR

( I

fI= Cur-OFF FREQUENCY. P, =FREQUENCY OF

NIGH ATTENUATION

Figure 39

TYPICAL LOW -PASS AND HIGH -PASS FILTERS, ILLUSTRATING SHUNT AND SERIES

DERIVATIONS

pass all frequencies above or below a certain

frequency while rejecting or attenuating all

others, or to pass only a certain band or bands

of frequencies while attenuating all others.

All of these things can be done by suitable

combinations of inductance, capacitance and

resistance. However, as whole books have

been devoted to nothing but electric filters, it

can be appreciated that it is possible only to

touch upon them superficially in a general

coverage book.

Filter Operation A filter acts by virtue of its

property of offering very high

impedance to the undesired frequencies, while

offering but little impedance to the desired

frequencies. This will also apply to d.c. with

a superimposed a -c component, as d.c. can

be considered as an alternating current of zero

frequency so far as filter discussion goes.

Basic Filters Filters are divided into four

classes, descriptive of the fre-

quency bands which they are designed to

transmit: high pass, low pass, band pass and

band elimination. Each of these classes of

filters is made up of elementary filter sections

called L sections which consist of a series

element (ZA) and a parallel element (ZR) as

illustrated in figure 38. A finite number of L

sections may be combined into basic filter

sections, called T networks or pi networks,

also shown in figure 38. Both the T and pi

networks may be divided in two to form half -

sections.

Filter Sections The most common filter sec-

tion is one in which the two

impedances ZA and Zg are so related that

their arithmetical product is a constant: ZA x

Zg = K2 at all frequencies. This type of filter

section is called a constant -K section.

A section having a sharper cutoff frequency

than a constant -K section, but less attenua-

tion at frequencies far removed from cutoff is

the M- derived section, so called because the

shunt or series element is resonated with a

reactance of the opposite sign. If the comple-

mentary reactance is added to the series arm,

the section is said to be shunt derived; if

added to the shunt arm, series derived. Each

impedance of the M- derived section is related

to a corresponding impedance in the constant -

K section by some factor which is a function

of the constant m. M, in turn, is a function of

the ratio between the cutoff frequency and

the frequency of infinite attenuation, and will

www.americanradiohistory.com

HANDBOOK Filter Design 65

TT- SECTION FILTER DESIGN

LOW PASS

R' LOAD RESISTANCE

2 =CUT-OFF FREQUENCY

f.= FREQUENCY OFVERY

HIGH ATTENUATION

Ln- R

CONSTANT K M=0 S

SECTIONS

LFSEC NG

I

°---/-i OTO t------0

C 0

2 T T C2

t yI 0

t

T ú Li

TiC2 o

-YO013y

-F-

O

C2 -O

Li i

T o o f

}SAME VALUES AS M o.6

SAME CURVE AS M=0_6

Lt=0.6Ln=MLK t-M2

Ci CK

z 0 j 2

<

Lt=Ln

Cz-Cn

f2

.

n f2

Cn - t

C2=o.6Cn=Cn

z 0 j

M

<

4M

A fm

--f2

ltf2 R

M .11-(--f-2-,. 2- 0 6

1 aC

FREQUENCY

H IGH PASS

R. LOAD RESISTANCE

fICUT-OFFFREQUENCY

f FREQUENCY OF VERY

HIGH ATTENUATION

LK=4

CI 1 I 1 }Lt iLt

-

2L2 - I

I .`..1 I 2L2 0

- 2L2 2L2- RL

2L2 21_2 t

SAME VALUES ASM=0.6

SAME CURVE AS M c 0.6

Lt=3.75L1t=t4M

Cl.

Lz=

z ó

z

.- <

xLK

S1= 1--k

0.6

la = 1

o.s . fm

t /f

z 0

<

ú i

C/=Cn

L2=LK

f,

r

Cn- t

4 / V I R

/¡ \

='-( f12'oe

)

FREQUENCY FREQUENCY

Figure 40

Through the use of the curves and equations which accompany the diagrams in the illustration above it is

possible to determine the correct values of inductance and capacitance for the usual types of pi- section

filters.

have some value between zero and one. As the

value of m approaches zero, the sharpness of

cutoff increases, but the less will be the

attenuation at several times cutoff frequency.

A value of 0.6 may be used for min most appli-

cations. The "notch" frequency is determined

by the resonant frequency of the tuned filter

element. The amount of attenuation obtained

at the "notch" when a derived section is used

is determined by the effective Q of the reso-

nant arm (figure 39).

Filter Assembly Constant -K sections and de-

rived sections may be cas-

caded to obtain the combined characteristics

of sharp cutoff and good remote frequency

attenuation. Such a filter is known as a com-

posite filter. The amount of attenuation will

depend upon the number of filter sections

used, and the shape of the transmission curve

depends upon the type of filter sections used.

All filters have some insertion loss. This

attenuation is usually uniform to all frequen-

cies within the pass band. The insertion loss

varies with the type of filter, the Q of the

components and the type of termination em-

ployed.

Electric Filter Electric wave filters have long

Design been used in some amateur sia-

tions in the audio channel to

reduce the transmission of unwanted high fre-

quencies and hence to reduce the bandwidth

occupied by a radiophone signal. The effec-

tiveness of a properly designed and properly

used filter circuit in reducing QRM and side -

band splatter should not be underestimated.

In recent years, high frequency filters have

become commonplace in TVI reduction. High -

pass type filters are placed before the input

stage of television receivers to reject the

fundamental signal of low frequency trans-

mitters. Low -pass filters are used in the out-

put circuits of low frequency transmitters to

prevent harmonics of the transmitter from

being radiated in the television channels.

www.americanradiohistory.com

66 Alternating Current Circuits

The chart of figure 40 gives design data

and procedure on the pi- section type of filter.

M- derived sections with an M of 0.6 will be

found to be most satisfactory as the input

section (or half- section) of the usual filter

since the input impedance of such a section

is most constant over the pass band of the

filter section.

Simple filters may use either L, T, or n sec-

tions. Since the rr section is the more com-

monly used type figure 40 gives design data

and characteristics for this type of filter.

A PUSH -PULL 250 -TH AMPLIFIER WITH TVI SHIELD REMOVED

Use of harmonic filters in power leads and antenna circuit reduces radiation of TVI- producing harmonics

of typical push -pull amplifier. Shielded enclosure completes harmonic reduction measures.

www.americanradiohistory.com

CHAPTE:R FOUR

Vacuum Tube Principles

In the previous chapters we have seen the

manner in which an electric current flows

through a metallic conductor as a result of an

electron drift. This drift, which takes place

when there is a difference in potential between

the ends of the metallic conductor, is in addi-

tion to the normal random electron motion

between the molecules of the conductor.

The electron may be considered as a minute

negatively charged particle, having a mass of

9 x 10 -7° gram, and a charge of 1.59 x 10 -19

coulomb. Electrons are always identical,

regardless of the source from which they are

obtained.

An electric current can be caused to flow

through other media than a metallic conductor.

One such medium is an ionized solution, such

as the sulfuric acid electrolyte in a storage

battery. This type of current flow is called

electrolytic conduction. Further, it was shown

at about the turn of the century that an elec-

tric current can be carried by a stream of free

electrons in an evacuated chamber. The flow

of a current in such a manner is said to take

place by electronic conduction. The study of

electron tubes (also called vacuum tubes, or

valves) is actually the study of the control and

use of electronic currents within an evacuated

or partially evacuated chamber.

Since the current flow in an electron tube

takes place in an evacuated chamber, there

must be located within the enclosure both a

source of electrons and a collector for the

electrons which have been emitted. The elec-

tron source is called the cathode, and the

electron collector is usually called the anode.

Some external source of energy must be ap-

plied to the cathode in order to impart suffi-

cient velocity to the electrons within the

cathode material to enable them to overcome

the surface forces and thus escape into the

surrounding medium. In the usual types of

67

electron tubes the cathode energy is applied

in the form of heat; electron emission from a

heated cathode is called tbermionic emission.

In another common type of electron tube, the

photoelectric cell, energy in the form of light

is applied to the cathode to cause photo-

electric emission.

4-1 Thermionic Emission

Electron Emission of electrons from the

Emission cathode of a thermionic electron

tube takes place when the cathode

of the tube is heated to a temperature suffi-

ciently high that the free electrons in the

emitter have sufficient velocity to overcome

the restraining forces at the surface of the

material. These surface forces vary greatly

with different materials. Hence different types

of cathodes must be raised to different temper-

atures to obtain adequate quantities of elec-

tron emission. The several types of emitters

found in common types of transmitting and

receiving tubes will be described in the fol-

lowing paragraphs.

Cathode Types The emitters or cathodes as

used in present -day thermi-

onic electron tubes may be classified into

two groups: the directly- heated or filament

type and the indirectly -heated or heater - cathode

type. Directly- heated emitters may be further

subdivided into three important groups, all

of which are commonly used in modern vacuum

tubes. These classifications are: the pure -

tungsten filament, the thoriated- tungsten

filament, and the oxide -coated filament.

The Pure Tung- Pure tungsten wire was used

sten Filament as the filament in nearly all

the earlier transmitting and

www.americanradiohistory.com

68 Vacuum Tube Principles THE RADIO

Figure 1

ELECTRON TUBE TYPES

The new General E l e c t r i c ceramic triode (68Y4) is shown alongside a con-

ventional miniature tube (6265) and an octal -based receiving tube (25L6). The ceramic

tube is designed for rugged service and features extremely low lead inductance.

receiving tubes. However, the thermionic effi-

ciency of tungsten wire as an emitter (the

number of milliamperes emission per watt of

filament heating power) is quite low, the fila-

ments become fragile after use, their life is

rather short, and they are susceptible to burn-

out at any time. Pure tungsten filaments must

be run at bright white heat (about 2500° Kel-

vin). For these reasons, tungsten filaments

have been replaced in all applications where

another type of filament could be used. They

are, however, still universally employed in

large water -cooled tubes and in certain large,

high -power air- cooled triodes where another

filament type would be unsuitable. Tungsten

filaments are the most satisfactory for high -

power, high -voltage tubes where the emitter

is subjected to positive ion bombardment

caused by the residual gas content of the

tubes. Tungsten is not adversely affected by

such bombardment.

The Thoriated- In the course of experi-

Tungsten Filament ments made upon tungsten

emitters, it was found that

filaments made from tungsten having a small

amount of thoria (thorium oxide) as an im-

purity had much greater emission than those

made from the pure metal. Subsequent develop-

ment has resulted in the highly efficient car-

burized thoriated- tungsten filament as used in

virtually all medium -power transmitting tubes

today.

Thoriated- tungsten emitters consist of a

tungsten wire containing from 1% to 2% thoria.

The activation process varies between dif-

ferent manufacturers of vacuum tubes, but

it is essentially as follows: (1) the tube is

evacuated; (2) the filament is burned for a

short period at about 2800° Kelvin to clean

the surface and reduce some of the thoria

within the filament to metallic thorium; (3)

the filament is burned for a longer period at

about 2100° Kelvin to form a layer of thor-

ium on the surface of the tungsten; (4) the

temperature is reduced to about 1600° Kelvin

and some pure hydrocarbon gas is admitted

to form a layer of tungsten carbide on the

surface of the tungsten. This layer of tungsten

carbide reduces the rate of thorium evapora-

tion from the surface at the normal operating

temperature of the filament and thus increases

the operating life of the vacuum tube. Thor-

ium evaporation from the surface is a natural

consequence of the operation of the thoriated-

tungsten filament. The carburized layer on the

tungsten wire plays another role in acting as

a reducing agent to produce new thorium from

the thoria to replace that lost by evapora-

tion. This new thorium continually diffuses to

the surface during the normal operation of

the filament. The last process, (5), in the

activation of a thoriated tungsten filament con-

sists of re- evacuating the envelope and then

burning or ageing the new filament for a con-

siderable period of time at the normal operat-

ing temperature of approximately 1900°K.

One thing to remember about any type of

filament, particularly the thoriated type, is

that the emitter deteriorates practically as

fast when "standing by" (no plate current) as

it does with any normal amount of emission

load. Also, a thoriated filament may be either

temporarily or permanently damaged by a

heavy overload which may strip the surface

layer of thorium from the filament.

Reactivating Thoriated- tungsten fila-

Thoriated- Tungsten ments (and only thoriated-

Filaments tungsten filaments) which

have lost emission as

a result of insufficient filament voltage, a

severe temporary overload, a less severe ex-

tended overload, or even normal operation

www.americanradiohistory.com

HANDBOOK Types of Emitters 69

Figure 2

V -H -F and U -H -F TUBE TYPES

The tube to the left In this photograph is a 955 "acorn" triode. The 6F4 acorn triode is very similar in

appearance to the 955 but has two leads brought out each for the grid and for the plate connection. The

second tubs Is a 446A "lighthouse" triode. The 2C40, 2C43, and 2C44 ore more recent examples of the

same type tube and are tially the same in external appearance. The third tube from the left is o

2C39 "oilcan" tube. This tube type is essentially the inverse of the lighthouse variety since the cathode

and heater connections come out the small end and the plats is the large finned radiator on the large end.

The use of the finned plate radiator makes the oilcan tube capable of approximately 10 times as much

plate dissipation as the lighthouse type. The tube to the right is the 4X 150A beam tetrads. This tube, a

comparatively recent release, is capable of somewhat greater power output than any of the other tube

types shown, and is rated for full output at 500 Mc. and of reduced output at frequencies greater than

1000 Mc.

may quite frequently be reactivated to their

original characteristics by a process similar

to that of the original activation. However,

only filaments which have not approached too

close to the end of their useful life may be

successfully reactivated.

The actual process of reactivation is rel-

atively simple. The tube which has gone

"flat" is placed in a socket to which only the

two filament wires have been connected. The

filament is then "flashed" for about 20 to 40

seconds at about 1% times normal rated volt-

age. The filament will become extremely bright

during this time and, if there is still some

thoria left in the tungsten and if the tube did

not originally fail as a result of an air leak,

some of this thoria will be reduced to metallic

thorium. The filament is then burned at 15 to

25 per cent overvoltage for from 30 minutes to

3 to 4 hours to bring this new thorium to the

surface.

The tube should then be tested to see if it

shows signs of renewed life. If it does, but is

still weak, the burning process should be con-

tinued at about 10 to 15 per cent overvoltage

for a few more hours. This should bring it

back almost to normal. If the tube checks still

very low after the first attempt at reactivation,

the complete process can be repeated as a

last effort.

The Oxide. The most efficient of all

Coated Filament modern filaments is the

oxide -coated type which con-

sists of a mixture of barium and strontium

oxides coated upon a nickel alloy wire or

strip. This type of filament operates at a dull -

red to orange -red temperature (1050' to 1170°

K) at which temperature it will emit large

quantities of electrons. The oxide -coated

filament is somewhat more efficient than the

thoriated- tungsten type in small sizes and it

is considerably less expensive to manufacture.

For this reason all receiving tubes and quite

a number of the low -powered transmitting

tubes use the oxide - coated filament. Another

advantage of the oxide -coated emitter is its

extremely long life - the average tube can be

expected to run from 3000 to 5000 hours, and

when loaded very lightly, tubes of this type

have been known to give 50,000 hours of life

before their characteristics changed to any

great extent.

Oxide filaments are unsatisfactory for use

at high continuous plate voltages because: (1)

their activity is seriously impaired by the

high temperature necessary to de -gas the high -

voltage tubes and, (2) the positive ion bom-

bardment which takes place even in the best

evacuated high -voltage tube causes destruc-

tion of the oxide layer on the surface of the

filament.

Oxide -coated emitters have been found cap-

able of emitting an enormously large current

pulse with a high applied voltage for a very

short period of time without damage. This

characteristic has proved to be of great value

www.americanradiohistory.com

70 Vacuum Tube Principles THE RADIO

Figure 3

CUTAWAY DRAWING OF A 6C4 TRIODE

in radar work. For example, the relatively

small cathode in a microwave magnetron may

be called upon to deliver 25 to 50 amperes at

an applied voltage of perhaps 25,000 volts for

a period in the order of one microsecond.

After this large current pulse has been passed,

plate voltage normally will be removed for

1000 microseconds or more so that the cathode

surface may be restored in time for the next

pulse of current. If the cathode were to be

subjected to a continuous current drain of this

magnitude, it would be destroyed in an ex-

ceedingly short period of time.

The activation of oxide- coated filaments

also varies with tube manufacturers but con-

sists essentially in heating the wire which has

been coated with a mixture of barium and

strontium carbonates to a temperature of about

1500° Kelvin for a time and then applying a

potential of 100 to 200 volts through a pro-

tective resistor to limit the emission current.

This process reduces the carbonates to oxides

thermally, cleans the filament surface of

foreign materials, and activates the cathode

surface.

Reactivation of oxide- coated filaments is

not possible since there is always more than

sufficient reduction of the oxides and diffusion

of the metals to the surface of the filament to

meet the emission needs of the cathode.

The Heater The heater type cathode was de-

Cathode veloped as a result of the re-

quirement for a type of emitter

which could be operated from alternating cur-

rent and yet would not introduce a -c ripple

modulation even when used in low -level stages.

It consists essentially of a small nickel -alloy

cylinder with a coating of strontium and bar-

ium oxides on its surface similar to the coat-

ing used on the oxide- coated filament. Inside

the cylinder is an insulated heater element

consisting usually of a double spiral of tung-

sten wire. The heater may operate on any volt-

age from 2 to 117 volts, although 6.3 is the

most common value. The heater is operated

at quite a high temperature so that the cathode

itself usually may be brought to operating

temperature in a matter of 15 to 30 seconds.

Heat -coupling between the heater and the

cathode is mainly by radiation, although there

is some thermal conduction through the in-

sulating coating on the heater wire, as this

coating is also in contact with the cathode

thimble.

Indirectly heated cathodes are employed in

all a-c operated tubes which are designed to

operate at a low level either for r -f or a -f use.

However, some receiver power tubes use

heater cathodes (6L6, 6V6, 6F6, and 6K6 -GT)

as do some of the low -power transmitter tubes

(802, 807, 815, 3E29, 2E26, 5763, etc.). Heater

cathodes are employed almost exclusively

when a number of tubes are to be operated in

series as in an a.c. -d.c. receiver. A heater

cathode is often called a uni- potential cathode

because there is no voltage drop along its

length as there is in the directly- heated or

filament cathode.

The Bombardment A special bombardment cath-

Cathode ode is employed in many of

the new high powered tele-

vision transmitting klystrons(Eimac 3K 20,000

LA). The cathode takes the form of a tantalum

diode, heated to operating temperature by the

bombardment of electrons from a directly

heated filament. The cathode operates at a

positive potential of 2000 volts with respect

to the filament, and a d -c bombardment cur-

rent of 0.66 amperes flows between filament

and cathode. The filament is designed to

operate under space -charge limited conditions.

Cathode temperature is varied by changing the

bombardment potential between the filament

and the cathode.

The Emission The emission of electrons from

Equation a heated cathode is quite sim-

ilar to the evaporation of mole-

cules from the surface of a liquid. The mole-

cules which leave the surface are those having

sufficient kinetic (heat) energy to overcome

the forces at the surface of the liquid. As the

temperature of the liquid is raised, the aver-

age velocity of the molecules is increased,

and a greater number of molecules will acquire

sufficient energy to be evaporated. The evapor-

ation of electrons from the surface of a ther-

mionic emitter is similarly a function of aver-

age electron velocity, and hence is a function

of the temperature of the emitter.

Electron emission per unit area of emitting

surface is a function of the temperature T

in degrees Kelvin, the work function of the

emitting surface b (which is a measure of the

www.americanradiohistory.com

HANDBOOK Thermionic Emission 71

/ PLATE

35LP?RS314R

ÌSCREEN frT.

4910 __-,

çArHOOE -

6CB6

L ww

HEATER

HEATER - - - -- ,

Figure 4

CUT -AWAY DRAWING OF A 6CB6 PENTODE

surface forces of the material and hence of

the energy required of the electron before it

may escape), and of the constant A which

also varies with the emitting surface. The re-

lationship between emission current in am-

peres per square centimeter, 1, and the above

quantities can be expressed as:

1 = AT'c"b'T

Secondary The bombarding of most metals

Emission and a few insulators by electrons

will result in the emission of other

electrons by a process called secondary emis-

sion. The secondary electrons are literally

knocked from the surface layers of the bom-

barded material by the primary electrons which

strike the material. The number of secondary

electrons emitted per primary electron varies

from a very small percentage to as high as

5 to 10 secondary electrons per primary.

The phenomena of secondary emission is

undesirable for most thermionic electron tubes.

However, the process is used to advantage in

certain types of electron tubes such as the

image orthicon (TV camera tube) and the

electron -multiplier type of photo -electric cell.

In types of electron tubes which make use of

secondary emission, such as the type 931

photo cell, the secondary- electron -emitting

surfaces are specially treated to provide a

high ratio of secondary to primary electrons.

Thus a high degree of current amplification in

the electron -multiplier section of the tube is

obtained.

The Space As a cathode is heated so that

Charge Effect it begins to emit, those elec-

trons which have been dis-

charged into the surrounding space form a

negatively charged cloud in the immediate

vicinity of the cathode. This cloud of electrons

around the cathode is called the space charge.

The electrons comprising the charge are con-

tinuously changing, since those electrons

making up the original charge fall back into

600

w ¢ W

400

i

F 200

á

TYPE 6W4-GT

Er' 63 VOLTS

10 20 30

D.C. PLATE VOLTS

Figure 5

ao so

AVERAGE PLATE CHARACTERISTICS

OF A POWER DIODE

the cathode and are replaced by others emitted

by it.

4 -2 The Diode

If a cathode capable of being heated either

indirectly or directly is placed in an evacuated

envelope along with a plate, such a two -

element vacuum tube is called a diode. The

diode is the simplest of all vacuum tubes and

is the fundamental type from which all the

others are derived.

Characteristics When the cathode within a

of the Diode diode is heated, it will be

found that a few of the elec-

trons leaving the cathode will leave with suf-

ficient velocity to reach the plate. If the plate

is electrically connected back to the cathode,

the electrons which have had sufficient veloc=

ity to arrive at the plate will flow back to the

cathode through the external circuit. This

small amount of initial plate current is an

effect found in all two -element vacuum tubes.

If a battery or other source of d -c voltage

is placed in the external circuit between the

plate and cathode so that it places a positive

potential on the plate, the flow of current from

the cathode to plate will be increased. This is

due to the strong attraction offered by the posi-

tively charged plate for any negatively charged

particles (figure 5).

Space- Charge Limited At moderate values of

Current plate voltage the cur-

rent flow from cathode

to anode is limited by the space charge of

electrons around the cathode. Increased values

www.americanradiohistory.com

72 Vacuum Tube Principles THE RADIO

DE COATED

ED TUNGSTEN

TUNGSTEN FILAMENT

POINT OF MAXIMUM SPACE -

CHARGE -LIMITED EMISSION

PLATE VOLTAGE +

Figure 6

MAXIMUM SPACE -CHARGE -LIMITED

EMISSION FOR DIFFERENT

TYPES OF EMITTERS

of plate voltage will tend to neutralize a

greater portion of the cathode space charge

and hence will cause a greater current to flow.

Under these conditions, with plate current

limited by the cathode space charge, the plate

current is not linear with plate voltage. In

fact it may be stated in general that the plate -

current flow in electron tubes does not obey

Ohm's Law. Rather, plate current increases as

the three -halves power of the plate voltage.

The relationship between plate voltage, E,

and plate current, 1, can be expressed as:

/ =K F3!2

where K is a constant determined by the

geometry of the element structure within the

electron tube.

Plate Current As plate voltage is raised to

Saturation the potential where the cath-

ode space charge is neutral-

ized, all the electrons that the cathode is cap-

able of emitting are being attracted to the

plate. The electron tube is said then to have

reached saturation plate current. Further in-

crease in plate voltage will cause only a

relatively small increase in plate current. The

initial point of plate current saturation is

sometimes called the point of Maximum Space -

Charge- Limited Emission (MSCLE).

The degree of flattening in the plate -voltage

plate- current curve after the MSCLE point will

vary with different types of cathodes. This ef-

fect is shown in figure 6. The flattening is

quite sharp with a pure tungsten emitter. With

thoriated tungsten the flattening is smoothed

somewhat, while with an oxide- coated cathode

the flattening is quite gradual. The gradual

saturation in emission with an oxide- coated

emitter is generally considered to result from

Figure 7

ACTION OF THE GRID IN A TRIODE

(A) shows the triode tube with cutoff bias on

the grid. Note that all the electrons emitted

by the cathode remain inside the grid mesh.

(B) shows the same tube with an intermediate

value of bias on the grid. Note the medium

value of plate current and the fact that there

is a reserve of electrons remaining within the

grid mesh. (C) shows the operation with a

relatively small amount of bias which with

certain tube types will allow substantially all

the electrons emitted by the cathode to reach

the plate. Emission is said to be saturated in

this case. In a majority of tube types a high

value of positive grid voltage is required be-

fore plate - current saturation takes place.

a lowering of the surface work function by the

field at the cathode resulting from the plate

potential.

Electron Energy The current flowing in the

plate- cathode space of a con-

ducting electron tube repre-

sents the energy required to accelerate elec-

trons from the zero potential of the cathode

space charge to the potential of the anode.

Then, when these accelerated electrons strike

the anode, the energy associated with their

velocity is immediately released to the anode

structure. In normal electron tubes this energy

release appears as heating of the plate or

anode structure.

Dissipation

4 -3 The Triode

If an element consisting of a mesh or spiral

of wire is inserted concentric with the plate

and between the plate and the cathode, such

an element will be able to control by electro-

static action the cathode -to -plate current of

the tube. The new element is called a grid, and

a vacuum tube containing a cathode, grid, and

plate is commonly called a triode.

Action of If this new element through which

the Grid the electrons must pass in their

course from cathode to plate is made

negative with respect to the cathode, the nega-

www.americanradiohistory.com

HANDBOOK Triode Characteristics 73

16

N TYPE 6J5

s! 4

A mom Er=6.5VOLTS

MI litiliI MM 1I 1111111

italar .111!

..01 /!!. ! !

; /I AMÍGlI pp.

A IM

o 100 200 300 400

PLATE VOLTS (EP)

Figure 8

NEGATIVE -GRID CHARACTERISTICS(Ip

VS. Ep CURVES) OF A TYPICAL

TRIODE

Average plate characteristics of this type

are most commonly used in determining the

Class A operating characteristics of a

triode amplifier stage.

500

cive charge on this grid will effectively repel

the negatively charged electrons (like charges

repel; unlike charges attract) back into the

space charge surrounding the cathode Hence,

the number of electrons which are able to pass

through the grid mesh and reach the plate will

be reduced, and the plate current will be re-

duced accordingly. If the charge on the grid

is made sufficiently negative, all the electrons

leaving the cathode will be repelled back to

it and the plate current will be reduced to zero.

Any d -c voltage placed upon a grid is called

a bias (especially so when speaking of a con-

trol grid). The smallest negative voltage which

will cause cutoff of plate current at a particu-

lar plate voltage is called the value of cutoff

bias (figure 7).

Amplification The amount of plate current in a

Factor triode is a result of the net field

at the cathode from interaction

between the field caused by the grid bias and

that caused by the plate voltage. Hence, both

grid bias and plate voltage affect the plate

current. In all normal tubes a small change in

grid bias has a considerably greater effect

than a similar change in plate voltage. The

ratio between the change in grid bias and the

change in plate current which will cause the

same small change in plate current is called

the amplification /actor or of the electron

tube. Expressed as an equation:

AE

- AE,

with i, constant (A represents a small incre-

ment).

The µ can be determined experimentally by

making a small change in grid bias, thus

slightly changing the plate current. The plate

current is then returned to the original value

by making a change in the plate voltage. The

ratio of the change in plate voltage to the

change in grid voltage is the µ of the tube

under the operating conditions chosen for the

test.

Current Flow In a diode it was shown that

in a Triode the electrostatic field at the

cathode was proportional to

the plate potential, Ep, and that the total

cathode current was proportional to the three -

halves power of the plate voltage. Similiarly,

in a triode it can be shown that the field at

the cathode space charge is proportional to

the equivalent voltage (Eg + Ep /ft), where

the amplification factor, µ, actually represents

the relative effectiveness of grid potential and

plate potential in producing a field at the

cathode.

It would then be expected that the cathode

current in a triode would be proportional to

the three -halves power of (E5 + Ep /µ). The

cathode current of a triode can be represented

with fair accuracy by the expression:

E 3/2

Cathode current = K (E5 +-=-) )

where K is a constant determined by element

geometry within the triode.

Plate Resistance The plate resistance of a

vacuum tube is the ratio of a

change in plate voltage to the change in plate

current which the change in plate voltage

produces. To be accurate, the changes should

be very small with respect to the operating

values. Expressed as an equation:

AE

Rp = P E,= constant, A = small

p increment

The plate resistance can also be determined

by the experiment mentioned above. By noting

the change in plate current as it occurs when

the plate voltage is changed (grid voltage

held constant), and by dividing the latter by

the former, the plate resistance can be deter-

mined. Plate resistance is expressed in Ohms.

Transconductance The mutual conductance,

also referred to as trans -

conductance, is the ratio of a change in the

plate current to the change in grid voltage

which brought about the plate current change,

the plate voltage being held constant. Ex-

pressed as an equation:

www.americanradiohistory.com

74 Vacuum Tube Principles THE RADIO

430

400

330

300

2

_ W 6 seo

6

W

C 100

110

130

o -0 -e -10 10 10 /0 40 10 b 70 e0 10 100

GRID VOLTAGE (E9)

Figure 9

POSITIVE -GRID CHARACTERISTICS

(Ip VS. E9) OF A TYPICAL TRIODE

Plate characteristics of this type are most

commonly used in determining the pulse -signal

operating characteristics of a triode amplifier

stage. Note the large emission capability of

the oxidecoated heater cathode in tubes of the

9 1 type of the 6J5.

Al

= Gm E = constant, A = small

AEg increment

The transconductance is also numerically

equal to the amplification factor divided by

the plate resistance. Gm =

Transconductance is most commonly ex-

pressed in microreciprocal -ohms or micro -

mhos. However, since transconductance ex-

presses change in plate current as a function

of a change in grid voltage, a tube is often

said to have a transconductance of so many

milliamperes- per -volt. If the transconductance

in milliamperes -per -volt is multiplied by 1000

it will then be expressed in micromhos. Thus

the transconductance of a 6A3 could be called

either 5.25 ma.!volt or 5250 micromhos.

Characteristic Curves The operating character -

of a Triode Tube istics of a triode tube

may be summarized in

three sets of curves: The Ip vs. Ep curve

(figure 8), the Ip vs. Eg curve (figure 9) and

the Ep vs. E curve (figure 10). The plate

resistance (Rj of the tube may be observed

from the Ip vs. Ep curve, the transconductance

(Gm) may be observed from the Ip vs. Eg curve,

and the amplification factor (pt) may be deter-

mined from the Ep vs. Eg curve.

The Load Line A load line is a graphical

representation of the voltage

on the plate of a vacuum tube, and the current

-20 -11 -10 -5 0 1-5

GRID VOLTS (Es)

Figure 10

CONSTANT CURRENT (Ep VS. E9)

CHARACTERISTICS OF A

TYPICAL TRIODE TUBE

This type of graphical representation is used

for Class C amplifier calculations since the

operating characteristic of a Class C amplifier

is a straight line when drawn upon a constant

current- graph.

passing through the plate circuit of the tube

for various values of plate -load resistance and

plate - supply voltage. Figure 11 illustrates a

triode tube with a resistive plate load, and a

supply voltage of 300 volts. The voltage at

the plate of the tube (ep) may be expressed

as: ep = Ep -(i x RL)

where Ep is the plate supply voltage, ip is the

plate current, and RL is the load resistance in

ohms.

Assuming various values of ip flowing in

the circuit, controlled by the internal resist-

ance of the tube, (a function of the grid bias)

values of plate voltage may be plotted as

shown for each value of plate current (ir). The

line connecting these points is called the

load line for the particular value of plate -load

resistance used. The slope of the load line is

equal to the ratio of the lengths of the vertical

and horizontal projections of any segment of

the load line. For this example it is:

.01 - .02 1

Slope = - - .0001 -

100 - 200 10,000

The slope of the load line is equal to

-1/11L. At point A on the load line, the volt-

age across the tube is zero. This would be

true for a perfect tube with zero internal volt-

age drop, or if the tube is short -circuited from

cathode to plate. Point B on the load line

corresponds to the cutoff point of the tube,

where no plate current is flowing. The op-

erating range of the tube lies between these

two extremes. For additional information re-

www.americanradiohistory.com

HANDBOOK Triode Load Line 75

EP=300v

IP(YA)i En I

300

S 250

10 200

IS I SO

20 100

25 50

30 0

EP

Figure 11

The static load line for a typical triode

tube with a plate load resistance of 10,000

ohms.

garding dynamic load lines, the reader is

referred to the Radiotron Designer's Handbook,

4th edition, distributed by Radio Corporation

of America.

Application of Tube As an example of the ap-

Characteristics plication of tube charac-

teristics, the constants

of the triode amplifier circuit shown in figure

12 may be considered. The plate supply is

RL=en

Figure 12

TRIODE TUBE CONNECTED FOR DETER-

MINATION OF PLATE CIRCUIT LOAD

LINE, AND OPERATING PARAMETERS

OF THE CIRCUIT

300 volts, and the plate load is 8,000 ohms.

If the tube is considered to be an open circuit

no plate current will flow, and there is no

voltage drop across the plate load resistor,

RL. The plate voltage on the tube is therefore

300 volts. If, on the other hand, the tube is

considered to be a short circuit, maximum

possible plate current flows and the full 300

volt drop appears across RL. The plate volt-

age is zero, and the plate current is 300/8,000,

or 37.5 milliamperes. These two extreme con-

ditions define the load line on the I, vs. Ep

characteristic curve, figure 13.

For this application the grid of the tube is

returned to a steady biasing voltage of -4

volts. The steady or quiescent operation of the

tube is determined by the intersection of the

load line with the -4 volt curve at point Q.

By projection from point Q through the plate

Figure 13

APPLICATION OF Ip VS. Ep

CHARACTERISTICS OF

VACUUM TUBE

40

37.5

3s

i 2

30

2

IP11145-410.2

0J.

LOGO LINO =

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- - --i - -

14N.

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N VOLT Pt.AT[ SWING

300 400

PLATE VOLTS (E.)

www.americanradiohistory.com

76 Vacuum Tube Principles THE RADIO

EG -4r \ . D.C. BIAS LEVEL (EC)

+ 18.23

EP

+is

T-

STEADY STATE //

PLATE CURRENT\i)1

STEADY STATE (EP)

PLATE VOLTAGE

Figure 14

POLARITY REVERSAL BETWEEN GRID

AND PLATE VOLTAGES

current axis it is found that the value of plate

current with no signal applied to the grid is

12.75 milliamperes. By projection from point

Q through the plate voltage axis it is found

that the quiescent plate voltage is 198 volts.

This leaves a drop of 102 volts across RL

which is borne out by the relation 0.01275 x

8,000 = 102 volts.

An alternating voltage of 4 volts maximum

swing about the normal bias value of -4 volts

is applied now to the grid of the triode ampli-

fier. This signal swings the grid in a positive

direction to 0 volts, and in a negative direction

to -8 volts, and establishes the operating

region of the tube along the load line between

points A and B. Thus the maxima and minima

of the plate voltage and plate current are

established. By projection from points A and

B through the plate current axis the maximum

instantaneous plate current is found to be

18.25 milliamperes and the minimum is 7.5

milliamperes. By projections from points A and

B through the plate voltage axis the minimum

instantaneous plate voltage swing is found to

be 154 volts and the maximum is 240 volts.

By this graphical application of the IP vs.

Ep characteristic of the 6SN7 triode the opera-

tion of the circuit illustrated in figure 12 be-

Figure 15

SCHEMATIC REPRESENTATION

OF INTERELECTRODE

CAPACITANCE

comes apparent. A voltage variation of 8 volts

(peak -to -peak) on the grid produces a variation

of 84 volts at the plate.

Polarity Inversion When the signal voltage ap-

plied to the grid has its

maximum positive instantaneous value the

plate current is also maximum. Reference to

figure 12 shows that this maximum plate cur-

rent flows through the plate load resistor RL,

producing a maximum voltage drop across it.

The lower end of RL is connected to the plate

supply, and is therefore held at a constant

potential of 300 volts. With maximum voltage

drop across the load resistor, the upper end of

RL is at a minimum instantaneous voltage.

The plate of the tube is connected to this end

of RL and is therefore at the same minimum

instantaneous potential.

This polarity reversal between instantaneous

grid and plate voltages is further clarified by

a consideration of Kirchhoff's law as it ap-

plies to series resistance. The sum of the IR

drops around the plate circuit must at all times

equal the supply voltage of 300 volts. Thus

when the instantaneous voltage drop across

RL is maximum, the voltage drop across the

tube is minimum, and their sum must equal

300 volts. The variations of grid voltage,plate

current and plate voltage about their steady

state values is illustrated in figure 14.

Interelectrode Capacitance always exists be-

Capacitance tween any two pieces of metal

separated by a dielectric. The

exact amount of capacitance depends upon the

size of the metal pieces, the dielectric be-

tween them, and the type of dielectric. The

electrodes of a vacuum tube have a similar

characteristic known as the interelectrode

capacitance, illustrated in figure 15. These

direct capacities in a triode are: grid -to-

cathode capacitance, grid -to -plate capacitance,

and plate -to- cathode capacitance. The inter -

electrode capacitance, though very small, has

a coupling effect, and often can cause un-

balance in a particular circuit. At very high

www.americanradiohistory.com

HANDBOOK Tetrodes and Pentodes 77

6

4

TYPE 24 -A

esc =so v.

200 300

VOLTS (Eel

Figure 16

TYPICAL Ip VS. Ep TETRODE

CHARACTERISTIC CURVES

S00

G=-3

u `

cr u O. 4

J

TYPE 6SK7

esc= too v.

esu =ov.

100 200 300 400 500

VOLTS (E)

Figure 17

TYPICAL IP VS. EP PENTODE

CHARACTERISTIC CURVES

frequencies (v -h -f), interelectrode capacities

become very objectionable and prevent the use

of conventional tubes at these frequencies.

Special v -h -f tubes must be used which are

characterized by very small electrodes and

close internal spacing of the elements of the

tube.

4 -4 Tetrode or Screen Grid Tubes

Many desirable characteristics can be ob-

tained in a vacuum tube by the use of more

than one grid. The most common multi -element

tube is the tetrode (four electrodes). Other

tubes containing as many as eight electrodes

are available for special applications.

The Tetrode The quest for a simple and easily

usable method of eliminating the

effects of the grid -to -plate capacitance of the

triode led to the development of the screen -

grid tube or tetrode. When another grid is

added between the grid and plate of a vacuum

tube the tube is called a tetrode, and because

the new grid is called a screen, as a result of

its screening or shielding action, the tube is

often called a screen -grid tube. The inter-

posed screen grid acts as an electrostatic

shield between the grid and plate, with the

consequence that the grid -to -plate capacitance

is reduced. Although the screen grid is main-

tained at a positive voltage with respect to

the cathode of the tube, it is maintained at

ground potential with respect to r.f. by means

of a by -pass capacitor of very low reactance

at the frequency of operation.

In addition to the shielding effect, the

screen grid serves another very useful purpose.

Since the screen is maintained at a positive

potential, it serves to increase or accelerate

the flow of electrons to the plate. There being

large openings in the screen mesh, most of

the electrons pass through it and on to the

plate. Due also to the screen, the plate cur-

rent is largely independent of plate voltage,

thus making for high amplification. When the

screen voltage is held at a constant value, it

is possible to make large changes in plate

voltage without appreciably affecting the plate

current, (figure 16).

When the electrons from the cathode ap-

proach the plate with sufficient velocity, they

dislodge electrons upon striking the plate.

This effect of bombarding the plate with high

velocity electrons, with the consequent dis-

lodgement of other electrons from the plate,

gives rise to the condition of secondary emis-

sion which has been discussed in a previous

paragraph. This effect can cause no particular

difficulty in a triode because the secondary

electrons so emitted are eventually attracted

back to the plate. In the screen -grid tube, how-

ever, the screen is close to the plate and is

maintained at a positive potential. Thus, the

screen will attract these electrons which have

been knocked from the plate, particularly when

the plate voltage falls to a lower value than

the screen voltage, with the result that the

plate current is lowered and the amplification

is decreased.

In the application of tetrodes, it is neces-

sary to operate the plate at a high voltage in

relation to the screen in order to overcome

these effects of secondary emission.

The Pentode The undesirable effects of sec-

ondary emission from the plate

can be greatly reduced if yet another element

is added between the screen and plate. This

additional element Is called a suppressor, and

tubes in which it is used are called pentodes.

The suppressor grid is sometimes connected

to the cathode within the tube; sometimes it is

brought out to a connecting pin on the tube

base, but in any case it is established nega-

www.americanradiohistory.com

78 Vacuum Tube Principles THE RADIO

Cÿ

GRiD

- C HODE

.L-

REMOTE CUT -OFF SHARP CUT -OFF

GRID GRID

Figure 18

REMOTE CUTOFF GRID STRUCTURE

- GRID VOLTS

Figure 19

ACTION OF A REMOTE CUTOFF

GRID STRUCTURE

tive with respect to the minimum plate volt-

age. The secondary electrons that would travel

to the screen if there were no suppressor are

diverted back to the plate. The plate current

is, therefore, not reduced and the amplifica-

tion possibilities are increased (figure 17).

Pentodes for audio applications are de-

signed so that the suppressor increases the

limits to which the plate voltage may swing;

therefore the consequent power output and

gain can be very great. Pentodes for radio -

frequency service function in such a manner

that the suppressor allows high voltage gain,

at the same time permitting fairly high gain

at low plate voltage. This holds true even if

the plate voltage is the same or slightly lower

than the screen voltage.

Remote Cutoff Remote cutoff tubes (variable

Tubes mu) are screen grid tubes in

which the control grid struc-

ture has been physically modified so as to

cause the plate current of the tube to drop off

gradually, rather than to have a well defined

cutoff point (figure 18). A non -uniform control

grid structure is used, so that the amplifica-

tion factor is different for different parts of the

control grid.

Remote cutoff tubes are used in circuits

where it is desired to control the amplification

by varying the control grid bias. The charac-

teristic curve of an ordinary screen grid tube

has considerable curvature near the plate cur-

rent cutoff point, while the curve of a remote

cutoff tube is much more linear (figure 19).

The remote cutoff tube minimizes cross-

talk interference that would otherwise be

produced. Examples of remote cutoff tubes

are: 6BD6, 6K7, 6SG7 and 6SK7.

Beam Power A beam power tube makes use

Tubes of another method for suppressing

secondary emission. In this tube

there are four electrodes: a cathode, a grid, a

screen, and a plate, so spaced and placed that

secondary emission from the plate is sup-

pressed without actual power loss. Because

of the manner in which the electrodes are

spaced, the electrons which travel to the

plate are slowed down when the plate voltage

is low, almost to zero velocity in a certain

region between screen and plate. For this

reason the electrons form a stationary cloud,

or space charge. The effect of this space

charge is to repel secondary electrons emitted

from the plate and thus cause them to return

to the plate. In this way, secondary emission

is suppressed.

Another feature of the beam power tube is

the low current drawn by the screen. The

screen and the grid are spiral wires wound so

that each turn in the screen is shaded from

the cathode by a grid turn. This alignment of

the screen and the grid causes the electrons

to travel in sheets between the turns of the

screen so that very few of them strike the

screen itself. This formation of the electron

stream into sheets or beams increases the

charge density in the screen -plate region and

assists in the creation of the space charge in

this region.

Because of the effective suppressor action

provided by the space charge, and because of

the low current drawn by the screen, the beam

power tube has the advantages of high power

output, high power -sensitivity, and high ef-

ficiency. The 6L6 is such a beam power tube,

designed for use in the power amplifier stages

of receivers and spec -h amplifiers or modulat-

ors. Larger tubes employing the beam -power

principle are being made by various manu-

facturers for use in the radio -frequency stages

of transmitters. These tubes feature extremely

high power- sensitivity (a very small amount

of driving power is required for a large out-

put), good plate efficiency, and low grid -to-

plate capacitance. Examples of these tubes

are 813, 4 -250A, 4X150A, etc.

Grid -Screen The grid - screen mu factor (Ass)

Mu Factor is analogous to the amplification

factor in a triode, except that

the screen of a pentode or tetrode is sub-

www.americanradiohistory.com

HANDBOOK Mixer and Converter Tubes 79

stituted for the plate of a triode. µ5g denotes

the ratio of a change in grid voltage to a

change in screen voltage, each of which will

produce the same change in screen current.

Expressed as an equation:

AEss

flag = Ise = constant, A = small

AEs increment

The grid- screen mu factor is important in

determining the operating bias of a tetrode

or pentode tube. The relationship between con-

trol -grid potential and screen potential deter-

mines the plate current of the tube as well as

the screen current since the plate current is

essentially independent of the plate voltage

in tubes of this type. In other words, when

the tube is operated at cutoff bias as deter-

mined by the screen voltage and the grid -

screen mu factor (determined in the same way

as with a triode, by dividing the operating

voltage by the mu factor) the plate current

will be substantially at cutoff, as will be the

screen current. The grid- screen mu factor is

numerically equal to the amplification factor

of the same tetrode or pentode tube when

it is triode connected.

Current Flow The following equation is the

in Tetrodes expression for total cathode cur -

and Pentodes rent in a triode tube. The ex-

pression for the total cathode

current of a tetrode and a pentode tube is the

same, except that the screen -grid voltage and

the grid- screen it-factor are used in place of

the plate voltage and it of the triode.

/ E 3/2

Cathode current = K 1 Es + $g )

Ilse

Cathode current, of course, is the sum of the

screen and plate current, plus control grid cur-

rent in the event that the control grid is posi-

tive with respect to the cathode. It will be

noted that total cathode current is independent

of plate voltage in a tetrode or pentode. Also,

in the usual tetrode or pentode the plate cur-

rent is substantially independent of plate

voltage over the usual operating range- which

means simply that the effective plate resist-

ance of such tubes is relatively high. How-

ever, when the plate voltage falls below the

normal operating range, the plate current

falls sharply, while the screen current rises to

such a value that the total cathode current

remains substantially constant. Hence, the

screen grid in a tetrode or pentode will almost

invariably be damaged by excessive dissipa-

tion if the plate voltage is removed while the

screen voltage is still being applied from a

low -impedance source.

The Effect of The current equations show how

Grid Current the total cathode current in

triodes, tetrodes, and pentodes

is a function of the potentials applied to the

various electrodes. If only one electrode is

positive with respect to the cathode (such as

would be the case in a triode acting as a

class A amplifier) all the cathode current goes

to the plate. But when both screen and plate

are positive in a tetrode or pentode, the cath-

ode current divides between the two elements.

Hence the screen current is taken from the

total cathode current, while the balance goes

to the plate. Further, if the control grid in a

tetrode or pentode is operated at a positive

potential the total cathode current is divided

between all three elements which have a posi-

tive potential. In a tube which is receiving a

large excitation voltage, it may be said that

the control grid robs electrons from the output

electrode during the period that the grid is

positive, making it always necessary to limit

the peak -positive excursion of the control

grid.

Coefficients of In general it may be stated

Tetrodes and that the amplification factor

Pentodes of tetrode and pentode tubes

is a coefficient which is not

of much use to the designer. In fact the ampli-

fication factor is seldom given on the design

data sheets of such tubes. Its value is usually

very high, due to the relatively high plate

resistance of such tubes, but bears little

relationship to the stage gain which actually

will be obtained with such tubes.

On the other hand, the grid -plate transcon-

ductance is the most important coefficient of

pentode and tetrode tubes. Gain per stage can

be computed directly when the Gm is known.

The grid -plate transconductance of a tetrode

or pentode tube can be calculated through use

of the 'expression:

Alp

Gm = AE e

with E5s and Es constant.

The plate resistance of such tubes is of

less importance than in the case of triodes,

though it is often of value in determining the

amount of damping a tube will exert upon the

impedance in its plate circuit. Plate resist-

ance is calculated from:

AEp

R

v

with Es and Esg constant.

4 -5 Mixer and Converter Tubes

The superheterodyne receiver always in-

www.americanradiohistory.com

80 Vacuum Tube Principles THE RADIO

CAT NODE

OSCILLATOR GRID

r_FSCREEN GRID

- PLATE

METAL SPELL

FILAMENT ` SUPPRESSOR AND SHELL

SIGNAL GRID

Figure 20

GRID STRUCTURE OF 6SA7

CONVERTER TUBE

eludes at least one stage for changing the

frequency of the incoming signal to the fixed

frequency of the main intermediate amplifier

in the receiver. This frequency changing

process is accomplished by selecting the

beat -note difference frequency between a

locally generated oscillation and the incoming

signal frequency. If the oscillator signal is

supplied by a separate tube, the frequency

changing tube is called a mixer. Alternatively,

the oscillation may be generated by additional

elements within the frequency changer tube.

In this case the frequency changer is common-

ly called a converter tube.

Conversion The conversion conductance(Ge)

Conductance is a coefficient of interest in the

case of mixer or converter tubes,

or of conventional triodes, tetrodes, or pen-

todes operating as frequency changers. The

conversion conductance is the ratio of a

change in the signal -grid voltage at the input

frequency to a change in the output current at

the converted frequency. Hence Gc in a mixer

is essentially the same as transconductance

in an amplifier, with the exception that the

input signal and the output current are on dif-

ferent frequencies. The value of G, in con-

ventional mixer tubes is from 300 to 1000

micromhos. The value of G, in an amplifier

tube operated as a mixer is approximately 0.3

the Gm of the tube operated as an amplifier.

The voltage gain of a mixer stage is equal to

GCZL where ZL is the impedance of the plate

load into which the mixer tube operates.

The Diode Mixer The simplest mixer tube is

the diode. The noise figure,

or figure of merit, for a mixer of this type is

not as good as that obtained with other more

complex mixers; however, the diode is useful

as a mixer in u -h -f and v -h -f equipment where

low interelectrode capacities are vital to cir-

cuit operation. Since the diode impedance is

Figure 21

SHOWING THE EFFECT OF CATHODE

LEAD INDUCTANCE

The degenerative action of cathode lead in-

ductance tends to reduce the effective grid-to-

cathode voltage with respect to the voltage

available across the input tuned circuit. Cath-

ode lead inductance also introduces undesir-

able coupling between the input and the out-

put circuits.

low, the local oscillator must furnish con-

siderable power to the diode mixer. A good

diode mixer has an overall gain of about 0.5.

The Triode Mixer A triode mixer has better

gain and a better noise figure

than the diode mixer. At low frequencies, the

gain and noise figure of a triode mixer closely

approaches those figures obtained when the

tube is used as an amplifier. In the u -h -f and

v -h -f range, the efficiency of the triode mixer

deteriorates rapidly. The optimum local oscil-

lator voltage for a triode mixer is about 0.7 as

large as the cutoff bias of the triode. Very

little local oscillator power is required by a

triode mixer.

Pentode Mixers and The most common multi -

Converter Tubes grid converter tube for

broadcast or shortwave

use is the penta grid converter, typified by

the 6SA7, 6SB7 -Y and 6BA7 tubes (figure 20).

Operation of these converter tubes and pentode

mixers will be covered in the Receiver Funda-

mentals Chapter.

4 -6 Electron Tubes at Very

High Frequencies

As the frequency of operation of the usual

type of electron tube is increased above about

20 Mc., certain assumptions which are valid

for operation at lower frequencies must be re-

examined. First, we find that lead inductances

from the socket connections to the actual

elements within the envelope no longer are

negligible. Second, we find that electron

www.americanradiohistory.com

HANDBOOK The Klystron 81

transit time no longer may be ignored; an

appreciable fraction of a cycle of input signal

may be required for an electron to leave the

cathode space charge, pass through the grid

wires, and travel through the space between

grid and plate.

Effects of The effect of lead induct -

Lead Inductance ance is two -fold. First, as

shown in figure 21, the

combination of grid -lead inductance, grid -

cathode capacitance, and cathode lead induct-

ance tends to reduce the effective grid- cathode

signal voltage for a constant voltage at the

tube terminals as the frequency is increased.

Second, cathode lead inductance tends to

introduce undesired coupling between the

various elements within the tube.

Tubes especially designed for v -h -f and

u -h -f use have had their lead inductances

minimized. The usual procedures for reducing

lead inductance are: (1) using heavy lead

conductors or several leads in parallel (ex-

amples are the 6SH7 and 6AK5), (2) scaling

down the tube in all dimensions to reduce

both lead inductances and interelectrode

capacitances (examples are the 6AK5, 6F4,

and other acorn and miniature tubes), and (3)

the use of very low inductance extensions of

the elements themselves as external connec-

tions (examples are lighthouse tubes such as

the 2C40, oilcan tubes such as the 2C29, and

many types of v -h -f transmitting tubes).

Effect of When an electron tube is op-

Transit Time erated at a frequency high

enough that electron transit

time between cathode and plate is an ap-

preciable fraction of a cycle at the input fre-

quency, several undesirable effects take place.

First, the grid takes power from the input

signal even though the grid is negative at all

times. This comes about since the grid will

have changed its potential during the time

required for an electron to pass from cathode

to plate. Due to interaction, and a resulting

phase difference between the field associated

with the grid and that associated with a mov-

ing electron, the grid presents a resistance to

an input signal in addition to its normal

"cold" capacitance. Further, as a result of

this action, plate current no longer is in phase

with grid voltage.

An amplifier stage operating at a frequency

high enough that transit time is appreciable:

(a) Is difficult to excite as a result of grid

loss from the equivalent input grid resistance,

(b) Is capable of less output since trans -

conductance is reduced and plate current is

not in phase with grid voltage.

The effects of transit time increase with the

square of the operating frequency, and they

increase rapidly as frequency is increased

above the value where they become just ap-

preciable. These effects may be reduced by

scaling down tube dimensions; a procedure

which also reduces lead inductance. Further,

transit -time effects may be reduced by the

obvious procedure of increasing electrode po-

tentials so that electron velocity will be in-

creased. However, due to the law of electron -

motion in an electric field, transit time is

increased only as the square root of the ratio

of operating potential increase; therefore this

expedient is of limited value due to other

limitations upon operating voltages of small

electron tubes.

4 -7 Special Microwave

Electron Tubes

Due primarily to the limitation imposed by

transit time, conventional negative -grid elec-

tron tubes are capable of affording worthwhile

amplification and power output only up to a

definite upper frequency. This upper frequency

limit varies from perhaps 100 Mc. for con-

ventional tube types to about 4000 Mc. for

specialized types such as the lighthouse tube.

Above the limiting frequency, the conventional

negative -grid tube no longer is practicable and

recourse must be taken to totally different

types of electron tubes in which electron

transit time is not a limitation to operation.

Three of the most important of such microwave

tube types are the klystron, the magnetron, and

the travelling wave tube.

The Power Klystron The klystron is a type

of electron tube in which

electron transit time is used to advantage,

Such tubes comprise, as shown in figure 22,

a cathode, a focussing electrode, a resonator

connected to a pair of grids which afford

velocity modulation of the electron beam

(called the "buncher "), a drift space, and

another resonator connected to a pair of grids

(called the "catcher "). A collector for the

expended electrons may be included at the

end of the tube, or the catcher may also per-

form the function of electron collection.

The tube operates in the following manner:

The cathode emits a stream of electrons which

is focussed into a beam by the focussing

electrode. The stream passes through the

buncher where it is acted upon by any field

existing between the two grids of the buncher

cavity. When the potential between the two

grids is zero, the stream passes through with-

out change in velocity. But when the potential

between the two grids of the buncher is in-

creasingly positive in the direction of electron

www.americanradiohistory.com

82 Vacuum Tube Principles THE RADIO

Figure 22

TWO- CAVITY KLYSTRON OSCILLATOR

A conventional two -cavity klystron is shown

with a feedback loop connected between the

two cavities so that the tube may be used as

an oscillator.

motion, the velocity of the electrons in the

beam is increased. Conversely, when the field

becomes increasingly negative in the direction

of the beam (corresponding to the other half

cycle of the exciting voltage from that which

produced electron acceleration) the velocity

of the electrons in the beam is decreased.

When the velocity- modulated electron beam

reaches the drift space, where there is no field,

those electrons which have been sped up on

one half -cycle overtake those immediately

ahead which were slowed down on the other

half- cycle. In this way, the beam electrons be-

come bunched together. As the bunched groups

pass through the two grids of the catcher

cavity, they impart pulses of energy to these

grids. The catcher grid -space is charged to

different voltage levels by the passing electron

bunches, and a corresponding oscillating field

is set up in the catcher cavity. The catcher is

designed to resonate at the frequency of the

velocity- modulated beam, or at a harmonic of

this frequency.

In the klystron amplifier, energy delivered

by the buncher to the catcher grids is greater

than that applied to the buncher cavity by the

input signal. In the klystron oscillator a feed-

back loop connects the two cavities. Coupling

to either buncher or catcher is provided by

small Loops which enter the cavities by way of

concentric lines.

The klystron is an electron -coupled device.

When used as an oscillator, its output voltage

is rich in harmonics. Klystron oscillators of

various types afford power outputs ranging

from less than I watt to many thousand watts.

Operating efficiency varies between 5 and 30

per cent. Frequency may be shifted to some

extent by varying the beam voltage. Tuning is

Figure 23

REFLEX KLYSTRON OSCILLATOR

A conventional reflex klystron oscillator of

the type commonly used as o local oscillator

in superheterodyne receivers operating above

about 2000 Mc. is shown above. Frequency

modulation of the output frequency of the oscil-

lator, or o-f -c operation in a receiver, may be

obtained by varying the negative voltage on the

repeller electrode.

carried on mechanically in some klystrons by

altering (by means of knob settings) the shape

of the resonant cavity.

The Reflex Klystron The two -cavity klystron

as described in the pre-

ceding paragraphs is primarily used as a trans-

mitting device since quite reasonable amounts

of power are made available in its output cir-

cuit. However, for applications where a much

smaller amount of power is required-power

levels in the milliwatt range - for low -power

transmitters, receiver local oscillators, etc.,

another type of klystron having only a single

cavity is more frequently used.

The theory of operation of the single- cavity

klystron is essentially the same as the multi -

cavity type with the exception that the veloc-

ity- modulated electron beam, after having left

the " buncher" cavity is reflected back into

the area of the buncher again by a repeller

electrode as illustrated in figure 23. The

potentials on the various electrodes are ad-

justed to the value such that proper bunching

of the electron beam will take place just as a

particular portion of the velocity -modulated

beam reenters the area of the resonant cavity.

Since this type of klystron has only one circuit

it can be used only as an oscillator and not as

an amplifier. Effective modulation of the fre-

quency of a single- cavity klystron for FM

work can be obtained by modulating the re-

peller electrode voltage.

www.americanradiohistory.com

HANDBOOK The Magnetron 83

GRID

TERMINAL

CATHODE

OR'D

ANODE

TERMINAL

II`

ANODE

GLASS

/ SEAL

ANODE

HEATER

EYELET

GLASS

SEAL

LEAD

TERMINAL EYELET TURULATiON

Figure 24

CUTAWAY VIEW OF

WESTERN ELECTRIC 416- B/6280

VHF PLANAR TRIODE TUBE

The 416 -B, designed by the Bell

Telephone Laboratories is intended

for amplifier or frequency multiplier

service in the 4000 me region. Em-

ploying grid wires having a diameter

equal to fifteen wavelengths of light,

the 416 -B has a transconductance of

50,000. Spacing between grid and

cathode is .0005', to reduce transit

time effects. Entire tube is gold plated.

The Magnetron The magnetron is an s -h -f

oscillator tube normally em-

ployed where very high values of peak power

or moderate amounts of average power are

required in the range from perhaps 700 Mc.

to 30,000 Mc. Special magnetrons were de-

veloped for wartime use in radar equipments

which had peak power capabilities of several

million watts (megawatts) output at frequen-

cies in the vicinity of 3000 Mc. The normal

duty cycle of operation of these radar equip-

ments was approximately 1 /10 of one per

cent (the tube operated about 1 /1000 of the

time and rested for the balance of the operat-

ing period) so that the average power output

of these magnetrons was in the vicinity of

1000 watts.

PLATE 1 MAGNET COIL

FIL

`PLATE 2

O

ANODE

IL FILAMENT

GLASS ENVELOPE

FILAMENT

VOLTAGE PLATE

VOLTAGE

Figure 25

SIMPLE MAGNETRON OSCILLATOR

An external tank circuit is used with this type

of magnetron oscillator for operation in the

lower u -h -f ronge.

In its simplest form the magnetron tube is a

filament -type diode with two half -cylindrical

plates or anodes situated coaxially with re-

spect to the filament. The construction is

illustrated in figure 25A. The anodes of the

magnetron are connected to a resonant circuit

as illustrated on figure 25B. The tube is sur-

rounded by an electromagnet coil which, in

turn, is connected to a low -voltage d -c ener-

gizing source through a rheostat R for control-

ling the strength of the magnetic field. The

field coil is oriented so that the lines of

magnetic force it sets up are parallel to the

axis of the electrodes.

Under the influence of the strong magnetic

field, electrons leaving the filament are de-

flected from their normal paths and move in

circular orbits within the anode cylinder. This

effect results in a negative resistance which

sustains oscillations. The oscillation fre-

quency is very nearly the value determined by

L and C. In other magnetron circuits, the fre-

quency may be governed by the electron rota-

tion, no external tuned circuits being em-

ployed. Wavelengths of less than 1 centi-

meter have been produced with such circuits.

More complex magnetron tubes employ no

external tuned circuit, but utilize instead one

or more resonant cavities which are integral

with the anode structure. Figure 26 shows a

magnetron of this type having a multi -cellular

www.americanradiohistory.com

84 Vacuum Tube Principles THE RADIO

CTNODE

ANODE ESSASS IIII

NODE BLOC

- CATNODE LEAOE

MAGNETRON

PE MANE NT

MAGNET

rT TING Outryt

Figure 26

MODERN MULTI- CAVITY MAGNETRON

Illustrated is an external -anode strapped mag-

netron of the type commonly used in radar equip-

ment for the 10 -cm. range. A permanent magnet

of the general type used with such a magnetron

Is shown in the right -hand portion of the drawing,

with the magnetron in place between the pole

pieces of the magnet.

anode of eight cavities. It will be noted, also,

that alternate cavities (which would operate at

the same polarity when the tube is oscillating)

are strapped together. Strapping was found to

improve the efficiency and stability of high -

power radar magnetrons. In most radar appli-

cations of magnetron oscillators a powerful

permanent magnet of controlled characteristics

is employed to supply the magnetic field

rather than the use of an electromagnet.

The Travelling The Travelling Wave Tube

Wave Tube (figure 27) consists of a helix

located within an evacuated

envelope. Input and output terminations are

affixed to each end of the helix. An electron

beam passes through the helix and interacts

with a wave travelling along the helix to pro-

duce broad band amplification at microwave

frequencies.

When the input signal is applied to the gun

end of the helix, it travels along the helix wire

at approximately the speed of light. However,

the signal velocity measured along the axis

of the helix is considerably lower. The elec-

trons emitted by the cathode gun pass axially

through the helix to the collector, located at

the output end of the helix. The average veloc-

ity of the electrons depends upon the potential

of the collector with respect to the cathode.

When the average velocity of the electrons is

greater than the velocity of the helix wave,

the electrons become crowded together in the

various regions of retarded field, where they

impart energy to the helix wave. A power gain

of 100 or more may be produced by this tube.

4 -8 The Cathode -Ray Tube

The Cathode -Ray Tube The cathode -ray tube

is a special type of

WAVE GUIDE

INPUT ELECTRON BEAM

WAVE GU IDE

OUTPUT

iii1!,;,'+ ANODE COLLECTOR

Figure 27

THE TRAVELLING WAVE TUBE

Operation of this tube is the result of inter.

action between the electron beam and wave

travelling along the helix.

electron tube which permits the visual observa-

tion of electrical signals. It may be incorpo-

rated into an oscilloscope for use as a test

instrument or it may be the display device for

radar equipment or a television receiver.

Operation of A cathode -ray tube always in-

the CRT cludes an electron gun for pro-

ducing a stream of electrons, a

grid for controlling the intensity of the elec-

tron beam, and a luminescent screen for con-

verting the impinging electron beam into visi-

ble light. Such a tube always operates in con-

junction with either a built -in or an external

means for focussing the electron stream into a

narrow beam, and a means for deflecting the

electron beam in accordance with an electrical

signal.

The main electrical difference between

types of cathode -ray tubes lies in the means

employed for focussing and deflecting the

electron beam. The beam may be focussed

and/or deflected either electrostatically or

magnetically, since a stream of electrons can

be acted upon either by an electrostatic or a

magnetic field. In an electrostatic field the

electron beam tends to be deflected toward the

positive termination of the field (figure 28).

In a magnetic field the stream tends to be

deflected at right angles to the field. Further,

an electron beam tends to be deflected so that

it is normal (perpendicular) to the equipotential

lines of an electrostatic field- and it tends to

be deflected so that it is parallel to the lines

of force in a magnetic field.

Large cathode -ray tubes used as kinescopes

in television receivers usually are both focused

and deflected magnetically. On the other hand,

the medium -size CR tubes used in oscillo-

scopes and small television receivers usually

are both focused and deflected electrostat-

ically. But CR tubes for special applications

may be focused magnetically and deflected

electrostatically or vice versa.

There are advantages and disadvantages to

www.americanradiohistory.com

HANDBOOK The Cathode Ray Tube 85

SASE ACCELERATING ANODE IN)

NEATE C Lr. LEG T OA DE IFI

r1

- NOR I LONTAL DEFLECTION

ICI

-ADUADAG SECONDARY

-1 COATING

CL

è.

CCTIIN NEEAu

ZATNODE (10

CONTROL ACCELERATI

GRID IG ANODE (A)

CATNODE (10 -VERTICAL DEFLECTION

PLATES IS)

RUORESCENT SCREEN

Figure 28

TYPICAL ELECTROSTATIC

CATHODE -RAY TUBE

both types of focussing and deflection. How-

ever, it may be stated that electrostatic deflec-

tion is much better than magnetic deflection

when high -frequency waves are to be displayed

on the screen; hence the almost universal use

of this type of deflection for oscillographic

work. But when a tube is operated at a high

value of accelerating potential so as to obtain

a bright display on the face of the tube as for

television or radar work, the use of magnetic

deflection becomes desirable since it is rela-

tively easier to deflect a high -velocity electron

beam magnetically than electrostatically.

However, an ion trap is required with mag-

netic deflection since the heavy negative ions

emitted by the cathode are not materially de-

flected by the magnetic field and hence would

burn an "ion spot" in the center of the lumi-

nescent screen. With electrostatic deflection

the heavy ions are deflected equally as well

as the electrons in the beam so that an ion

spot is not formed.

Construction of The construction of a typical

Electrostatic CRT electrostatic- focus, electro-

static- deflection cathode -ray

tube is illustrated in the pictorial diagram of

figure 28. The indirectly heated cathode K re-

leases free electrons when heated by the

enclosed filament. The cathode is surrounded

by a cylinder G, which has a small hole in its

front for the passage of the electron stream.

Although this element is not a wire mesh as

is the usual grid, it is known by the same

name because its action is similar: it controls

the electron stream when its negative potential

is varied.

Next in order, is found the first accelerating

anode, H, which resembles another disk or

cylinder with a small hole in its center. This

electrode is run at a high or moderately high

positive voltage, to accelerate the electrons

towards the far end of the tube.

The focussing electrode, F, is a sleeve

which usually contains two small disks, each

with a small hole.

After leaving the focussing electrode, the

electrons pass through another accelerating

anode, A, which is operated at a high positive

potential. In some tubes this electrode is oper-

atcd at a higher potential than the first accel-

erating electrode, H, while in other tubes both

accelerating electrodes are operated at the

same potential.

The electrodes which have been described

up to this point constitute the electron gun,

which produces the free electrons and focusses

them into a slender, concentrated, rapidly -

traveling stream for projecting onto the view-

ing screen.

Electrostatic To make the tube useful, means

Deflection must be provided for deflecting

the electron beam along two axes

at right angles to each other. The more corn -

mon tubes employ electrostatic deflection

plates, one pair to exert a force on the beam

in the vertical plane and one pair to exert a

force in the horizontal plane. These plates

are designated as B and C in figure 28.

Standard oscilloscope practice with small

cathode -ray tubes calls for connecting one of

the B plates and one of the C plates together

and to the high voltage accelerating anode.

With the newer three -inch tubes and with five -

inch tubes and larger, all four deflecting plates

are commonly used for deflection. The positive

high voltage is grounded, instead of the nega-

tive as is common practice in amplifiers, etc.,

in order to permit operation of the deflecting

plates at a d -c potential at or near ground.

An Aquadag coating is applied to the inside

of the envelope to attract any secondary elec-

trons emitted by the flourescent screen.

In the average electrostatic -deflection CR

tube the spot will be fairly well centered if all

four deflection plates are returned to the po-

tential of the second anode (ground). How -

ever, for accurate centering and to permit mov-

ing the entire trace either horizontally or

vertically to permit display of a particular

waveform, horizontal and vertical centering

controls usually are provided on the front of

the oscilloscope.

After the spot is once centered, it is neces-

sary only to apply a positive or negative volt-

age (with respect to ground) to one of the

ungrounded or "free" deflector plates in order

to move the spot. If the voltage is positive

with respect to ground, the beam will be

attracted toward that deflector plate, while if

negative the beam and spot will be repulsed.

The amount of deflection is directly propor-

tional to the voltage (with respect to ground)

that is applied to the free electrode.

With the larger- screen higher -voltage tubes

it becomes necessary to place deflecting volt-

age on both horizontal and both vertical plates.

This is done for two reasons: First, the amount

of deflection voltage required by the high-

www.americanradiohistory.com

86 Vacuum Tube Principles THE RADIO

FOCUS COIL T-TERMINAL

BASE FIRST \ DEFLECTION COILS

ANODE

/CONTROL ® 10

GRID ID)

/CATHODE (R)

SECOND

(ADYADA.,Ï

_ _ _ ECEETFOÑ R[AM

FLUORESCENT SCREEN

Figure 29

TYPICAL ELECTROMAGNETIC

CATHODE -RAY TUBE

voltage tubes is so great that a transmitting

tube operating from a high voltage supply

would be required to attain this voltage with-

out distortion. By using push -pull deflection

with two tubes feeding the deflection plates,

the necessary plate supply voltage for the de-

flection amplifier is halved. Second, a certain

amount of de- focussing of the electron stream

is always present on the extreme excursions in

deflection voltage when this voltage is applied

only to one deflecting plate. When the de-

flecting voltage is fed in push -pull to both

deflecting plates in each plane, there is no de-

focussing because the average voltage acting

on the electron stream is zero, even though the

net voltage (which causes the deflection)

acting on the stream is twice that on either

plate. The fact that the beam is deflected by a

magnetic field is important even in an oscillo-

scope which employs a tube using electro-

static deflection, because it means that pre-

cautions must be taken to protect the tube from

the transformer fields and sometimes even the

earth's magnetic field. This normally is done

by incorporating a magnetic shield around the

tube and by placing any transformers as far

from the tube as possible, oriented to the posi-

tion which produces minimum effect upon the

electron stream.

Construction of Electro- The electromagnetic

magnetic CRT cathode -ray tube allows

greater definition than

does the electrostatic tube. Also, electro-

magnetic definition has a number of advan-

tages when a rotating radial sweep is required

to give polar indications.

The production of the electron beam in an

electromagnetic tube is essentially the same

as in the electrostatic tube. The grid structure

is similar, and controls the electron beam in

an identical manner. The elements of a typical

electromagnetic tube are shown in figure 29.

The focus coil is wound on an iron core which

may be moved along the neck of the tube to

focus the electron beam. For final adjustment,

-v

. 11111

- _ iirr- A

1tI1.-o

R R

.1m

Jew

.1n1R

Figure 30

Two pairs of coils arranged for electro-

magnetic deflection in two directions.

the current flowing in the coil may be varied.

A second pair of coils, the deflection coils

are mounted at right angles to each other

around the neck of the tube. In some cases,

these coils can rotate around the axis of the

tube. Two anodes are used for accelerating the

electrons from the cathode to the screen. The

second anode is a graphite coating (Aquadag)

on the inside of the glass envelope. The func-

tion of this coating is to attract any secondary

electrons emitted by the flourescent screen,

and also to shield the electron beam.

In some types of electromagnetic tubes, a

first, or accelerating anode is also used in

addition to the Aquadag.

Electromagnetic A magnetic field will deflect

Deflection an electron beam in a direc-

tion which is at right angles

to both the direction of the field and the direc-

tion of motion of the beam.

In the general case, two pairs of deflection

coils are used (figure 30). One pair is for

horizontal deflection, and the other pair is for

vertical deflection. The two coils in a pair

are connected in series and are wound in such

directions that the magnetic field flows from

one coil, through the electron beam to the

other coil. The force exerted on the beam by

the field moves it to any point on the screen

by application of the proper currents to these

coils.

The Trace The human eye retains an image

for about one - sixteenth second

after viewing. In a CRT, the spot can be

moved so quickly that a series of adjacent

spots can be made to appear as a line, if the

beam is swept over the path fast enough. As

www.americanradiohistory.com

HANDBOOK Gas Tubes 87

long as the electron beam strikes in a given

place at least sixteen times a second, the

spot will appear to the human eye as a source

of continuous light with very little flicker.

Screen Materials - At least five types of lumi-

"Phosphors" nescent screen materials

are commonly available on

the various types of CR tubes commercially

available. These screen materials are called

phosphors; each of the five phosphors is best

suited to a particular type of application. The

P -1 phosphor, which has a green flourescence

with medium persistence, is almost invariably

used for oscilloscope tubes for visual observa-

tion. The P -4 phosphor, with white fluores-

cence and medium persistence, is used on

television viewing tubes ( "Kinescopes "). The

P -5 and P -11 phosphors, with blue fluores-

cence and very short persistence, are used

primarily in oscilloscopes where photographic

recording of the trace is to be obtained. The

P -7 phosphor, which has a blue flash and a

long -persistence greenish -yellow persistence,

is used primarily for radar displays where

retention of the image for several seconds

after the initial signal display is required.

4 -9 Gas Tubes

The space charge of electrons in the vicinity

of the cathode in a diode causes the plate -to-

cathode voltage drop to be a function of the

current being carried between the cathode and

the plate. This voltage drop can be rather high

when large currents are being passed, causing

a considerable amount of energy loss which

shows up as plate dissipation.

Action of The negative space charge can

Positive Ions be neutralized by the presence

of the proper density of positive

ions in the space between the cathode and

anode. The positive ions may be obtained by

the introduction of the proper amount of gas or

a small amount of mercury into the envelope of

the tube. Then the voltage drop across the

tube reaches the ionization potential of the

gas or mercury vapor, the gas molecules will

become ionized to form positive ions. The

positive ions then tend to neutralize the space

charge in the vicinity of the cathode. The volt-

age drop across the tube then remains constant

at the ionization potential of the gas up to a

current drain equal to the maximum emission

capability of the cathode. The voltage drop

varies between 10 and 20 volts, depending

upon the particular gas employed, up to the

maximum current rating of the tube.

Mercury Vapor

Tubes Mercury -vapor tubes, although

very widely used, have the

disadvantage that they must be

operated within a specific temperature range

(25° to 70°C.) in order that the mercury vapor

pressure within the tube shall be within the

proper range. If the temperature is too low,

the drop across the tube becomes too high

causing immediate overheating and possible

damage to the elements. If the temperature is

too high, the vapor pressure is too high, and

the voltage at which the tube will "flash back"

is lowered to the point where destruction of

the tube may take place. Since the ambient

temperature range specified above is within

the normal room temperature range, no trouble

will be encountered under normal operating

conditions. However, by the substitution of

xenon gas for mercury it is possible to pro-

duce a rectifier with characteristics comparable

to those of the mercury -vapor tube except that

the tube is capable of operating over the range

from approximately -70° to 90° C. The 3B25

rectifier is an example of this type of tube.

Thyratron If a grid is inserted between the ca-

Tubes thode and plate of a mercury -vapor

gaseous- conduction rectifier, a neg-

ative potential placed upon the added element

will increase the plate -to- cathode voltage drop

required before the tube will ionize or "fire."

The potential upon the control grid will have

no effect on the plate -to- cathode drop after the

tube has ionized. However, the grid voltage

may be adjusted to such a value that conduc-

tion will take place only over the desired

portion of the cycle of the a -c voltage being

impressed upon the plate of the rectifier.

Voltage Regulator In a glow -discharge gas tube

Tubes the voltage drop across the

electrodes remains constant

over a wide range of current passing through

the tube. This property exists because the

degree of ionization of the gas in the tube

varies with the amount of current passing

through the tube. When a large current is

passed, the gas is highly ionized and the

internal impedance of the tube is low. When a

small current is passed, the gas is lightly

ionized and the internal impedance of the tube

is high. Over the operating range of the tube,

the product (IR) of the current through the tube

and the internal impedance of the tube is very

nearly constant. Examples of this type of tube

are VR -150, VR -105 and the old 874.

Vacuum Tube Vacuum tubes are grouped into

Classification three major classifications:

commercial, ruggedized, and

premium (or reliable). Any one of these three

groups may also be further classified for

www.americanradiohistory.com

88 Vacuum Tube Principles THE RADIO

military duty (JAN classification). To qualify

for JAN classification, sample lots of the

particular tube must have passed special

qualification tests at the factory. It should not

be construed that a JAN-type tube is better

than a commercial tube, since some commercial

tests and specifications are more rigid than

the corresponding JAN specifications. The

JAN -stamped tube has merely been accepted

under a certain set of conditions for military

service.

Ruggedized or Radio tubes are being used in

Premium Tubes increasing numbers for indus-

trial applications, such as

computing and control machinery, and in avia-

tion and marine equipment. When a tube fails

in a home radio receiver, it is merely incon-

venient, but a tube failure in industrial appli-

cations may bring about stoppage of some vital

process, resulting in financial loss, or even

danger to life.

To meet the demands of these industrial

applications, a series of tubes was evolved

incorporating many special features designed

to ensure a long and pre- determined operating

life, and uniform characteristics among similar

tubes. Such tubes are known as ruggedized or

premium tubes. Early attempts to select re-

TRIODE PLATE , `FLUORESCENT ANODE

TRIODE GRID

CATHODES

RAY CONTROL

ELECTRODE

Figure 31

SCHEMATIC REPRESENTATION

OF "MAGIC EYE" TUBE

liable specimens of tubes from ordinary stock

tubes proved that in the long run the selected

tubes were no better than tubes picked at

random. Long life and ruggedness had to be

built into the tubes by means of proper choice

and 100% inspection of all materials used in

the tube, by critical processing inspection and

assembling, and by conservative ratings of the

tube. Pure tungsten wire is used for heaters in

preference to alloys of lower tensile strength.

Nickel tubing is employed around the heater

wires at the junction to the stem wires to

reduce breakage at this point. Element struc-

tures are given extra supports and bracing.

Finally, all tubes are given a 50 hour test run

under full operating conditions to eliminate

early failures. When operated within their

ratings, ruggedized or premium tubes should

provide a life well in excess of 10,000 hours.

Ruggedized tubes will withstand severe

impact shocks for short periods, and will

100-

eo

60

L 0

20

0 0

IP=2.5IAA.

10 20 30 40

EP VOLTS)

50 60

Figure 32

AMPLIFICATION FACTOR OF TYPICAL MODE

TUBE DROPS RAPIDLY AS PLATE VOLTAGE

IS DECREASED BELOW 20 VOLTS

operate under conditions of vibration for many

hours. The tubes may be identified in many

cases by the fact that their nomenclature in-

cludes a "W" in the type number, as in 807W,

5U4W, etc. Some ruggedized tubes are included

in the "5000" series nomenclature. The 5654

is a ruggedized version of the 6AK5, the 5692

is a ruggedized version of the 6SN7, etc.

4 -10 Miscellaneous Tube Types

E lectron The electron -ray tube or magic eye

Ray Tubes contains two sets of elements, one

of which is a triode amplifier and

the other a cathode -ray indicator. The plate of

the triode section is internally connected to

the ray- control electrode (figure 31), so that

as the plate voltage varies in accordance with

the applied signal the voltage on the ray -control

electrode also varies. The ray -control electrode

is a metal cylinder so placed relative to the

cathode that it deflects some of the electrons

emitted from the cathode. The electrons which

strike the anode cause it to fluoresce, or give

off light, so that the deflection caused by the

ray -control electrode, which prevents electrons

from striking part of the anode, produces a

wedge- shaped electrical shadow on the fluores-

cent anode. The size of this shadow is deter-

mined by the voltage on the ray -electrode. When

this electrode is at the same potential as the

fluorescent anode, the shadow disappears; if

the ray -electrode is less positive than the

anode, a shadow appears the width of which

is proportional to the voltage on the ray -elec-

trode. Magic eye tubes may be used as tuning

indicators, and as balance indicators in electri-

cal bridge circuits. If the angle of shadow is

calibrated, the eye tube may be used as a volt-

meter where rough measurements suffice.

www.americanradiohistory.com

HANDBOOK Miscellaneous 89

Ec1=,z.ev

Figure 33

CHARACTERISTIC CURVES OF 12AK5

SFACE- CHARGE TRIODE

Ecz (VOLTS)

Controlled Series heater strings are employed

Warm -up in ac -dc radio receivers and tele-

Tubes vision sets to reduce the cost,

size, and weight of the equipment.

Voltage surges of great magnitude occur in

series operated filaments because of variations

in the rate of warm -up of the various tubes.

As the tubes warm up, the heater resistance

changes. This change is not the same between

tubes of various types, or even between tubes

of the same type made by different manu-

facturers. Some 6 -volt tubes show an initial

surge as high as 9 -volts during warm -up, while

slow -heating tubes such as the 25BQ6 are

underheated during the voltage surge on the

6 -volt tubes.

Standardization of heater characteristics in

a new group of cubes designed for series heater

strings has eliminated this trouble. The new

tubes have either 600 ma. or 400 ma. heaters,

with a controlled warm -up time of approximately

11 seconds. The 5U8, 6CG7, and 12BH7 -A are

examples of controlled warm -up tubes.

Low Introduction of the 12 -volt ignition

Plate system in American automobiles

Potential has brought about the design of a

Tubes series of tubes capable of operation

with a plate potential of 12 -14

volts. Standard tubes perform poorly at low

plate potentials, as the amplification factor

of the tube drops rapidly as the plate voltage

is decreased (figure 32). Contact potential

effects, and change of characteristics with

variations of filament voltage combine to make

operation at low plate potentials even more

erratic.

By employing special processing techniques

and by altering the electrode geometry a series

of low voltage tubes has been developed by

Tung -Sol that effectively perform with all elec-

trodes energized by a 12 -volt system. With a

suitable power output transistor, this makes

possible an automobile radio without a vibrator

power supply. A special space- charge tube

(12K5) has been developed that delivers 40

milliwatts of audio power with a 12 volt plate

supply (figure 33).

Foreign

Tubes The increased number of imported

radios and high- fidelity equipment

have brought many foreign vacuum

tubes into the United States. Many of these

tubes are comparable to, or interchangeable

with standard American tubes. A complete

listing of the electrical characteristics and

hase connection diagrams of all general -

purpose tubes made in all tube -producing

countries outside the "Iron Curtain" is con-

tained in the Radio Tube Vade Mecum (World's

Radio Tubes) available at most larger radio

parts jobbers or by mail from the publishers

of this Handbook. The Equivalent Tubes Vade

Mecum (World's Equivalent Tubes) gives all

replacement tubes for a given type, both exact

and near -equivalents (with points of differ-

ence detailed). (Data on TV and special -

purpose tubes if needed is contained in a com-

panion volume Television Tubes Vade Mecum).

www.americanradiohistory.com

CHAPTER FIVE

Transistors and

Semi -Conductors

One of the earliest detection devices used

in radio was the galena crystal, a crude ex-

ample of a semiconductor. More modern ex-

amples of semiconductors are the copper -

oxide rectifier, the selenium rectifier and the

germanium diode. All of these devices offer

the interesting property of greater resistance

to the flow of electrical current in one direc-

tion than in the opposite direction. Typical

conduction curves for these semiconductors

are shown in Figure 1. The copper oxide recti-

fier action results from the function of a thin

film of cuprous oxide formed upon a pure cop-

per disc. This film offers low resistance for

positive voltages, and high resistance for

negative voltages. The same action is ob-

served in selenium rectifiers, where a film of

selenium is deposited on an iron surface.

s 1

O CD0IIT*L

1N3 DIODI

TYPICAL

1

STATIC CHARACTERISTICS

00

w

o

I. I

1.1 I.1 -

-00 -.0 -SO -20 - 0 0

VOLTS

2

Figure lA

TYPICAL CHARACTERISTIC CURVE

OF SEMI -CONDUCTOR DIODE

a

90

5 -1 Atomic Structure of

Germanium and Silicon

It has been previously stated that the elec-

trons in an element having a large atomic

number are grouped into rings, each ring hav-

ing a definite number of electrons. Atoms in

which these rings are completely filled are

called inert gases, of which helium and argon

are examples. All other elements have one or

more incomplete rings of electrons. If the in-

complete ring is loosely bound, the electrons

may be easily removed, the element is called

metallic, and is a conductor of electric current.

If the incomplete ring is tightly bound, with

only a few missing electrons, the element is

called non - metallic and is an insulator of elec-

tric current. Germanium and silicon fall be-

tween these two sharply defined groups, and

exhibit both metallic and non -metallic char-

acteristics. Pure germanium or silicon may be

considered to be a good insulator. The addition

of certain impurities in carefully controlled

amounts to the pure germanium will alter the

conductivity of the material. In addition, the

choice of the impurity can change the direction

of conductivity through the crystal, some im-

purities increasing conductivity to positive volt-

ages, and others increasing conductivity to neg-

ative voltages.

5 -2 Mechanism of

Conduction

As indicated by their name, semiconductors

are substances which have a conductivity

intermediate between the high values observed

for metals and the low values observed for in-

sulating materials. The mechanism of conduc-

tion in semiconductors is different from that

www.americanradiohistory.com

Transistors 91

ANODES

SCHEMATIC REPRESENTATION

-1=011 _

Color BEOI

M11

¡ Calor l. r Bao ft

-- Wrba

TUBE. GERMANIUM. SILICON

AND SELENIUM DIODES

CATB000

Figure 1 -B

COMMON DIODE COLOR CODES

AND MARKINGS ARE SHOWN

IN ABOVE CHART

observed in metallic conductors. There exist

in semiconductors both negatively charged

electrons and positively charged particles,

called holes, which behave as though they

had a positive electrical charge equal in mag-

nitude to the negative electrical charge on

the electron. These holes and electrons drift

in an electrical field with a velocity which is

proportional to the field itself:

VAN = µnE

where VAN = drift velocity of hole

E = magnitude of electric field

= mobility of hole

In an electric field the holes will drift in a

direction opposite to that of the electron and

with about one -half the velocity, since the

hole mobility is about one -half the electron

mobility. A sample of a semiconductor, such as

germanium or silicon, which is both chemically

pure and mechanically perfect will contain in it

approximately equal numbers of holes and elec-

trons and is called an intrinsic semiconductor.

The intrinsic resistivity of the semiconductor

depends strongly upon the temperature, being

about 50 ohm /cm. for germanium at room

temperature. The intrinsic resistivity of silicon

is about 65,000 ohm /cm. at the same temper-

ature.

If, in the growing of the semiconductor crys-

tal, a small amount of an impurity, such as

phosphorous, arsenic or antimony is included

in the crystal, each atom of the impurity con-

tributes one free electron. This electron is

available for conduction. The crystal is said

to be doped and has become electron- conduct-

.320

PLASTIC CASE

P - TYPE GERMANIUM

N- TYPE CRYSTAL LAYER

GERMANIUM COLLECTOR

CRYSTAL LAYER

BASE CONNECTION

EMITTER

SMALL 3 -PIN

BASE

ASE CONNECTION

COLLECTOR

EMITTER LI

Figure 2A

CUT -AWAY VIEW OF JUNCTION

TRANSISTOR, SHOWING PHYSICAL

ARRANGEMENT

.MS'

Pe- Nb JUNCTION

L Jw Z

P y1!

Nb

o o

b-PC JUNCTION

Pt 4-

Figure 2B

PICTORIAL EQUIVALENT OF

P -N -P JUNCTION TRANSISTOR

SIGN Z

www.americanradiohistory.com

92 Transistors and Semi- Conductors THE RADIO

Figure 3

CONSTRUCTION DETAIL OF A

POINT CONTACT TRANSISTOR

ing in nature and is called N (negative) type

germanium. The impurities which contribute

electrons are called donors. N -type germanium

has better conductivity than pure germanium in

one direction, and a continuous stream of elec-

trons will flow through the crystal in this direc-

tion as long as an external potential of the

correct polarity is applied across the crystal.

Other impurities, such as aluminum, gal-

lium or indium add one hole to the semicon-

ducting crystal by accepting one electron for

each atom of impurity, thus creating additional

holes in the semiconducting crystal. The ma-

terial is now said to be hole- conducting, or P

(positive) type germanium. The impurities

which create holes are called acceptors. P -type

germanium has better conductivity than pure

germanium in one direction. This direction is

opposite to that of the N -type material. Either

the N -type or the P -type germanium is called

extrinsic conducting type. The doped materials

have lower resistivities than the pure materials,

and doped semiconductors in the resistivity

range of .01 to 10 ohm /cm. are normally used

in the production of transistors.

5 -3 The Transistor

In the past few years an entire new tech-

nology has been developed for the application

of certain semiconducting materials in produc-

tion of devices having gain properties. These

gain properties were previously found only in

vacuum tubes. The elements germanium and

silicon are the principal materials which ex-

hibit the proper semiconducting properties per-

mitting their application in the new ampli-

fying devices called transistors. However,

other semiconducting materials, including the

compounds indium antimonide and lead sulfide

have been used experimentally in the produc-

tion of transistors.

EMITTER COLLECTOR EMITTER

BISE

P -N -P TRANSISTOR OR

POINT CONTACT TRANSISTOR

COLLECTOR

BASE-I

N-P-N TRANSISTOR

Figure 4

ELECTRICAL SYMBOLS

FOR TRANSISTORS

Types of Transistors There are two basic

types of transistors, the

point -contact type and the junction type (fig-

ure 2) . Typical construction detail of a point -

contact transistor is shown in Figure 3, and

the electrical symbol is shown in Figure 4. The

emitter and collector electrodes make contact

with a small block of germanium, called the

base. The base may be either N -type or P -type

germanium, and is approximately .05" long

and .03" thick. The emitter and collector elec-

trodes are fine wires, and are spaced about

.005" apart on the germanium base. The com-

plete assembly is usually encapsulated in a

small, plastic case to provide ruggedness and

to avoid contaminating effects of the atmos-

phere. The polarity of emitter and collector

voltages depends upon the type of germanium

employed in the base, as illustrated in figure 4.

The junction transistor consists of a piece

of either N -type or P -type germanium between

two wafers of germanium of the opposite type.

Either N -P -N or P -N -P transistors may be

made. In one construction called the grown

crystal process, the original crystal, grown

from molten germanium or silicon, is created

in such a way as to have the two closely spaced

junctions imbedded in it. In the other con-

struction called the fusion process, the crystals

are grown so as to make them a single con-

ductivity type. The junctions are then pro-

duced by fusing small pellets of special metal

alloys into minute plates cut from the original

crystal. Typical construction detail of a junction

transistor is shown in figure 2A.

The electrical schematic for the P -N -P junc-

tion transistor is the same as for the point -

contact type, as is shown in figure 4.

Transistor Action Presently available types of

transistors have three es-

sential actions which collectively are called

transistor action. These are: minority carrier

injection, transport, and collection. Figure 2B

shows a simplified drawing of a P -N -P junc-

tion -type transistor, which can illustrate this

www.americanradiohistory.com

HANDBOOK Transistors 93

collective action. The P -N -P transistor con-

sists of a piece of N -type germanium on op-

posite sides of which a layer of P -type mate-

rial has been grown by the fusion process.

Terminals are connected to the two P- sections

and to the N -type base. The transistor may be

considered as two P -N junction rectifiers

p!aced in close juxaposition with a semi -

conduction crystal coupling the two rectifiers

together. The left -hand terminal is biased in

the forward (or conducting) direction and is

called the emitter. The right -hand terminal is

biased in the back (or reverse) direction and

is called the collector The operating potentials

are chosen with respect to the base terminal,

which may or may not be grounded. If an

N -P -N transistor is used in place of the P -N -P,

the operating potentials are reversed.

The P. - Nb junction on the left is biased

in the forward direction and holes from the

P region are injected into the Nb region, pro-

ducing therein a concentration of holes sub-

stantially greater than normally present in the

material. These holes travel across the base

region towards the collector, attracting neigh-

boring electrons, finally increasing the avail-

able supply of conducting electrons in the

collector loop. As a result, the collector loop

possesses lower resistance whenever the emit-

ter circuit is in operation. In junction tran-

sistors this charge transport is by means of

diffusion wherein the charges move from a

region of high concentration to a region of

lower concentration at the collector. The col-

lector, biased in the opposite direction, acts

as a sink for these holes, and is said to col-

lect them.

It is known that any rectifier biased in the

forward direction has a very low internal im-

pedance, whereas one biased in the back direc-

tion has a very high internal impedance. Thus,

current flows into the transistor in a low im-

pedance circuit, and appears at the output as

current flowing in a high impedance circuit.

The ratio of a change in collector current to

a change in emitter current is called the current

amplification, or alpha:

a = ie

where a = current amplification

ie = change in collector current

i. = change in emitter current

Values of alpha up to 3 or so may be ob-

tained in commercially available point- contact

transistors, and values of alpha up to about

0.95 are obtainable in junction transistors.

Alpha Cutoff The alpha cutoff frequency of

Frequency a transistor is that frequency

at which the grounded base

current gain has decreased to 0.7 of the gain

obtained at 1 kc. For audio transistors, the

alpha cutoff frequency is in the region of 0.7

Mc. to 1.5 Mc. For r -f and switching transis-

tors, the alpha cutoff frequency may be 5 Mc.

or higher. The upper frequency limit of oper-

ation of the transistor is determined by the

small but finite time it takes the majority car-

rier to move from one electrode to another.

Drift Transistors As previously noted, the

signal current in a con-

ventional transistor is transmitted across the

base region by a diffusion process. The transit

time of the carriers across this region is, there-

fore relatively long. RCA has developed a

technique for the manufacture of transistors

which does not depend upon diffusion for

transmission of the signal across the base re-

gion. Transistors featuring this new process are

known as drift transistors. Diffusion of charge

carriers across the base region is eliminated and

the carriers are propelled across the region by

a "built in electric field. The resulting reduc-

tion of transit time of the carrier permits drift

transistors to be used at much higher fre-

quencies than transistors of conventional de-

sign. The "built in" electric field is in the base

region of the drift transistor. This field is

achieved by utilizing an impurity density

which varies from one side of the base to the

other. The impurity density is high next to

the emitter and low next to the collector. Thus,

there are more mobile electrons in the region

near the emitter than in the region near the

collector, and they will try to diffuse evenly

throughout the base. However, any displace-

ment of the negative charge leaves a positive

charge in the region from which the electrons

came, because every atom of the base material

was originally electrically neutral. The dis-

placement of the charge creates an electric

field that tends to prevent further electron dif-

fusion so that a condition of equilibrium is

reached. The direction of this field is such as

to prevent electron diffusion from the high

density area near the emitter to the low density

area near the collector. Therefore, holes enter-

ing the base will be accelerated from the emit-

ter to the collector by the electric field. Thus

the diffusion of charge carriers across the base

region is augmented by the built -in electric

field. A potential energy diagram for a drift

transistor is shown in figure 5.

www.americanradiohistory.com

94 Transistors and Semi -Conductors THE RADIO

DECREASING

POTENTIAL

OF ENERGY

MAJORITY CARRIER

DRIFT

REGION

S

h 7

CC

_Aiii/1111

:i1, I \ r_=/

Ifi/Ciioill111E.1

WW

6

S

34 % u IMTINEE:011101

I

CASE I

REGION REGION

I

ION

I

I COLLECTOR

I

o1 u l'/. ri_

EMITTER '/_!= \ 10 20 50 40 s0

COLLECTOR VOLTS

0

DISTANCE

Figure 5

POTENTIAL ENERGY DIAGRAM

FOR DRIFT TRANSISTOR (2N247)

5 -4 Transistor

Characteristics

The transistor produces results that may be

comparable to a vacuum tube, but there is a

basic difference between the two devices. The

vacuum tube is a voltage controlled device

whereas the transistor is a current controlled

device. A vacuum tube normally operates with

its grid biased in the negative or high resist-

ance direction, and its plate biased in the

positive or low resistance direction. The tube

conducts only by means of electrons, and has

its conducting counterpart in the form of the

N -P -N transistor, whose majority carriers are

also electrons. There is no vacuum tube equiv-

alent of the P -N -P transistor, whose majority

carriers are holes.

The biasing conditions stated above provide

the high input impedance and low output im-

pedance of the vacuum tube. The transistor is

biased in the positive or low resistance direc-

tion in the emitter circuit, and in the negative,

or high resistance direction in the collector

circuit resulting in a low input impedance

and a high output impedance, contrary to and

opposite from the vacuum tube. A comparison

of point- contact transistor characteristics and

vacuum tube characteristics is made in figure 6.

The resistance gain of a transistor is ex-

pressed as the ratio of output resistance to

input resistance. The input resistance of a

typical transistor is low, in the neighborhood

of 300 ohms, while the output resistance is

relatively high, usually over 20,000 ohms. For

a point- contact transistor, the resistance gain

is usually over 60.

The voltage gain of a transistor is the

product of alpha times the resistance gain,

and for a point- contact transistor is of the

M

IMI .

,1\I/I

FAlBM!lPA

IIIPA lf

I/ICJ 1

25 so 75 100 125

PLATE VOLTS

ISO 175

Figure 6

COMPARISON OF POINT -CONTACT

TRANSISTOR AND VACUUM TUBE

CHARACTERISTICS

200

order of 3 X 60 = 180. A junction transistor

which has a value of alpha less than unity

nevertheless has a resistance gain of the order

of 2000 because of its extremely high output

resistance, and the resulting voltage gain is

about 1800 or so. For both types of transistors

the power gain is the product of alpha squared

times the resistance gain and is of the order

of 400 to 500.

The output characteristics of the junction

transistor are of great interest. A typical ex-

ample is shown in figure 7. It is seen that the

junction transistor has the characteristics of

an ideal pentode vacuum tube. The collector

current is practically independent of the col-

lector voltage. The range of linear operation

extends from a minimum voltage of about 0.2

volts up to the maximum rated collector volt-

age. A typical load line is shown, which il-

lustrates the very high load impedance that

would be required for maximum power trans-

fer. A grounded emitter circuit is usually used,

since the output impedance is not as high as

when a grounded base circuit is used.

www.americanradiohistory.com

HANDBOOK Transistor Characteristics 95

-1 0 -0.S 0 +5 +10 + 5

COLLECTOR VOLTS

+20 +25

Figure 7

OUTPUT CHARACTERISTICS OF

TYPICAL JUNCTION TRANSISTOR

The output characteristics of a typical point -

contact transistor are shown in figure 6. The

pentode characteristics are less evident, rind the

output impedance is much lower, with the

range of linear operation extending down to

a collector voltage of 2 or 3. Of greater prac-

tical interest, however, is the input character-

istic curve with short -circuited, or nearly short-

circuited input, as shown in figure 8. It is

this point -contact transistor characteristic of

having a region of negative impedance that

lends the unit to use in switching circuits. The

transistor circuit may be made to have two,

one or zero stable operating points, depending

upon the bias voltages and the load impedance

used.

Equivalent Circuit As is known from net -

of a Transistor work theory, the small

signal performance of

any device in any network can be represented

by means of an equivalent circuit. The most

EMITTER MILLIAMPERES (te)

Figure 8

EMITTER CHARACTERISTIC CURVE

FOR TYPICAL POINT CONTACT

TRANSISTOR

EMITTER

d le

CASE

VALUES OF THE EQUIVALENT CIRCUIT

COLLECTOR

POINT- CONTACT

ISTOR

Vs..iMA VC15V.)

JUNCTION

ISTOR

(LE IMA VCR SV.)

re -EMITTER

RESISTANCE 1O0ß SOA

Cb -SASE

RESISTANCE 300A SOOA

RESCS ÁL10EOR 20000A 1 MEGONM

c4- CURRENT

AMPLIFICATION 2.0 0.57

Figure 9

LOW FREQUENCY EQUIVALENT

(Common Bose) CIRCUIT FOR POINT

CONTACT AND JUNCTION

TRANSISTOR

convenient equivalent circuit for the low fre-

quency small signal performance of both point-

contact and junction transistors is shown in

figure 9. r., rN, and rT, are dynamic resistances

which can be associated with the emitter, base

and collector regions of the transistor. The

current generator aI., represents the transport

of charge from emitter to collector. Typical

values of the equivalent circuit are shown in

figure 9.

Transistor

Configurations There are three basic transis-

tor configurations: grounded

base connection, grounded

emitter connection, and grounded collector

connection. These correspond roughly to

grounded grid, grounded cathode, and ground-

ed plate circuits in vacuum tube terminology

(figure 10) .

The grounded base circuit has a low input

impedance and high output impedance, and no

phase reversal of signal from input to output

circuit. The grounded emitter circuit has a

higher input impedance and a lower output

impedance than the grounded base circuit, and

a reversal of phase between the input and out-

put signal occurs. This circuit usually provides

maximum voltage gain from a transistor. The

grounded collector circuit has relatively high

input impedance, low output impedance, and

no phase reversal of signal from input to out-

put circuit. Power and voltage gain are both

low. Figure 11 illustrates some practical vacuum

tube circuits, as applied to transistors.

www.americanradiohistory.com

96 Transistors and Semi -Conductors THE RADIO

GROUNDED BASE

CONNECTION GROUNDED EMITTER

CONNECTION GROUNDED COLLECTOR

CONNECTION

Figure 10

COMPARISON OF BASIC VACUUM TUBE AND TRANSISTOR CONFIGURATIONS

5 -5 Transistor Circuitry

To establish the correct operating parameters

of the transistor, a bias voltage must be estab-

lished between the emitter and the base. Since

transistors are temperature sensitive devices,

and since some variation in characteristics usu-

ally exists between transistors of a given type,

attention must be given to the bias system to

overcome these difficulties. The simple self -bias

system is shown in figure 12A. The base is

simply connected to the power supply through

a large resistance which supplies a fixed value

of base current to the transistor. This bias

system is extremely sensitive to the current

transfer ratio of the transistor, and must be ad-

justed for optimum results with each transistor.

When the supply voltage is fairly high and

FLIP -FLOP COUNTER

CRYSTAL OSCILLATOR

RFC

R. F. OSCILLATOR

BLOCKING OSCILLATOR

DIRECT -COUPLED AMPLIFIER

Figure 11

TYPICAL TRANSISTOR CIRCUITS

ONE -STAGE RECEIVER

AUDIO AMPLIF ER

www.americanradiohistory.com

HANDBOOK Transistor Circuitry 97

BIAS

RESISTOR

E

LOAD

RESISTOR

-E

BIAS

RESISTOR LOAD

RESISTOR

2

R2= lo Re

Re = soo- +000 n

SO Li

+ (REVERSE POLARITY

FOR NPN TRANSISTOR)

O e O

Figure 12

BIAS CONFIGURATIONS FOR TRANSISTORS.

The voltage divider system of C is recommended for general transistor use. Ratio of R1 /R:

establishes base bias, and emitter bias is provided by voltage drop across Re.

Battery Polarity is reversed for N -P -N transistors.

wide variations in ambient temperature do not

occur, the bias system of figure 12B may be

used, with the bias resistor connected from

base to collector. When the collector voltage

is high, the base current is increased, moving

the operating point of the transistor down the

load line. If the collector voltage is low. the

operating point moves upwards along the load

line, thus providing automatic control of the

base bias voltage. This circuit is sensitive to

changes in ambient temperature, and may per-

mit transistor failure when the transistor is

operated near maximum dissipation ratings.

A better bias system is shown in figure 12C,

where the base bias is obtained from a voltage

divider, (R1, R2 ), and the emitter is forward

biased. To prevent signal degeneration, the

emitter bias resistor is bypassed with a large

capacitance. A high degree of circuit stability

is provided by this form of bias, providing the

emitter capacitance is of the order of 50 t+fd.

for audio frequency applications.

Audio Circuitry A simple voltage amplifier

is shown in figure 13. Di-

rect current stabilization is employed in the

enitter circuit. Operating parameters for the

amplifier are given in the drawing. In this case,

the input impedance of the amplifier is quite

low. When used with a high impedance driv-

ing source such as a crystal microphone a step

down input transformer should be employed

as shown in figure 13B. The grounded collec-

tor circuit of figure 13C provides a high input

impedance and a low output impedance, much

as in the manner of a vacuum tube cathode

follower.

The circuit of a two stage resistance coupled

amplifier is shown in figure 14A. The input

impedance is approximately 1100 ohms. Feed-

back may be placed around this amplifier from

the emitter of the second stage to the base of

the first stage, as shown in figure 14B. A

direct coupled version of the r -c amplifier is

shown in figure 14C. The input impedance is

of the order of 15,000 ohms, and an overall

voltage gain of 80 may be obtained with a

supply potential of 12 volts.

It is possible to employ N -P -N and P -N -P

transistors in complementary symmetry circuits

which have no equivalent in vacuum tube de-

sign. Figure 15A illustrates such a circuit. A

symmetrical push -pull circuit is shown in

-t2V

VOLTAGE GAIN = 80

INPUT IMPEDANCE L 1200 11

Figure 13

P -N -P TRANSISTOR VOLTAGE AMPLIFIERS

A resistance coupled amplifier employing an inexpensive CK -722 transistor is shown in A. For use with

a high impedance crystal microphone, a step -down transformer matches the low input impedance of the

transistor, as shown in B. The grounded collector configuration of C provides an input impedance of

about 300,000 ohms.

VOLTAGE GAI N . 0 97

INPUT IMPEDANCE 'L 300 /l 11

www.americanradiohistory.com

98 Transistors and Semi -Conductors THE RADIO

4.7N IOUF

6A/N'L R2

Figure 14

TWO STAGE TRANSISTOR AUDIO AMPLIFIERS

The feedback loop of B may be added to the r -c amplifier to reduce distortion, or to control the

audio response. A direct coupled amplifier is shown in C.

figure 15B. This circuit may be used to di-

rectly drive a high impedance loudspeaker,

eliminating the output transformer. A direct

coupled three stage amplifier having a gain

figure of 80 db is shown in figure 15C.

The transistor may also be used as a class

A power amplifier, as shown in figure 16A.

Commercial transistors are available that will

provide five or six watts of audio power when

operating from a 12 volt supply. The smaller

units provide power levels of a few milli -

watts. The correct operating point is chosen so

that the output signal can swing equally in the

positive and negative directions, as shown in

the collector curves of figure 16B.

The proper primary impedance of the out-

put transformer depends upon the amount of

power to be delivered to the load:

R- E:

P 2R.

The collector current bias is:

213"

Ic-= E,

In a class A output stage, the maximum a -c

power output obtainable is limited to 0.5 the

allowable dissipation of the transistor. The

product I, E, determines the maximum collector

dissipation, and a plot of these values is shown

in figure 16B. The load line should always lie

under the dissipation curve, and should encom-

pass the maximum possible area between the

axes of the graph for maximum output condi-

tion. In general, the load line is tangent to the

dissipation curve and passes through the supply

voltage point at zero collector current. The d -c

operating point is thus approximately one -half

the supply voltage.

The circuit of a typical push -pull class B

transistor amplifier is shown in figure 17A.

Push -pull operation is desirable for transistor

operation, since the even -order harmonics are

largely eliminated. This permits transistors to

be driven into high collector current regions

without distortion normally caused by non -

linearity of the collector. Cross -over distortion

is reduced to a minimum by providing a slight

forward base bias in addition to the normal

emitter bias. The base bias is usually less than

0.5 volt in most cases. Excessive base bias will

boost the quiescent collector current and there-

by lower the overall efficiency of the stage.

NPN PNP

O

+E

2N78

NPN 2N78

NPN 2N77

PNP

só_11

SPEAKER

O

Figure 15

COMPLEMENTARY SYMMETRY AMPLIFIERS.

N -P -N and P -N -P transistors may be combined in circuits which have no equivalent in vacuum tube

design. Direct coupling between cascaded stages using a single power supply source may be employed, as

in C. Impedance of power supply should be extremely low.

www.americanradiohistory.com

HANDBOOK Transistor Circuitry 99

Figure 16

TYPICAL CLASS -A

AUDIO POWER

TRANSISTOR CIRCUIT.

The correct operating point is

:hosen so that output signal can

swing equally in a positive or

negative direction, without ex-

:ceding maximum collector dis-

sipation.

2N 187A

I MAX MAXIMUM COLLECTOR

DISSIPATION (IC X EC)

OPERATING POINT

Ec 2Ec

COLLECTOR VOLTAGE

The operating point of the class B ampli-

fier is set on the I. =O axis at the point where

the collector voltage equals the supply voltage.

The collector to collector impedance of the

output transformer is:

2Er'

Rc-. = Po

In the class B circuit, the maximum a -c

power input is approximately equal to five

times the allowable collector dissipation of

each transistor. Power transistors, such as the

2N301 have collector dissipation ratings of

5.5 watts and operate with class B efficiency

of about 67%. To achieve this level of opera-

tion the heavy duty transistor relies upon ef-

ficient heat transfer from the transistor case

to the chassis, using the large thermal capacity

of the chassis as a heat sink. An infinite heat

sink may be approximated by mounting the

transistor in the center of a 6" x 6" copper or

aluminum sheet. This area may be part of a

'arger chassis.

The collector of most power transistors is

electrically connected to the case. For appli-

cations where the collector is not grounded a

thin sheet of mica may be used between the

case of the transistor and the chassis.

Power transistors such as the Philco T -1041

may be used in the common collector class B

configuration (figure 17C) to obtain high

power output at very low distortions compar-

able with those found in quality vacuum tube

circuits having heavy overall feedback. In ad-

dition, the transistor may be directly bolted to

the chassis, assuming a negative grounded

power supply Power output is of the order of

10 watts, with about 0.5% total distortion.

R -F Circuitry Transistors may be used for

radio frequency work provided

the alpha cutoff frequency of the units is

sufficiently higher than the operating fre-

quency. Shown in figure 18A is a typical i -f

amplifier employing an N -P -N transistor. The

collector current is determined by a voltage

divider on the base circuit and by a bias re-

sistor in the emitter leg. Input and output are

coupled by means of tuned i -f transformers.

Bypass capacitors are placed across the bias

resistors to prevent signal frequency degener-

ation. The base is connected to a low im-

pedance untuned winding of the input trans-

former, and the collector is connected to a tap

on the output transformer to provide proper

matching, and also to make the performance of

the stage relatively independent of variations

between transistors of the same type. With a

rate -grown N -P -N transistor such as the G.E.

2N293, it is unnecessary to use neutralization

to obtain circuit stability. When P -N -P alloy

ZS'3000CT.

a.7N 12V.

ZP-soonc.T.

200 MW

2N109

V

z

LOAD LINE

NO SIGNAL

OPERATING

POINT

J

COLLECTOR VOLTAGE EC

ZS= BOO n

2N225 T-104_1

R1

SO -T

í13v.

ADJUST Ri FOR 1 X ICC - 0.3 AMP.

O.4 V. BASE BIAS ICC (NAB.)',.35A.

PO' 10 WATTS

Figure 17

CLASS -B AUDIO AMPLIFIER CIRCUITRY.

The common collector circuit of C permits the transistor to be bolted directly to the chassis for efficient

heat transfer from the transistor case to the chassis.

www.americanradiohistory.com

100 Transistors and Semi- Conductors THE RADIO

TO

MIXER

OR

CONVERTER

2N293

N PN

r9v

p OUT. TO

MIXER

OR

CONVERTER

2N135 e

Figure 18

TRANSISTORIZED I -F AMPLIFIERS.

Typical P -N -P transistor must be neutralized because of high collector capacitance. Rate

grown N -P -N transistor does not usually require external neutralizing circuit.

T N

I P.

OUT.

ro

MIXER

OR

CONVERTER

1N64

Figure 19

AUTOMATIC VOLUME

CONTROL CIRCUIT

FOR TRANSISTORIZED

I -F AMPLIFIER.

transistors are used, it is necessary to neutra-

lize the circuit to obtain stability (figure 18B).

The gain of a transistor i -f amplifier will

decrease as the emitter current is decreased.

This transistor property can be used to control

the gain of an i -f amplifier so that weak and

strong signals will produce the same audio

output. A typical i -f strip incorporating this

automatic volume control action is shown in

figure 19.

R -f transistors may be used as mixers or

autodyne converters much in the same manner

as vacuum tubes The autodyne circuit is shown

in figure 20. Transformer T, feeds back a

Figure 20

THE AUTODYNE CONVERTER CIRCUIT

USING A 2N168A AS A MIXER.

signal from the collector to the emitter caus-

ing oscillation. Capacitor C, tunes thé oscillator

circuit to a frequency 455 kc. higher than that

of the incoming signal. The local oscillator

signal is inductively coupled into the emitter

circuit of the transistor. The incoming signal

is resonated in T_ and coupled via a low im-

pedance winding to the base circuit. Notice

that the base is biased by a voltage divider

circuit much the same as is used in audio fre-

quency operation. The two signals are mixed

in this stage and the desired beat frequency of

455 kc. is selected by i -f transformer T1 and

passed to the next stage. Collector currents of

0.6 ma. to 0.8 ma. are common, and the local

oscillator injection voltage at the emitter is in

the range of 0.15 to 0.25 volts, r.m.s.

A complete receiver "front end" capable of

operation up to 23 Mc. is shown in figure 21.

The RCA 2N247 drift transistor is used for

the r -f amplifier (TRI ), mixer (TR2), and

high frequency oscillator (TR3) . The 2N247

incorporates an interlead shield, cutting the

interlead capacitance to .003 q fd. If proper

shielding is employed between the tuned cir-

cuits of the r -f stage, no neutralization of the

stage is required. The complete assembly ob-

tains power from a 9 -volt transistor battery.

Note that input and output circuits of the tran-

sistors are tapped at low impedance points on

the r -f coils to achieve proper impedance match.

www.americanradiohistory.com

HANDBOOK Transistor Circuitry 101

Figure 21

RF AMPLIFIER, MIXER,

AND OSCILLATOR

STAGES FOR

TRANSISTORIZED i

HIGH FREQUENCY

RECEIVER. THE RCA

2N247 DRIFT L

TRANSISTOR IS

CAPABLE OF

EFFICIENT OPERATION

UP TO 23 Mc.

Transistor Sufficient coupling of the proper

Oscillators phase between input and output

circuits of the transistor will per-

mit oscillation up to and slightly above the

alpha cutoff frequency. Various forms of tran-

sistor oscillators are shown in figure 22. A

simple grounded emitter Hartley oscillator hav-

ing positive feedback between the base and the

collector (22A) is compared to a grounded

base Hartley oscillator (22B) . In each case

the resonant tank circuit is common to the in-

put and output circuits of the transistor. Self -

bias of the transistor is employed in both these

circuits A more sophisticated oscillator em-

ploying a 2N247 transistor and utilizing a

voltage divider -type bias system (figure 22C)

is capable of operation up to 50 Mc. or so.

The tuned circuit is placed in the collector,

with a small emitter -collector capacitor provid-

ing feedback to the emitter electrode.

A P -N -P and an N -P -N transistor may be

combined to form a complementary Hartley

oscillator of high stability ( figure 23). The

collector of the P -N -P transistor is directly

coupled to the base of the N -P -N transistor,

and the emitter of the N -P -N transistor furn-

ishes the correct phase reversal to sustain os-

cillation. Heavy feedback is maintained be-

tween the emitter of the P -N -P transistor and

the collector of N -P -N transistor. The degree

of feedback is controlled by R1. The emitter

resistor of the second transistor is placed at the

+9V.

PwP

2N247 ion NPN

1N81 1N81 2N78

Figure 23

COMPLEMENTARY HARTLEY

OSCILLATOR

P -N -P and N -P -N transistors form high sta-

bility oscillator. Feedback between P -N -P

emitter and N -P -N collector is controlled by

R,. 1N81 diodes are used as amplitude lim-

iters. Frequency of oscillation is determined

by L, C, -C :.

RFC

Figure 22

TYPICAL TRANSISTOR OSCILLATOR CIRCUITS

A- Grounded Emitter Hartley

B- Grounded Base Hartley

C -2N247 Oscillator Suitable for SO Mc. operation.

RFC

www.americanradiohistory.com

102 Transistors and Semi -Conductors THE RADIO

2N33

POINT -CONTACT TRANSISTOR

RFC

V.

Figure 24

NEGATIVE RESISTANCE OF

POINT -CONTACT TRANSISTOR

PERMITS HIGH FREQUENCY

OSCILLATION (50 Mc) WITHOUT

WITHOUT NECESSITY OF

EXTERNAL FEEDBACK PATH.

center of the oscillator coil to eliminate load-

ing of the tuned circuit.

Two germanium diodes are employed as

amplitude limiters, further stabilizing ampli-

fier operation. Because of the low circuit im-

pedances, it is permissible to use extremely

high -C in the oscillator tank circuit, effectively

limiting oscillator temperature stability to var-

iations in the tank inductance.

The point- contact transistor exhibits nega-

tive input and output resistances over part of

its operaing range, due to its unique ability

to multiply the input current. This character-

istic affords the use of oscillator circuitry hav-

ing no external feedback paths ( figure 24).

A high impedance resonant circuit in the base

lead produces circuit instability and oscillation

at the resonant frequency of the L -C circuit.

Positive emitter bias is used to insure thermal

circuit stability.

POiNT-CONTA.T

TRANS.STON LE CI.ARGN

PER OO

Figure 25

RELAXATION OSCILLATOR USING

POINT- CONTACT OR SURFACE

BARRIER TRANSISTORS.

:E

Relaxation Transistors have almost unlimit-

Oscillators ed use in relaxation acid R -C os-

cillator service. The negative re-

sistance characteristic of the point contact tran-

sistor make it well suited to such application.

Surface barrier transistors are also widely used

in this service, as they have the highest alpha

cutoff frequency among the group of -alpha-

less-than-unity- transistors. Relaxation oscilla-

tors used for high speed counting require tran-

sistors capable of operation at repetition rates

of 5 Mc. to 10 Mc.

A simple emitter controlled relaxation os-

cillator is shown in figure 25, together with

its operating characteristic. The emitter of the

transistor is biased to cutoff at the start of the

cycle (point I) . The charge on the emitter ca-

pacitor slowly leaks to ground through the

emitter resistor, R1. Discharge time is deter-

mined by the time constant of RICI. When the

emitter voltage drops sufficiently low to permit

the transistor to reach the negative resistance

region (point 2) the emitter and collector re-

sistances drop to a low value, and the collector

*E

e.

EPUL 5E

OUT

NPN

PO5ITiwE

TRi.,,ER

PJLSE

lON

1/111

M -

N P P N P

Figure 26

TRANSISTORIZED BLOCKING OSCILLATOR (A) AND ECCLES -JORDAN

BI- STABLE MULTIVIBRATOR (B).

High -alpha transistors must be employed in counting circuits to reduce effects of

storage time caused by transit lag in transistor base.

www.americanradiohistory.com

HANDBOOK Transistor Circuitry 103

2n

PHONE`,

C1- 123LLF, .W. M /LLER # 2110

C2- 791/11F, PART OF C I

L1 - "LOOPSTIC K. COIL, J.W MILLER #2003

La- OSCILLATOR COIL, J W. MILLER 4 2002

TI -4SS RC. I.F. TRANSFORMER, J.W MILLER2031

T2 -455 PI C. I F. TRANSFORMER, J.W. MILLER 2032

Figure 28

SCHEMATIC, TRANSISTORIZED BROADCAST BAND (S00 - 1600 KC.) SUPERHETERO-

DYNE RECEIVER.

" L OOP ST ICI

COIL

Figure 27

"WRIST RADIO" CAN BE MADE

WITH LOOPSTICK, DIODE, AND

INEXPENSIVE CK -722 TRANSISTOR.

A TWENTY FOOT ANTENNA WIRE

WILL PROVIDE GOOD RECEPTION

IN STRONG SIGNAL AREAS.

current is limited only by the collector resistor,

R. The collector current is abruptly reduced

by the charging action of the emitter capacitor

CI (point 3), bringing the circuit back to the

original operating point. The "spike" of col-

lector current is produced during the charging

period of C. The duration of the pulse and the

pulse repetition frequency (p.r.f.) are con-

trolled by the values of C, R1, R_, and R.

Transistors may also be used as blocking

oscillators (figure 26A) . The oscillator may

be synchronized by coupling the locking signal

to the base circuit of the transistor. An oscil-

lator of this type may be used to drive a flip -

flop circuit as a counter. An Eccles -Jordan

bi- stable flip -flop circuit employing surface -

barrier transistors may be driven between "off"

and "on" positions by an exciting pulse as

shown in figure 26B. The first pulse drives

the "on" transistor into saturation. This tran-

sistor remains in a highly conductive state until

the second exciting pulse arrives. The transis-

tor does not immediately return to the cut -off

state, since a time lapse occurs before the out-

put waveform starts to decrease. This storage

time is caused by the transit lag of the minority

carriers in the base of the transistor. Proper cir-

cuit design and the use of high -alpha transis-

tors can reduce the effects of storage time to a

minimum. Driving pulses may be coupled to

the multivibrator through steering diodes as

shown in the illustration.

5 -6 Transistor Circuits

With the introduction of the dollar tran-

sistor, many interesting and unusual experi-

ments and circuits may be built up by the be-

ginner in the transistor field. One of the most

interesting is the "wrist watch" receiver, illus-

trated in figure 27. A diode and a transistor

amplifier form a miniature broadcast receiver,

which may be built in a small box and carried

on the person. A single 1.5 -volt penlite cell

provides power for the transistor, and a short

length of antenna wire will suffice in the vi-

cinity of a local broadcasting station.

A transistorized superhetrodyne for broad-

cast reception is shown in figure 28. No an-

tenna is required, as a ferrite "loop-stick" is

used for the r -f input circuit of the 2N136

mixer transistor. A miniature magnetic "hear-

ing aid" type earphone may be employed with

this receiver.

A simple phonograph amplifier designed

for use with a high impedance crystal pickup

is shown in figure 29. Two stages of amplifi-

cation using 2N109 transistors are used to

drive two 2N109 transistors in a class B con-

figuration. Approximately 200 milliwatts of

www.americanradiohistory.com

104 Transistors and Semi -Conductors

220N

CRYSTAL f0 K

PICKUP

2N1O9

0 i5M 2UF 2N109

0 27M

2N109

Figure 29

HIGH GAIN, LOW DISTORTION AUDIO AMPLIFIER, SUITABLE FOR USE

WITH A CRYSTAL PICKUP. POWER OUTPUT IS 250 MILLIWATTS.

power may be obtained with a battery supply

of 12 volts. Peak current drain under maxi-

mum signal conditions is 40 ma.

Shown in figure 30 is an inexpensive and

compact 25 watt transistorized modulator suit-

able for mobile use with an automobile having

a 12 volt ignition system. This unit may be

used to modulate a 6146 r.f. amplifier stage

running at 400 volts and 125 milliamperes

plate power input. The two DS -501 power

transistors (Delco) are mounted on a heat

sink made of a 6" x 6" x 1/8" aluminum plate.

The components are mounted on the sink

which serves as the chassis. Output transformer

T., consists of a 6.3 volt filament transformer

with the ends of the low voltage winding con-

nected to the collectors of the output transis-

tors. Resting modulator current is about 0.7

amperes, rising to nearly 2 amperes on full

modulation peaks.

The modulator should be positioned so that

motor heat and warm air is deflected from the

unit, or the efficiency of the aluminum heat

sink will be impaired and damage to the out-

put transistors may result. A good location for

the unit is under the dash against the firewall.

Microphone gain may be adjusted by chang-

ing the value of the 100 ohm, 2 watt series

resistor.

Figure 30.

TRANSISTORIZED 25 WATT MOBILE MODULATOR.

T1 -I50 ohm primary, 490 ohm secondary (center tap on primary not used), Thordarson

TR -S.

T2-400 ohm primary, 4 and 16 ohm secondary. Stancor TA -41.

T3-6.3 volt center tap, 3a. (See text.)

Note-Output transistors are insulated from heat -sink by Delco mica insulators

(s: 1221264).

NOTE: CONTROL C /ACV /T RETURNS ro

-Nor- BATTERY TERMINAL. CONTROL +12.e V.

CIRCUIT -1z.e V.

R.F.

LOAD

www.americanradiohistory.com

Zener Diodes 105

5 -7 Zener Diodes

The Zener Diode is a semiconductor device

that can be used as a constant voltage refer-

ence, or as a control element. Zener diodes

are available in ratings to 50 watts, with zener

voltages of approximately 4 volts to 200 volts.

The zener diode has electrical character-

istics that are derived from a rectifying junc-

tion which operates at a reverse bias condition

not normally used. The zener knee (figure 31)

and constant voltage plateau are obtained

when this rectifying junction is back -biased

above the junction breakdown voltage. The

break from non -conductance to conductance is

very sharp. At applied voltages greater than

the breakdown point, the voltage drop across

the diode junction becomes essentially con-

stant for a relatively wide range of currents.

This is the zener control region.

Thermal dissipation is obtained by mount-

ing the zener diode to a heat sink composed

of a large area of metal having free access

to ambient air.

Zener Diode The zener diode may be em-

Applications ployed as a shunt regulator

(figure 32A) in the manner

of a typical "VR- tube." Two zener diodes may

be employed in the circuit of figure 32B to

supply very low values of regulated voltage.

Two opposed zener diodes can be used to pro-

vide a.c. clipping of both halves of the cycle

(figure 32C) . Zener diodes may also be used

to protect meter movements as they provide a

very low resistance shunt across the move-

ment when the applied voltage exceeds a

certain critical level.

IN ti

10 v

DIODE

UNREG.

VOLT

2 VOLT

REG.

VOLT.

FIGURE 32

A -ZENER DIODE FUNCTIONS AS

VOLTAGE REGULATOR OVER

RANGE OF CONSTANT VOLTAGE

PLATEAU.

B -TWO ZENER DIODES OF DIFFER -

1 LOUT ENT VOLTAGE CAN PROVIDE

SMALL REGULATED VOLTAGE.

C- OPPOSED ZENER DIODES CLIP

BOTH HALVES OF CYCLE OF A.C.

WAVE.

- MUM= ,. qM/iI

2 10 MMMMM

AMMO

CHARACTERISTIC

REVERSE VOLTAGE

30 20 10

ZENER KNEE MM

MlAT 5 MA.

iM=

Mi _ MAX.ZENER

M CURRENT

1.S AM.

CONSTANT

VOLTAGE

OS 1S

VP (VOLTS)

1.0 u

S=

{M N

2

REVERSe

CHARACTERISTIC

FIGURE 31

BETWEEN .ZENER KNEE- AND POINT OF MAXIMUM ZENER CUR -

RENT, THE ZENER VOLTAGE IS ESSENTIALLY CONSTANT AT SOVOLTS.

www.americanradiohistory.com

CHAPTER SIX

Vacuum Tube Amplifiers

6 -1 Vacuum Tube Parameters

The ability of the control grid of a vacuum

tube to control large amounts of plate power

with a small amount of grid energy allows the

vacuum tube to be used as an amplifier. It is

this ability of vacuum tube s to amplify an

extremely small amount of energy up to almost

any level without change in anything except

amplitude which makes the vacuum tube such

an extremely valuable adjunct to modern elec-

tronics and communication.

Symbols for As an assistance in simplify -

Vacuum -Tube ing and shortening expressions

Parameters involving vacuum -tube param-

eters, the following symbols

will be used throughout this book:

Tube Constants

ft- amplification factor

R, - plate resistance

Gm -transconductance

/tu - grid- screen mu factor

Gc - conversion transconductance(mixer tube)

Intere!ectrode Capacitances

C¡gk - grid- cathode capacitance

`Bp - grid -plate capacitance

Cpi, - plate- cathode capacitance

Cm - input capacitance (tetrode or pentode)

Cou, - output capacitance (tetrode or pentode)

106

Electrode Potentials

Ebb -d-c plate supply voltage (a positive

quantity)

E« -d -c grid supply voltage (a negative

quantity)

Egm - peak grid excitation voltage (1/2 total

peak -to -peak grid swing)

Epm -peak plate voltage (! i total peak -to -peak

plate swing)

ep - instantaneous plate potential

eg - instantaneous grid potential

epmin - minimum instantaneous plate voltage

egmp - maximum positive instantaneous grid

voltage

Ep - static plate voltage

Eg - static grid voltage

eco - cutoff bias

Electrode Currents

lb - average plate current

I, - average grid current

fpm -peak fundamental plate current

ipmax - maximum instantaneous plate current

igmaz - maximum instantaneous grid current

Ip - static plate current

Ig - static grid current

Other Symbols

Pi - plate power input

P.-plate power output

Pp - plate dissipation

Pd - grid driving power (grid plus bias losses)

www.americanradiohistory.com

Classes of Amplifiers 107

r--1

Ccv:7: 17-T-

.7.-.7. Clan

CcKi_ :=.

1 _J L __

r--

Cw --- ST

I - -,Cour

-F-_, -

CiN :t: I

TRIODE PENTODE OR TETRODE

Figure 1

STATIC INTERELECTRODE CAPACI-

TANCES WITHIN A TRIODE, PENTODE,

OR TETRODE

135-grid dissipation

Np plate efficiency(expressed as a decimal)

9p - one -half angle of plate current flow

9g - one -half angle of grid current flow

R1 -load resistance

Z1 - load impedance

Vacuum -Tube The relationships between cer-

Constants tain of the electrode potentials

and currents within a vacuum

tube are reasonably constant under specified

conditions of operation. These relationships

are called vacuum -tube constants and are

listed in the data published by the manufac-

turers of vacuum tubes. The defining equations

for the basic vacuum -tube constants are given

in Chapter Four.

Interelectrode The values of interelectrode

Capacitances and capacitance published in

Miller Effect vacuum -tube tables are the

static values measured, in

the case of triodes for example, as shown in

figure 1. The static capacitances are simply

as shown in the drawing, but when a tube is

operating as amplifier there is another con-

sideration known as Miller Effect which causes

the dynamic input capacitance to be different

from the static value. The output capacitance

of an amplifier is essentially the same as the

static value given in the published tube tables.

The grid -to -plate capacitance is also the same

as the published static value, but since the

Cgp acts as a small capacitance coupling en-

ergy back from the pl ate circuit to the grid

circuit, the dynamic input capacitance is equal

ro the static value plus an amount (frequently

much greater in the case of a triode) deter-

mined by the gain of the stage, the plate load

impedance, and the Cgp feedback capacitance.

The total value for an audio amplifier stage

can be expressed in the following equation:

(dynamic) _ ( static) + (A + 1) Cgp

where Co is the grid -to- cathode capacitance,

Cgp is the grid -to -plate capacitance, and A is

the stage gain. This expression assumes that

the vacuum tube is operating into a resistive

load such as would be the case with an audio

stage working into a resistance plate load in

the middle audio range.

The more complete expression for the input

admittance (vector sum of capacitance and

resistance) of an amplifier operating into any

type of plate load is as follows:

Input capacitance = Cgk + (1 + A cos (9) Cgp

Input resistance

A sin O

Where: Cgk = grid -to- cathode capacitance

Cgp = grid -to -plate capacitance

A = voltage amplification of the tube

alone

O = phase angle of the plate load im-

pedance, positive for inductive

loads, negative for capacitive

It can be seen from the above that if the

plate load impedance of the stage is capaci-

tive or inductive, there will be a resistive com-

ponent in the input admittance of the stage.

The resistive component of the input admit-

tance will be positive (tending to load the

circuit feeding the grid) if the load impedance

of the plate is capacitive, or it will be negative

(tending to make the stage oscillate) if the

load impedance of the plate is inductive.

Neutralization Neutralization of the effects

of Interelectrode of interelectrode capacitance

Capacitance is employed most frequently

in the case of radio fre-

quency power amplifiers. Before the introduc-

tion of the tetrode and pentode tube, triodes

were employed as neutralized Class A ampli-

fiers in receivers. This practice has been

largely superseded in the present state of the

art through the use of tetrode and pentode

tubes in which the Cge or feedback capaci-

tance has been reduced to such a low value

that neutralization of its effects is not neces-

sary to prevent oscillation and instability.

6 -2 Classes and Types of

Vacuum -Tube Amplifiers

Vacuum -tube amplifiers are grouped into

various classes and sub -classes according to

the type of work they are intended to perform.

The difference between the various classes is

determined primarily by the value of average

grid bias employed and the maximum value of

www.americanradiohistory.com

108 Vacuum Tube Amplifiers THE RADIO

the exciting signal to he impressed upon the

grid.

Class A A Class A amplifier is an amplifier

Amplifier biased and supplied with excitation

of such amplitude that plate cur-

rent flows continuously (360° of the exciting

voltage waveshape) and grid current does not

flow at any time. Such an amplifier is normally

operated in the center of the grid- voltage

plate- current transfer characteristic and gives

an output waveshape which is a substantial

replica of the input waveshape.

Class Al This is another term applied to the

Amplifier Class A amplifier in which grid

current does not flow over any

portion of the input wave cycle.

Class A2 This is a Class A amplifier oper-

Amplifier ated under such conditions that the

grid is driven positive over a por-

tion of the input voltage cycle, but plate cur-

rent still flows over the entire cycle.

Class AB1 This is an amplifier operated under

Amplifier such conditions of grid bias and

exciting voltage that plate current

flows for more than one -half the input voltage

cycle but for less than the complete cycle. In

other words the operating angle of plate cur-

rent flow is appreciably greater than 180° but

less than 360°. The suffix 1 indicates that grid

current does not flow over any portion of the

input cycle.

Class AB2 A Class AB2 amplifier is operated

Amplifier under essentially the same condi-

tions of grid bias as the Class

AB t amplifier mentioned above, but the excit-

ing voltage is of such amplitude that grid cur-

rent flows over an appreciable portion of the

input wave cycle.

Class B A Class B amplifier is biased sub -

Amplifier stantially to cutoff of plate current

(without exciting voltage) so that

plate current flows essentially over one -half

the input voltage cycle. The operating angle

of plate current flow is essentially 180 °. The

Class B amplifier is almost always excited

to such an extent that grid current flows.

Class C A Class C amplifier is biased to a

Amplifier value greater than the value re-

quired for plate current cutoff and

is excited with a signal of such amplitude

that grid current flows over an appreciable

period of the input voltage waveshape. The

angle of plate current flow in a Class C am-

plifier is appreciably less than 180 °, or in

other words, plate current flows appreciably

Figure 2

TYPES OF BIAS SYSTEMS

A Grid bias

B - Cathode bias

C - Grid leak bias

less than one -half the time. Actually, the con-

ventional operating conditions for a Class C

amplifier are such that plate current flows for

120° to 150° of the exciting voltage wave-

shape.

Types of There are three general types of

Amplifiers amplifier circuits in use. These

types are classified on the basis

of the return for the input and output circuits.

Conventional amplifiers are called cathode re-

turn amplifiers since the cathode is effectively

grounded and acts as the common return for

both the input and output circuits. The second

type is known as a plate return amplifier or

cathode follower since the plate circuit is ef-

fectively at ground for the input and output

signal voltages and the output voltage or

power is taken between cathode and plate. The

third type is called a grid -return or grounded -

grid amplifier since the grid is effectively at

ground potential for input and output signals

and output is taken between grid and plate.

6 -3 Biasing Methods

The difference of potential between grid and

cathode is called the grid bias of a vacuum

tube. There are three general methods of

providing this bias voltage. In each of these

methods the purpose is to establish the grid

at a potential with respect to the cathode

which will place the tube in the desired opera-

ting condition as determined by its charac-

teristics.

Grid bias may be obtained from a source of

voltage especially provided for this purpose,

as a battery or other d -c power supply. This

method is illustrated in figure 2A, and is

known as fixed bias.

A second biasing method is illustrated in

figure 2B which utilizes a cathode resistor

across which an IR drop is developed as a

result of plate current flowing through it. The

www.americanradiohistory.com

HANDBOOK Amplifier Distortion 109

cathode of the tube is held at a positive po-

tential with respect to ground by the amount of

the IR drop because the grid is at ground po-

tential. Since the biasing voltage depends

upon the flow of plate current the tube cannot

be held in a cutoff condition by means of the

cat bode bias voltage developed across the

cathode resistor. The value of this resistor is

determined by the bias required and the plate

current which flows at this value of bias, as

found from the tube characteristic curves.

A capacitor is shunted across the bias resistor

to provide a low impedance path to ground for

the a -c component of the plate current which

results from an a -c input signal on the grid.

The third method of providing a biasing

voltage is shown in figure 2C, and is called

grid -leak bias. During the portion of the input

cycle which causes the grid to be positive

with respect to the cathode, grid current flows

from cathode to grid, charging capacitor C,.

When the grid draws current, the grid -to- cathode

resistance of the tube drops from an infinite

value to a very low value, on the order of

1,000 ohms or so, making the charging time

constant of the capacitor very short. This en-

ables Cs to charge up to essentially the full

value of the positive input voltage and results

in the grid (which is connected to the low po-

tential plate of the capacitor) being held es-

sentially at ground potential. During the nega-

tive swing of the input signal no grid current

flows and the discharge path of Cg is through

the grid resistance which has a value of

500,000 ohms or so. The discharge time con-

stant for C5 is, therefore, very long in com-

parison to the period of the input signal and

only a small part of the charge on C5 is lost.

Thus, the bias voltage developed by the dis-

charge of Cs is substantially constant and the

grid is not permitted to follow the positive

portions of the input signal.

6 -4 Distortion in Amplifiers

There are three main types of distortion that

may occur in amplifiers: frequency distortion,

phase distortion and amplitude distortion.

Frequency Frequency distortion may occur

Distortion when some frequency components

of a signal are amplified more than

others. Frequency distortion occurs at low

frequencies if coupling capacitors between

stages are too small, or may occur at high fre-

quencies as a result of the shunting effects of

the distributed capacities in the circuit.

Phase In figure 3 an input signal con -

Distortiob sisting of a fundamental and a

third harmonic is passed through

OUTPUT

SIGNAL

Figure 3

Illustration of the effect of phase distortion

on input wave containing o third harmonic

signal

a two stage amplifier. Although the amplitudes

of both components are amplified by identical

ratios, the output waveshape is considerably

different from the input signal because the

phase of the third harmonic signal has been

shifted with respect to the fundamental signal.

This phase shift is known as phase distortion,

and is caused principally by the coupling cir-

cuits between the stages of the amplifier.

Most coupling circuits shift the phase of a

sine wave, but this has no effect on the shape

of the output wave. However, when a complex

wave is passed through the same coupling

circuit, each component frequency of the wave-

shape may be shifted in phase by a different

amount so that the output wave is not a faith-

ful reproduction of the input waveshape.

Amplitude If a signal is passed through a vac -

Distortion uum tube that is operating on any

non -linear part of its characteristic,

amplitude distortion will occur. In such a re-

gion, a change in grid voltage does not result

in a change in plate current which is directly

proportional to the change in grid voltage. For

example, if an amplifies is excited with a sig-

nal that overdrives the tubes, the resultant

signal is distorted in amplitude, since the

tubes operate over a non -linear portion of their

characteristic.

6 -5 Resistance -

Capacitance Coupled

Audio -Frequency Amplifiers

Present practice in the design of audio -fre-

quency voltage amplifiers is almost exclusively

to use resistance -capacitance coupling be-

tween the low -level stages. Both triodes and

www.americanradiohistory.com

1 1 0 Vacuum Tube A m p l i f i e r s T H E R A D I O

Figure 4

STANDARD CIRCUIT FOR RESISTANCE -

CAPACITANCE COUPLED TRIODE AM-

PLIFIER STAGE

pentodes are used; triode amplifier stages

will be discussed first.

R -C Coupled Figure 4 illustrates the stand -

Triode Stages and circuit for a resistance -

capacitance coupled amplifier

stage utilizing a triode tube with cathode bias.

In conventional audio -frequency amplifier de-

sign such stages are used at medium voltage

levels (from 0.01 to 5 volts peak on the grid

of the tube) and use medium -p triodes such

as the 6J5 or high -p triodes such as the 6SF5

or 6SL7 -GT. Normal voltage gain for a single

stage of this type is from 10 to 70, depending

upon the tube chosen and its operating con-

ditions. Triode tubes are normally used in the

last voltage amplifier stage of an R -C ampli-

fier since their harmonic distortion with large

output voltage (25 to 75 volts) is less than

with a pentode tube.

Voltage Gain The voltage gain per stage of

per Stage a resistance -capacitance cou-

pled triode amplifier can be cal-

culated with the aid of the equivalent circuits

and expressions for the mid -frequency, high -

frequency, and low- frequency range given in

figure 5.

A triode R -C coupled amplifier stage is

normally operated with values of cathode re-

sistor and plate load resistor such that the

actual voltage on the tube is approximately

one -half the d -c plate supply voltage. To

L=-LEG

E 11EG

G

MID FREQUENCY RANGE

E= -L EG

CGN

(DYNAMIC,

NEXT STAGE)

HIGH FREQUENCY RANGE

G

LOW FREQUENCY RANGE

A_ A) RL RG

RP (RL+RC)+RL RG

A HIGH FREE). - 1

A MID FREE). Ni 1+ (REQ /XS)2

R CO RL

RL RL

1+ RG Rn

Xs ' 2TTF (CPN+CGN (orNAMlc)

A LOW FREQ. =

A MID FREQ. 1+ (XC /R)2

Xc - 1

2 TTFCC

R = RG+ RL RP

RL+ RP

Figure 5

Equivalent circuits and gain equations for a triode R -C coupled amplifier stage. In using these

equations, be sure to select the values of mu and RP which are proper for the static current and

voltages with which the tube will operate. These values may be obtained from curves published

in the RCA Tube Handbook RC -16.

www.americanradiohistory.com

'7IANDBOOK R -C Coupled Amplifiers 111

Figure 6

STANDARD CIRCUIT FOR RESISTANCE

CAPACITANCE COUPLED PENTODE AM-

PLIFIER STAGE

assist the designer of such stages, data on

operating conditions for commonly used tubes

is published in the RCA Tube Handbook RC -16.

It is assumed, in the case of the gain equations

of figure 5, that the cathode by -pass capacitor,

Ck, has a reactance that is low with respect

to the cathode resistor at the lowest frequency

to be passed by the amplifier stage.

R -C Coupled Figure 6 illustrates the stand -

Pentode Stages and circuit for a resistance -

capacitance coupled pentode

amplifier stage. Cathode bias is used and the

screen voltage is supplied through a dropping

resistor from the plate voltage supply. In con-

ventional audio -frequency amplifier design

such stages are normally used at low voltage

levels (from 0.00001 to 0.1 volts peak on the

grid of the tube) and use moderate -Gm pentodes

such as the 6SJ7. Normal voltage gain for a

stage of this type is from 60 to 250, depend-

ing upon the tube chosen and its operating

conditions. Pentode tubes are ordinarily used

the first stage of an R -C amplifier where the

high gain which they afford is of greatest ad-

vantage and where only a small voltage output

is required from the stage.

The voltage gain per stage of a resistance -

capacitance coupled pentode amplifier can be

calculated with the aid of the equivalent cir-

cuits and expressions for the mid -frequency,

high- frequency, and low- frequency range given

in figure 7.

To assist the designer of such stages, data

on operating conditions for commonly used

types of tubes is published in the RCA Tube

Handbook RC -16. It is assumed, in the case of

the gain equations of figure 7, that the cathode

by -pass capacitor, Ck, has a reactance that is

low with respect to the cathode resistor at the

lowest frequency to be passed by the stage. It

is additionally assumed that the reactance of

the screen by -pass capacitor Cd, is low with

respect to the screen dropping resistor, Rd, at

the lowest frequency to be passed by the am-

plifier stage.

Cascade Voltage When voltage amplifier stages

Amplifier Stages are operated in such a manner

that the output voltage of the

first is fed to the grid of the second, and so

forth, such stages are said to be cascaded.

The total voltage gain of cascaded amplifier

Figure 7

Equivalent circuits and

gain equations for a pen-

tode R -C coupled amplifier

stage. In using these equa-

tions be sure to select the

values of Gm and Rp which

are proper for the static

currents and voltages with

which the tube will oper-

ate. These values may be

obtained from curves pub-

lished in the RCA Tube

Handbook RC -16.

t= -GMEc

c

MID FREQUENCY RANGE

I= -GMEc

HIGH FREQUENCY RANGE

LOW FREQUENCY RANGE

A = GM REO

REO RL

RG Re

A HIGH FRED.

AMID FOtO. }(REO /Xs)2

R CO - RL 1

Xs'

1+ Rc+RP

277F (CPK+CGK (DYNAMIC)

A LOW FREQ. _

A MID FOCO. I+(XCF R)2

Xc 277r CC

R = RO + RL RP

RL+RP

www.americanradiohistory.com

112 Vacuum Tube Amplifiers THE RADIO

100-

z30

á u

1. RL 500000 ONMs

2-RL= 100000OHMS

3. RL= so 000 oHMs

4. RL' 20000 01-1M3

100 1000 10000 100000

FREQUENCY (C.P.S

1004000

Figure 8

The variation of stage gain with frequency

in an r-c coupled pentode amplifier for vari-

ous values of plate load resistance

stages is obtained by taking the product of the

voltage gains of each of the successive stages.

Sometimes the voltage gain of an amplifier

stage is rated in decibels. Voltage gain is

converted into decibels gain through the use

of the following expression: db = 20 log A,

where A is the voltage gain of the stage. The

total gain of cascaded voltage amplifier stages

can be obtained by adding the number of

decibels gain in each of the cascaded stages.

R -C Amplifier A typical frequency response

Response curve for an R -C coupled audio

amplifier is shown in figure 8.

It is seen that the amplification is poor for the

extreme high and low frequencies. The reduced

gain at the low frequencies is caused by the

loss of voltage across the coupling capacitor.

In some cases, a low value of coupling capaci-

tor is deliberately chosen to reduce the re-

sponse of the stage to hum, or to attenuate

the lower voice frequencies for communication

purposes. For high fidelity work the product of

the grid resistor in ohms times the coupling

capacitor in microfarads should equal 25,000.

(ie.: 500,000 ohms x 0.05 µfd = 25,000).

The amplification of high frequencies falls

off because of the Miller effect of the sub-

sequent stage, and the shunting effect of resi-

dual circuit capacities. Both of these effects

may be minimized by the use of a low value of

plate load resistor.

Grid Leak Bias The correct operating bias

for High Mu Triodes for a high -mu triode such

as the GSL7, is fairly crit-

ical, and will be found to be highly variable

from tube to tube because of minute variations

in contact potential within the tube itself. A

satisfactory bias method is to use grid leak

bias, with a grid resistor of one to ten meg-

MI0-EREQUENCr GAIN = GMV, RL

NIGN- FREOUCNCY GAIN a Gm*, ¿COUPLING NETWORK

C e CouT V,.CINr2 T C DISTRIBUTED

FOR COMPROMISE HIGH REOOCNCV EQUALIZATION

XLL - 0 S XC AT fC

RL . XC AT fC

WHERE f C e CUTOFF FRCQUENC, OF AMPLIFIER

LL I PEAKING INDUCTOR

POR COMPROMISE LOW PRCOUENC EQUALISATION

Re' Ro (Goo vi RL)

Rs Ce °RACK

Co a 25 TO SO Of0 IN PARALLEL WITH D01 MICA

C = CAPACITANCE FROM AsOAC WITH 001 MICA IN PARALLEL

Figure 9

SIMPLE COMPENSATED VIDEO

AMPLIFIER CIRCUIT

Resistor RL in conjunction with coil LL

serves to flatten the high -frequency response

of the stage, while CB and R serve to equal-

ize the low- frequency response of this sim-

ple video amplifier stage.

ohms connected directly between grid and

cathode of the tube. The cathode is grounded.

Grid current flows at all times, and the effec-

tive input resistance is about one -half the

resistance value of the grid leak. This circuit

is particularly well suited as a high gain

amplifier following low output devices, such

as crystal microphones, or dynamic micro-

phones.

R -C Amplifier A resistance- capacity

General Characteristics coupled amplifier can

be designed to provide

a good frequency response for almost any

desired range. For instance, such an amplifier

can be built to provide a fairly uniform ampli-

fication for frequencies in the audio range of

about 100 to 20,000 cycles. Changes in the

values of coupling capacitors and load re-

sistors can extend this frequency range to

cover the very wide range required for video

service. However, extension of the range can

only be obtained at the cost of reduced over-

all amplification. Thus the R -C method of

coupling allows good frequency response with

minimum distortion, but low amplification.

Phase distortion is less with R -C coupling

www.americanradiohistory.com

HANDBOOK Video Frequency Amplifiers 113

than with other types, except direct coupling.

The R -C amplifier may exhibit tendencies to

"motorboat" or oscillate if it is used with a

high impedance plate supply.

6 -6 Video -Frequency

Amplifiers

A video -frequency amplifier is one which

has been designed to pass frequencies from

the lower audio range (lower limit perhaps 50

cycles) to the middle r -f range (upper limit

perhaps 4 to 6 megacycles). Such amplifiers,

in addition to passing such an extremely wide

frequency range, must be capable of amplify-

ing this range with a minimum of amplitude,

phase, and frequency distortion. Video ampli-

fiers are commonly used in television, pulse

communication, and radar work.

Tubes used in video amplifiers must have

a high ratio of Gm to capacitance if a usable

gain per stage is to be obtained. Commonly

available tubes which have been designed for

or are suitable for use in video amplifiers are:

6AU6, 6AG5, 6AK5, 6CB6, 6AC7, 6AG7, and

6K6 -GT. Since, at the upper frequency limits

of a video amplifier the input and output

shunting capacitances of the amplifier tubes

have rather low values of reactance, low

values of coupling resistance along with

peaking coils or other special interstage cou-

pling impedances are usually used to flatten

out the gain /frequency and hence the phase/

frequency characteristic of the amplifier.

Recommended operating conditions along with

expressions for calculation of gain and circuit

values are given in figure 9. Only a simple

two -terminal interstage coupling network is

shown in this figure.

The performance and gain -per -stage of a

video amplifier can be improved by the use

of increasingly complex two -terminal inter -

stage coupling networks or through the use

of four -terminal coupling networks or filters

between successive stages. The reader is re-

ferred to Terman's "Radio Engineer's Hand-

book" for design data on such interstage

coupling networks.

6 -7 Other Interstage

Coupling Methods

Figure 10 illustrates, in addition to resist-

ance- capacitance interstage coupling, seven

additional methods in which coupling between

two successive stages of an audio -frequency

amplifier may be accomplished. Although

resistance -capacitance coupling is most com-

monly used, there are certain circuit condi-

tions wherein coupling methods other than

resistance capacitance are more effective.

Transformer Transformer coupling, as illus-

Coupling trated in figure 1013, is seldom

used at the present time between

two successive single -ended stages of an

audio amplifier. There are several reasons why

resistance coupling is favored over transformer

coupling between two successive single -ended

stages. These are: (1) a transformer having

frequency characteristics comparable with a

properly designed R -C stage is very expensive;

(2) transformers, unless they are very well

shielded, will pick up inductive hum from

nearby power and filament transformers; (3)

the phase characteristics of step -up interstage

transformers are poor, making very difficult

the inclusion of a transformer of this type

within a feedback loop; and (4) transformers

are heavy.

However, there is one circuit application

where a step -up interstage transformer is of

considerable assistance to the designer; this

is the case where it is desired to obtain a

large amount of voltage to excite the grid of a

cathode follower or of a high -power Class A

amplifier from a tube operating at a moderate

plate voltage. Under these conditions it is pos-

sible to obtain apeak voltage on the secondary

of the transformer of a value somewhat greater

than the d -c plate supply voltage of the tube

supplying the primary of the transformer.

Push -Pull Transformer Push -pull transformer

Interstage Coupling coupling between two

stages is illustrated in

figure 10C. This interstage coupling arrange-

ment is fairly commonly used. The system is

particularly effective when it is desired, as in

the system just described, to obtain a fairly

high voltage to excite the grids of a high -

power audio stage. The arrangement is also

very good when it is desired to apply feed-

back to the grids of the push -pull stage by

applying the feedback voltage to the low -

potential sides of the two push -pull second-

aries.

Impedance Impedance coupling between two

Coupling stages is shown in figure 10D.

This circuit arrangement is seldom

used, but it offers one strong advantage over

R -C interstage coupling. This advantage is

the fact that, since the operating voltage on

the tube with the impedance in the plate cir-

cuit is the plate supply voltage, it is possible

to obtain approximately twice the peak volt-

age output that it is possible to obtain with

R -C coupling. This is because, as has been

www.americanradiohistory.com

114 Vacuum Tube Amplifiers THE RADIO

pA RESISTANCE- CAPACITANCE COUPLING

+e

+5

E© IMPEDANCE -TRANSFORMER COUPLING 0 RESISTANCE- TRANSFORMER COUPLING

+5

pH DIR- ECT COUPLING

+5

Figure 10

INTERSTAGE COUPLING METHODS FOR AUDIO FREQUENCY VOLTAGE AMPLIFIERS

mentioned before, the d -c plate voltage on an

R -C stage is approximately one -half the plate

supply voltage.

Impedance -Transformer These two circuit ar-

and Resistance -Trans- rangements, illustrated

former Coupling in figures 10E and 10F,

are employed when it is

desired to use transformer coupling for the

reasons cited above, but where it is desired

that the d -c plate current of the amplifier

stage be isolated from the primary of the cou-

pling transformer. With most types of high -

permeability wide -response transformers it is

necessary that there be no direct -current flow

through the windings of the transformer. The

impedance- transformer arrangement of figure

10E will give a higher voltage output from

the stage but is not often used since the plate

coupling impedance (choke) must have very

high inductance and very low distributed ca-

pacitance in order not to restrict the range of

www.americanradiohistory.com

HANDBOOK Phase Inverters 115

Gw = - GM 2G+1

RP' = RP G+11

G

G+1

G= RK GM (1+ )

RR = CATHODE RESISTOR

GM = GM OF EACH TUBE

L = Al OF EACH TUBE

RP = RP OF EACH TUBE

EQUIVALENT FACTORS INDICATED ABOVE BY (I) ARE

THOSE OBTAINED BY USING AN AMPLIFIER WITH A PAIR

OF SIMILAR TUBE TYPES IN CIRCUIT SHOWN ABOVE.

Figure 11

Equivalent factors for a pair of similar tri-

odes operating as a cathode -coupled audio -

frequency voltage amplifier.

the transformer which it and its associated

tube feed. The resistance -transformer arrange-

ment of figure 10F is ordinarily quite satis-

factory where it is desired to feed a trans-

former from a voltage amplifier stage with no

d.c.in the transformer primary.

Cathode The cathode coupling arrangement

Coupling of figure 10G has been widely used

only comparatively recently. One

outstanding characteristic of such a circuit is

that there is no phase reversal between the

grid and the plate circuit. All other common

types of interstage coupling are accompanied

by a 180° phase reversal between the grid

circuit and the plate circuit of the tube.

Figure 11 gives the expressions for deter-

mining the appropriate factors for an equiva-

lent triode obtained through the use of a pair

of similar triodes connected in the cathode -

coupled circuit shown. With these equivalent

triode factors it is possible to use the ex-

pressions shown in figure 5 to determine the

gain of the stage at different frequencies. The

input capacitance of such a stage is less than

that of one of the triodes, the effective grid -

to -plate capacitance is very much less (it is

so much less that such a stage may be used

as an r -f amplifier without neutralization), and

the output capacitance is approximately equal

to the grid -to -plate capacitance of one of the

triode sections. This circuit is particularly

effective with tubes such as the 6J6, 6N7, and

6SN7 -GT which have two similar triodes in

one envelope. An appropriate value of cathode

resistor to use for such a stage is the value

which would be used for the cathode resistor

cf a conventional amplifier using one of the

same type tubes with the values of plate volt-

age and load resistance to be used for the

cathode -coupled stage.

Inspection of the equations in figure 11

shows that as the cathode resistor is made

smaller, to approach zero, the Gm approaches

zero, the plate resistance approaches the Rp

of one tube, and the mu approaches zero. As

the cathode resistor is made very large the Gm

approaches one half that of a single tube of

the same type, the plate resistance approaches

twice that of one tube, and the mu approaches

the same value as one tube. But since the Gm

of each tube decreases as the cathode resistor

is made larger (since the plate current will

decrease on each tube) the optimum value of

cathode resistor will be found to be in the

vicinity of the value mentioned in the previous

paragraph.

Direct Coupling Direct coupling between suc-

cessive amplifier stages (plate

of first stage connected directly to the grid of

the succeeding stage) is complicated by the

fact that the grid of an amplifier stage must

be operated at an average negative poten-

tial with respect to the cathode of that stage.

However, if the cathode of the second ampli-

fier stage can be operated at a potential more

positive than the plate of the preceding stage

by the amount of the grid bias on the second

amplifier stage, this direct connection between

the plate of one stage and the grid of the suc-

ceeding stage can be used. Figure 10H il-

lustrates an application of this principle in

the coupling of a pentode amplifier stage to

the grid of a "hot- cathode" phase inverter. In

this arrangement the values of cathode, screen,

and plate resistor in the pentode stage are

chosen such that the plate of the pentode is at

approximately 0. 3 times the plate supply po-

tential. The succeeding phase- inverter stage

then operates with conventional values of

cathode and plate resistor (same value of re-

sistance) in its normal manner. This type of

phase inverter is described in more detail in

the section to follow.

6 -8 Phase Inverters

It is necessary in order to excite the grids

of a push -pull stage that voltages equal in

amplitude and opposite in polarity be applied

to the two grids. These voltages may be ob-

tained through the use of a push -pull input

transformer such as is shown in figure 10C.

It is possible also, without the attendant bulk

and expense of a push -pull input transformer,

to obtain voltages of the proper polarity and

www.americanradiohistory.com

116 Vacuum Tube Amplifiers THE RADIO

phase through the use of a so- called phase -

inverter stage. There are a large number of

phase inversion circuits which have been de-

veloped and applied Elut the three shown in

figure 12 have been found over a period of

time to be the most satisfactory from the point

of view of the number of components required

and from the standpoint of the accuracy with

which the two out -of -phase voltages are held

to the same amplitude with changes in supply

voltage and changes in tubes.

All of these vacuum tube phase inverters

are based upon the fact that a 180° phase

shift occurs within a vacuum tube between the

grid input voltage and the plate output voltage.

In certain circuits, the fact that the grid input

voltage and the voltage appearing across the

cathode bias resistor are in phase is used for

phase inversion purposes.

"Hot- Cathode" Figure 12A illustrates the hot -

Phase Inverter cathode type of phase in-

verter. This type of phase in-

verter is the simplest of the three types since

it requires only one tube and a minimum of

circuit components. It is particularly simple

when directly coupled from the plate of a

pentode amplifier stage as shown in figure

10H. The circuit does, however, possess the

following two disadvantages: (1) the cathode

of the tube must run at a potential of approxi-

mately 0.3 times the plate supply voltage

above the heater when a grounded common

heater winding is used for this tube as well

as the other heater -cathode tubes in a receiver

or amplifier: (2) the circuit actually has a

loss in voltage from its input to either of the

output grids-about 0.9 times the input volt-

age will be applied to each of these grids.

This does represent a voltage gain of about

1.8 in total voltage output with respect to in-

put (grid -to -grid output voltage) but it is still

small with respect to the other two phase

inverter circuits shown.

Recommended component values for use

with a 6J5 tube in this circuit are shown in

figure 12A. If it is desired to use another tube

in this circuit, appropriate values for the oper-

ation of that tube as a conventional amplifier

can be obtained from manufacturer's tube data.

The value of RL obtained should be divided by

two, and this new value of resistance placed

in the circuit as RL. The value of Rk from

tube manual tables should then be used as

Rkl in this circuit, and then the total of Rkl

and Rk2 should be equal to RL.

"Floating Paraphase" An alternate type of

Phase Inverter phase inverter some-

times called the "float-

ing paraphase" is illustrated in figure 12B.

This circuit is quite often used with a 6N7

OA "HOT CATHODE, PHASE INVERTER

® "FLOATING PARAPHAS6'PHASE INVERTER

RL Cc ma RG

47 11 22011

G=

CC.02 22011

Figure 12

THREE POPULAR PHASE -INVERTER CIR-

CUITS WITH RECOMMENDED VALUES FOR

CIRCUIT COMPONENTS

tube, and appropriate values for the 6N7 tube

in this application are shown. The circuit

shown with the values given will give a volt-

age gain of approximately 21 from the input

grid to each of the grids of the succeeding

stage. It is capable of approximately 70 volts

peak output to each grid.

The circuit inherently has a small unbalance

in output voltage. This unbalance can be elim-

inated, if it is required for some special appli-

cation, by making the resistor Rgl a few per

cent lower in resistance value than RB3.

Cathode -Coupled The circuit shown in figure

Phase Inverter 12C gives approximately one -

half the voltage gain from the

input grid to either of the grids of the suc-

ceeding stage that would be obtained from a

single tube of the same type operating as a

conventional R -C amplifier stage. Thus, with

a 6SN7 -GT tube as shown (two 6J5's in one

www.americanradiohistory.com

HANDBOOK Vacuum Tube Voltmeter 117

.01 R5 R6

Figure 13

VOLTAGE DIVIDER PHASE

INVERTER

envelope) the voltage gain from the input

grid to either of the output grids will be ap-

proximately 7-the gain is, of course, 14 from

the input to both output grids. The phase

characteristics are such that the circuit is

commonly used in deriving push -pull deflec-

tion voltage for a cathode -ray tube from a

signal ended input signal.

The first half of the 6SN7 is used as an

amplifier to increase the amplitude of the ap-

plied signal to the desired level. The second

half of the 6SN7 is used as an inverter and

amplifier to produce a signal of the same

amplitude but of opposite polarity. Since the

common cathode resistor, Rk, is not by- passed

the voltage across it is the algebraic sum of

the two plate currents and has the same shape

and polarity as the voltage applied to the in-

put grid of the first half of the 6SN7. When a

signal, e, is applied to the input circuit, the

effective grid- cathode voltage of the first

section is Ae /2, when A is the gain of the

first section. Since the grid of the second

section of the 6SN7 is grounded, the effect of

the signal voltage across Rk (equal to e/2 if

Rk is the proper value) is the same as though

a signal of the same amplitude but of opposite

polarity were applied to the grid. The output

of the second section is equal to Ae /2 if the

plate load resistors are the same for both tube

sections.

Voltage Divider A commonly used phase in-

Phase Inverter verter is shown in figure 13.

The input section (V,) is con-

nected as a conventional amplifier. The out-

put voltage from V, is impressed on the volt-

age divider R, -R,. The values of R, and R,

are in such a ratio that the voltage impressed

upon the grid of V2 is 1/A times the output

voltage of V where A is the amplification

factor of V,. The output of Vt is then of the

D.C.

INPUT Ec

o _i11-

RP1

Figure 14

DIRECT COUPLED

D -C AMPLIFIER

I.R

EP

same amplitude as the output of V but of

opposite phase.

6 -9 D -C Amplifiers

Direct current amplifiers are special types

used where amplification of very slow varia-

tions in voltage, or of d -c voltages is desired.

A simple d -c amplifier consists of a single

tube with a grid resistor across the input

terminals, and the load in the plate circuit.

Basic D -C A simple d -c amplifier

Amplifier Circuit circuit is shown in

figure 14, wherein the

the grid of one tube is connected directly to

the plate of the preceding tube in such a

manner that voltage changes on the grid of

the first tube will be amplified by the system.

The voltage drop across the plate coupling

resistor is impressed directly upon the grid

of the second tube, which is provided with

enough negative grid bias to balance out the

excessive voltage drop across the coupling

resistor. The grid of the second tube is thus

maintained in a slightly negative position.

The d -c amplifier will provide good low fre-

quency response, with negligible phase dis-

tortion. high frequency response is limited

by the shunting effect of the tube capacitances,

as in the normal resistance coupled amplifier.

A common fault with d -c amplifiers of all

types is static instability. Small changes in

the filament, plate, or grid voltages cannot

be distinguished from the exciting voltage.

Regulated power supplies and special balanc-

ing circuits have been devised to reduce the

effects of supply variations on these ampli-

fiers. A successful system is to apply the

plate potential in phase to two tubes, and to

apply the exciting signal to a push -pull grid

www.americanradiohistory.com

118 Vacuum Tube Amplifiers THE RADIO

Figure 15

LOFTIN -WHITE

D -C AMPLIFIER

circuit configuration. If the two tubes are

identical, any change in electrode voltage is

balanced out. The use of negative feedback

can also greatly reduce drift problems.

The "Loftin -Whiter Two d -c amplifier

Circuit stages may be arranged,

so that their plate

supplies are effectively in series, as illus-

trated in figure 15. This is known as a Loftin -

White amplifier. All plate and grid voltages

may be obtained from one master power supply

instead of separate grid and plate supplies.

A push -pull version of this amplifier (figure 16)

can be used to balance out the effects of slow

variations in the supply voltage.

6 -10 Single -ended Triode

Amplifiers

Figure 17 illustrates five circuits for the

operation of Class A triode amplifier stages.

Since the cathode current of a triode Class Al

(no grid current) amplifier stage is constant

with and without excitation, it is common

practice to operate the tube with cathode

bias. Recommended operating conditions in

regard to plate voltage, grid bias, and load

impedance for conventional triode amplifier

stages are given in the RCA Tube Manual,

RC -16.

Extended Class A It is possible, under certain

Operation conditions to operate single-

ended triode amplifier stages

(and pentode and tetrode stages as well) with

grid excitation of sufficient amplitude that

grid current is taken by the tube on peaks.

This type of operation is called Class A2 and

Figure 16

PUSH -PULL D -C AMPLIFIER

WITH EITHER SINGLE -ENDED

OR PUSH -PULL INPUT

BALANCE

CONTROL

is characterized by increased plate -circuit

efficiency over straight Class A amplification

without grid current. The normal Class A1

amplifier power stage will operate with a plate

circuit efficiency of from 20 per cent to perhaps

35 per cent. Through the use of Class A2

operation it is possible to increase this plate

circuit efficiency to approximately 38 to 45

per cent. However, such operation requires

careful choice of the value of plate load im-

pedance, a grid bias supply with good regula-

tion (since the tube draws grid current on

peaks although the plate current does not

change with signal), and a driver tube with

moderate power capability to excite the grid

of the Class A2 tube.

Figures 17D and 17E illustrate two methods

of connection for such stages. Tubes such as

the 845, 849, and 304TL are suitable for such

a stage. In each case the grid bias is approxi-

mately the same as would be used for a Class

Al amplifier using the same tube, and as

mentioned before, fixed bias must be used

along with an audio driver of good regulation -

preferably a triode stage with a 1:1 or step -

down driver transformer. In each case it will

be found that the correct value of plate load

impedance will be increased about 40 per cent

over the value recommended by the tube manu-

facturer for Class A1 operation of the tube.

Operation Character-

istics of a Triode

Power Amplifier

A Class A power amplifier

operates in such a way as

to amplify as faithfully as

possible the waveform ap-

plied to the grid of the tube. Large power out-

put is of more importance than high voltage

amplification, consequently gain character-

istics may be sacrificed in power tube design

to obtain more important power handling capa-

bilities. Class A power tubes, such as the 45,

2A3 and 6ÁS7 are characterized by a low

amplification factor, high plate dissipation

and relatively high filament emission.

The operating characteristics of a Class A

www.americanradiohistory.com

HANDBOOK Triode Amplifier Characteristics 119

pA IMPEDANCE COUPLING

® TRANSFORMER COUPLING

Zsz RL

-e1AS = +e

0 TRANSFORMER COUPLING FOR At OPERATION

AUTO -

TRANSFORMER

TO CLASS C

-DIAS e+e LOAD

FORMER COUPLING

Figure 17

Output coupling arrangements for single -ended

Class A triode audio -frequency power amplifiers.

triode amplifier employing an output trans-

former- coupled load may be calculated from

the plate family of curves for the particular

tube in question by employing the following

steps:

1- The load resistance should be approxi-

mately twice the plate resistance of the

tube for maximum undistorted power out-

put. Remember this fact for a quick check

on calculations.

2- Calculate the zero -signal bias voltage

(Eg).

E5 -(0. 68 x Ebb)

ll

There Ebb is the actual plate voltage of

the Class A stage, and µ is the amplifi-

cation factor of the tube.

3- Locate the E5 bias point on the IP vs.

Ep graph where the E5 bias line crosses

the plate voltage line, as shown in figure

18. Call this point P.

4- Locate on the plate family of curves the

value of zero -signal plate current, I

corresponding to the operating point, P.

5- Locate 2 x 1, (twice the value of 1p) on

the plate current axis (Y- axis). This

point corresponds to the value of maxi-

mum signal plate current, imu.

6- Locate point x on the d -c bias curve at

zero volts (Eg = 0), corresponding to the

value of imax.

7- Draw a straight line(x - y) through points

x and P. This line is the load resistance

line. Its slope corresponds to the value

of the load resistance.

8- Load Resistance, (in ohms)

R L emu - Cain

- .

¡max - tmin

where e is in volts, i is in amperes, and

RL is in ohms.

9- Check: Multiply the zero - signal plate

current, Ii,, by the operating plate volt-

age, Ep. If the plate dissipation rating

of the tube is exceeded, it is necessary

to increase the bias (E5) on the tube so

that the plate dissipation falls within

the maximum rating of the tube. If this

step is taken, operations 2 through 8

must be repeated with the new value of

E5.

10- For maximum power output, the peak a -c

grid voltage on the tube should swing to

2E5 on the negative cycle, and to zero -

bias on the positive cycle. At the peak

of the negative swing, the plate volt-

age reaches emu and the plate current

drops to iain On the positive swing of

the grid signal, the plate voltage drops

to envia and the plate current reaches

ima. The power output of the tube is:

Power Output (watts)

P (imax - ¡min) x (emaz - emin)

o - 8

where i is in amperes and e is in volts.

11- The second harmonic distortion generated

in a single -ended Class A triode ampli-

fier, expressed as a percentage of the

fundamental output signal is:

www.americanradiohistory.com

120 Vacuum Tube Amplifiers THE RADIO

250

to W200

cc

ä

,50

_ 7

MN111 a1 W . 11

SOW. I71

gIf/

11 I

ai11 \

a : f ou

IMIN - -

- v I 1

0 100 200 EG 300 1 400

EMIN. PLATE VOLTS EMAX.

AVERAGE PLATE CHARACTERISTICS - 2A3

p. =4.2 Rp = BOO OHMS

PLATE DISSIPATION =15 WATTS

I.1 xi

/ /

LOAD RESISTANCE

RL - EMAZ - EMIN. OHMS

I MAX.' IMIN.

POWER OUTPUT

Po (IMAX. -IMIN) IEMAX- Ei,) WATTS

8

SECOND HARMONIC DISTORTION

IMAX.+ IMiN.) - IP

2

'MAX. X 100 PERCENT

Figure 18

Formulas for determining the operating con-

ditions for a Class A triode single -ended audio -

frequency power output stage. A typical load

line has been drawn on the average plate char-

acteristics of a type 2A3 tube to illustrate

the procedure.

% 2d harmonic =

(imax - imin) Ip

2

iman -imin

Figure 18 illustrates the above steps as

applied to a single Class A 2A3 ampli-

fier stage.

(x 100)

6 -11 Single -ended Pentode

Amplifiers

Figure 19 illustrates the conventional cir-

cuit for a single -ended tetrode or pentode am-

Figure 19

Normal single -ended pentode or beam tetrad.

audio- frequency power output stage.

plifier stage. Tubes of this type have largely

replaced triodes in the output stage of re-

ceivers and amplifiers due to the higher plate

efficiency (30 % -40 %) with which they operate.

Tetrode and pentode tubes do, however, intro-

duce a considerably greater amount of harmonic

distortion in their output circuit, particularly

odd harmonics. In addition, their plate circuit

impedance (which acts in an amplifier to damp

loudspeaker overshoot and ringing, and acts

in a driver stage to provide good regulation) is

many times higher than that of an equivalent

triode. The application of negative feedback

acts both to reduce distortion and to reduce

the effective plate circuit impedance of these

tubes.

Operating Character- The operating character-

istics of a Pentode istics of pentode power

Power Amplifier amplifiers may be obtained

from the plate family of

curves, much as in the manner applied to

triode tubes. A typical family of pentode plate

curves is shown in figure 20. It can be seen

from these curves that the plate current of the

tube is relatively independent of the applied

plate voltage, but is sensitive to screen volt-

age. In general, the correct pentode load re-

sistance is about

0. 9 Ep

IP

and the power output is somewhat less than

Ep x Ip

2

These formulae may be used for a quick check

on more precise calculations. To obtain the

operating parameters for Class A pentode am-

plifiers, the following steps are taken:

1- The imax point is chosen so as to fall on

the zero -bias curve, just above the

"knee" of the curve (point A, figure 20) .

2- A preliminary operating point, P, is deter-

mined by the intersection of the plate

voltage line, Fp, and the line of imaz /2.

www.americanradiohistory.com

'HANDBOOK Push -Pull Amplifiers 121

. .

,iAA .OAD_iNE CHOOSE

t4/. ,OA^I¡vE SOrHAT AP=PB

e MIN P(STATIC VALUE) eMA%

PLATE VOLTS

Figure 20

GRAPHIC DETERMINATION OF OPERAT-

ING CHARACTERISTICS OF A PENTODE

POWER AMPLIFIER

"V" is the negative control grid voltage at

the operating point P

The grid voltage curve that this point

falls upon should be one that is about a

the value of ER required to cut the plate

current to a very low value (Point B).

Point B represents imin on the plate cur-

rent axis (y- axis). The line ima /2 should

be located half -way between ima and

twin

3- A trial load line is constructed about

point P and point A in such a way that

the lengths A -P and P -B are approxi-

mately equal.

4- hen the most satisfactory load line has

been determined, the load resistance may

calculated:

R L emax - envia

imax - imin

5- The operating bias (ER) is the bias at

point P.

6- The power output is:

Power Output (watts)

Po (¡max - ¡min) + 1.41 (Ix -4)2 X RL

0 32

Where l is the plate current at the point

on the load line where the grid voltage,

eR, is equal to: ER - 0. 7 F.R; and where

Iy is the plate current at the point where

eR is equal to: ER + 0. 7 F.R.

7- The percentage harmonic distortion is:

% 2d harmonic distortion

¡max - min - 2 1,

= x100

¡max - ¡min + 1.41 (Ix - Iy)

Where IP is the static plate current of

of the tube.

% 3d harmonic distortion

¡max -imin -1.41 (Ix - Ty) x100

¡max - tmin + 1.41 (1x - ly)

6 -12 Push -Pull Audio

Amplifiers

A number of advantages are obtained through

the use of the push -pull connection of two or

four tubes in an audio -frequency power am-

plifier. Two conventional circuits for the use

of triode and tetrode tubes in the push -pull

connection are shown in figure 21. The two

main advantages of the push -pull circuit ar-

rangement are: (1) the magnetizing effect

of the plate currents of the output tubes is

cancelled in the windings of the output trans-

former; (2) even harmonics of the input signal

(second and fourth harmonics primarily) gen-

erated in the push -pull stage are cancelled

when the tubes are balanced.

The cancellation of even harmonics gener-

ated in the stage allows the tubes to be oper-

PUSH -PULL TRIODE AND TETRODE

FIGURE 21

www.americanradiohistory.com

122 Vacuum Tube Amplifiers THE RADIO

300

o.

1111NN1111NNN11NNlli' NN111111.

J111111rrI e..

itdNii, iGGiiGiiGiiG

GGG táiGGiiGiï rM\RJ111N.

I.a r11\ I,11

N..llrCiC '

1111N1C::NII::..Ci

1111I r ,

I I%

.rv

11111111NII , Ci7 Ci

11I.1111AIi i

11 IIM I.M

II

iiii:iii:' :

11N 11 / I

. . W/N

M w1'CCl

1/ 74VU'A

Y

II M. :íRG

i Cil:i=l. 'MIN

l.... _

N/ t1111P.. . /N!!:í ..\!

VALUE Of

ZERO SIGNAL

PLATE CUR

00 50

PLATE VOLTS 300

(EP)

Ilun

"um= A

250

-. 11

1111111111 . 1 11111

111/111/11 1!

!!I!!1!!1!!IIlII!I!!a 111111111111111

111111111111111111,¡m1111111111N111N

11111111111111111i1111111111111111111111

1111111111111111i/1111111111.1111111111

Iiill iiiiiii%1/N1111111111111111111

1111111111 1111111111111

IIIIIIN11/I IIIINIIII111111

i'Il:il:?'i11NN11111111111111111111111

11111 P;11161111N111111111111111111111

1l:í1111D41N1 111111111111111111111111

-60 70 e0 -30 -<0 30 -20 - 0 0

GRID VOLTS (EG)

Figure 22

DETERMINATION OF OPERATING PARAMETERS FOR PUSH-PULL CLASS A

TRIODE TUBES

ated Class AB -in other words the tubes may

be operated with bias and input signals of

such amplitude that the plate current of alter-

nate tubes may be cut off during a portion of

the input voltage cycle. If a tube were operated

in such a manner in a single -ended amplifier

the second harmonic amplitude generated would

be prohibitively high.

Push -pull Class AB operation allows a plate

circuit efficiency of from 45 to 60 per cent to

be obtained in an amplifier stage depending

upon whether or not the exciting voltage is

of such amplitude that grid current is drawn

by the tubes. If grid current is taken on input

voltage peaks the amplifier is said to be oper-

ating Class AB2 and the plate circuit effi-

ciency can be as high as the upper value just

mentioned. If grid current is not taken by the

stage it is said to be operating Class AB1 and

the plate circuit efficiency will be toward the

lower end of the range just quoted. In all Class

AB amplifiers the plate current will increase

from 40 to 150 per cent over the no- signal

value when full signal is applied.

Operating Characteristics The operating char-

of Push -Pull Class A acteristics of push -

Triode Power Amplifier pull Class A ampli-

fiers may also be

determined from the plate family of curves for

a particular triode tube by the following steps:

1- Erect a vertical line from the plate volt-

age axis (x -axis) at 0.6 Ep (figure 22),

which intersects the Eg = 0 curve. This

point of intersection (P), interpolated to

the plate current axis (y -axis) may be

taken as imp. It is assumed for simplifi-

cation that imaz occurs at the point of

the zero -bias curve corresponding to

0.6 Ep.

2- The power output obtainable from the two

tubes is:

i x Ep

Power output (Po) -

5

where PO is expressed in watts, imax in

amperes, and Ep is the applied plate

voltage.

3- Draw a preliminary load line through

point P to the Ep point located on the

x -axis (the zero plate current line). This

load line represents % of the actual plate -

to -plate load of the Class A tubes. There-

fore:

Ep 0.6 Ep

RL (plate -to- plate) = 4 x

¡Max

1.6 ED

max

www.americanradiohistory.com

I-ANDBOOK Class B Audio Amplifiers 123

where RL is expressed in ohms, Ep in

volts, and imu in amperes.

Figure 22 illustrates the above steps ap-

plied to a push -pull Class A amplifier using

two 2A3 tubes.

4- The average plate current is 0.636 Imp,

and, multiplied by the plate voltage, Ep,

will give the average watts input to the

plates of the two tubes. The power out-

put should be subtracted from this value

to obtain the total operating plate dis-

sipation of the two tubes. If the plate

dissipation is excessive, a slightly

higher value of RL should be chosen to

limit the plate dissipation.

5- The correct value of operating bias, and

the static plate current for the push -pull

tubes may be determined from the Eg vs.

1p curves, which are a derivation of the

Ep vs. Ip curves for various values of

Eg.

6- The Fg vs. Ip curve may be constructed

in this manner: Values of grid bias are

read from the intersection of each grid

bias curve with the load line. These

points are transferred to the Eg vs. Ip

graph to produce a curved line, A -B. If

the grid bias curves of the Ep vs. Ip

graph were straight lines, the lines of

the Eg vs. 1p graph would also be straight

This is usually not the case. A tangent

to this curve is therefore drawn, starting

at point A', and intersecting the grid

voltage abscissa (x- axis). This inter-

section (C) is the operating bias point

for fixed bias operation.

7- This operating bias point may now be

plotted on the original Eg vs. 1p family

of curves (C'), and the zero -signal cur-

rent produced by this bias is determined.

This operating bias point (C') does not fall

on the operating load line, as in the case of a

single -ended amplifier.

8- Under conditions of maximum power out-

put, the exciting signal voltage swings

from zero -bias voltage to zero -bias volt-

age for each of the tubes on each half

of the signal cycle. Second harmonic

distortion is largely cancelled out.

6 -13 Class B Audio Frequency

Power Amplifiers

The Class B audio - frequency power ampli-

fier (figure 23) operates at a higher plate -

circuit efficiency than any of the previously

described types of audio power amplifiers.

Full- signal plate- circuit efficiencies of 60 to

Bt DRIVER - BIAS

(GROUND FOR

ZERO WAS

OPERATING

CONDITION)

Bt MOD

Figure 23

CLASS B AUDIO FREQUENCY

POWER AMPLIFIER

70 per cent are readily obtainable with the

tube types at present available for this type

of work. Since the plate circuit efficiency is

higher, smaller tubes of lower plate dissipa-

tion may be used in a Class B power amplifier

of a given power output than can be used in

any other conventional type of audio amplifier.

An additional factor in favor of the Class B

audio amplifier is the fact that the power in-

put to the stage is relatively low under no-

signal conditions. It is for these reasons that

this type of amplifier has largely superseded

other types in the generation of audio -frequency

levels from perhaps 100 watts on up to levels

of approximately 150,000 watts as required for

large short -wave broadcast stations.

Disadvantages of There are attendant dis -

Class B Amplifier advantageous features to the

Operation operation of a power ampli-

fier of this type; but all

these disadvantages can be overcome by proper

design of the circuits associated with the

power amplifier stage. These disadvantages

are: (1) The Class B audio amplifier requires

driving power in its grid circuit; this dis-

advantage can be overcome by the use of an

oversize power stage preceding the Class B

stage with a step -down transformer between

the driver stage and the Class -B grids. De-

generative feedback is sometimes employed

to reduce the plate impedance of the driver

stage and thus to improve the voltage regula-

tion under the varying load presented by the

Class B grids. (2) The Class B stage requires

a constant value of average grid bias to be

supplied in spite of the fact that the grid cur-

rent on the stage is zero over most of the

cycle but rises to values as high as one -third

of the peak plate current on the peak of the

exciting voltage cycle. Special regulated bias

supplies have been used for this application,

or B batteries can be used. However, a number

www.americanradiohistory.com

124 Vacuum Tube Amplifiers THE RADIO

of tubes especially designed for Class B audio

amplifiers have been developed which require

zero average grid bias for their operation. The

811A, 838, 805, 809, HY -5514, and TZ -40 are

examples of this type of tube. All these so-

called "zero- bias" tubes have rated operating

conditions up to moderate plate voltages

wherein they can be operated without grid

bias. As the plate voltage is increased to

to their maximum ratings, however, a small

amount of grid bias, such as could be obtained

from several 4 1/2-volt C batteries, is required.

(3), A Class B audio -frequency power ampli-

fier or modulator requires a source of plate

supply voltage having reasonably good regula-

tion. This requirement led to the development

of the swinging choke. The swinging choke is

essentially a conventional filter choke in

which the core air gap has been reduced. This

reduction in the air gap allows the choke to

have a much greater value of inductance with

low current values such as are encountered

with no signal or small signal being applied

to the Class B stage. With a higher value of

current such as would be taken by a Class B

stage with full signal applied the inductance

of the choke drops to a much lower value.

With a swinging choke of this type, having

adequate current rating, as the input inductor

in the filter system for a rectifier power sup-

ply, the regulation will be improved to a point

which is normally adequate for a power supply

for a Class B amplifier or modulator stage.

Calculation of Operating The following proce-

Conditions of Class B dure can be used for

Power Amplifiers the calculation of the

operating conditions

of Class B power amplifiers when they are to

operate into a resistive load such as the type

of load presented by a Class C power ampli-

fier. This procedure will be found quite satis-

factory for the application of vacuum tubes as

Class B modulators when it is desired to

operate the tubes under conditions which are

not specified in the tube operating character-

istics published by the tube manufacturer. The

same procedure can be used with equal effec-

tiveness for the calculation of the operating

conditions of beam tetrodes as Class AB2

amplifiers or modulators when the resting

plate current on the tubes (no signal condi-

tion) is less than 25 or 30 per cent of the

maximum -signal plate current.

1- With the average plate characteristics

of the tube as published by the manu-

facturer before you, select a point on

the Ep = E& (diode bend) line at about

twice the plate current you expect the

tubes to kick to under modulation. If

beam tetrode tubes are concerned, select

a point at about the same amount of plate

current mentioned above, just to the

right of the region where the Ib line

takes a sharp curve downward. This will

be the first trial point, and the plate

voltage at the point chosen should be

not more than about 20 per cent of the

d -c voltage applied to the tubes if good

plate- circuit efficiency is desired.

2- Note down the value of ipp. and cp.,¡, at

this point.

3- Subtract the value of epm¡ from the d -c

plate voltage on the tubes.

4- Substitute the values obtained in the

following equations:

= pmau(Ebb epmin)

P0 = Power output

from 2 tubes

(Ebb emu.)

RL_4

i pma:

= Plate -to -plate load for 2 tubes

Full signal efficiency (Nu)

78.5 Cl_evm Ebb /I

Effects of Speech All the above equations are

Clipping true for sine -wave operating

conditions of the tubes con-

cerned. However, if a speech clipper is being

used in the speech amplifier, or if it is desired

to calculate the operating conditions on the

basis of the fact that the ratio of peak power

to average power in a speech wave is approxi-

mately 4 -to-1 as contrasted to the ratio of

2 -to -1 in a sine wave-in other words, when

non- sinusoidal waves such as plain speech or

speech that has passed through a clipper are

concerned, we are no longer concerned with

average power output of the modulator as far

as its capability of modulating a Class -C ampli-

fier is concerned; we are concerned with its

peak -power- output capability.

Under these conditions we call upon other,

more general relationships. The first of these

is: It requires a peak power output equal to

the Class -C stage input to modulate that input

fully. The second one is: The average power out-

put required of the modulator is equal to the

shape factor of the modulating wave multi-

plied by the input to the Class -C stage. The

shape factor of unclipped speech is approxi-

mately 0. 25. The shape factor of a sine wave

is 0. 5. The shape factor of a speech wave that

www.americanradiohistory.com

HANDBOOK Class B Parameters 125

ï

.u.

ó

U1

U Ò

Figure 24

Typical Class 8 o -f amplifier

load line. The load line has

been drawn on the overage

characteristics of o type 811

tube.

eoo

600

400

200

NA

:Ma

-111 W! d'

N' N(Ong_.

H-201111.

s 11iáse

m ,C

ma. ..s/tll

Ts Ecc - +6a -

p10 vaL o n

agarla...

EF e 6.3 VOLTS O.C.

I

-O=e=sr = _

400 600 1200 1800

PLATE VOLTS (Ebb)

AVERAGE PLATE CHARACTERISTICS TYPE 811 AND 811 -A

2000 2400

has been passed through a clipper -filter ar-

rangement is somewhere between 0. 25 and 0. 9

depending upon the amount of clipping that

has taken place. With 15 or 20 db of clipping

the shape factor may be as high as the figure

of 0.9 mentioned above. This means that the

audio power output of the modulator will be

90% of the input to the Class -C stage. Thus

with a kilowatt input we would be putting

900 watts of audio into the Class -C stage for

100 per cent modulation as contrasted to per-

haps 250 watts for unclipped speech modula-

tion of 100 per cettt.

Sample Calculation Figure 24 shows a set of

for 811A Tubes plate characteristics for

a type 811A tube with a

load line for Class B operation. Figure 25

lists a sample calculation for determining the

proper operating conditions for obtaining ap-

proximately 185 watts output from a pair of

the tubes with 1000 volts d -c plate potential.

Also shown in figure 25 is the method of de-

termining the proper ratio for the modulation

transformer to couple between the 811's or

811A's and the anticipated final amplifier

which is to operate at 2000 plate volts and

175 ma. plate current.

Modulation Transformer The method illustrated

Calculation in figure 25 can be used

in general for the deter-

mination of the proper transformer ratio to

couple between the modulator tube and the

amplifier to be modulated. The procedure can

be stated as follows: (1) Determine the proper

plate -to -plate load impedance for the modulator

tubes either by the use of the type of calcula-

Lion shown in figure 25. or by reference to the

published characteristics on the tubes to be

used. (2) Determine the load impedance which

will be presented by the Class C amplifier

stage to be modulated by dividing the operating

plate voltage on that stage by the operating

value of plate current in amperes. (3) Divide

the Class C load impedance determined in (2)

SAMPLE CALCULATION

CONDITION: 2 TYPE 811 TUBES, Ebb, = 1000

INPUT TO FINAL STAGE, 350 W.

PEAR POWER OUTPUT NEEDED. 350 IS% = 370 W.

FINAL AMPLIFIER Ebb = 2000 V.

FINAL AMPLIFIER Ib = .175 A.

FINAL AMPLIFIER ZL = -22SISL = 11400 R

.175

EXAMPLE CHOSE POINT ON 811 CHARACTERISTICS JUST

TO RIGHT OF Ebb' Ecc. (PO /NT X. F /G. 24 )

IP MAX. =.410 A. EP MIN. = +100

IG MAX. _ .100 A. EG MAX. _ 80

PEAK PO = .410 0 (1000 -10o) _ .410 X 900 = 369 W.

RL = 4 X :9000 = 8800 n.

NP = 78.5 (1 - 1 ) = 76.5 (.9) = 70.5 "b

WO (AVERAGE WITH SINE WAVE) = POIPEAR)_I813W

WIN = Ió.5 - 260 W.

Ib (MAXIMUM WITH SINE WAVE) = 260 MA

WO PEAR = 100 X80 = e W

DRIVING POWER = WZ PR - W.

TRANSFORMER:

114 - 1.29

ZP sew

TURNS RATIO = LA = 1 29 = 1.14 STEP UP

ZP

Figure 25

Typical calculation of operating conditions for

a Class B a -f power amplifier using a pair of

type 811 or 811A tubes. Plate characteristics

and load line shown in figure 24.

www.americanradiohistory.com

126 Vacuum Tube Amplifiers THE RADIO

above by the plate -to -plate load impedance for

the modulator tubes determined in (1) above.

The ratio determined in this way is the sec -

ondary-to- primary impedance ratio. (4) Take

the square root of this ratio to determine the

secondary -to- primary turns ratio. If the turns

ratio is greater than one the use of a step -up

transformer is required. If the turns ratio as

determined in this way is less than one a step -

down transformer is called for.

If the procedure shown in figure '25 has

been used to calculate the operating conditions

for the modulator tubes, the transformer ratio

calculation can be checked in the following

manner: Divide the plate voltage on the mod-

ulated amplifier by the total voltage swing on

the modulator tubes: 2 (Ebb - e, 0). This ratio

should be quite close numerically to the trans-

former turns ratio as previously determined.

The reason for this condition is that the ratio

between the total primary voltage and the d -c

plate supply voltage on the modulated stage

is equal to the turns ratio of the transformer,

since a peak secondary voltage equal to the

plate voltage on the modulated stage is re-

quired to modulate this stage 100 per cent.

Use of Clipper Speech When a clipper speech

Amplifier with Tetrode amplifier is used in

Modulator Tubes conjunction with a Class

B modulator stage, the

plate current on that stage will kick to a

higher value with modulation(due to the greater

average power output and input) but the plate

dissipation on the tubes will ordinarily be

less than with sine -wave modulation. However,

when tetrode tubes are used as modulators,

the screen dissipation will be much greater

than with sine -wave modulation. Care must

be taken to insure that the screen dissipa-

tion rating on the modulator tubes is not ex-

ceeded under full modulation conditions with

a clipper speech amplifier. The screen dissipa-

tion is equal to screen voltage times screen

current.

Practical Aspects of As stated previously, a

Class B Modulators Class B audio amplifier

requires the driving stage

to supply well -regulated audio power to the

grid circuit of the Class B stage. Since the

performance of a Class B modulator may easily

be impaired by an improperly designed driver

stage, it is well to study the problems in-

curred in the design of the driver stage.

The grid circuit of a Class B modulator may

be compared to a variable resistance which

decreases in value as the exciting grid volt-

age is increased. This variable resistance ap-

pears across the secondary terminals of the

driver transformer so that the driver stage is

called upon to deliver power to a varying load.

For best operation of the Class 13 stage, the

grid excitation voltage should not drop as the

power taken by the grid circuit increases.

These opposing conditions call for a high or-

der of voltage regulation in the driver stage

plate circuit. In order to enhance the voltage

regulation of this circuit, the driver tubes must

have low plate resistance, the driver trans-

former must have as large a step -down ratio

as possible, and the d -c resistance of both

primary and secondary windings of the driver

transformer should be low.

The driver transformer should reflect into

the plate circuit of the driver tubes a load of

such value that the required driving power is

just developed with full excitation applied to

the driver grid circuit. If this is done, the

driver transformer will have as high a step -

down ratio as is consistent with the maximum

drive requirements of the Class B stage. If

the step -down ratio of the driver transformer is

too large, the driver plate load will be so

high that the power required to drive the Class

B stage to full output cannot be developed.

If the step -down ratio is too small the regula-

tion of the driver stage will be impaired.

Driver Stage The parameters for the driver

Calculations stage may be calculated from

the plate characteristic curve, a

sample of which is shown in figure 24. The

required positive grid voltage (eg -m,$) for the

811A tubes used in the sample calculation is

found at point X, the intersection of the load

line and the peak plate current as found on the

y -axis. This is + 80 volts. If a vertical line

is dropped from point X to intersect the dotted

grid current curves, it will be found that the

grid current for a single 811A at this value of

grid voltage is 100 milliamperes (point Y).

The peak grid driving power is therefore

80 x 0.100 = 8 watts. The approximate average

driving power is 4 watts. This is an approxi-

mate figure because the grid impedance is not

constant over the entire audio cycle.

A pair of 2A3 tubes will be used as drivers,

operating Class A, with the maximum excita-

tion to the drivers occuring just below the

point of grid current flow in the 2A3 tubes.

The driver plate voltage is 300 volts, and the

grid bias is -62 volts. The peak power devel-

oped in the primary winding of the driver

transformer is:

Peak Power (Pe) = 2R1 I PE& ,a

(watts) `Rp + RI-

where it is the amplification factor of the

driver tubes (4.2 for 2A3). Eg is the peak grid

swing of the driver stage (62 volts). Rp is the

www.americanradiohistory.com

HANDBOOK Cathode Follower Amplifier 127

plate resistance of one driver tube (800 ohms).

RL is % the plate -to -plate load of the driver

stage, and Pp is 8 watts.

Solving the above equation for RL, we

obtain a value of 14,500 ohms load, plate -to-

plate for the 2A3 driver tubes.

The peak primary voltage is:

epri = 2RL x g 493 volts

Ft, +RL

and the turns ratio of the driver transformer

(primary to % secondary) is:

epri 493

= -= 6.15:1

eg(ma:) 80

Plate Circuit One of the commonest causes of

Impedance distortion in a Class B modu-

Matching lator is incorrect load impedance

in the plate circuit. The purpose

of the Class B modulation transformer is to

take the power developed by the modulator

(which has a certain operating impedance) and

transform it to the operating impedance im-

posed by the modulated amplifier stage.

If the transformer in question has the same

number of turns on the primary winding as it

has on the secondary winding, the turns ratio

is 1:1, and the impedance ratio is also 1:1. If

a 10,000 ohm resistor is placed across the

secondary terminals of the transformer, a re-

flected load of 10,000 ohms would appear

across the primary terminals. If the resistor

is changed to one of 2376 ohms, the reflected

primary impedance would also be 2376 ohms.

If the transformer has twice as many turns

on the secondary as on the primary, the turns

ratio is 2:1. The impedance ratio is the square

of the turns ratio, or 4:1. If a 10,000 ohm

resistor is now placed across the secondary

winding, a reflected load of 2,500 ohms will

appear across the primary winding.

Effects of Plate It can be seen from the

Circuit Mis -match above paragraphs that the

Class B modulator plate

load is entirely dependent upon the load

placed upon the secondary terminals of the

Class B modulation transformer. If the second-

ary load is incorrect, certain changes will

take place in the operation of the Class B

modulator stage.

When the modulator load impedance is too

low, the efficiency of the Class B stage

is reduced and the plate dissipation of the

tubes is increased. Peak plate current of

the modulator stage is increased, and satura-

tion of the modulation transformer core may

result. "Talk -back" of the modulation trans-

former may result if the plate load impedance

of the modulator stage is too low.

When the modulator load impedance is too

high, the maximum power capability of the

stage is reduced. An attempt to increase the

output by increasing grid excitation to the

stage will result in peak -clipping of the audio

wave. In addition, high peak voltages may be

built up in the plate circuit that may damage

the modulation transformer.

6 -14 Cathode- Follower

Power Amplifiers

The cathode -follower is essentially a power

output stage in which the exciting signal is

applied between grid and ground. The plate is

maintained at ground potential with respect to

input and output signals, and the output signal

is taken between cathode and ground.

Types of Cathode- Figure 26 illustrates four

Follower Amplifiers types of cathode - follower

power amplifiers in com-

mon usage and figure 27 shows the output

impedance (Ro), and stage gain (A) of both

triode and pentode(or tetrode) cathode- follower

stages. It will be seen by inspection of the

equations that the stage voltage gain is always

less than one, that the output impedance of

the stage is much less than the same stage

operated as a conventional cathode -return

amplifier. The output impedance for con-

ventional tubes will be somewhere between

100 and 1000 ohms, depending primarily on

the transconductance of the tube.

This reduction in gain and output imped-

ance for the cathode -follower comes about

since the stage operates as though it has 100

per cent degenerative feedback applied between

its output and input circuit. Even though the

voltage gain of the stage is reduced to a value

less than one by the action of the degenerative

feedback, the power gain of the stage (if it is

operating Class A) is not reduced. Although

more voltage is required to excite a cathode -

follower amplifier than appears across the load

circuit, since the cathode "follows" along

with the grid, the relative grid -to- cathode volt-

age is essentially the same as in a con-

ventional amplifier.

Use of Cathode- Although the cathode -fol-

Follower Amplifiers lower gives no voltage

gain, it is an effective

power amplifier where it is desired to feed a

low- impedance load, or where it is desired to

feed a load of varying impedance with a signal

having good regulation. This latter capability

www.americanradiohistory.com

128 Vacuum Tube Amplifiers THE RADIO

Figure 26

CATHODE-FOLLOWER OUTPUT

CIRCUITS FOR AUDIO OR

VIDEO AMPLIFIERS

makes the cathode follower particularly effec-

tive as a driver for the grids of a Class B

modulator stage.

The circuit of figure 26A is the type of am-

plifier, either single -ended or push -pull, which

may be used as a driver for a Class B modu-

lator or which may be used for other applica-

tions such as feeding a loudspeaker where un-

usually good damping of the speaker is de-

sired. If the d -c resistance of the primary of

the transformer T2 is approximately the correct

value for the cathode bias resistor for the am-

TRIODE ucr -U

J,1 +1

Re (CATHODE +

PENTODE: Ro(cAr.,00E

A = G.. Rea

GM

A

RL

L RL

RL(.U+I ) +Rp

(Rn,+Rea) Ri

RK, +Rn2+ RL'

R

Rao 1+RL Gu

Figure 27

Equivalent factors for pentode (or tetrad.)

cathode- follower power amplifiers.

plifier tube, the components Rk and Ck need

not be used. Figure 26B shows an arrangement

which may be used to feed directly a value of

load impedance which is equal to or higher

than the cathode impedance of the amplifier

tube. The value of Cc must be quite high,

somewhat higher than would be used in a con-

ventional circuit, if the frequency response of

the circuit when operating into a low- imped-

ance load is to be preserved.

Figures 26C and 26D show cathode -follower

circuits for use with tetrode or pentode tubes.

Figure 26C is a circuit similar to that shown

in 26A and essentially the same comments

apply in regard to the components Rk and Ck

and the primary resistance of the transformer

T2. Notice also that the screen of the tube is

maintained at the same signal potential as the

cathode by means of coupling capacitor Cd.

This capacitance should be large enough so

that at the lowest frequency it is desired to

pass through the stage its reactance will be

low with respect to the dynamic screen -to-

cathode resistance in parallel with Rd T2 in

this stage as well as in the circuit of figure

26A should have the proper turns (or imped-

ance) ratio to give the desired step -down or

step -up from the cathode circuit to the load.

Figure 26D is an arrangement frequently used

in video systems for feeding a coaxial cable of

relatively low impedance from a vacuum -tube

amplifier. A pentode or tetrode tube with a

cathode imped*tce as a cathode follower

(1 /G,a) approximately the same as the cable

impedance should be chosen. The 6AG7 and

6AC7 have cathode impedances of the same

order as the surge impedances of certain types

of low- capacitance coaxial cable. An arrange-

ment such as 26D is also usable for feeding

coaxial cable with audio or r -f energy where

it is desired to transmit the output signal

over moderate distances. The resistor Rk is

added to the circuit as shown if the cathode

impedance of the tube used is lower than the

www.americanradiohistory.com

HANDBOOK Feedback Amplifiers 129

characteristic impedance of the cable. If the

output impedance of the stage is higher than

the cable impedance a resistance of appro-

priate value is sometimes placed in parallel

with the input end of the cable. The values

of Cd and Rd should be chosen with the same

considerations in mind as mentioned in the

discussion of the circuit of figure 26C above.

The Cathode -Follower The cathode follower

in R -F Stages may conveniently be

used asa method of cou-

pling r -f or i -f energy between two units sepa-

rated a considerable distance. In such an

application a coaxial cable should be used to

carry the r -f or i -f energy. One such applica-

tion would be for carrying the output of a v -f -o

to a transmitter located a considerable dis-

tance from the operating position. Another

application would be where it is desired to

feed a single -sideband demodulator, an FM

adaptor, or another accessory with inter-

mediate frequency signal from a communica-

tions receiver. A tube such as a 6CB6 con-

nected in a manner such as is shown in figure

26D would be adequate for the i -f amplifier

coupler, while a 6L6 or a 6AG7 could be used

in the output stage of a v -f -o as a cathode

follower to feed the coaxial line which carries

the v -f -o signal from the control unit to the

transmitter proper.

6 -15 Feedback Amplifiers

It is possible to modify the characteristics

of an amplifier by feeding back a portion of

the output to the input. All components, cir-

cuits and tubes included between the point

where the feedback is taken off and the point

where the feedback energy is inserted are

said to be included within the feedback loop.

An amplifier containing a feedback loop is

said to be a feedback amplifier. One stage or

any number of stages may be included within

the feedback loop. However, the difficulty of

obtaining proper operation of a feedback am-

plifier increases with the bandwidth of the

amplifier, and with the number of stages and

circuit elements included within the feedback

loop.

Gain and Phase -shift The gain and phase

in Feedback Amplifiers shift of any amplifier

are functions of fre-

quency. For any amplifier containing a feed-

back loop to be completely stable the gain of

such an amplifier, as measured from the input

back to the point where the feedback circuit

connects to the input, must be less than one

INPUT SIGNAL ES AMPLIFIER

GAIN= A uTPUT E

FEEDBACK OR B PATH

VOLTAGE AMPLIFICATION WITH FEEDBACK

1 -A B

A = GAIN IN ABSENCE OF FEEDBACK

A

B = FRACTION OF OUTPUT VOLTAGE FED BACK

8 IS NEGATIVE FOR NEGATIVE FEEDBACK

FEEDBACK IN DECIBELS = 20 LOG (1 -A8)

- 20 L0G MID FRED- GAIN WITHOUT FEEDBACK

MIDFREO. GAIN WITH FEEDBACK

DISTORTION WITH FEEDBACK _ DISTORTION WITHOUT FEEDBACK

(1 -A 8)

RD _ RN

WHERE

1 -Aa (1+ -)

RD =OUTPUT IMPEDANCE OF AMPLIFIER WITH FEEDBACK

RNA OUTPUT IMPEDANCE OF AMPLIFIER WITHOUT FEEDBACK

RL = LOAD IMPEDANCE INTO WHICH AMPLIFIER OPERATES

Figure 28

FEEDBACK AMPLIFIER RELATIONSHIPS

at the frequency where the feedback voltage

is in phase with the input voltage of the am-

plifier. If the gain is equal to or more than

one at the frequency where the feedback volt-

age is in phase with the input the amplifier

will oscillate. This fact imposes a limitation

upon the amount of feedback which may be

employed in an amplifier which is to remain

stable. If the reader is desirous of designing

amplifiers in which a large amount of feed-

back is to be employed he is referred to a

book on the subject by H. W. Bode.

Types of Feedback may be either negative

Feedback or positive, and the feedback volt-

age may be proportional either to

output voltage or output current. The most

commonly used type of feedback with a -f or

video amplifiers is negative feedback pro-

portional to output voltage. Figure 28 gives

the general operating conditions for feedback

amplifiers. Note that the reduction in distor-

tion is proportional to the reduction in gain of

the amplifier, also that the reduction in the

output impedance of the amplifier is somewhat

greater than the reduction in the gain by an

amount which is a function of the ratio of the

H. W. Bode, "Network Analysis and Feedback Ampli-

fier Dsign,' D. Van Nostrand Co., 250 Fourth Ave.,

New York 3, N. Y.

www.americanradiohistory.com

130 Vacuum Tube Amplifiers THE RADIO

O° FEEDBACK e 20 LOG I R2 * R. (G..V2 RO)

¡l

RZ

a 20L04 I Ra +RR (VOLTAGE CAIN OrV2))

R2 )

GAIN OP BOTH sTNGES = [ Goo, ( ::.!-:1)

RN ; R(G..vz Ro)

111

WHERE. RN- R, %RD

R, +R2

Rz

R° GN.z Ro

RD = RCrLECTt0 LOAD IMPEDANCE ON V2

R2 PEED°ACN RESISTOR (USUALLY ABOUT S00 R)

OUTPUT .NEDPNCE - RN R2

iRZ+RN(GwV2RO))(.+ R )

RN = PLATE IMPEDANCE or V2

Figure 29

SHUNT FEEDBACK CIRCUIT

FOR PENTODES OR TETRODES

This circuit requires only the addition of

one resistor, R2, to the normal circuit for

such an application. The plate impedance

and distortion Introduced by the output

stage are materially reduced.

output impedance of the amplifier without

feedback to the load impedance. The reduction

in noise and hum in those stages included

within the feedback loop is proportional to the

reduction in gain. However, due to the reduc-

tion in gain of the output section of the ampli-

fier somewhat increased gain is required of

the stages preceding the stages included with-

in the feedback loop. Therefore the noise and

hum output of the entire amplifier may or may

not be reduced dependent upon the relative

contributions of the first part and the latter

part of the amplifier to hum and noise. If most

of the noise and hum is coming from the stages

included within the feedback loop the un-

desired signals will be reduced in the output

from the complete amplifier. It is most fre-

quently true in conventional amplifiers that

the hum and distortion come from the latter

stages, hence these will be reduced by feed-

back, but thermal agitation and microphonic

noise come from the first stage and wilt not

be reduced but may be increased by feedback

unless the feedback loop includes the first

stage of the amplifier.

Figure 29 illustrates a very simple and ef-

fective application of negative voltage feed-

back to an output pentode or tetrode amplifier

stage. The reduction in hum and distortion

may amount to 15 to 20 db. The reduction in

the effective plate impedance of the stage will

be by a factor of 20 to 100 dependent upon the

operating conditions. The circuit is commonly

used in commercial equipment with tubes such

as the 6SJ7 for VI and the 6V6 or 6L6 for V2.

6 -16 Vacuum -Tube Voltmeters

The vacuum -tube voltmeter may be considered

to be a vacuum -tube detector in which the

rectified d -c current is used as an indication

of the magnitude of the applied alternating

voltage. The vacuum tube voltmeter (v.t.v.m.)

consumes little or no power and it may be

calibrated at 60 cycles and used at audio or

radio frequencies with little change in the

calibration.

Basic D -C Vacuum - A si mple v.t.v.m. is

Tube Voltmeter shown in figure 30.

The plate load may be

a mechanical device, such as a relay or a

meter, or the output voltage may be developed

across a resistor and used for various con-

trol purposes. The tube is biased by Ec and

a fixed value of plate current flows, causing

a fixed voltage drop across the plate load

resistor, Rp. When a positive d -c voltage is

applied to the input terminals it cancels part

of the negative grid bias, making the grid

more positive with respect to the cathode.

This grid voltage change permits a greater

amount of plate current to flow, and develops

a greater voltage drop across the plate load

resistor. A negative input voltage would de-

crease the plate current and decrease the

voltage drop across Rp, The varying voltage

drop across Rp may be employed as a control

voltage for relays or other devices. When it is

desired to measure various voltages, a voltage

Figure 30

SIMPLE VACUUM TUBE

VOLTMETER

www.americanradiohistory.com

HANDBOOK Vacuum Tube Voltmeters 131

ZERO- ADJUST

Figure 31

D -C VACUUM TUBE VOLTMETER

range switch (figure 31) may precede the v.t.

v.m. The voltage to be measured is applied to

voltage divider, R1, R2, R3, by means of the

"voltage range" switch. Resistor R4 is used

to protect the meter from excessive input

voltage to the v.t.v.m. In the plate circuit of

the tube an additional battery and a variable

resistor ( "zero adjustment ") are used to

balance out the meter reading of the normal

plate current of the tube. The zero adjustment

potentiometer can be so adjusted that the

meter M reads zero current with no input volt-

age to the v.t.v.m. When a d -c input voltage

is applied to the circuit, current flows through

the meter, and the meter reading is proportion-

al to the applied d -c voltage.

The Bridge -type Another important use

V.T.V.M. of a d -c amplifier is to

show the exact point of

balance between two d -c voltages. This is

done by means of a bridge circuit with two

d -c amplifiers serving as two legs of the

bridge (figure 32). With no input signal, and

with matched triodes, no current will be read

on meter M, since the IR drops across Ri and

R2 are identical. When a signal is applied to

one tube, the IR drops in the plate circuits

become unbalanced, and meter M indicates

the unbalance. In the same way, two d -c volt-

ages may be compared if they are applied to

the two input circuits. When the voltages are

equal, the bridge is balanced and no current

flows through the meter. If one voltage changes,

the bridge becomes unbalanced and indication

of this will be noted by a reading of the meter.

A Modern VTVM For the purpose of analysis,

the operation of a modern

v.t.v.m. will be described. The lleatbkit V -7A

isa fit instrument for such adescription, since

it is able to measure positive or negative d -c

potentials, a -c r -m -s values, peak -to-peak

values, and resistance. The circuit of this

unit is shown in figure 33. A sensitive 200 d -c

Figure 32

BRIDGE -TYPE VACUUM TUBE

VOLTMETER

microammeter is placed in the cathode circuit

of a 12AU7 twin triode. The zero adjust control

sets up a balance between the twosections of

the triode such that with zero input voltage

applied to the first grid, the voltage drop across

each portion of the zero adjust control is the

same. Under this condition of balance the

meter will read zero. When a voltage is applied

to the first grid, the balance in the cathode

circuits is upset and the meter indicates the

degree of unbalance. The relationship between

the applied voltage on the first grid and the

meter current is linear and therefore the meter

can be calibrated with a linear scale. Since

the tube is limited in the amount of current

it can draw, the meter movement is elec-

tronically protected.

The maximum test voltage applied to the

12AU7 tube is about 3 volts. Higher applied

voltages are reduced by a voltage divider

which has a total resistance of about 10

megohms. An additional resistance of 1- megohm

is located in the d -c test prod, thereby per-

mitting measurements to be made in high im-

pedance circuits with minimum disturbance.

The rectifier portion of the v.t.v.m. is shown

in figure 34. When a -c measurements are de-

sired, a 6AL5 double diode is used as a full

wave rectifier to provide a d -c voltage pro -

portionalto the applied a -c voltage. This d -c

voltage is applied through the voltage divider

string to the 12AU7 tube causing the meter to

indicate in the manner previously described.

The a -c voltage scales of the meter are cali-

brated in both RMS and peak -to -peak values.

In the 1.5, 5, 15, 50, and 150 volt positions

of the range switch, the full a -c voltage being

measured is applied to the input of the 6AL5

full wave rectifier. On the 500 and 1500 volt

positions of the range switch, a divider net-

work reduces the applied voltage in order to

limit the voltage input to the 6AL5 to a safe

recommended level.

www.americanradiohistory.com

132 THE RADIO

HEATHKIT PEAKTO -PEAK VTVM

MODEL V -7A

Figure 33

www.americanradiohistory.com

HANDBOOK Vacuum Tube Voltmeters 133

02

A -C INPUT 0---I

150v OM5

MAX

6AL5

22 MEG TO VTVM

ff DC INPUT

JACK

Figure 34

FULL -WAVE RECTIFIER

FOR V.T.V.M.

The a -c calibrate control (figure 33) is used

to obtain the proper meter deflection for the

applied a -c voltage. Vacuum tubes develop a

contact potential between tube elements. Such

contact potential developed in the diode would

cause a slight voltage to be present at all

times. This voltage is cancelled out by proper

application of a bucking voltage. The amount

of bucking voltage is controlled by the a -c

balance control. This eliminates zero shift

of the meter when switching from a -c to d -c

readings.

For resistance measurements, a 1.5 volt

battery is connected through a string of multi-

pliers and the external resistance to be meas-

ured, thus forming a voltage divider across

the battery, and a resultant portion of the

battery voltage is applied to the 12AU7 twin

SNIELDED PROBE CASE

COAA,L LINE A 7 MEG OOSi =PROBE TIP

Ce-70 S

Figure 35

R -F PROBE SUITABLE

FOR USE IN IKC -100 MC

RANGE

triode. The meter scale is calibrated in re-

sistance (ohms) for this function.

Test Probes Auxiliary test probes may

be used with the v.t.v.m.

to extend the operating range, or to measure

radio frequencies with high accuracy. Shown

in figure 35 is a radio frequency probe which

provides linear response to over 100 mega-

cycles. A crystal diode is used as a rectifier,

and d -c isolation is provided by a .005 uufd

capacitor. The components of the detector are

mounted within a shield at the end of a length

of coaxial line, which terminates in the d -c

input jack of the v.t.v.m. The readings ob-

tained are RM1S, and should be multiplied by

1.414 to convert to peak readings.

www.americanradiohistory.com

CHAPTER SEVEN

High Fidelity Techniques

The art and science of the reproduction of

sound has steadily advanced, following the

major audio developments of the last decade.

Public acceptance of home music reproduction

on a "high fidelity" basis probably dates from

the summer of 1948 when the Columbia L -P

microgroove recording techniques were intro-

duced.

The term high fidelity refers to the repro-

duction of sound in which the different dis-

tortions of the electronic system are held below

limits which are audible to the majority of

listeners. The actual determination, therefore,

of the degree of fidelity of a music system is

largely psychological as it is dependent upon

the ear and temperament of the listener. By and

large, a rough area of agreement exists as to

what boundaries establish a "hi -fi" system. To

enumerate these boundaries it is first necessary

to examine sound itself.

7 -1 The Nature of Sound

Experiments with a simple tuning fork in

the seventeenth century led to the discovery

that sound consists of a series of condensations

and rarefactions of the air brought about by

movement of air molecules. The vibrations of

the prongs of the fork are communicated to

the surrounding air, which in turn transmits

the agitation to the ear drums, with the result

that we hear a sound. The vibrating fork pro-

duces a sound of extreme regularity, and this

regularity is the essence of music, as opposed to

noise which has no such regularity.

134

As shown in figure 1, the sound wave of

the fork has frequency, period, and pitch. The

frequency is a measure of the number of vi-

brations per second of the sound. A fork tuned

to produce 261 vibrations per second is tuned

to the musical note of middle -C. It is of in-

terest to note that any object vibrating, moving,

or alternating 261 times per second will pro-

duce a sound having the pitch of middle -C.

The pitch of a sound is that property which is

determined by the frequency of vibration of

the source, and not by the source itself. Thus

an electric dynamo producing 261 c.p.s. will

have a hum -pitch of middle -C, as will a siren,

a gasoline engine, or other object having the

same period of oscillation.

))111I) ))' III

TUNING FORK

Figure 1

VIBRATION OF TUNING FORK PRO-

DUCES A SERIES OF CONDENSATIONS

AND RAREFACTIONS OF AIR MOLE-

CULES. THE DISPLACEMENT OF AIR

MOLECULES CHANGES CONTINUALLY

WITH RESPECT TO TIME, CREATING

A SINE WAVE OF MOTION OF THE

DENSITY VARIATIONS.

www.americanradiohistory.com

Nature of Sound 135

FREQUENCY (CYCLES PER SECOND)

NOTE C D E F G A B -I C'

I

EQUAL-

TEMPERED

SCALE 261.0 293.7 329.6 349.2 392.0 440.0 493.9 523.21

I

Figure 2

THE EQUAL- TEMPERED SCALE CON-

TAINS TWELVE INTERVALS, EACH OF

WHICH IS 1.06 TIMES THE FREQUEN-

CY OF THE NEXT LOWEST. THE HALF-

TONE INTERVALS INCLUDE THE

ABOVE NOTES PLUS FIVE ADDITION-

AL NOTES: 277.2, 311.1, 370, 415.3,

466.2 REPRESENTED BY THE BLACK

KEYS OF THE PIANO.

The Musical The musical scale is composed

Scale of notes or sounds of various

frequencies that bear a pleasing

aural relationship to one another. Certain com-

binations of notes are harmonious to the ear

if their frequencies can be expressed by the

simple ratios of 1:2, 2:3, 3:4, and 4:5. Notes

differing by a ratio of 1:2 are said to be sep-

arated by an octave.

The frequency interval represented by an

octave is divided into smaller intervals, form-

ing the musical scales. Many types of scales

have been proposed and used, but the scale of

the piano has dominated western music for the

last hundred or so years. Adapted by J. S.

Bach, the equal- tempered scale ( figure 2) has

twelve notes, each differing from the next by

the ratio 1:1.06. The reference frequency, or

American Standard Pitch is A, or 440.0 cycles.

Harmonics and The complex sounds pro -

Overtones duced by a violin or a wind

instrument bear little resem-

blance to the simple sound wave of the tuning

fork. A note of a clarinet, for example (when

viewed on an oscilloscope) resembles figure 3.

Vocal sounds are even more complex than this.

In 1805 Joseph Fourier advanced his monu-

mental theorem that made possible a mathe-

matical analysis of all musical sounds by show-

ing that even the most complex sounds are

made up of fundamental vibrations plus har-

monics, or overtones. The tonal qualities of

any musical note may be expressed in terms of

the amplitude and phase relationship between

the overtones of the note.

To produce overtones, the sound source must

be vibrating in a complex manner, such as is

shown in figure 3. The resulting vibration is

a combination of simple vibrations, producing

a rich tone having fundamental, the octave

tone, and the higher overtones. Any sound -

W C 7 H

-J a. 2

T/ME

Figure 3

THE COMPLEX SOUND OF A MUSICAL

INSTRUMENT IS A COMBINATION OF

SIMPLE SINE -WAVE SOUNDS, CALLED

HARMONICS. THE SOUND OF LOWEST

FREQUENCY IS TERMED THE FUNDA-

MENTAL. THE COMPLEX VIBRATION

OF A CLARINET REED PRODUCES A

SOUND SUCH AS SHOWN ABOVE.

no matter how complex - can be analyzed

into pure tones, and can be reproduced by a

group of sources of pure tones. The number

and degree of the various harmonics of a tone

and their phase relationship determine the

quality of the tone.

For reproduction of the highest quality, these

overtones must be faithfully reproduced. A mu-

sical note of 523 cycles may be rich in twen-

tieth order overtones. To reproduce the origi-

nal quality of the note, the audio system must

be capable of passing overtone frequencies of

the order of 11,000 cycles. Notes of higher

fundamental frequency demand that the audio

system be capable of good reproduction up to

the maximum response limit of the human

ear, in the region of 15,000 cycles.

Reproduction Many factors enter into the

Limitations problem of high quality audio

reproduction. Most important

of these factors influence the overall design of

the music system. These are:

1- Restricted frequency range.

2- Nonlinear distortions.

3- Transient distortion.

4- Nonlinear frequency response.

5 -Phase distortion.

6- Noise, "wow ", and "flutter ".

A restricted frequency range of reproduction

will tend to make the music sound "tinny"

and unrealistic. The fundamental frequency

range covered by the various musical instru-

ments and the human voice lies between 15

cycles and 9,000 cycles. Overtones of the in-

struments and the voice extend the upper

audible limit of the music range to 15,000

cycles or so. In order to fully reproduce the

musical tones falling within this range of fre-

quencies the music system must be capable of

flawlessly reproducing all frequencies within

the range without discrimination.

www.americanradiohistory.com

136 High Fidelity Techniques THE RADIO

BASIC LIMITS FOR HIGH FIDELITY AND

"GOOD QUALITY+ REPRODUCTION

TYPE OF

DISTORTION

LIMIT

HIGH FIDELITY "GOODY REPRODUCTI I

RESTRICTED

FREQUENCY

RANGE 20 -IS000 CPS SO -10000 CPS

INTER MODULATION

DISTORTION AT

FULL OUTPUT 4 % 10 W.

HARMONIC

DISTORTION AT

FULL OUTPUT 2 % S %

WOW" 0.1% 1W.

HUM AND NOISE -70 DB BELOW

FULL OUTPUT -50 OB BELOW

FULL OUTPUT

Figure 4

Nonlinear qualities such as harmonic and

intermodulation (IM) distortion are extremely

objectionable and are created when the output

of the music system is not exactly proportional

to the input signal. Nonlinearity of any part

of the system produces spurious harmonic fre-

quencies, which in turn lead to unwanted beats

and resonances. The combination of harmonic

frequencies and intermodulation products pro-

duce discordant tones which are disagreeable

to the ears.

The degree of intermodulation may be meas-

ured by applying two tones f1 and f: of known

amplitude to the input of the amplifier under

test. The relative amplitude of the difference

tone (f2 -f1) is considered a measure of the

intermodulation distortion. Values of the order

of 4% IM or less define a high fidelity music

amplifier.

Response of the music system to rapid tran-

sient changes is extremely important. Tran-

sient peaks cause overloading and shock- excita-

tion of resonant circuits, leaving a "hang- over"

effect that masks the clarity of the sound. A

system having poor transient response will not

sound natural to the ear, even though the dis-

tortion factors are acceptably low.

Linear frequency response and good power

handling capability over the complete audio

range go hand in hand. The response should

be smooth, with no humps or dips in the curve

over the entire frequency range. This require-

ment is particularly important in the electro-

mechanical components of the music system,

such as the phonograph pickup and the loud-

speaker.

Phase distortion is the change of phase an-

gle between the fundamental and harmonic

frequencies of a complex tone. The output

wave envelope therefore is different from the

envelope of the input wave. In general, phase

distortion is difficult to hear in sounds having

complex waveforms and may be considered to

be sufficiently low in value if the IM figure of

the amplifier is acceptable.

Noise and distortions introduced into the

program material by the music system must

be kept to a minimum as they are particularly

noticeable. Record scratch, turntable "rumble ",

and "flutter" can mar an otherwise high qual-

ity system. Inexpensive phonograph motors do

not run at constant speed, and the slight varia-

tions in speed impart a variation in pitch

( wow) to the music which can easily be heard.

Vibration of the motor may be detected by the

pickup arm, superimposing a low frequency

rumble on the music.

The various distortions that appear in a

music system are summarized in figure 4, to-

gether with suggested limits within which the

system may truly be termed "high fidelity."

7 -2 The Phonograph

The modern phonograph record is a thin

disc made of vinylite or shellac material. Disc

rotation speeds of 78.3, 33 1/3, and 45 r.p.m.

are in use, with the older 78.3 r.p.m. speed

gradually being replaced by the lower speeds.

A speed of 16 2/3 r.p.m. is used for special

"talking book" recordings. A continuous groove

is cut in the record by the stylus of the record-

ing machine, spiralling inward towards the

center of the record. Amplitude variations in

this groove proportional to the sound being

recorded constitute the means of placing the

intelligence upon the surface of the record.

The old 78.3 r.p.m. recordings were cut ap-

proximately 100 grooves per inch, while the

newer "micro- groove" recordings are cut ap-

proximately 250 grooves per inch. Care must

be taken to see that the amplitude excursions

of one groove do not fall into the adjoining

groove. The groove excursions may be con-

trolled by the system of recording, and by

equalization of the recording equipment.

Recording The early commercial phono-

Techniques graph records were cut with a

mechanical- acoustic system that

produced a constant velocity characteristic with

the amplitude of cut increasing as the recorded

frequency decreased ( figure 5A) . When the

recording technique became advanced enough

to reproduce low audio frequencies, it was

necessary to reduce the amplitude of the lower

frequencies to prevent overcutting the record.

A crossover point near 500 cycles was chosen,

www.americanradiohistory.com

HANDBOOK High Fidelity Amplifier 137

L__ 4 4

SO 100

SURFACE NO /SE

f 1 4

200 500 1000

FREQUENCY (cvs)

CONSTANT VELOCITY RECORDING

SURFACE NOISE

2000 5000

V-- CROSSOVER

FREQUENCY

50 100 200 500 1000

FREQUENCY (,P-)

2000

CONSTANT AMPLITUDE BELOW CROSSOVER

FREQUENCY, NIGH FREQUENCY PRE -EMPHASIS

ABOVE CROSSOVER FREQUENCY

5000

"RIAA" PLAYBACK CURVE

10 20 40 70100 300 500 1000

FREQUENCY. CPS

3000 10000 30000

and a constant amplitude groove was cut below

this frequency ( figure 5B). This system does

not reproduce the higher audio notes, since the

recording level rapidly drops into the surface

noise level of the record as the cutting fre-

quency is raised. The modern record employs

pre -emphasis of the higher frequencies to boost

them out of the noise level of the record

(figure 5C) . When such a record is played

back on properly compensated equipment, the

audio level will remain well above the back-

ground noise level, as shown in figure 5D.

CONSTANT AMPLITUDE,

SURFACE NO /SE

CROSSOVER

FREQUENCY

CONSTANT

VELOCITY

-1- t I \

50 100 200 500 1000 2000

FREQUENCY (CPS)

CONSTANT AMPLITUDE BELOW CROSSOVER

FREQUENCY. CONSTANT VELOCITY ABOVE

CROSSOVER FREQUENCY

SO 100 200

' 5000

SURFACE NO /SE

500 1000 2000

FREQUENCY (cps)

RESPONSE OF RECORD OF SC PLAYED ON

PROPERLY COMPENSATED EQUIPMENT

Figure 5

MODERN PHONOGRAPH RECORD EM-

PLOYS CONSTANT AMPLITUDE CUT

BELOW CROSSOVER POINT AND HIGH

FREQUENCY PRE- EMPHASIS (BOOST)

ABOVE CROSSOVER FREQUENCY.

solo°

The Phonograph The most popular types

Pickup of pickup cartridges in

use today are the high

impedance crystal unit, and the low im-

pedance variable reluctance cartridge. The

crystal pickup consists of a Rochelle salt

element which is warped by the action of

the phonograph needle, producing an elec-

trical impulse whose frequency and ampli-

tude are proportional to the modulation of

the record groove. One of the new "transducer"

crystal cartridges is shown in figure 6. When

working into a high impedance load, the out-

put of a high quality crystal pickup is of the

www.americanradiohistory.com

138 High Fidelity Techniques THE RADIO

Figure 6

NEW CRYSTAL "TRANSDUCER"

CARTRIDGE PROVIDES HIGH -

FIDELITY OUTPUT AT RELATIVELY

HIGH LEVEL

order of one -half volt or so. Inexpensive crystal

units used in 78 r.p.m. record changers and

ac -dc phonographs may have as much as two

or three volts peak output. The frequency re-

sponse of a typical high quality crystal pickup

is shown in figure 7.

The variable reluctance pickup is shown in

figure 8. The reluctance of the air gap in a

magnetic circuit is changed by the movement

of the phonograph needle, creating a variable

voltage in a small coil coupled to the magnetic

lines of force of the circuit. The output im-

pedance of the reluctance cartridge is of the

order of a few hundred ohms, and the output

is approximately 10 millivolts.

For optimum performance, an equalized pre-

amplifier stage is usually employed with the

reluctance pickup. The circuit of a suitable unit

is shown in figure 9. Equalization is provided

by R5, R,, and C., with a low frequency cross-

over at about 500 cycles. Total equalization is

15 db. High frequency response may be limited

by reducing the value of R. to 5,000 - 15,000

ohms.

The standard pickup stylus for 78 r.p.m.

records has a tip radius of .0025 inch, whereas

the microgroove (33 1/3 and 45 r.p.m.) stylus

has a tip radius of .001 inch. Many pickups,

a +10

w +5-

Z 0

-5

a -10

Lai

¢ 20 50 too 200 500 1000 2000 5000 10000

FREQUENCY (cps)

Figure 7

FREQUENCY RESPONSE OF HIGH -

QUALITY CRYSTAL PHONOGRAPH

CARTRIDGE. (ELECTROVOICE 56 -DS

POWER POINT TRANSDUCER)

Figure 8

"RELUCTANCE" CARTRIDGE IS

STANDARD PICK -UP FOR

MUSIC SYSTEM.

Low stylus pressure of four grams insures

minimum record wear. Dual stylus is used

having two needle tip diameters for long

playing and 78 R.P.M. recordings.

therefore, are designed to have interchangeable

cartridges or needles to accomodate the dif-

ferent groove widths.

7 -3 The High Fidelity Amplifier

A block diagram of a typical high fidelity

system is shown in figure 10. A preamplifier

is used to boost the output level of the phono-

graph pickup, and to permit adjustment of in-

put selection, volume, record compensation,

and tone control. The preamplifier may be

mounted directly at the phonograph turntable

position, permitting the larger power amplifier

to be placed in an out of the way position.

The power amplifier is designed to operate

from an input signal of a volt or so derived

from the preamplifier, and to build this signal

to the desired power level with a minimum

amount of distortion. Maximum power output

levels of ten to twenty watts are common for

home music systems.

The power supply provides the smoothed,

d -c voltages necessary for operation of the pre-

amplifier and power amplifier, and also the

SNORT

LEADS TO

RELUCTANCE

CARTRIDGE 4

6SC7

OUTPUT

Rts Ce

66K .0

Rz e

33M r =

TC3

_ .01

Re 68K

R3 RT

66 ,33K

C4 C5

T- e Rs 33K

e+ 100 v

Figure 9

PREAMPLIFIER SUITABLE FOR USE WITH

LOW LEVEL RELUCTANCE CARTRIDGE.

www.americanradiohistory.com

HANDBOOK High Fidelity Amplifier 139

!PHONOGRAPH i f

PREAMPLIFIER+ POWER RI

AMPLIFIER

POWER

SUPPLY

TWEETERI

I

LOUD 1

I SPEAKER

SPEAKER

ENCLOSURE

S v I.

Figure 10

BLOCK DIAGRAM OF HIGH FIDELITY

MUSIC SYSTEM.

BASS BOOST TREBLE BOOST

AND ATTENUATION AND ATTENUATION

INPUT

RI

OUTPUT

80ó$r

OUTPUT

eóosr C,

Rz

ATTENUATE ArrENUATE

C2

Rn

EQUIVALENT CIRCUITS

ATTENUATE

IN.

Ri

OUT.

C2 Rz C1 our

EQUIVALENT CIRCUITS

BOOST ATTENUATE

OUT.

Figure 11

SIMPLE R -C CIRCUITS MAY BE

USED FOR BASS AND TREBLE

BOOST OR ATTENUATION.

O

filament voltages (usually a -c) for the heaters

of the various amplifier tubes.

The loudspeaker is a device which couples

the electrical energy of the high fidelity sys-

tem to the human ear and usually limits the

overall fidelity of the complete system. Great

advances in speaker design have been made in

the past years, permitting the loudspeaker to

+20

BASS BOOST TREBLE BOOST

-t 1

SPONSE

M N

TREBLE

-20 20 SO 100 200 SOO 1000 2000 5000 10000

FREQUENCY (CeS)

Figure 12

FREQUENCY RESPONSE CURVES FOR

THE BASS AND TREBLE BOOST

AND ATTENUATION CIRCUITS

OF FIGURE 11.

hold its own in the race for true fidelity.

Speaker efficiency runs from about 10% for

cone units to nearly 40% for high frequency

tweeters. The frequency response of any speak-

er is a function of the design and construction

of the speaker enclosure or cabinet that mounts

the reproducer.

Tone Equalizer networks are em-

Compensaton ployed in high fidelity equip-

ment to 1)- tailor the re-

sponse curve of the system to obtain the correct

overall frequency response, 2) -to compensate

for inherent faults in the program material,

3 )-or merely to satisfy the hearing preference

of the listener. The usual compensation net-

works are combinations of RC and RL net-

works that provide a gradual attenuation over

a given frequency range. The basic RC net-

works suitable for equlizer service are shown

in figure 11. Shunt capacitance is employed

for high frequency attenuation, and series

capacitance is used for low frequency atten-

uation. A combination of these simple a -c

voltage dividers may be used to provide almost

any response, as shown in figure 12. It is

common practice to place equalizers between

two vacuum tubes in the low level stages of

the preamplifier, as shown in figure 13. Bass

and treble boost and attenuation of the order

INPUT

Figure 13

BASS AND TREBLE LEVEL

CONTROLS, AS

EMPLOYED IN THE

HEATHKIT WA -P2

PREAMPLIFIER.

+1zAx7 .04 E. OUTPUT

BASS CONTROL TREBLE CONTROL

220 K

B+

www.americanradiohistory.com

140 High Fidelity Techniques THE RADIO

VARIABLE

e 3 9K or;

2N190 2N190 12V

IRK 1K 2N 190

RELUCTANCE

PICKUP

BASS TREBLE

OUTPUT

Figure 14

TRANSISTORIZED HIGH- FIDELITY PREAMPLIFIER FOR USE WITH

RELUCTANCE PHONOGRAPH CARTRIDGE.

f100-

+90

+BO

+70

+60

+50

+40

PAIN L EVEL

+30

+20

+10

0

10

LOWER LIMIT OF

HEARING

20 50 200 500 1d00 2000 5000 10000

FREQUENCY (CPS) 200

Figure 15

THE "FLETCHER -MUNSON" CURVE

ILLUSTRATING THE INTENSITY

RESPONSE OF THE HUMAN EAR.

100

of 15 db may be obtained from such a circuit.

A simple transistorized preamplifier using

this type of equalizing network is shown in

figure 14.

RI -R2 -R 31 THREE SECTION POTENTIOMETER, /RC

TYPE, BUILT OF THE FOLLOWING,

R1 - lAC P011 -135

R2 -/RC MOLT /SECT /ON M13 -137

R3 - /RC MOLT /SECT /ON M13-128

Figure 16

VARIABLE LOUDNESS CONTROL FOR

USE IN LOW IMPEDANCE PLATE

CIRCUITS. MAY BE PURCHASED

AS IRC TYPE LC -1 LOUDNESS

CONTROL.

Loudness The minimum threshold of

Compensaron hearing and the maximum

threshold of pain vary great-

ly with the frequency of the sound as shown

in the Fletcher- Munson curves of figure 15.

To maintain a reasonable constant tonal bal-

ance as the intensity of the sound is changed

it is necessary to employ extra bass and treble

boost as the program level is decreased. A

simple variable loudness control is shown in

figure 16 which may be substituted for the

ordinary volume control used in most audio

equipment.

The Power The power amplifier stage of

Amplifier the music system must supply

driving power for the loud-

speaker. Commercially available loudspeakers

are low impedance devices which present a

mVIO

o

w +5

z 0

o a O1 -5

ß:

R -10

w

aw

o")

SPEAKER

RESPONSE

V 14

'

-RESONANT FREQUENCY

- BAFFLE COMBINATION

OF

O -SPEAKER Ir

a iiI SPEAKER 4

Illi

20 a 10 m_

20 50 100 200 500 1000 2000 5000 10000

FREQUENCY (CPS) 20000

Figure 17

IMPEDANCE AND FREQUENCY

RESPONSE OF "4 -OHM" 12 -INCH

SPEAKER PROPERLY MOUNTED IN

MATCHING BAFFLE.

www.americanradiohistory.com

HANDBOOK High Fidelity Amplifier 141

Figure 18

TYPICAL TRIODE

AMPLIFIER WITH

FEEDBACK LOOP.

FEEDBACK RESISTOR

VJ'

* ° MATCHED PAIR RESISTORS

varying load of two to nearly one hundred

ohms to the output stage ( figure 17) . It is

necessary to employ a high quality output

transformer to match the loudspeaker load to

the relatively high impedance plate circuit of

the power amplifier stage. In general, push -

pull amplifiers are employed for the output

stage since they have even harmonic cancelling

properties and permit better low frequency

response of the output transformer since there

is no d -c core saturation effect present.

To further reduce the harmonic distortion

and intermodulation inherent in the amplifier

system a negative feedback loop is placed

around one or more stages of the unit. Fre-

quency response is thereby improved, and the

output impedance of the amplifier is sharply

reduced, providing a very low source im-

pedance for the loudspeaker.

Shown in figure 18 is a basic push -pull

triode amplifier, using inverse feedback around

the power output and driver stage. A simple

triode inverter is used to provide 180- degree

phase reversal to drive the grid circuit of the

power amplifier stage. Maximum undistorted

power output of this amplifier is about 8 watts.

A modification of the basic triode amplifier

is the popular Williamson circuit (figure 19)

developed in England in 1947. This circuit

rapidly became the "standard of comparison"

in a few short years. Pentode power tubes are

connected as triodes for the output stage, and

negative feedback is taken from the secondary

of the output transformer to the cathode of

the input stage. Only the most linear portion

of the tube characteristic curve is used. Al-

though that portion has been extended by

higher than normal plate supply voltage, it

INPUT

FEEDBACK RESISTOR

65N7GT 6SN7GT

SK-ISK

0.25

807

30KK{

= 1

10 20

22K

NJ

4-400V.

AT 140 MA

OUTPUT

Figure 19

U. S. VERSION OF BRITISH "WILLIAMSON" AMPLIFIER PROVIDES 10 WATTS POWER

OUTPUT AT LESS THAN 2 °o INTERMODULATION DISTORTION. 6SN7 STAGE USES

DIRECT COUPLING.

www.americanradiohistory.com

142 High Fidelity Techniques THE RADIO

FROM

05N7GT

PHASE

INVERTER

0 25

807/5881 TO FEEDBACK

CIRCUIT

0.25

807/5881

NOTE, P/N CONNECT IONS ARE

POR 807 TUBES f00V

OUTPUT

Figure 20

"ULTRA- LINEAR" CONFIGURATION OF WILLIAMSON AMPLIFIER DOUBLES POWER OUT-

PUT, AND REDUCES IM LEVEL. SCREEN TAPS ON OUTPUT TRANSFORMER PERMIT

"SEMI -TETRODE" OPERATION.

is only a fraction of the curve normally used

in amplifiers. Thus a comparatively low output

power level is obtained with tubes capable of

much more efficient operation under less

stringent requirements. With 400 volts ap-

plied to the output stage, a power output of 10

watts may be obained wtih less than 2% inter -

modulation distortion.

A recent variation of the Williamson cir-

cuit involves the use of a tapped output trans-

former. The screen grids of the push -pull am-

plifier stage are connected to the primary taps,

allowing operating efficiency to approach that

of the true pentode. Power output in excess

of 25 watts at less than 2% intermodulation dis-

Figure 21

"BABY HI -FI" AMPLIFIER IS DWARFED

BY 12 -INCH SPEAKER ENCLOSURE

This miniature music system is capable of ex-

cellent performance in the small home or

apartment. Preamplifier, bass and treble con-

trols, and volume control are all incorporated

in the unit. Amplifier provides 4 watts output

at 4 IM distortion.

tortion may be obtained with this circuit

(figure 20) .

7 -4 Amplifier Construction

Wiring Assembly and layout of high

Techniques fidelity audio amplifiers fol-

lows the general technique des-

cribed for other forms of electronic equipment.

Extra care, however, must be taken to insure

that the hum level of the amplifier is extreme-

ly low. A good hi -fi system has excellent re-

sponse in the 60 cycle region, and even a

minute quantity of induced a -c voltage will be

disagreeably audible in the loudspeaker. Spur-

ious eddy currents produced in the chassis by

the power transformer are usually responsible

for input stage hum.

To insure the lowest hum level, the power

transformer should be of the "upright" type

instead of the "half -shell" type which can

couple minute voltages from the windings to

a steel chassis. In addition, part of the windings

of the half -shell type project below the chassis

where they are exposed to the input wiring of

the amplifier. The core of the power trans-

former should be placed at right angles to the

core of a nearby audio transformer to reduce

spurious coupling between the two units to a

minimum.

It is common practice in amplifier design

to employ a ground bus return system for all

audio tubes. All grounds are returned to a

single heavy bus wire, which in turn is

grounded at one point to the metal chassis.

This ground point is usually at the input jack

of the amplifier. When this system is used,

a -c chassis currents are not coupled into the

amplifying stages. This type of construction is

illustrated in the amplifiers described later in

this chapter.

www.americanradiohistory.com

HANDBOOK Amplifier Construction 143

PHONO INPUT

(HIGH LEVEL)

M 3

1.9K

12AÚ7

.05 220

BASE

CONTROL TREBLE

CONTROL

K

VOLUME

CONTROL

12AU7 .05 470K

J, Sp MMr

NO RI=e.2 K^

FOR e n SPEAKER W

R110

6AQ5

e len

1,T2 SPKR SPKR

100

SISOK 1.e 10011

M

115V

ti

TI 6X5

10 1211

450V I

CH1

+210V

R 1,7

OS 70 K

+203 V.

5

6AQ5

3K 2W

2 IxÇ,A .20C111 C1C

TO

NOTES

1. ALL RESISTORS 0.3 WATT

UNLESS OTHERWISE SPECIFIED

2. ALL CAPACITOR VALUES IN MF

UNLESS OTHERWISE SPECIFIED

=4,5 =3 3 = 3. RESISTORS MARKED A ARE

MATCHED PAIRS

9 ? 4 4

12AÚ7 124Ú7 RAO5 6.405

Figure 22

SCHEMATIC, "BABY HI -FI" AMPLIFIER

T,- 260 -0 -260 volts at 90 ma., 6.3 volts at 4.0 CH,-1.5 henry at 200 ma. Chicago Standard C -2327.

amp., upright mounting. Chicago -Standard

PC -8420.

T,-10 K, CT. to 8, 16 ohms. Peerless (Altec)

S -510F.

C,A- B- C- 30 -20 -10 Aid. 350 volt. Mallory Fp -330.7

NOTE -Feedback loop returns to 8 ohm tap on T,

when 8 ohm speaker is used.

Care should be taken to reduce the capaci-

tance to the chassis of high impedance circuits,

or the high frequency response of the unit will

suffer. Shielded "bath -tub" type capacitors

should not be used for interstage coupling ca-

pacitors. Tubular paper capacitors are satis-

factory. These should be spaced well away from

the chassis.

It is a poor idea to employ the chassis as a

common filament return, especially for low

level audio stages. The filament center -tap of

the power transformer should be grounded,

and twisted filament wires run to each tube

socket. High impedance audio components and

wiring should be kept clear of the filament

lines, which may even be shielded in the vicin-

ity of the input stage. In some instances, the

filament center tap may be taken from the arm

of a low resistance, wirewound potentiometer

placed across the filament pins of the input

tube socket. The arm of this potentiometer is

grounded, and the setting of the control is ad-

justed for minimum speaker hum.

7 -5 The "Baby Hi Fi"

A definite need exists for a compact, high

fidelity audio amplifier suitable for use in the

small home or apartment. Listening tests have

shown that an average power level of less than

one watt in a high efficiency speaker will pro-

vide a comfortable listening level for a small

room, and levels in excess of two or three watts

are uncomfortably loud to the ear. The "Baby

Hi -Fi" amplifier has been designed for use

in the small home, and will provide excellent

quality at a level high enough to rattle the

windows.

Designed around the new Electro -Voice min-

iature ceramic cartridge, the amplifier will pro-

vide over 4 watts power, measured at the sec-

ondary of the output transformer. At this level,

the distortion figure is below 1 %, and the IM

figure is 4 %. At normal listening levels, the IM

is much lower, as shown in figure 24.

The Amplifier The schematic of the ampli -

Circuit fier is shown in figure 22.

Bass and treble boost con-

trols are incorporated in the circuit, as is the

volume control. A dual purpose 12AU7 double

triode serves as a voltage amplifier with cath-

ode degeneration. A simple voltage divider

network is used in the grid circuit to prevent

amplifier overloading when the ceramic cart-

ridge is used. The required input signal for

maximum output is of the order of 0.3 volts.

The output level of the Electro -Voice cartridge

is approximately twice this, as shown in figure

7. The use of the high -level cartridge elimin-

www.americanradiohistory.com

144 High Fidelity Techniques THE RADIO

Figure 23

UNDER -CHASSIS

VIEW OF

"BABY HI -FI"

Low level audio stages are at

upper left, with components

mounted between socket pins

and potentiometer controls.

6X5 socket is at lower cen-

ter of photo with filter

choke CH, at right. Feed-

back resistor R, is at left

of rectifier socket.

ates the necessity of high gain amplifiers re-

quired when low level magnetic pickup heads

are used. Problems of hum and distortion in-

troduced by these extra stages are thereby

eliminated, greatly simplifying the amplifier.

The second section of the 12AU7 is used for

bass and treble boost. Simple R -C networks

are placed in the grid circuit permitting gain

boost of over 12 db at the extremities of the

response range of the amplifier.

A second 12AU7 is employed as a direct

coupled "hot-cathode" phase inverter, capaci-

tively coupled to two 6AQ5 pentode connected

s-

t

o a 3

EQUIVALENT SINE WAVE WATTS

Figure 24

INTERMODULATION CURVE FOR

"BABY HI -Fl" AS MEASURED ON

HEATHKIT INTERMODULATION

ANALYZER.

output tubes. The feedback loop is run from

the secondary of the output transformer to the

cathode of the input section of the phase in-

verter.

The power supply of the "Baby Hi -Fi" con-

sists of a 6X5 -GT rectifier and a capacitor in-

put filter. A second R -C filter section is used

to smooth the d -c voltage applied to the

12AU7 tubes. A cathode -type rectifier is used

in preference to the usual filament type to pre-

vent voltage surges during the warm -up period

of the other cathode -type tubes.

Amplifier The complete amplifier is

Construction built upon a small "amplifier

foundation" chassis and cover

measuring 5 "x7 "x6" (Bud CA- 1754). Height

of the amplifier including dust cover is 6 ".

The power transformer (T,) and output trans-

former (Tl) are placed in the rear corners of

the chassis, with the'6X5 -GT rectifier socket

placed between them. The small filter choke

(CH,) is mounted to the wall of the chassis

and may be seen in the under -chassis photo-

graph of figure 23. The four audio tubes are

placed in a row across the front of the chassis.

Viewed from the front, the 12AU7 tubes are

to the left, and the 6AQ5 tubes are to the right.

The three section filter capacitor (C,A, B, C)

is a chassis mounting unit, and is placed be-

tween the rectifier tube and the four audio

tubes. Since the chassis is painted, it is im-

portant that good grounding points be made at

each tube socket. The paint is cleared away

www.americanradiohistory.com

HANDBOOK "Baby Hi -Fi" 145

Figure 25

TYPICAL

INTERMODULATION

TEST OF AUDIO

AMPLIFIER.

Audio tones of two fre-

quencies are applied to in-

put of amplifier under testi

and amplitude of "sum" or

"difference" frequency is

measured, providing relative

inter -modulation figure.

beneath the socket bolt heads, and lock nuts

are used beneath the socket retaining nuts to

insure a good ground connection. All ground

leads of the first 12AU7 tube are returned to

the socket, whereas all grounds for the rest of

the circuit are returned to a ground lug of

filter capacitor G.

Since the input level to the amplifier is of

the order of one -half volt, the problem of

chassis ground currents and hum is not so

prevalent, as is the case with a high gain input

stage. Phonograph -type coaxial receptacles are

mounted on the rear apron of the chassis, serv-

ing as the input and output connections. The

four panel controls (bass boost, treble boost,

volume, and a -c on) are spaced equidistant

across the front of the chassis.

Amplifier

Wiring The filament wiring should be

done first. The center -tap of the

filament winding is grounded to

a lug of the 6X5 -GT socket ring, and the 6.3

volt leads from the transformer are attached

to pins 2 and 7 of the same socket. A twisted

pair of wires run from the rectifier socket to

the right -hand 6AQ5 socket (figure 23). The

filament leads then proceed to the next 6AQ5

socket and then to the two 12AU7 sockets in

turn. The 12AU7 preamplifier stage is wired next.

A two terminal phenolic tie -point strip is

mounted to the rear of the chassis, holding the

12K decoupling resistor and the positive lead

of the 10 µfd., 450 -volt filter capacitor. All

B -plus leads are run to this point. Most of the

components of the bass and treble boost system

may be mounted between the tube socket ter-

minals and the terminals of the two potentio-

meters. The feedback resistor R, is mounted

between the terminal of the coaxial output

connector and a phenolic tie -point strip placed

beneath an adjacent socket bolt.

When the wiring has been completed and

checked, the amplifier should be turned on,

and the various voltages compared with the

values given on the schematic. It is important

that the polarity of the feedback loop is correct.

The easiest way to reverse the feedback polarity

is to cross -connect the two plate leads of the

6AQ5 tubes. If the feedback polarization is

incorrect, the amplifier will oscillate at a super-

sonic frequency and the reproduced signal will

sound fuzzy to the ear. The correct connection

may be determined with the aid of an oscillo-

scope, as the oscillation will be easily found.

The builder might experiment with different

values of feedback resistor RI, especially if a

speaker of different impedance is employed.

Increasing the value of R1 will decrease the

degree of feedback. For an 8 -ohm speaker, Rt

should be decreased in value to maintain the

same amount of feedback.

This amplifier was used in conjunction with

a General Electric S-1201A 12 -inch speaker

mounted in an Electro -Voice KD6 Aristocrat

speaker enclosure which was constructed from

www.americanradiohistory.com

146 High Fidelity Techniques THE RADIO

a kit. The reproduction was extremely smooth,

with good balance of bass and treble.

7 -6 A Transformerless

25 Watt Music

Amplifier

Because the output transformer is usually

the weakest link in both frequency response

and power output of an audio amplifier,

several methods have been used to drive loud-

speakers directly from the output tubes. These

have either used non -conventional high -impe-

dance loudspeakers, have been very inefficient,

or have had low power output capabilities.

The amplifier described in this section

drives a conventional 16 ohm loudspeaker

with normal class A amplifier efficiency, and

supplies 25 watts of low distortion output

throughout the audio range (figure 26). The

amplifier requires an input signal of approxi-

mately one volt to drive it to maximum out-

put. The unit attains its high performance

through the use of 40 db of inverse feedback.

Because of the relatively simple power supply

requirements and the absence of expensive

power and audio transformers it is more eco-

nomical than conventional amplifiers of similar

performance.

The Amplifier The output stage of this un-

Circuit usual amplifier is the single

ended, push -pull type as

shown in figure 27. The quiescent current is

equal in both tubes with no d.c. current flow-

ing through the speaker load. The absence of

an output transformer allows 40 db of feed-

back to be app'ied by connecting the voice

coil of the speaker directly to the cathode of

the 12ÁT7 phase- inverter driver. In addition

to its distortion reducing characteristic, the

application of feedback serves to reduce the

hum voltage which might otherwise be present.

As the gain within the feedback loop is essen-

tially unity, an additional voltage amplifier

is used (with separate feedback) to build the

input voltage up to the voice coil level.

The power supply is a double half -wave se-

lenium rectifier circuit developing +140 volts

and -140 volts with respect to ground. The

supply uses large filter capacitors, and no

Figure 26.

25 -WATT

TRANS FORMERLESS

AMPLIFIER PRO-

VIDES ULTIMATE IN

LISTENING

PLEASURE FOR THE

"GOLDEN EAR."

Amplifier employs three

6082 triode tubes in

single-ended push-pull

configuration for maxi-

mum fidelity. The output

tubes are placed across

right end of chassis.

65N7 phase inverter is at

rear, center; and low

level stages are at the

front, center. To the left

are the power supply fil-

ter capacitors. In this

particular amplifier, the

40 ohm, 20 watt fila-

ment dropping resistor

was eliminated and a

small iron core reactor

was used in its place

(left, rear corner of the

chassis). Across front of

chassis are (I. to r.):

power switch, input lack,

and output stage balanc-

ing potentiometer.

www.americanradiohistory.com

HANDBOOK Transformerless Amplifier 147

INPUT

. IM

12 AT 7

L 40

+ 150

T

2 A T7

40 ± I 40

150 + + 150

loo 150

K 47K K a 0.1

56K 0.1

4- 40

150

5 I M

1.8

10K K 1K 1M

12 AT7

9 4,5

9

12AT7

115V1.,

39K 10 K

1.2

M

56K

6SN7 -GT

5M

* Sl

680

1.5M

M 4

5 56 K

;ff-6

+

2 6K*

5

4

8K

6082 6082 6082

100

4

100

00

4

2

00

1611

OUTPUT

+250 VOLTS

+,o VOLTS

65147 4062 6082 6062 40

S6W

40/150

11Hf--`

SR i 1.5Kz

7 6 7 6 7 6 7

SI

n r

NOTE: ONE SIDE OP LINE GROUNDED

70 CHASSIS.

1. ALL RESISTORS I -WATT UNLESS

OTHERWISE NOTED.

2. RESISTORS MARRED R ARE

MATCHED PA /RS.

3. CAPACITOR VALUES GIVEN IN //FD.

4.SRI. SR 4 75 MA., I50 VOLT.

5.SRz, SR3 =500MA., 150 VOLT.

A.

2.5 K

10 fOWW

W

10

560

roo o

SR2 +aoo

ó

P300500

S RS ï 5S

I

*140 VOLTS

-140 VOLTS

15

150 SR4 8.29 10K 27 K 12K

Figure 27.

SCHEMATIC OF 25 -WATT MUSIC AMPLIFIER

I I

O%

of

Qx

A1° A % T %

REACTANCE

NETWORKS

160 0.5

extra filtering is required. The output impe-

dance of the supply is extremely low. To ob-

tain higher voltage for the low level stages,

additional selenium rectifiers are used in a

voltage- adding configuration to obtain +250

and -250 volts.

Circuit Details The complete schematic of

this amplifier is given in

figure 27. Three type 6082 double triodes are

employed in the output stage. These are 26.5

volt versions of the popular 6AS7G. These

tubes are capable of 700 milliamperes of peak

plate current per triode section at the plate

voltage employed. The choice of the 6082 is

an economy measure to allow the use of a

series heater string. These tubes cut off at

about -70 volts, but for this class of service

the bias is held at -60 volts. A bias control

is provided for one set of tubes so that the

d.c. current flowing in the tubes may be

equalized, and to insure that no d.c. current

flows through the speaker voice coil.

The type 6082 tube is not rated for use

with fixed bias unless a limiting resistor is

added in either the plate or the cathode cir-

cuit. Although this circuit does not use such

resistors, their omission is feasible only be-

cause the tubes are used under quiescent con-

ditions well below maximum ratings. With

tubes of this type, it may be expected that the

average current through the voice coil will

drift with time but the presence of this un-

www.americanradiohistory.com

148 High Fidelity Techniques THE RADIO

balance current will generally be of little

concern. In any event, the circuit has been

designed so that the output stage can be con-

veniently rebalanced.

The Voltage The low level stages are all

Amplifier operated Class A with con-

ventional circuitry. A separate

driver is needed for each side of the output

circuit, as insufficient output is obtained from

the phase inverter to drive the output tubes

directly. One side of the phase inverter has a

larger load than the other, since the input to

the lower group of output tubes has the

speaker impedance in the cathode. This causes

degeneration and necessitates higher input

Figure 28.

UNDER -CHASSIS VIEW OF

TRANSFORMERLESS AMPLIFIER

Output tube sockets are at left, with power

supply components at right. Components of

preamplifier stages are grouped about the

center sockets, mounted between socket pins

and phenolic tie -point strips. Line fuse is

mounted on rear apron of chassis.

voltage than the upper group.

In the first voltage amplifier, bias is ob-

tained from unbypassed cathode resistors since

the loss of gain can easily be tolerated. The

phase inverter- driver, however, has fixed bias

applied to the grid from the -140 volt

supply, since maximum gain is desired within

the main feedback loop.

The Power Supply The high current power

supply uses 300 µµfd.

filter capacitors and 5 ohm protective resis-

tors. R -C decoupling is used to minimize hum

in the low level audio stages. As with all

"power- transformerless" equipment, care must

be taken when connecting this amplifier to

other pieces of equipment to ensure that the

grounded side of the power line is connected

to the chassis. This may be achieved by the

use of a polarized line plug, or a small isola-

tion transformer may be employed.

The Equalizing As would be expected, 40

Circuit db of feedback can only be

applied within a loop hav-

ing a minimum of phase shift or circuit in-

stability will result. Since the loudspeaker

O www.americanradiohistory.com

HANDBOOK Transformerless Amplifier 149

o

-10

20

.0

0.8

0 6

0.

0.2

10 100 1000 10 KC 100 KC 1000 KC o 10 15 20

Figure 29.

FREQUENCY POWER OUTPUT (wnrrs)

A- Overall frequency response of amplifier 0

B- Distortion versus power output of amplifier

25

impedance becomes inductive above the audio

range it causes an increase in phase shift and

loop gain. To avoid instability an impedance

can be shunted across the voice coil to prevent

the output reactance from rising at the higher

audio frequencies. Three networks that have

been used successfully for this purpose are

shown in figure 27. The 180 ohm res for

merely limits the maximum impedance of the

output system and thus preven , excessive

feedback. The 0.5 pfd. capacitor places a low

impedance across the inductive load which is

effective at the higher audio frequencies. The

series 16 ohm resistor and 0.01 pfd. capacitor

places a resistance across the speaker at the

higher frequencies and an open circuit at the

lower frequencies. This serves to provide con-

stant impedance and feedback over the fre-

quency range of the amplifier.

The balance adjustment for zero d.c. cur-

rent through the speaker voice coil can be

made with a milliammeter in series with the

coil, or by measuring the voltage across the

coil with a sensitive voltmeter.

Amplifier The amplifier is built upon

Construction an aluminum chassis measur-

ing 8" x 10" x 2 ". Perfor-

ated end pieces and 1/4 -inch holes drilled

around the 6082 tube sockets insure adequate

ventilation. Layout of the major components

is shown in figure 26, and placement of the

under -chassis components is shown in figure

28. As no a.c. power transformer is used,

ground currents are of small concern, and the

ground bus wiring technique need not be em-

ployed. In its place, a tinned copper wire is

run between the various chassis ground points.

Ground connections may now be made to the

socket grounding lugs, or to terminal strip

ground points. A.c. filament and power leads

are twisted wherever possible, and are run

around the outer edges of the chassis.

Point -to -point wiring technique is used,

with small capacitors and resistors mounted

to socket pins or to phenolic tie -point strips

placed near the sockets. The small silicon rec-

tifiers are mounted to tie -point strips placed

near the upright filter capacitors.

Several of the filter capacitors do not have

their negative terminal at ground potential.

It is therefore necessary to mount the capacitor

on a phenolic plate and to slip a fiber insu-

lating jacket over the metal shell.

Amplifier The frequency response of

Performance the amplifier is flat within

one db from 10 cycles to over

100 kilocycles. Since R -C coupled circuits are

used throughout, there is no serious limitation

on frequency response, and the response is

down only 4 db at 250,000 cycles. The inter -

stage coupling networks limit the low fre-

quency response below 10 cycles.

Harmonic distortion and intermodulation at

full rated output are exceptionally low and vir-

tually independent of frequency. The ability

to deliver 25 watts at 20 cycles and below

with negligible distortion is practically impos-

sible in a transformer -type circuit of similar

mid -frequency power rating. Square wave re-

sponse of the amplifier as measured between

20 cycles and 50 kilocycles is extremely good.

www.americanradiohistory.com

LO-russ T seCT,oM LOIS r4ss i/ seCT,OM

FREQUENCY SCALE

LOW -PASS HIGH -PASS

.00 25.0

90

80 20.0

70

60 15.0

50

40 10.0

9.0

e.0

7.0

6.0

20 5.0

4.0

3.0

--IilIF- te

j-----;

M1oM-491 T seCT,ON 011411 ..,f 7T 5e0110M

VALUES

L C LOAD RESISTANCE

5 .5

6 6

r .7

e .6

9 .9

10 1.0

4.s

O 2.0

3.0

1000 10000

1100 9000

200 e000

1300

1400 Z- 1000

1500

1600 6000

1700 - W

1600 -r -1

W 1900 u

t 2000 5000 n

u M

W V Z 2500

< r u

O Z

LC

40 .0 Ñ 2

50 5.0 u 4000

3000

60 e.0 1- h.

70 I 7.0 W J

e0 I 8.0 =

90 r

9.0

100 10.0 3

ta Ñ 1000 -- E-

A -5.- 1000

150 --i- 15.0

00 20.0

300 30.0

10 2.5 10000 -- 1000

4000

3000

5000_4- - 2000

C 1000

8000 -

J9000

Courler, rrcdic e.0,9 9ni, n ing CO.

FILTER DESIGN CHART

For both Pi -type and T -type Sections

To find L, connect cut -off frequency on left -hand scale (using left -side scale for low -pass and right -

side scale for high -pass) with load on left -hand side of right -hand scale by means of a straight -edge.

Then read the value of L from the point where the edge intersects the left side of the center scale. Read-

ings are in henries for frequencies in cycles per second.

To find C, connect cut -off frequency on left -hand scale (using left -side scale for low -pass and right -

side scale for high pass) with the load on the right -hand side of the right -hand scale. Then read the

value of C from the point where the straightedge cuts the right side of the center scale. Readings are

in microfarads for frequencies in cycles per d.

For frequencies in kilocycles, C is expressed in thousands of micromicrofarads, L is expressed in

mlllihenries. For frequencies in megacycles, L is expressed in microhenries and C is expressed in microml-

cro farads.

For each tenfold increase In the value of load resistance multiply L by 10 and divide C by IO.

For each ten fold decrease in frequency multiply L by 10 end multiply C by 10.

150

www.americanradiohistory.com

CHAPTER EIGHT

Radio Frequency

Vacuum Tube Amplifiers

TUNED RF VACUUM TUBE AMPLIFIERS

Tuned r -f voltage amplifiers are used in re-

ceivers for the amplification of the incoming

r -f signal and for the amplification of inter-

mediate frequency signals after the incoming

frequency has been converted to the intermed-

iate frequency by the mixer stage. Signal fre-

quency stages are normally called tuned r -f

amplifiers and intermediate -frequency stages

are called i -f amplifiers. Both tuned r -f and

i -f amplifiers are operated Class A and nor-

mally operate at signal levels from a fraction

of a microvolt to amplitudes as high as 10 to

50 volts at the plate of the last i -f stage in a

receiver.

8 -1 Grid Circuit

Considerations

Since the full amplification of a receiver fol-

lows the first tuned circuit, the operating con-

ditions existing in that circuit and in its cou-

pling to the antenna on one side and to the

grid of the first amplifier stage on the other

are of greatest importance in determining the

signal -to -noise ratio of the receiver on weak

signals.

First Tuned

Circuit It is obvious that the highest

ratio of signal -to -noise be im-

pressed on the grid of the first

r -f amplifier tube. Attaining the optimum ratio

is a complex problem since noise will be gen-

erated in the antenna due to its equivalent

radiation resistance (this noise is in addition

to any noise of atmospheric origin) and in the

151

first tuned circuit due to its equivalent cou-

pled resistance at resonance. The noise volt-

age generated due to antenna radiation resist-

ance and to equivalent tuned circuit resistance

is similar to that generated in a resistor due

to thermal agitation and is expressed by the

following equation:

En' = 4kTRAf

Where: E° = r -m -s value of noise voltage over

the interval .1f

k = Boltzman's constant = 1374

X 10-22 joule per °K.

T = Absolute temperature °K.

R = Resistive component of imped-

ance across which thermal noise

is developed.

1f = Frequency band across which

voltage is measured.

In the above equation \f is essentially the

frequency band passed by the intermediate fre-

quency amplifier of the receiver under consid-

eration. This equation can be greatly simpli-

fied for the conditions normally encountered

in communications work. If we assume the fol-

lowing conditions: T = 300° K or 27° C or

80.5° F, room temperature; 1f = 8000 cycles

(the average pass band of a communications

receiver or speech amplifier) the equation re-

duces to: Et.m.s. = 0.0115 microvolts. Ac-

cordingly, the thermal -agitation voltage ap-

pearing in the center of half -wave antenna (as-

suming effective temperature to be 300° K)

having a radiation resistance of 73 ohms is

www.americanradiohistory.com

152 R -F Vacuum Tube Amplifiers THE RADIO

approximately 0.096 microvolts. Also, the ther-

mal agitation voltage appearing across a 500,-

000 -ohm grid resistor in the first stage of a

speech amplifier is approximately 8 microvolts

under the conditions cited above. Further, the

voltage due to thermal agitation being im-

pressed on the grid of the first r -f stage in a

receiver by a first tuned circuit whose reson-

ant resistance is 50,000 ohms is approximately

2.5 microvolts. Suffice to say, however, that

the value of thermal agitation voltage appear-

ing across the first tuned circuit when the an-

tenna is properly coupled to this circuit will

be very much less than this value.

It is common practice to match the imped-

ance of the antenna transmission line to the

input impedance of the grid of the first r -f am-

plifier stage in a receiver. This is the condi-

tion of antenna coupling which gives maximum

gain in the receiver. However, when u -h -f tubes

such as acorns and miniatures are used at fre-

quencies somewhat less than their maximum

capabilities, a significant improvement in sig-

nal -to -noise ratio can be attained by increas-

ing the coupling between the antenna and first

tuned circuit to a value greater than that which

gives greatest signal amplitude out of the re-

ceiver. In other words, in the 10, 6, and 2 me-

ter bands it is possible to attain somewhat im-

proved signal -to -noise ratio by increasing an-

tenna coupling to the point where the gain of

the receiver is slightly reduced.

It is always possible, in addition, to obtain

improved signal -to -noise ratio in a v -h -f re-

ceiver through the use of tubes which have

improved input impedance characteristics at

the frequency in question over conventional

types.

Noise Factor The limiting condition for sen-

sitivity in any receiver is the

thermal noise generated in the antenna and in

the first tuned circuit. However, with proper

coupling between the antenna and the grid of

the tube, through the first tuned circuit, the

noise contribution of the first tuned circuit

can be made quite small. Unfortunately, though,

the major noise contribution in a properly de-

signed receiver is that of the first tube. The

noise contribution due to electron flow and

due to losses in the tube can be lumped into

an equivalent value of resistance which, if

placed in the grid circuit of a perfect tube hav-

ing the same gain but no noise would give the

same noise voltage output in the plate load.

The equivalent noise resistance of tubes such

as the 6SK7, 6SG7, etc., runs from 5000 to

10,000 ohms. Very high Gm tubes such as the

6AC7 and 6AK5 have equivalent noise resist-

ances as low as 700 to 1500 ohms. The lower

the value of equivalent noise resistance, the

lower will be the noise output under a fixed

set of conditions.

The equivalent noise resistance of a tube

must not be confused with the actual input

loading resistance of a tube. For highest sig-

nal -to -noise ratio in an amplifier the input

loading resistance should be as high as possi-

ble so that the amount of voltage that can be

developed from grid to ground by the antenna

energy will be as high as possible. The equi-

valent noise resistance should be as low as

possible so that the noise generated by this

resistance will be lower than that attributable

to the antenna and first tuned circuit, and the

losses in the first tuned circuit should be as

low as possible.

The absolute sensitivity of receivers has

been designated in recent years in government

and commercial work by an arbitrary dimension-

less number known as "noise factor" or N.

The noise factor is the ratio of noise output

of a "perfect" receiver having a given amount

of gain with a dummy antenna matched to its

input, to the noise output of the receiver under

measurement having the same amount of gain

with the dummy antenna matched to its input.

Although a perfect receiver is not a physically

realizable thing, the noise factor of a receiver

under measurement can be determined by cal-

culation from the amount of additional noise

(from a temperature -limited diode or other cali-

brated noise generator) required to increase

the noise power output of a receiver by a pre-

determined amount.

Tube Input As has been mentioned in a pre -

Loading vious paragraph, greatest gain

in a receiver is obtained when

the antenna is matched, through the r -f cou-

pling transformer, to the input resistance of

the r -f tube. However, the higher the ratio of

tube input resistance to equivalent noise re-

sistance of the tube the higher will be the sig-

nal -to -noise ratio of the stage -and of course,

the better will be the noise factor of the over-

all receiver. The input resistance of a tube

is very high at frequencies in the broadcast

band and gradually decreases as the frequency

increases. Tube input resistance on conven-

tional tube types begins to become an import-

ant factor at frequencies of about 25 Mc. and

above. At frequencies above about 100 Mc. the

use of conventional tube types becomes im-

practicable since the input resistance of the

tube has become so much lower than the equi-

valent noise resistance that it is impossible

to attain reasonable signal -to -noise ratio on

any but very strong signals. Hence, special

v -h -f tube types such as the 6AK5, 6ÁG5, and

6CB6 must be used.

The lowering of the effective input resist-

www.americanradiohistory.com

HANDBOOK R -F Amplifiers 153

ance of a vacuum tube at higher frequencies

is brought about by a number of factors. The

first, and most obvious, is the fact that the

dielectric loss in the internal insulators, and

in the base and press of the tube increases

with frequency. The second factor is due to

the fact that a finite time is required for an

electron to move from the space charge in the

vicinity of the cathode, pass between the grid

wires, and travel on to the plate. The fact that

the electrostatic effect of the grid on the mov-

ing electron acts over an appreciable portion

of a cycle at these high frequencies causes a

current flow in the grid circuit which appears

to the input circuit feeding the grid as a re-

sistance. The decrease in input resistance of

a tube due to electron transit time varies as

the square of the frequency. The undesirable

effects of transit time can be reduced in cer-

tain cases by the use of higher plate voltages.

Transit time varies inversely as the square

root of the applied plate voltage.

Cathode lead inductance is an additional

cause of reduced input resistance at high fre-

quencies. This effect has been reduced in cer-

tain tubes such as the 6S117 and the 6AK5 by

providing two cathode leads on the tube base.

One cathode lead should be connected to the

input circuit of the tube and the other lead

should be connected to the by -pass capacitor

for the plate return of the tube.

The reader is referred to the Radiation Labo-

ratory Series, Volume 23: "Microwave Receiv-

ers" (McGraw -Hill, publishers) for additional

information on noise factor and input loading

of vacuum tubes.

8 -2 Plate- Circuit

Considerations

Noise is generated in a vacuum tube by the

fact that the current flow within the tube is not

a smooth flow but rather is made up of the con-

tinuous arrival of particles (electrons) at a

very high rate. This shot effect is a source of

noise in the tube, but its effect is referred

back to the grid circuit of the tube since it is

included in the equivalent noise resistance

discussed in the preceding paragraphs.

Plate Circuit For the purpose of this section,

Coupling it will be considered that the

function of the plate load cir-

cuit of a tuned vacuum -tube amplifier is to de-

liver energy to the next stage with the greatest

efficiency over the required band of frequen-

cies. Figure 1 shows three methods of inter -

stage coupling for tuned r -f voltage amplifiers.

In figure IA omega (w) is 2n times the reso-

nant frequency of the circuit in the plate of

OA AMPLIFICATION AT RESONANCE (APPROX.) =GMWLQ

OB AMPLIFICATION AT RESONANCE (APPROX ) =GWMQ

K2t 1

QP S

WHERE 1. PRI. ANO SEC. RESONANT AT SAME FREQUENCY

2 K IS COEFFICIENT OF COUPLING

IF FRI. AND SEC. Q ARE APPROXIMATELY THE SAME.

TOTAL BANDWIDTH

CENTER FREQUENCY 1.2 K

MAXIMUM AMPLITUDE OCCURS AT CRITICAL COUPLING -

WHEN K- QP

Figure 1

Gain equations for pentode r -f amplifier

stages operating into a tuned load

the amplifier tube, and L and Q are the induct-

ance and Q of the inductor L. In figure 1B the

notation is the same and M is the mutual in-

ductance between the primary coil and the sec-

ondary coil. In figure 1C the notation is again

the same and k is the coefficient of coupling

between the two tuned circuits. As the co-

efficient of coupling between the circuits is

increased the bandwidth becomes greater but

the response over the band becomes progres-

sively more double -humped. The response over

the band is the most flat when the Q's of pri-

mary and secondary are approximately the same

and the value of each Q is equal to 1.75/k.

www.americanradiohistory.com

154 R -F Vacuum Tube Amplifiers THE RADIO

Variable -Mu Tubes It is common practice to

in R -F Stages control the gain of a suc-

cession of r -f or i -f am-

plifier stages by varying the average bias on

their control grids. H)wever, as the bias is

raised above the operating value on a conven-

tional sharp- cutoff tube the tube becomes in-

creasingly non -linear in operation as cutoff of

plate current is approached. The effect of such

non -linearity is to cause cross modulation be-

tween strong signals which appear on the grid

of the tube. When a tube operating in such a

manner is in one of the first stages of a re-

ceiver a number of signals are appearing on its

grid simultaneously and cross modulation be-

tween them will take place. The result of this

effect is to produce a large number of spurious

signals in the output of the receiver -in most

cases these signals will carry the modulation

of both the carriers which have been cross

modulated to produce the spurious signal.

The undesirable effect of cross modulation

can be eliminated in most cases and greatly

reduced in the balance through the use of a

variable -mu tube in all stages which have a -v -c

voltage or other large negative bias applied to

their grids. The variable -mu tube has a char-

acteristic which causes the cutoff of plate cur-

rent to be gradual with an increase in grid

bias, and the reduction in plate current is ac-

companied by a decrease in the effective am-

plification factor of the tube. Variable -mu tubes

ordinarily have somewhat reduced Gm as com-

pared to a sharp- cutoff tube of the same group.

Hence the sharp- cutoff tube will perform best

in stages to which a -v -c voltage is not applied.

RADIO- FREQUENCY POWER AMPLIFIERS

All modern transmitters in the medium -fre-

quency range and an increasing percentage of

those in the v -h -f and u -h -f ranges consist of

a comparatively low -level source of radio-fre-

quency energy which is multiplied in frequency

and successively amplified to the desired power

level. Microwave transmitters are still predom-

inately of the self- excited oscillator type, but

when it is possible to use r -f amplifiers in

s -h -f transmitters the flexibility of their ap-

plication will be increased. The following por-

tion of this chapter will be devoted, however,

to the method of operation and calculation of

operating characteristics of r -f power ampli-

fiers for operation in the range of approximate-

ly 3.5 to 500 Mc.

8 -3 Class C R -F

Power Amplifiers

The majority of r -f power amplifiers fall into

the Class C category since such stages can

be made to give the best plate circuit efficien-

cy of any present type of vacuum -tube ampli-

fier. Hence, the cost of tubes for such a stage

and the cost of the power to supply that stage

is least for any given power output. Neverthe-

less, the Class C amplifier gives less power

gain than either a Class A or Class B ampli-

fier under similar conditions since the grid of

a Class C stage must be driven highly posi-

tive over the portion of the cycle of the excit-

ing wave when the plate voltage on the ampli-

fier is low, and must be at a large negative

potential over a large portion of the cycle so

that no plate current will flow except when

plate voltage is very low. This, in fact, is the

fundamental reason why the plate circuit effi-

ciency of a Class C amplifier stage can be

made high -plate current is cut off at all times

except when the plate -to- cathode voltage drop

across the tube is at its lowest value. Class

C amplifiers almost invariably operate into a

tuned tank circuit as a load, and as a result

are used as amplifiers of a single frequency

or of a comparatively narrow band of frequen-

cies.

Relationships in Figure 2 shows the relation -

Class C Stage ships between the various

voltages and currents over

one cycle of the exciting grid voltage for a

Class C amplifier stage. The notation given in

figure 2 and in the discussion to follow is the

same as given at the first of Chapter Six un-

der "Symbols for Vacuum -Tube Parameters."

The various manufacturers of vacuum tubes

publish booklets listing in adequate detail al-

ternative Class C operating conditions for the

tubes which they manufacture. In addition,

operating condition sheets for any particular

type of vacuum tube are available for the ask-

ing from the different vacuum -tube manufac-

turers. It is, nevertheless, often desirable to

determine optimum operating conditions for a

tube under a particular set of circumstances.

To assist in such calculations the following

paragraphs are devoted to a method of calcu-

lating Class C operating conditions which is

moderately simple and yet sufficiently accu-

rate for all practical purposes.

www.americanradiohistory.com

HANDBOOK Class C R -F Amplifiers 155

EPM

EBB

P

PLATE

VOLTAGE

O1._.L.L- r

eP --'-- -1-- ---

I I 1

PEAK

PLATE

CURRENT

_ I

--I-- If--t- - -I -

I19P-.1.-9P+11 I I

I

I FUNDAMENTAL COMPONENT

I 1 OF PLATE CURRENT

7- -i I

I II I

I II

I 'I

t VI I I

11 I I I

IG MAS. I I I I I GRID I

I - V - - FI-CURRENT

1 r t1 -- 1------ I-

I III I I11 I

III III I

I

11 111 I

e III I

11i I

EGM 0i- d { : Ecc -1-,,,* I -

'-Ieca-1--

III GRID

III VOLTAGE

11-- -I

G I

Figure 2

Instantaneous electrode and tank circuit

voltages and currents for a Class C r -f power

amplifier

Calculation of Class Although Class C op-

C Amplifier Operating crating conditions can

Characteristics be determined with the

aid of the more conven-

tional grid voltage -plate current operating

curves, the calculation is considerably sim-

plified if the alternative "constant- current

curve" of the tube in question is used. This is

true since the operating line of a Class C am-

plifier is a straight line on a set of constant-

current curves. A set of constant -current curves

on the 250TH tube with a sample load line

drawn thereon is shown in figure 5.

In calculating and predicting the operation

of a vacuum tube as a Class C radio -frequency

amplifier, the considerations which determine

the operating conditions are plate efficiency,

power output required, maximum allowable

plate and grid dissipation, maximum allowable

plate voltage and maximum allowable plate

current. The values chosen for these factors

will depend both upon the demands of a par-

ticular application and upon the tube chosen.

The plate and grid currents of a Class C

amplifier tube are periodic pulses, the dura-

tions of which are always less than 180 de-

grees. For this reason the average grid cur-

rent, average plate current, power output, driv-

ing power, etc., cannot be directly calculated

but must be determined by a Fourier analysis

from points selected at proper intervals along

the line of operation as plotted upon the con-

stant- current characteristics. This may be done

either analytically or graphically. While the

Fourier analysis has the advantage of accu-

racy, it also has the disadvantage of being

tedious and involved.

The approximate analysis which follows

has proved to be sufficiently accurate for most

applications. This type of analysis also has

the advantage of giving the desired informa-

tion at the first trial. The system is direct in

giving the desired information since the im-

portant factors, power output, plate efficiency,

and plate voltage are arbitrarily selected at

the beginning.

Method of The first step in the method to

Calculation be described is to determine the

power which must be delivered

by the Class C amplifier. In making this deter-

mination it is well to remember that ordinarily

from 5 to 10 per cent of the power delivered

by the amplifier tube or tubes will be lost in

well- designed tank and coupling circuits at

frequencies below 20 Mc. Above 20 Mc. the

tank and circuit losses are ordinarily some-

what above 10 per cent.

The plate power input necessary to produce

the desired output is determined by the plate

efficiency: Pin = Pout/Np.

For most applications tt is desirable to oper-

ate at the highest practicable efficiency. High -

efficiency operation usually requires less ex-

pensive tubes and power supplies, and the

www.americanradiohistory.com

156 R -F Vacuum Tube Amplifiers THE RADIO

7.0

.0

40

30 e tt

RATIO iáu

,.

Figure 3

Relationship between the peak value of the

fundamental component of the tube plate cur-

rent, and average plate current; as compared

to the ratio of the instantaneous peak value

of tube plate current, and average plate

current

t

O 7.0

.0

3.0 -10

MENE

E i' \ \1 E

\ \ E

\\ agi E E EE

\EE

u EEI \

=\ EEE IMEN

I -20

RATIO

.1 -10

Figure 4

Relationship between the ratio of the peak

value of the fundamental component of the

grid excitation voltage, and the overage grid

bias; as compared to the ratio between in-

stantaneous peak grid current and average

grid current

amount of artificial cooling required is fre-

quently less than for low- efficiency operation.

On the other hand, high- efficiency operation

usually requires more driving power and in-

volves the use of higher plate voltages and

higher peak tube voltages. The better types

of triodes will ordinarily operate at a plate

efficiency of 75 to 85 per cent at the highest

rated plate voltage, and at a plate efficiency

of 65 to 75 per cent at intermediate values of

plate voltage.

The first determining factor in selecting a

tube or tubes for a particular application is

the amount of plate dissipation which will be

required of the stage. The total plate dissipa-

tion rating for the tube or tubes to be used in

the stage must be equal to or greater than that

calculated from: Pp = Pin - Pout.

After selecting a tube or tubes to meet the

power output and plate dissipation require-

ments it becomes necessary to determine from

the tube characteristics whether the tube se-

lected is capable of the desired operation and,

if so, to determine the driving power, grid

bias, and grid dissipation.

The complete procedure necessary to deter-

mine a set of Class C amplifier operating con-

ditions is given in the following steps:

1. Select the plate voltage, power output,

and efficiency.

2. Determine plate input from: Pin =

Pout/Np.

3 Determine plate dissipation from:

Pp= Pin - Pout Pp must not exceed

maximum rated plate dissipation for tube

or tubes selected.

4. Determine average plate current from:

lb = Pin /Ebb

5. Determine approximate ;p.a. from:

tpmax = 4.9 lb for Np = 0.85

tpmas = 4.5 lb for Np = 0.80

tpmas = 4.0 'b for N = 0.75

tpmax= 3.51b for Np =0.70

6. Locate the point on constant -current

characteristics where the constant plate

current line corresponding to the ap-

proximate ipmax determined in step 5

crosses the line of equal plate and grid

voltages (diode line). Read epmin at this

point. In a few cases the lines of con-

stant plate current will inflect sharply

upward before reaching the diode line.

In these cases epmin should not be read

at the diode line but at the point where

the plate current line intersects a line

drawn from the origin through these

points of inflection.

www.americanradiohistory.com

HANDBOOK Constant Current Calculations 157

FIRST TRIAL POINT FINAL POINT

EIMAC 250TH

CONSTANT CURRENT

CHARACTERISTICS

N -s, .... sa:

pP: .

Om r-;pze ,

o

ó

_

Ti -_-.

ERE 7b.

, ....... .........

00

EGO= - 240

MOO XOD x00

LOAD LINE PLATE VOLTAGE -VOLTS Ebb =4-3500

FIGURE 5

Active portion of the operating load line for an Eimoc 250TH Class C r -f power amplifier,

showing first trial point and the final operating point

7. Calculate Epm from: Epm = Ebb - epmin

8. Calculate the ratio Ipm /lb from:

1pm 2 Np Ebb

lb Epm

9. From the ratio of Ipm /Ib calculated in

step 8 determine the ratio ipmax /Ib from

figure 3.

10. Calculate a new value for ipmax from

the ratio found in step 9.

tpm as = (ratio from step 9) lb

11. Read egmp and igmax from the constant -

current characteristics for the values of

epmin and ipmax determined in steps 6

and 10.

12. Calculate the cosine of one -half the

angle of plate current flow from:

cos f3p =2.32( Ipm

Ib - 1.57)

13. Calculate the grid bias voltage from:

1

Ecc - X

1 - cos Op

Epm

\ µ - egmp)

Fos Op

for triodes.

Ecc 1

X

1- cos 6p L

Ebb

fi

En,

camp cos O - J

1112

for tetrodes, where tt is the grid- screen

amplification factor, and Ec2 is the d -c

screen voltage.

14. Calculate the peak fundamental grid ex-

citation voltage from:

Earn = egmp - Ecc

15. Calculate the ratio Egm /Ecc for the val-

www.americanradiohistory.com

158 R -F Vacuum Tube Amplifiers THE RADIO

ues of Ecc and Egm found in steps 13

and 14.

16. Read igmax /Ic from figure 4 for the ratio

Egm /Ecc found in step 15.

17. Calculate the average grid current from

the ratio found in step 16, and the value

of igmax found in step 11:

tgmax

IC Ratio from step 16

18. Calculate approximate grid driving pow-

er from:

Pd = 0.9 Egmlc

19. Calculate grid dissipation from:

Pa = Pa + Eccic

Pg must not exceed the maximum rated

grid dissipation for the tube selected.

Sample A typical example of a Class C

Calculation amplifier calculation is shown

in the example below. Reference

is made to figures 3, 4 and 5 in the calcula-

tion.

1. Desired power output -800 watts.

2. Desired plate voltage -3500 volts.

Desired plate efficiency -80 per cent

(Np = 0.80)

Pin = 800 /0.80 = 1000 watts

3. Pp = 1000 - 800 = 200 watts

Use 250TH; max. Pp = 250w;µ = 37.

4. lb = 1000/3500 = 0.285 ampere (285 ma.)

Max. Ib for 250TH is 350 ma.

5. Approximate ipmax = 0.285 X 4.5

= 1.28 ampere

6. epmin = 260 volts (see figure 5 first

trial point)

7. Epm = 3500 - 260 = 3240 volts

8. Ipm /lb = 2X0.80X3500/3240 =

5600/3240 = 1.73

9. ipmax /Ib = 4.1 (from figure 3)

10. ipmax = 0.285X4.1 = 1.17

11. egmp = 240 volts

igmax = 0.430 amperes

(Both above from final point on figure 5)

12. cos Bp=2.32(1.73-1.57)=0.37

(9p = 68.3°)

1

13. Ecc - X

1 - 0.37

3240

[0.37 (_ - 240)

37

= - 240 volts

3500

37

14. Egm = 240 - ( -240) = 480 volts grid

swing

15. Egm /Ecc = 480/ - 240 = - 2

16. igmax /Ic = 5.75 (from figure 4)

17. le = 0.430/5.75 = 0.075 amp. (75 ma.

grid current)

18. Pd = 0.9X480X0.075 = 32.5 watts

driving power

19. Pg = 32.5 - (- 240X0.75) = 14.5 watts

grid dissipation

Max. Pg for 250TH is 40 watts

The power output of any type of r -f ampli-

fier is equal to:

IpmEpm /2 = Po

Ipm can be determined, of course, from the

ratio determined in step 8 above (in this type

of calculation) by multiplying this ratio times

Ib It is frequently of importance to know the

value of load impedance into which a Class

C amplifier operating under a certain set of

conditions should operate. This is simply R L_

Epm /Ipm. In the case of the operating condi-

tions just determined for a 250TH amplifier

stage the value of load impedance is:

Epm 3240

RL _ -- -- 6600 ohms

Ipm .495

Ipm

Ipm = -Xlb

lb

Q of Amplifier In order to obtain good plate

Tank Circuit tank circuit tuning and low

radiation of harmonics from

an amplifier it is necessary that the plate tank

circuit have the correct Q. Charts giving com-

promise values of Q for Class C amplifiers

are given in the chapter, Generation o/ R -F

Energy. However, the amount of inductance

required for a specified tank circuit Q under

specified operating conditions can be calcu-

lated from the following expression:

www.americanradiohistory.com

HANDBOOK Class B R -F Amplifiers 159

Ri,

Q

= 2 ir X operating frequency

L = Tank inductance

RL = Required tube load impedance

Q = Effective tank circuit Q

A tank circuit Q of 12 to 20 is recommended

for all normal conditions. However, if a bal-

anced push -pull amplifier is employed the tank

receives two impulses per cycle and the cir-

cuit Q may be lowered somewhat from the

above values.

Quick Method of The plate circuit effi-

Calculating Amplifier ciency of a Class B or

Plate Efficiency Class C r -f amplifier

can be determined from

the following facts. The plate circuit efficiency

of such an amplifier is equal to the product of

two factors, F which is equal to the ratio of

Epm to Ebb (F, = Epm /Ebb) and F which is

proportional to the one -half angle of plate cur-

rent flow, 0p. A graph of F, against both 0,

and cos Bp is given in figure 6. Either 0p or

cos Bp may be used to determine F,. Cos 0p

may be determined either from the procedure

previously given for making Class C amplifier

computations or it may be determined from the

following expression:

Ecc + Ebb

cos Bp = - µEBm-Epm

Example of It is desired to know the one -half

Method angle of plate current flow and

the plate circuit efficiency for

an 812 tube operating under the following con-

ditions which have been assumed from inspec-

tion of the data and curves given in the RCA

Transmitting Tube Handbook HB -3:

1. Ebb = 1100 volts

Ecc = -40 volts

= 29

Ea. = 120 volts

Epm = 1000 volts

2. F, = Epm/Ebb = 0.91

- 29 X 40 + 1100

3. cos 0p= -

29 X 120 - 1000

60

- - 0.025

2480

4. F2= 0.79 (by reference to figure 6)

F2

DN,IIII.`....MM....

MN R\......

Ism

aee1

aee ....,....

.......,....

o.

D

.......-....

awl

ore ...\\...

1.' i.........,,..

ore I..........,.

ar211..........,

a M .... .....

0 10 20 30 40 50 e0 70 e0 90 100 110 120

PP IN ELECTRICAL DEGREES I I

, . .». <.. .... . . .:,. ... <.,. >... -..,»

cos AP

Figure 6

Relationship between Factor F_ and the

half -angle of plate current flow in an ampli-

fier with sine -wave input and output voltage,

operating at a grid -bias voltage greater than

cut -off

5. Np= F, X F, = 0.91 X 0.79 = 0.72

(72 per cent efficiency)

F, could be called the plate -voltage -swing

efficiency factor, and F2 can be called the

operating -angle efficiency factor or the maxi-

mum possible efficiency of any stage running

with that value of half -angle of plate current

flow. Np is, of course, only the ratio between

power output and power input. If it is desired

to determine the power input, exciting power,

and grid current of the stage, these can be ob-

tained through the use of steps 7, 8, 9, and 10

of the previously given method for power in-

put and output; and knowing that is

0.095 ampere the grid circuit conditions can

be determined through the use of steps 15, 16,

17, 18 and 19.

8 -4 Class B Radio

Frequency Power Amplifiers

Radio frequency power amplifiers operating

under Class B conditions of grid bias and ex-

citation voltage are used in two general types

of applications in transmitters. The first gen-

eral application is as a buffer amplifier stage

where it is desired to obtain a high value of

power amplification in a particular stage. A

particular tube type operated with a given

plate voltage will be capable of somewhat

greater output for a certain amount of excita-

tion power when operated as a Class B ampli-

www.americanradiohistory.com

160 R -F Vacuum Tube Amplifiers THE RADIO

l Ecz= 7-400 V.

Ec3=0v.

Eci= }s0

II

___ __A Ec=+Go

Eci=+4o

1 rd. i

I

1 I Eci= +20

i

\ t

I

I Eci=

iii c2,Ec=+ioo

Icz Ec=+w Fri, -20

/ J-- _,--.-- Eci=-4o

l00 200 300 400 500 G00 700 G0o am 000 um, .Hr, v. A..... ._.... ..,.,. ...

PLATE VOLTS

Figure 7

AVERAGE PLATE CHARACTERISTICS OF 813 TUBE

fier than when operated as a Class C ampli-

fier.

Calculation of Calculation of the operating

Operating conditions for this type of

Characteristics Class B r -f amplifier can be

carried out in a manner simi-

lar to that described in the previous para-

graphs, except that the grid bias voltage is set

on the tube before calculation at the value:

Ecc = Ebb /IL. Since the grid bias is set at

cutoff the one -half angle of plate current flow

is 900; hence cos angle is fixed at 0.00. The

plate circuit efficiency for a Class B r -f am-

plifier operated in this manner can be deter-

mined in the following manner:

Epm

NP =78.5 ( -1 Ebb //

The "Class B The second type of Class B

Linear" r -f amplifier is the so- called

Class Il linear amplifier which

is often used in transmitters for the amplifica-

tion of a single - sideband signal or a conven-

tional amplitude- modulated wave. Calculation

of operating conditions may be carried out in

a manner similar to that previously described

with the following exceptions: The first trial

operating point is chosen on the basis of the

100 per cent positive modulation peak of the

modulated exciting wave. The plate circuit

and grid peak voltages and currents can then

be determined and the power input and output

calculated. Then, with the exciting voltage

reduced to one -half for the no- modulation con-

dition of the exciting wave, and with the same

value of load resistance reflected on the tube,

the plate input and plate efficiency will drop

to approximately one -half the values at the

100 per cent positive modulation peak and the

power output of the stage will drop to one -

fourth the peak- modulation value. On the nega-

tive modulation peak the input, efficiency, and

output all drop to zero.

In general, the proper plate voltage, bias

voltage, load resistance and power output

listed in the tube tables for Class B audio

work will also apply to Class B linear r -f ap-

plication.

Calculation of Oper- Figure 7 illustrates

oting Parameters for a the characteristic

Class B Linear Amplifier curves for an 813

tube. Assume the

plate supply to be 2000 volts, and the screen

supply to be 400 volts. To determine the oper-

ating parameters of this tube as a Class B lin-

ear r -f amplifier, the following steps should

be taken:

1. The grid bias is chosen so that the rest-

ing plate current will produce approxi-

mately 1/3 of the maximum plate dissi-

pation of the tube. The maximum dissi-

pation of the 813 is 125 watts, so the

bias is set to allow one -third of this

value, or 42 watts of resting dissipation.

At a plate potential of 2000 volts, a

www.americanradiohistory.com

HANDBOOK Linear Amplifier Parameters 161

plate current of 21 milliamperes will

produce this figure. Referring to figure

7, a grid bias of -45 volts is approxi-

mately correct.

2. A practical Class 13 linear r -f amplifier

runs at an efficiency of about 66% at full

output, the efficiency dropping to about

33% with an unmodulated exciting sig-

nal. In the case of single- sideband sup-

pressed carrier excitation, a no- excita-

tion condition is substituted for the un-

modulated excitation case, and the lin-

ear amplifier runs at the resting or qui-

escent input of 42 watts with no exciting

signal. The peak allowable power input

to the 813 is:

Input Peak Power (Wp) _

(watts)

Plate Dissipation X 100

(100 -% plate efficiency)

125

- x 100 = 379 watts

33

3. The maximum signal plate current is:

Wp 379

tpmax = - _ -= 0.189 ampere

Ep 2000

4. The plate current flow of the linear am-

plifier is 1800, and the plate current

pulses have a peak of 3.14 times the

maximum signal current:

3.14 x 0.189 = 0.595 ampere

5. Referring to figure 7, a current of 0.605

ampere (Point A) will flow at a positive

grid potential of 60 volts and a minimum

plate potential of 420 volts. The grid is

biased at -45 volts, so a peak r -f grid

voltage of 60+45 volts = 105 volts is re-

quired.

6. The grid driving power required for the

Class B linear stage may be found by the

aid of figure 8. It is one -quarter the pro-

duct of the peak grid current times the

peak grid voltage:

0.02 X 105

Pp = - 0.53 watt

4

7. The single tone power output of the 813

stage is:

Pp = 78.5 (Ep - epmin) x Ip

Pp = 78.5 (2000 - 420) x .189 = 235 watts

BO

GO

ECO +400 V.

Ec3=ov.

40 Eci=+ioov.

ci=rsov.,

¡ Ec-raov.

Rib_ Eu=+sov

- - _

xo ` Ecr+20

o

.

100 200 300

PLATE VOLTS E

Figure 8

Eg VS. E P CHARACTERISTICS OF 813

TUBE

400

8. The plate load resistance is:

Ep - epmin 1580

RL 0.5ipmax 0.5 x .189

= 6000 ohms

9. If a loaded plate tank

desired, the reactance

capacitor at the r e s

should be:

RL

Reactance (ohms) = -- Q

circuit Q of 12 is

of the plate tank

on an t frequency

6000

= 500 ohms

12

10. For an operating frequency of 4.0 Mc.,

the effective resonant capacity is:

106

C = = 80 µµtd.

6.28 x 4.0 x 500

11. The inductance required to resonate at

4.0 Mc. with this value of capacity is:

500

L = 19.9 microhenries

6.28 x 4.0

Grid Circuit 1. The maximum positive grid

Considerations potential is 60 volts, and

the peak r -f grid voltage is

105 volts. Required driving power is 0.53 watt.

The equivalent grid resistance of this stage is:

www.americanradiohistory.com

162 R -F Vacuum Tube Amplifiers THE RADIO

(e5)2 1052

Rig - - -

2XPg 2X0.53

10,400 ohms

2. As in the case of the Class B audio am-

plifier the grid resistance of the linear

amplifier varies from infinity to a low

value when maximum grid current is

drawn. To decrease the effect of this

resistance excursion, a swamping resis-

tor should be placed across the grid tank

circuit. The value of the resistor should

be dropped until a shortage of driving

power begins to be noticed. For this ex-

ample, a resistor of 3,000 ohms is used.

The grid circuit load for no grid current

is now 3,000 ohms instead of infinity,

and drops to 2400 ohms when maximum

grid current is drawn.

3. A circuit Q of 15 is chosen for the grid

tank. The capacitive reactance required

is:

2400

X = -= 160 ohms

15

4. At 4.0 Mc. the effective capacity is:

C= 106

6.28x4X154 = 248 µµEd.

5. The inductive reactance required to reso-

nate the grid circuit at 4.0 Mc. is:

160

L= 6.28 x 4.0 = 6.4 microhenries

6. By substituting the loaded grid resist-

ance figure in the formula in the first

paragraph, the grid driving power is now

found to be approximately 2.3 watts.

Screen Circuit By reference to the plate

Considerations characteristic curve of the

813 tube, it can be seen that

at a minimum plate potential of 500 volts, and

a maximum plate current of 0.6 ampere, the

screen current will be approximately 30 milli-

amperes, dropping to one or two milliamperes

in the quiescent state. It is necessary to use

a well -regulated screen supply to hold the

screen voltage at the correct potential over

this range of current excursion. The use of an

electronic regulated screen supply is recom-

mended.

8 -5 Special R -F Power

Amplifier Circuits

The r -f power amplifier discussions of Sec-

tions 8 -4 and 8 -5 have been based on the as-

sumption that a conventional grounded- cathode

or cathode -return type of amplifier was in ques-

tion. It is possible, however, as in the case of

a -f and low -level r -f amplifiers to use circuits

in which electrodes other than the cathode are

returned to ground insofar as the signal poten-

tial is concerned. Both the plate- return or

cathode -follower amplifier and the grid- return

or grounded -grid amplifier are effective in cer-

tain circuit applications as tuned r -f power

amplifiers.

Disadvantages of An undesirable aspect of

Grounded -Cothode the operation of cathode -

Amplifiers return r -f power amplifiers

using triode tubes is that

such amplifiers must be neutralized. Princi-

ples and methods of neutralizing r -f power am-

plifiers are discussed in the chapter Genera-

tion of R -F Energy. As the frequency of opera-

tion of an amplifier is increased the stage be-

comes more and more difficult to neutralize

due to inductance in the grid and plate leads

of the tubes and in the leads to the neutraliz-

ing capacitors. In other words the bandwidth

of neutralization decreases as the frequency

is increased. In addition the very presence of

the neutralizing capacitors adds additional

undesirable capacitive loading to the grid and

plate tank circuits of the tube or tubes. To

look at the problem in another way, an ampli-

fier that may be perfectly neutralized at a fre-

quency of 30 Mc. may be completely out of

neutralization at a frequency of 120 Mc. There-

fore, if there are circuits in both the grid and

plate circuits which offer appreciable imped-

ance at this high frequency it is quite possi-

ble that the stage may develop a "parasitic

oscillation" in the vicinity of 120 Mc.

Grounded -Grid This condition of restricted -

R-F Amplifiers range neutralization of r -f

power amplifiers can be great-

ly alleviated through the use of a cathode -

return or grounded -grid r -f stage. The grounded -

grid amplifier has the following advantages:

1. The output capacitance of a stage is re-

duced to approximately one -half the value

which would be obtained if the same tube

or tubes were operated as a conventional

neutralized amplifier.

2. The tendency toward parasitic oscillations

in such a stage is greatly reduced since

the shielding effect of the control grid be-

www.americanradiohistory.com

HANDBOOK Grounded Grid Amplifier 163

tween the filament and the plate is effec-

tive over a broad range of frequencies.

3. The feedback capacitance within the stage

is the plate -to- cathode capacitance which

is ordinarily very much less than the grid -

to -plate capacitance. Hence neutralization

is ordinarily not required. If neutralization

is required the neutralizing capacitors are

very small in value and are cross con-

nected between plates and cathodes in a

push -pull stage, or between the opposite

end of a split plate tank and the cathode

in a single -ended stage.

The disadvantages of a grounded -grid am-

plifier are:

1. A large amount of excitation energy is re-

quired. However, only the normal amount

of energy is lost in the grid circuit of the

amplifier tube; all additional energy over

this amount is delivered to the load cir-

cuit as useful output.

2. The cathode of a grounded -grid amplifier

stage is "hot" to r.f. This means that the

cathode must be fed through a suitable im-

pedance from the filament supply, or the

secondary of the filament transformer must

be of the low- capacitance type and ade-

quately insulated for the r -f voltage which

will be present.

3. A grounded -grid r -f amplifier cannot be

plate modulated 100 per cent unless the

output of the exciting stage is modulated

also. Approximately 70 per cent modula-

tion of the exciter stage as the final stage

is being modulated 100 per cent is recom-

mended. However, the grounded -grid r -f

amplifier is quite satisfactory as a Class

B linear r -f amplifier for single sideband

or conventional amplitude modulated waves

or as an amplifier for a straight c -w or

FM signal.

Figure 9 shows a simplified representation

of a grounded -grid triode r -f power amplifier

stage. The relationships between input and

out put power and the peak fundamental com-

ponents of electrode voltages and currents are

given below the drawing. The calculation of

the complete operating conditions for a

grounded -grid amplifier stage is somewhat more

complex than that for a conventional amplifier

because the input circuit of the tube is in

series with the output circuit as far as the

load is concerned. The primary result of this

effect is, as stated before, that considerably

more power is required from the driver stage.

The normal power gain for a g -g stage is from

3 to 15 depending upon the grid circuit con-

ditions chosen for the output stage. The higher

the grid bias and grid swing required on the

PONEN OUTPUT TO LOAD {E4NEnr) IPr IP EPr r E4r i IPr

c-

POWER DEL,ORSED n OUTPUT TUBE Err Irr

POWER PROM DO UER TO LOAD E4 =IPr

TOTAL PONE, DELIVERED BY DIVE, EON (IPI+ I4r)

o, E Irr Or E4r lc

POOR, ,SORKD 5Y OUTPUT TUNE MD NO .!S SUPPLY

Z. (PP,o..N

E4r I4r O,OrE4NIt

E4r

I.N I lc

Figure 9

GROUNDED -GRID CLASS B OR CLASS C

AMPLIFIER

The equations in the above figure give the

relationships between the fundamental com-

ponents of grid and plate potential and cur-

rent, and the power input and power output

of the stage. An expression for the approxi-

mate cathode impedance is given

output stage, the higher will be the require-

ment from the driver.

Calculation of Operating It is most convenient

Conditions of Grounded to determine the op-

Grid R -F Amplifiers erating conditions for

a Class B or Class C

grounded -grid r -f power amplifier in a two -step

process. The first step is to determine the

plate- circuit and grid- circuit operating condi-

tions of the tube as though it were to operate

as a conventional cathode- return amplifier

stage. The second step is then to add in the

additional conditions imposed upon the oper-

ating conditions by the fact that the stage is

to operate as a grounded -grid amplifier.

For the first step in the calculation the pro-

cedure given in Section 8 -3 is quite satisfac-

tory and will be used in the example to follow.

Suppose we take for our example the case of a

type 304TL tube operating at 2700 plate volts

at a kilowatt input. Following through the pro-

cedure previously given:

1. Desired power output -850 watts

Desired Plate voltage -2700 volts

Desired plate efficiency -85 per cent

(Np = 0.85)

www.americanradiohistory.com

164 R -F Vacuum Tube Amplifiers THE RADIO

2. P1° = 850/0.85 = 1000 watts

3. P, = 1000 - 850 = 150 watts

Type 304TL chosen; max. P, = 300

watts, /I= 12.

4. Ib = 1000/2700 = 0.370 ampere

(370 ma.)

5. Approximate ipma, = 4.9 X 0.370 = 1.81

ampere

6. epm;n= 140 volts (from 30411. con-

stant- current curves)

7. Epm = 2700 - 140 = 2560 volts

8. 1./lb = 2 X 0.85 X 2700/2560 = 1.79

9. igmax/Ib = 4.65 (from figure 3)

10. igmax = 4.65 X 0.370 = 1.72 amperes

11. egmp = 140 volts

igma, = 0.480 amperes

12. Cos Op = 2.32 (1.79 -1.57) = 0.51

Op = 59°

13.

1

Ecc- 1- 0.51 X

[0.51 ( `

2560 140) 2700

- - -

12 J

_ -385 volts

14. Egm = 140 -( -385) = 525 volts

15. Egm /Ecc = -1.36

16. igmax /Ia = approx. 8.25 (extrapolated

from figure 4)

17. la = 0.480/8.25 = 0.058 (58 ma. d -c

grid current)

18. Pd = 0.9 X 525 X 0.058 = 27.5 watts

19. Pg = 27.5 -( -385 X 0.058) = 5.2 watts

Max. P, for 304TL is 50 watts

We can check the operating plate efficiency

of the stage by the method described in Sec-

tion 8 -4 as follows:

F, = Epm /Ebb = 2560/2700 = 0.95

F2 for Op of 59° (from figure 6) = 0.90

Np = F, X F2 = 0.95 X 0.90 = Approx.

0.85 (85 per cent plate efficiency)

Now, to determine the operating conditions

as a grounded -grid amplifier we must also know

the peak value of the fundamental components

of plate current. This is simply equal to

(Ipm /Ib) lb, or:

Ipm = 1.79 X 0.370 = 0.660 amperes (from 4

and 8 above)

The total average power required of the

driver (from figure 9) is equal to Egmlpm /2

(since the grid is grounded and the grid swing

appears also as cathode swing) plus Pd which

is 27.5 watts from 18 above. The total is:

525 X 0.660

Total drive = - 172.5 watts

2

plus 27.5 watts or 200 watts

Therefore the total power output of the stage

is equal to 850 watts (contributed by the

304TL) plus 172.5 watts (contributed by the

driver) or 1022.5 watts. The cathode driving

impedance of the 30411. (again referring to

figure 7) is approximately:

Zk = 525/(0.660 + 0.116) = approximately 675

ohms.

Plate- Return or Circuit diagram, elec-

Cathode- Follower R -F trodepotentials and cur-

Power Amplifier rents, an d operating

conditions for a cath-

ode- follower r -f power amplifier are given in

figure 10. This circuit can be used, in addi-

tion to the grounded -grid circuit just dis-

cussed, as an r -f amplifier with a triode tube

and no additional neutralization circuit. How-

ever, the circuit will oscillate if the imped-

ance from cathode to ground is allowed to be-

come capacitive rather than inductive or re-

sistive with respect to the operating frequen-

cy. The circuit is not recommended except for

v -h -f or u -h -f work with coaxial lines as tuned

circuits since the peak grid swing required

on the r -f amplifier stage is approximately

equal to the plate voltage on the amplifier

tube if high- efficiency operation is desired.

This means, of course, that the grid tank must

be able to withstand slightly more peak volt-

age than the plate tank. Such a stage may not

be plate modulated unless the driver stage is

modulated the same percentage as the final

amplifier. However, such a stage may be used

as an amplifier or modulated waves (Class B

linear) or as a c -w or FM amplifier.

www.americanradiohistory.com

HANDBOOK G -G Amplifier 165

POWER OUTPUT TO LOAD - EpM ( PM. IGM)

POWER DELIVERED BV OUTPUT TUBE - EPM IPM

2

POWER FROM DRIVER TO LOAD T. EPM IOM

2

TOTAL POWER FROM DRIVER_ ECU IGM

z 2

AppROA (EPM + eGMP) I e IC

ASSUMING IGM y 1.0 IC

(EPM +eGMP) IGM

POWER ABSORBED Or OUTPUT TUBE GRID AND BIAS SUPPLY.

c APPROA O S (Ecc eoup) lc

ZG - APPROA. ( EPM+ eGMP

I GM I I

Figure 10

CATHODE -FOLLOWER R -F POWER

AMPLIFIER

Showing the relationships between the tube

potentials and currents and the input and

output power of the stage. The approximate

grid impedance also is given.

The design of such an amplifier stage is

essentially the same as the design of a

grounded -grid amplifier stage as far as the

first step is concerned. Then, for the second

step the operating conditions given in figure

10 are applied to the data obtained in the first

step. As an example, take the 304TL stage

previously described. The total power required

of the driver will be (from figure 10) approxi-

mately (2700X0.58);1.8) /2 or 141 watts. Of

this 141 watts 27.5 watts (as before) will be

lost as grid dissipation and bias loss and the

balance of 113.5 watts will appear as output.

The total output of the stage will then be ap-

proximately 963 watts.

Cathode Tank for The cathode tank circuit

G -G or C -F for either a grounded -grid

Power Amplifier or cathode -follower r -f

power amplifier may be a

conventional tank circuit if the filament trans-

former for the stage is of the low -capacitance

high- voltage type. Conventional filament trans-

formers, however, will not operate with the

high values of r -f voltage present in such a

circuit. If a conventional filament transformer

is to be used the cathode tank coil may con-

sist of two parallel heavy conductors (to carry

the high filament current) by- passed at both

the ground end and at the tube socket. The

tuning capacitor is then placed between fila-

ment and ground.lt is possible in certain cases

to use two r -f chokes of special design to feed

the filament current to the tubes, with a con-

ventional tank circuit between filament and

ground. Coaxial lines also may be used to

serve both as cathode tank and filament feed

to the tubes for v -h -f and u -h -f work.

Control Grid Dissipation Tetrode tubes maybe

in Grounded -Grid Stages operated as ground-

ed grid (cathode

driven) amplifiers by tying the grid and screen

together and operating the tube as a high -u

triode (figure 11). Combined grid and screen

current, however, is a function of tube geo-

metry and may reach destructive values under

conditions of full excitation. Proper division

of excitation between grid and screen should

be as the ratio of the screen -to -grid amplifi-

cation, which is approximately 5 for tubes

such as the 4 -250A, 4 -400A, etc. The proper

ratio of grid /screen excitation may be achiev-

ed by tapping the grid at some point on the

filament choke, as shown. Grid dissipation is

reduced, Lut the overall level of excitation is

increased about 30% over the value required

for simple grounded -grid operation.

ORIVe

4 -200A, 4 -400A, fFt'

RFC- RFC

rl

FIGURE II

TAPPED FILAMENT CHORE REDUCES EXCESSIVE

GRID DISSIPATION IN G -G CIRCUIT.

RFC- -TWOP WINDINGS OP 191C WIRE, ES TURN!

CAC., I- DIAM. TOTAL LENGTH IS SIR INCHES. GRID

TAP II TURNS PROM GROUND CND OP ONE WINDING.

71- e.a vOLTS AT 1 AMPERE!. (VOLTAGE DROP ACROSS

RFC Is I.a VOLTS )

www.americanradiohistory.com

166 R -F Vacuum Tube ,'Amplifiers THE RADIO

8 -6 Class ABI Radio Frequency

Power Amplifiers

Class Aß1 r -f amplifiers operate under

such conditions of bias and excitation that

grid current does not flow over any portion of

the input cycle. This is desirable, since

distortion caused by grid current loading is

absent, and also because the stage is capable

of high power gain. Stage efficiency is about

58% when a plate current operating angle of

2100 is chosen, as compared to 62% for Class

B operation.

The level of static (quiescent) plate current

for lowest distortion is quite critical for

Class AB1 tetrode operation. This value is

determined by the tube characteristics, and is

not greatly affected by the circuit parameters

or operating voltages. The maximum d -c plate

potential is therefore limited by the static

dissipation of the tube, since the resting plate

current figure is fixed. The static plate current

of a tetrode tube varies as the 3/2 power of

the screen voltage. For example, raising the

screen voltage from 300 to 500 volts will

double the plate current. The optimum static

plate current for minimum distortion is also

doubled, since the shape of the Eg -Ip curve

does not change.

In actual practice, somewhat lower static

plate current than optimum may be employed

without raising the distortion appreciably,

and values of static plate current of 0.6 to

0.8 of optimum may be safely used, depending

upon the amount of nonlinearity that can be

tolerated.

As with the class B linear stage, the mini-

mum plate voltage swing of the class AB1

amplifier must be kept above the d -c screen

potential to prevent operation in the nonlinear

portion of the characteristic curve. A low value

of screen voltage allows greater r -f plate

voltage swing, resulting in improvement in

plate efficiency of the tube. A balance be-

tween plate dissipation, plate efficiency, and

plate voltage swing must be achieved for best

linearity of the amplifier.

The 5 -Curve The perfect linear ampli-

fier delivers a signal

that is a replica of the input signal. Inspection

of the plate characteristic curve of a typical

tube will disclose the tube linearity under

class A operating conditions (figure 12). The

curve is usually of exponential shape, and

the signal distortion is held to a small value

by operating the tube well below its maximum

output, and centering operation over the most

linear portion of the characteristic curve.

o MOSTLINRWA,-P

PORTIONOF i

EG-tP .Ì*n »

CURVE

yll i A,

iiiiiis

vis w

15111111111114 111

w1i

MIN / j IN

1-!ßr

91JJiT -

SIGNAL

Figure 12

Eg -Ip CURVE

Amplifier operation is confined to

most linear portion of characteristic

curve.

The relationship between exciting voltage

in a Class ABI amplifier and the r -f plate

circuit voltage is shown in figure 13. With a

small value of static plate current the lower

portion of the line is curved. Maximum un-

distorted output is limited by the point on the

line (A) where the instantaneous plate voltage

down to the screen voltage. This "hook" in

the line is caused by current diverted from

the plate to the grid and screen elements of

the tube. The characteristic plot of the usual

linear amplifier takes the shape of an S- curve.

The lower portion of the curve is straightened

out by using the proper value of static plate

current, and the upper portion of the curve is

avoided by limiting minimum plate voltage

swing to a point substantially above the value

of the screen voltage.

Operating Parameters The approximate oper-

for the Class ABI acing parameters may be

Linear Amplifier obtained from the Con-

stant Current curves

(Eg -Ep) or the Eg -Ip curves of the tube in

question. An operating load line is first

approximated. One end of the load line is

determined by the d -c operating voltage of the

tube, and the required static plate current.

As a starting point, let the product of the plate

voltage and current equal the plate dissipation

of the tube. Assuming we have a 4 -400A

tetrode, this end of the load line will fall on

point A (figure 14). Plate power dissipation

is 360 watts (3000V @ 120 ma). The opposite

end of the load line will fall on a point deter-

mined by the minimum instantaneous plate volt-

age, and by the maximum instantaneous plate

current. The minimum plate voltage, for best

linearity should be considerably higher than

www.americanradiohistory.com

HANDBOOK ABI R.F. Power Amplifiers 167

R F.

E our

R.F. E iN

Figure 13

LINEARITY CURVE OF

TYPICAL TETRODE AMPLIFIER

At point "A" the instantaneous plate

voltage is swinging down to the value

of screen voltage. At point "B" it is

swinging well below the screen and is

approaching the point where saturation,

or plate current limiting takes place.

the screen voltage. In this case, the screen

voltage is 500, so the minimum plate voltage

excursion should be limited to 600 volts.

Class AB1 operation implies no grid current,

therefore the load line cannot cross the Eg =0

line. At the point Ep =600, Eg =0, the maximum

plate current is 580 ma (Point "B ").

Each point the load line crosses a grid volt-

age axis may be taken as a point for construc-

tion of the Eg -Ip curve, just as was done in

figure 22, chapter 6. A constructed curve

shows that the approximate static bias voltage

is -74 volts, which checks closely with point A

of figure 14. In actual practice, the bias volt-

age is set to hold the actual dissipation slightly

below the maximum figure of the tube.

The single tone power output is:

Emax -Eminx Ipmax. or 3000 -600 x .58=348 watts

4 4

The plate current -angle efficiency factor for

this class of operation is 0.73, and the actual

plate circuit efficiency is:

Np =Emax -Emin x0.73, or 3000- 600 X 0.73 = 58.4%

Emax 3000

The power input to the stage is therefore

Pox 100 or, 348 = 595 watts

Np 58.4

The plate dissipation is: 595 -348 =247 watts.

Pi- network

. 4k1RtIoN

TOP VIEW OF A 4 -250A AMPLIFIER

tetrode amplifier may be operated Class AB, Class 8, or Class C by varying potentials

applied to tube. Same general physical and mechanical design applies in each cose.

www.americanradiohistory.com

168 R -F Vacuum Tube Amplifiers THE RADIO

w

r,

r

4

2

oo. 4 -400A rues

E SCR = 900 VOLTS

'S

I0'

t5L-

S

POINT

t B

,000 2000 woo oC

o

is -

SO .E LOAD LINE POINT

A

00 m,_ I

zs

VALUE

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VALUE

MAX.

DISSIPATION

(3000

OF

V. X 0.71 A

SO 11.2 .300 wArrs)

75

nni

hiclre 14

OPERATING PARAMETERS FOR TETRODE LINEAR

AMPLIFIER ARE OBTAINED FROM CONSTANT- CURRENT

CURVES.

ER

.s

.4

.2

o

It can be seen that the limiting factor for

this class of operation is the static plate dissi-

pation, which is quite a bit higher than the

operating dissipation level. It is possible, at

the expense of a higher level of distortion, to

drop the static plate dissipation and to increase

the screen voltage to obtain greater power out-

put. If the screen voltage is set at 800, and

the bias increased sufficiently to drop the

static plate current to 90 ma, the single tone

d -c plate current may rise to 300 ma, for a

power input of 900 watts. The plate circuit

efficiency is 55.6 %, and the power output is

500 watts. Static plate dissipation is 270 watts.

At a screen potential of 500 volts, the maxi-

mum screen current is less than 1 ma, and under

certain loading conditions may be negative.

When the screen potential is raised to 800 volts

maximum screen current is 18 ma. The per-

formance of the tube depends upon the voltage

fields set up within the tube by the cathode,

control grid, screen grid, and plate. The quantity

of current flowing in the screen circuit is only

incidental to the fact that the screen is main-

tained at a positive potential with respect to

the electron stream surrounding it.

The tube will perform as expected so long as

the screen current, in either direction, does

not create undesirable changes in the screen

voltage, or cause excessive screen dissipation.

Good regulation of the screen supply is there-

fore required. Screen dissipation is highly

responsive to plate loading conditions, and the

plate circuit should always be adjusted so as

to keep the screen current below the maximum

dissipation level as established by the applied

voltage.

G -G Class B Linear Certain tetrode and pentode

Tetrode Amplifier tubes, such as the 6AG7,

837, and 803 perform well

as grounded grid class B linear amplifiers. In

this configuration both grids and the suppressor

are grounded, and excitation is applied to the

cathode circuit of the tube. So connected, the

tubes take on characteristics of high -mu triodes.

No bias or screen supplies are required for

this type of operation, and reasonably linear

-r

DRIVER

6AG7

B4-300-700 v.

Figure 15

SIMPLE GROUNDED -GRID

LINEAR AMPLIFIER

www.americanradiohistory.com

HANDBOOK 169

VI

837 500 V2

837 V3

837 500

V4

803 VS

803 500

250 250 250 250

INPUT

2 War TS

PEAK

.001 = .001 =

2 N 5KV

Ti

Figure 16

3 -STAGE KILOWATT LINEAR

AMPLIFIER FOR 80 OR 40

METER OPERATION

E0 ACH

115V ti 2500

An open frame filament transformer may

be used for TI. Cathode taps are ad-

justed for proper excitation of following

stage.

operation can be had with a very minimum of

circuit components (figure 15). The input im-

pedance of the g -g stage falls between 100

and 250 ohms, eliminating the necessity of

swamping resistors, even though considerable

power is drawn by the cathode circuit of the

g -g stage.

Power gain of a g -g stage varies from ap-

proximately 20 when tubes of the 6AG7 type

are used, down to five or six for the 837 and

803 tubes. One or more g -g stages may be

cascaded to provide up to a kilowatt of power,

as illustrated in figure 16.

The input and output circuits of cascaded

g -g stages are in series, and a variation in

load impedance of the output stage reflects

back as a proportional change on the input

circuit. If the first g -g stage is driven by a high

impedance source, such as a tetrode amplifier,

any change in gain will automatically be com-

pensated for. If the gain of V4 -V5 drops, the

input impedance to that stage will rise. This

change will reflect through V2 -V3 so that the

load impedance of VI rises. Since V1 has a

high internal impedance the output voltage

will rise when the load impedance rises. The

increased output voltage will raise the output

voltage of each g -g stage so that the overall

output is nearly up to the initial value before

the drop in gain of V4 -V5.

The tank circuits, therefore, of all g -g stages

must be resonated with low plate voltage and

excitation applied to the tubes. Tuning of one

stage will affect the ocher stages, and the in-

put and coupling of each stage must be ad-

justed in turn until the proper power limit is

reached.

Operating Dota for

4 -400A Grounded

Grid Linear

Amplifier

Experiments have been

conducted by Collins Ra-

dio Co. on a grounded grid

linear amplifier stage

using a 4 -400A tube for the h.f. region. The

operating characteristics of the amplifier are

summarized in figure 17. It can be noted that

unusually low screen voltage is used on the

tube. The use of lower screen voltage has the

adverse effect of increasing the driving power,

but at the same time the static plate current

of the stage is decreased and linearity is im-

improved. For grounded grid operation of the

4 -400A, a screen voltage of 300 volts (filament

to screen) gives a reasonable compromise be-

tween these factors.

OPERATING DATA FOR 4- 400A/4 -250A

G -G. LINEAR AB, AMPLIFIER

(SINGLE CONE)

D -C SCREEN VOLTAGE +300 +300

D -C PLATE VOLTAGE +3000 +3500

STATIC PLATE CURRENT 60 MA. 60 MA.

D -C GRID BIAS -60 V. -59 V.

PEAK CATHODE SWING 67 V. 113 V.

MINIMUM PLATE VOLTAGE 660 V. S00 V.

MAXIMUM SIGNAL GRID CURRENT 3.6 MA. 10 MA.

MAXIMUM SIGNAL SCREEN CURRENT .1 MA. 20 MA.

MAXIMUM SIGNAL PLATE CURRENT 195 MA. 267 MA.

MAXIMUM SIGNAL PLATE DISSIPATION 235 W. 235 W.

STATIC PLATE DISSIPATION 160 W. 210W.

GRID DRIVING POWER 0.63 W. 3.4 W.

FEEDTHRU POWER 6.55 W. ,s.e W.

POWER OUTPUT (MAXIMUM) 350 W. 700 W.

POWER INPUT (MAX /MUM) 565 W. 935 W.

Figure 17

www.americanradiohistory.com

CHAPTER NINE

The Oscilloscope

The cathode -ray oscilloscope (also called

oscillograph) is an instrument which permits

visual examination of various electrical phe-

nomena of interest to the electronic engineer.

Instantaneous changes in voltage, current and

phase are observable if they take place slow-

ly enough for the eye to follow, or if they are

periodic for a long enough time so that the eye

can obtain an impression from the screen of

the cathode -ray tube. In addition, the cathode -

ray oscilloscope may be used to study any

variable (within the limits of its frequency

response characteristic) which can be con-

verted into electrical potentials. This conver-

sion is made possible by the use of some type

of transducer, such as a vibration pickup unit,

pressure pickup unit, photoelectric cell, mi-

crophone, or a variable impedance. The use

of such a transducer makes the oscilloscope

a valuable tool in fields other than electronics.

9 -1 A Typical Cathode -Ray

Oscilloscope

For the purpose of analysis, the operation

of a simple oscilloscope will be described.

The Du Mont type 274 -A unit is a fit instru-

ment for such a description. The block diagram

of the 274 -A is shown in figure 1. The elec-

tron beam of the cathode -ray tube can be moved

vertically or horizontally, or the vertical and

horizontal movements may be combined to pro-

duce composite patterns on the tube screen.

As shown in figure 1, the cathode -ray tube is

the recipient of signals from two sources: the

vertical and horizontal amplifiers. The opera-

tion of the cathode -ray tube itself has been

covered in Chapter 4; the auxiliary circuits

pertaining to the cathode -ray tube will be

covered here.

The Vertical The incoming signal which is

Amplifier to be examined is applied to

the terminals marked Vertical

Input and Ground. The Vertical Input terminal

is connected through capacitor CE (figure 2) so

that the a -c component of the input signal ap-

pears across the vertical amplifier gain con-

trol potentiometer, R,. Thus the magnitude of

the incoming signal may be controlled to pro-

vide the desired deflection on the screen of

170

'ERTICAL

NF T

GND O

To FLIP. ATE INTENSITY

MOD.

- -So -- ro HORIZONTAL

DEFLECTION

EAT.

RE T, STNC.

SI

-- -rsa - - - -

/

NoNIZONTAL/ SWEEPS

IUT / R-R}- - DIR7 DIR

111111000111111 1NFVT

AM.

Figure 1

BLOCK DIAGRAM, TYPE 274 -A

CATHODE -RAY OSCILLOSCOPE

www.americanradiohistory.com

Time Base Generator 171

CI

0.25 LF

INPUT O--{

VERT. AMP. Ra

CONTROL 11A

6AC7

R30

ee M

Cz R2

5100 1K

Figure 2

TYPICAL AMPLIFIER SCHEMATIC

RT.15aK

OUTPUT

the cathode -ray tube. Also, as shown in figure

1, S, has been incorporated to by -pass the

vertical amplifier and capacitively couple the

input signal directly to the vertical deflection

plate if so desired.

In figure 2, V, is a 6AC7 pentode tube which

is used as the vertical amplifier. As the sig-

nal variations appear on the grid of V varia-

tions in the plate current of V, will take place.

Thus signal variations will appear in opposite

phase and greatly amplified across the plate

resistor, R,. Capacitor C, has been added a-

cross R, in the cathode circuit of V, to flatten

the frequency response of the amplifier at the

high frequencies. This capacitor because of

its low value has very little effect at low in-

put frequencies, but operates more effectively

as the frequency of the signal increases. The

amplified signal delivered by V, is now ap-

plied through the second half of switch S, and

capacitor C, to the free vertical deflection

plate of the cathode -ray tube (figure 3).

The Horizontal The circuit of the horizontal

Amplifier amplifier and the circuit of

the vertical amplifier, de-

scribed in the above paragraph, are similar. A

switch in the input circuit makes provision for

the input from the Horizontal Input terminals

to be capacitively coupled to the grid of the

horizontal amplifier or to the free horizontal

deflection plate thus by- passing the amplifier,

or for the output of the sweep generator to be

capacitively coupled to the amplifier, as shown

in figure 1.

The Time Bose Investigation of e l e c t r i c al

Generator wave forms by the use of a

cathode -ray tube frequently

requires that some means be readily available

to determine the variation in these wave forms

with respect to time. When such a time base

is required, the patterns presented on the cath-

ode -ray tube screen show the variation in am-

plitude of the input signal with respect to

Figure 3

SCHEMATIC OF CATHODE -RAY TUBE

CIRCUITS

A 5BPIA cathode -ray tube is used in this

instrument. As shown, the necessary poten-

tials for operating this tube are obtained

from a voltage divider mode up of resistors

R21 through R26 inclusive. The intensity of

the beam is adjusted by moving the contact

on R21. This adjusts the potential on the

cathode more or less negative with respect

to the grid which is operated at the full neg-

ative voltage -1200 volts. Focusing to the

desired sharpness is accomplished by ad-

justing the contact on R23 to provide the

correct potential for anode no. 1. Interde-

pendency between the focus and the, inten-

sity controls is inherent in all electrostati-

cally focused cathode -ray tubes. In short,

there is an optimum setting of the focus con-

trol for every setting of the intensity con-

trol. The second anode of the 5BP IA is oper-

ated at ground potential in this instrument.

Also one of each pair of deflection plates is

operated at ground potential.

The cathode is operated at a high negative

potential (approximately 1200 volts) so that

the total overall accelerating voltage of this

tube is regarded os 1200 volts since the sec-

ond anode is operated at ground potential.

The vertical and horizontal positioning con-

trols which are connected to their respective

deflection plates are capable of supplying

either a positive or negative d -c potential

to the deflection plates. This permits the

spot to be positioned at any desired place

on the entire screen.

time. Such an arrangement is made possible

by the inclusion in the oscilloscope of a Time

Base- Generator. The function of this genera-

tor is to move the spot across the screen at a

constant rate from left to right between two

selected points, to return the spot almost in-

stantaneously to its original position, and to

www.americanradiohistory.com

172 The Oscilloscope THE RADIO

Figure 4

SAWTOOTH WAVE FORM

repeat this procedure at a specified rate. This

action is accomplished by the voltage output

from the time base (sweep) generator. The

rate at which this voltage repeats the cycle

of sweeping the spot across the screen is re-

ferred to as the sweep frequency. The sweep

voltage necessary to produce the motion de-

scribed above must be of a sawtooth wave-

form, such as that shown in figure 4.

The sweep occurs as the voltage varies

from A to B, and the return trace as the volt-

age varies from B to C. If A -B is a straight

line, the sweep generated by this voltage will

be linear. It should be realized that the saw -

tooth sweep signal is only used to plot varia-

tions in the vertical axis signal with respect

to time. Specialized studies have made neces-

sary the use of sweep signals of various

shapes which are introduced from an external

source through the Horizontal Input terminals.

The Sawtooth The sawtooth voltage neces-

Generator sary to obtain the linear time

base is generated by the cir-

cuit of figure 5, which operates as follows:

A type 884 gas triode (V3) is used for the

sweep generator tube. This tube contains an

inert gas which ionizes when the voltage be-

tween the cathode and the plate reaches a cer-

tain value. The ionizing voltage depends upon

the bias voltage of the tube, which is deter-

mined by the voltage divider resistors R12 -R17.

With a specific negative bias applied to the

884 tube, the tube will ionize (or fire) at a

specific plate voltage.

Capacitors C3.-C24 are selectively connected

in parallel with the 884 tube. Resistor R

limits the peak current drain of the gas triode.

The plate voltage on this tube is obtained

through resistors R22, R and R11. The voltage

applied to the plate of the 884 tube cannot reach

the power supply voltage because of the charg-

ing effect this voltage has upon the capacitor

which is connected across the tube. This ca-

pacitor charges until the plate voltage becomes

high enough to ionize the gas in the tube. At

this time, the 884 tube starts to conduct and

the capacitor discharges through the tube until

its voltage falls to the extinction potential of

the tube. When the tube stops conducting, the

capacitor voltage builds up until the tube fires

again. As this action continues, it results in

the sawtooth wave form of figure 4 appearing

at the junction of R and R77.

Synchronization Provision has been made so

the sweep generator may be

synchronized from the vertical amplifier or

from an external source. The switch S, shown

in figure 5 is mounted on the front panel to be

easily accessible to the operator.

If no synchronizing voltage is applied, the

discharge tube will begin to conduct when the

plate potential reaches the value of F.t (Firing

Potential). When this breakdown takes place

and the tube begins to conduct, the capacitor

is discharged rapidly through the tube, and the

plate voltage decreases until it reaches the

extinction potential E1. At this point conduc-

tion ceases, and the plate potential rises slow-

ly as the capacitor begins to charge through

R7, and R25. The plate potential will again

reach a point of conduction and the circuit

will start a new cycle. The rapidity of the

plate voltage rise is dependent upon the circuit

constants R77 R25, and the capacitor selected,

Rr ISO

6AC7

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T SIGNAL TO

DEFLECTION PLATE

470

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CONTROL

Figure 5

SCHEMATIC OF SWEEP GENERATOR

www.americanradiohistory.com

HANDBOOK The Oscilloscope 173

EPVSEg EP+

STATIC CONTROL

f CHARACTERISTIC

I

D.C.GRID

BIAS t+

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SYNCHRONIZE

PERIOD

Er - - - FIRING POTENTIAL

(D.C. BIAS)

Eex

EXTINCTION

POTENTIAL

SYNC. SIGNAL

APPLIED TO GRID

Figure 6

ANALYSIS OF SYNCHRONIZATION OF TIME -BASE GENERATOR

C10 -C14, as well as the supply voltage Eb

The exact relationship is given by:

t

Ec Eb(1_erc )

Where E,----=Capacitor voltage at time t

Eb =Supply voltage (B+ supply - cathode

bias)

Er---Firing potential or potential at which

time -base gas triode fires

Ex =Extinction potential or potential at

which time -base gas triode ceases

to conduct

e= Base of natural logarithms

t= Time in seconds

r =Resistance in ohms (R=T + Rzs)

c =Capacity in farads (C10, II, 12, ,,, or 14)

The frequency of oscillation will be approx-

imately:

Ebl

1

f=- rc Et-Ex

Under this condition (no synchronizing sig-

nal applied) the oscillator is said to be /ree

running.

When a positive synchronizing voltage is

applied to the grid, the firing potential of the

tube is reduced. The tube therefore ionizes at

a lower plate potential than when no grid sig-

nal is applied. Thus the applied snychroniz-

ing voltage fires the gas -filled triode each

time the plate potential rises to a sufficient

value, so that the sweep recurs at the same

or an integral sub -multiple of the synchroniz-

ing signal rate. This is illustrated in figure 6.

Power Supply Figure - shows the power sup-

ply to be made up of two defi-

nite sections: a low voltage positive supply

which provides power for operating the ampli-

fiers, the sweep generator, and the positioning

circuits of the cathode -ray tube; and the high

voltage negative supply which provides the

potentials necessary for operating the various

Figure 7

SCHEMATIC OF POWER SUPPLY

www.americanradiohistory.com

174 The Oscilloscope THE RADIO

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www.americanradiohistory.com

HANDBOOK Display of Waveforms 175

TIME -+

4 SEC.

I-

Figure 9

PROJECTION DRAWING OF A SINEWAVE

APPLIED TO THE VERTICAL AXIS AND A

SAWTOOTH WAVE OF THE SAME FRE-

QUENCY APPLIED SIMULTANEOUSLY ON

THE HORIZONTAL AXIS

electrodes of the cathode -ray tube, and for

certain positioning controls.

The positive low voltage supply consists

of full -wave rectifier (V,), the output of which

is filtered by a capacitor input filter (20 -20 µfd.

and 8 II). It furnishes approximately 400 volts.

The high voltage power supply employs a half

wave rectifier tube, V,. The output of this rec-

tifier is filtered by a resistance -capacitor fil-

ter consisting of 0.5 -0.5 pfd. and .18 M. A

voltage divider network attached from the out-

put of this filter obtains the proper operating

potentials for the various electrodes of the

cathode -ray tube. The complete schematic of

the Du Mont 274 -A Oscilloscope is shown in

figure 8.

9 -2 Display of Waveforms

Together with a working knowledge of the

controls of the oscilloscope, an understanding

of how the patterns are traced on the screen

must be obtained for a thorough knowledge of

oscilloscope operation. With this in mind a

careful analysis of two fundamental waveform

patterns is discussed under the following

headings:

a. Patterns plotted against time (using the

sweep generator for horizontal deflection).

b. Lissajous Figures (using a sine wave for

horizontal deflection).

Patterns Plotted A sine wave is typical of

Against Time such a pattern and is con-

venient for this study. This

Figure 10

PROJECTION DRAWING SHOWING THE RE-

SULTANT PATTERN WHEN THE FRE-

QUENCY OF THE SAWTOOTH IS ONE -HALF

OF THAT EMPLOYED IN FIGURE 9

wave is amplified by the vertical amplifier

and impressed on the vertical (Y -axis) deflec-

tion plates of the cathode -ray tube. Simultane-

ously the sawtooth wave from the time base

generator is amplified and impressed on the

horizontal (X -axis) deflection plates.

The electron beam moves in accordance

with the resultant of the sine and sawtooth

signals. The effect is shown in figure 9 where

the sine and sawtooth waves are graphically

represented on time and voltage axes. Points

on the two waves that occur simultaneously

are numbered similarly. For example, point 2

on the sine wave and point 2 on the sawtooth

wave occur at the same instant. Therefore the

position of the beam at instant 2 is the result-

ant of the voltages on the horizontal and ver-

tical deflection plates at instant 2. Referring

to figure 9, by projecting lines from the two

point 2 positions, the position of the electron

beam at instant 2 can be located. If projec-

tions were drawn from every other instantane-

ous position of each wave to intersect on the

circle representing the tube screen, the inter-

sections of similarly timed projections would

trace out a sine wave.

In summation, figure 9 illustrates the prin-

ciples involved in producing a sine wave trace

on the screen of a cathode -ray tube. Each in-

tersection of similarly timed projections rep-

resents the position of the electron beam act-

ing under the influence of the varying voltage

waveforms on each pair of deflection plates.

Figure 10 shows the effect on the pattern of

decreasing the frequency of the sawtooth

www.americanradiohistory.com

176 The Oscilloscope THE RADIO

Figure 11

PROJECTION DRAWING SHOWING THE RE-

-SULTANT LISSAJOUS PATTERN WHEN A

SINE WAVE APPLIED TO THE HORIZON-

TAL AXIS IS THREE TIMES THAT AP-

PLIED TO THE VERTICAL AXIS

wave. Any recurrent waveform plotted against

time can be displayed and analyzed by the

same procedure as used in these examples.

The sine wave problem just illustrated is

typical of the method by which any waveform

can be displayed on the screen of the cathode -

ray tube. Such waveforms as square wave,

sawtooth wave, and many more irregular recur-

rent waveforms can be observed by the same

method explained in the preceding paragraphs.

9 -3 Lissajous Figures

Another fundamental pattern is the Lissajous

figure, named after the 19th century French

scientist. This type of pattern is of particular

use in determining the frequency ratio between

two sine wave signals. If one of these signals

is known, the other can be easily calculated

from the pattern made by the two signals upon

the screen of the cathode -ray tube. Common

practice is to connect the known signal to the

horizontal channel and the unknown signal to

the vertical channel.

The presentation of Lissajous figures can

be analyzed by the same method as previously

used for sine wave presentation. A simple ex-

ample is shown in figure 11. The frequency

ratio of the signal on the horizontal axis to the

signal on the vertical axis is 3 to 1. If the

known signal on the horizontal axis is 60 cy-

cles per second, the signal on the vertical

axis is 20 cycles.

B

Figure 12

METHOD OF CALCULATING FREQUENCY

RATIO OF LISSAJOUS FIGURES

Obtaining a Lissalous 1. The horizontal am-

Pattern on the screen plifier should be discon-

Oscilloscope Settings nected from the sweep

oscillator. The signal

to be examined should be connected to the

horizontal amplifier of the oscilloscope.

2. An audio oscillator signal should be con-

nected to the vertical amplifier of the oscillo-

scope.

3. By adjusting the frequency of the audio

oscillator a stationary pattern should be ob-

tained on the screen of the oscilloscope. It is

not necessary to stop the pattern, but merely

to slow it up enough to count the loops at the

side of the pattern.

4. Count the number of loops which intersect

an imaginary vertical line AB and the number

of loops which intersect the imaginary hori-

zontal line BC as in figure 12. The ratio of

the number of loops which intersect AB is to

O RATIO I I O RATIO 2I

O RATIO 5

Figure 13

OTHER LISSAJOUS PATTERNS

www.americanradiohistory.com

HANDBOOK Lissajous Figures 177

HHASE DIFFERENCE =O PHASE DIFFERENCE -5 PHASE DIFFERENCE .90. PHASE DIFFERENCE=135

PHASE DIFFERENCE 190 PHASE OIFFERENCE'225 PHASE DIFFERENCE 270 PHASE DIFFERENCE 315

Figure 14

LISSAJOUS PATTERNS OBTAINED FROM THE MAJOR PHASE DIFFERENCE ANGLES

the number of loops which intersect BC as the

frequency of the horizontal signal is to the

frequency of the vertical signal.

Figure 13 shows other examples of Lissa -

jous figures. In each case the frequency ratio

shown is the frequency ratio of the signal on

the horizontal axis to that on the vertical v e r t i c

axis.

Phase Differ- Coming under the heading of

once Patterns Lissajous figures is the method

used to determine the phase

difference between signals of the same fre-

quency. The patterns i n vol v e d take on the

form of ellipses with different degrees of ec-

centricity.

The following steps should be taken to ob-

tain a phase -difference pattern:

1. With no signal input to the oscilloscope,

the spot should be centered on the screen

of the tube.

2. Connect one signal to the vertical ampli-

fier of the oscilloscope, and the other

signal to the horizontal amplifier.

3. Connect a common ground between the

two frequencies under investigation and

the oscilloscope.

4. Adjust the vertical amplifier gain so as

to give about 3 inches of deflection on a

5 inch tube, and adjust the calibrate d

scale of the oscilloscope so that the ver-

tical axis of the scale coincides precise-

ly with the vertical deflection of the spot.

5. Remove the signal from the vertical am-

plifier, being careful not to change the

setting of the vertical gain control.

6. Increase the gain of the horizontal am-

plifier to give a deflection exactly the

same as that to which the vertical am-

plifier control is adjusted (3 inches). Re-

connect the signal to the vertical ampli-

fier.

The resulting pattern will give an accurate

picture of the exact phase difference between

the two waves. If these two patterns are ex-

actly the same frequency but different in phase

and maintain that difference, the pattern on

the screen will remain stationary. If, 'however,

one of these frequencies is drifting slightly,

the pattern will drift slowly through 360°. The

phase angles of 0 °, 45 °, 90 °, 135 °, 180 °,

225 °, 270 °, 315° are shown in figure 14.

Each of the eight patterns in figure 14 can

be analyzed separately by the previously used

TIME -

Figure 15

PROJECTION DRAWING SHOWING THE RE-

SULTANT PHASE DIFFERENCE PATTERN

OF TWO SINE WAVES 45° OUT OF PHASE

www.americanradiohistory.com

178 The Oscilloscope THE RADIO

Y MAXIMUM =I

Y INTERCEPT =O / Y INTERCEPT =.5

SINE s' o

Y MAXIMUM = I Y MAXIMUM= I

¿;-:. SINE e=

= So _ S

Y MAXIMUM. I 'I MA IMU 'I

YINTERCEil,

SINE= SINEe',

s=so s=150-

INTERCEPT'.S

Figure 16

EXAMPLES SHOWING THE USE OF THE FORMULA FOR DETERMINATION OF PHASE

DIFFERENCE

projecti ',n method. Figure 15 shows two sine

waves which differ in phase being projected

on to the screen of the cathode -ray tube. These

signals represent a phase difference of 45 °.

It is extremely important: (1) that the spot

has been centered on the screen of the cathode -

ray tube, (2) that both the horizontal and ver-

tical amplifiers have been adjusted to give

exactly the same gain, and (3) that the cali-

brated scale be originally set to coincide with

the displacement of the signal along the ver-

tical axis. If the amplifiers of the oscilloscope

are not used for conveying the signal to the

deflection plates of the cathode -ray tube, the

coarse frequency switch should be set to hori-

zontal input direct and the vertical input

switch to direct and the outputs of the two

signals must be adjusted to result in exactly

the same vertical deflection as horizontal de-

flection. Once this deflection has been set by

either the oscillator output controls or the am-

plifier gain controls in the oscillograph, it

should not be changed for the duration of the

measurement.

Determination of

the Phase Angle The relation commonly used

in determining the phase

angle between signals is:

Y intercept

Sine 9 Y maximum

Figure 17

TRAPEZOIDAL MODULATION PATTERN

Figure 18

MODULATED CARRIER WAVE PATTERN

Figure 19

PROJECTION DRAWING SHOWING TRAPE-

ZOIDAL PATTERN

www.americanradiohistory.com

HANDBOOK Trapezoidal Pattern 179

MODULATED

CARRIER

TIME

- /SAW TOOTH

SWEEP

Figure 20

PROJECTION DRAWING SHOWING MODO-

LATED CARRIER WAVE PATTERN

where 9 = phase angle between signals

Y intercept = point where ellipse crosses ver-

tical axis measured in tenths of

inches. (Calibrations on the

calibrated screen)

Y maximum = highest vertical point on ellipse

in tenths of inches

Several examples of the use of the formula are

given in figure 16. In each case the Y inter-

cept and Y maximum are indicated together

with the sine of the angle and the angle itself.

For the operator to observe these various pat-

terns with a single signal source such as the

test signal, there are many types of phase

shifters which can be used. Circuits can be

obtained from a number of radio text books.

The procedure is to connect the original sig-

nal to the horizontal channel of the oscillo-

scope and the signal which has passed through

the phase shifter to the vertical channel of

the oscilloscope, and follow the procedure set

forth in this discussion to observe the various

phase shift patterns.

9 -4 Monitoring Transmitter

Performance with the Oscilloscope

The oscilloscope may be used as an aid for

the proper operation of a radiotelephone trans-

mitter, and may be used as an indicator of the

overall performance of the transmitter output

signal, and as a modulation monitor.

Waveforms There are two types of patterns

that can serve as indicators, the

trapezoidal pattern (figure 17) and the modu-

R F. POWER AMPLIFIER

1-0 ANTENNA

EACH 1M, 1

MODULATOR

STAGE 500.11ÁF

10000 V.

TV CAPACITOR

e+

CRO

LC TUNES TO OP-

ERATING FREQUENCY

C L _

NOTE' IF R F. PICKUP IS INSUFFICIENT,

A TUNED CIRCUIT MAY BE USED

AT THE OSCILLOSCOPE AS SHOWN.

Figure 21

MONITORING CIRCUIT FOR TRAPEZOI-

DAL MODULATION PATTERN

laced wave pattern (figure 18). The trapezoidal

pattern is presented on the screen by impress-

ing a modulated carrier wave signal on the ver-

tical deflection plates and the s i g n a l t h a t

modulates the carrier wave signal (the modu-

lating signal) on the horizontal deflection

plates. The trapezoidal pattern can be ana-

lyzed by the method used previously in analy-

zing waveforms. Figure 19 shows how the sig-

nals cause the electron beam to trace out the

pattern.

The modulated wave pattern is accomplished

by presenting a modulated carrier wave on the

vertical deflection plates and by using the

time -base generator for horizontal deflection.

The modulated wave pattern also can be used

for analyzing waveforms. Figure 20 shows how

the two signals cause the electron beam to

trace out the pattern.

The Trapezoidal The oscilloscope connec-

Pattern tions for obtaining a trape-

zoidal pattern are shown in

figure 21. A portion of the audio output of the

transmitter modulator is applied to the hori-

zontal input of the oscilloscope. The vertical

amplifier of the oscilloscope is disconnected,

and a small amount of modulated r -f energy is

coupled directly to the vertical d e f l e c t i o n

plates of the oscilloscope. A small pickup

loop, loosely coupled to the final amplifier

tank circuit and connected to the vertical de-

www.americanradiohistory.com

180 The Oscilloscope T H E RADIO

T EMIN

Figure 22

i E MAX

1

(L ESS THAN 100^; MODULATION)

TRAPEZOIDAL WAVE PATTERN

Figure 23

(100 MODULATION)

Figure 24

(OVER MODULATION)

flection plates by a short length of coaxial

line will suffice. The amount of excitation to

the plates of the oscilloscope may be adjusted

to provide a pattern of convenient size. Upon

modulation of the transmitter, the trapezoidal

pattern will appear. By changing the degree of

modulation of the carrier wave the shape of

the pattern will change. Figures 22 and 23

show the trapezoidal pattern for various de-

grees of modulation. The percentage of modu-

lation may be determined by the following for-

mula:

Modulation percentage =

Emax - Emin x 100

Emax t Emin

where Emax and Emin are defined as in

figure 22.

An overmodulated signal is shown in figure

24.

The Modulated

Wove Pattern The oscilloscope connections

for obtaining a modulated

wave pattern are shown in

R F. POWER AMPLIFIER

TO ANTENNA

CRO

USE INTERNAL

SWEE

a FROM

MODUL ATOR

Figure 25

LC TUNES TO OP-

ERATING FREQUENCY

MONITORING CIRCUIT FOR

MODULATED WAVE PATTERN

figure 25. The internal sweep circuit of the

oscilloscope is applied to the horizontal

plates, and the modulated r -f signal is applied

to the vertical plates, as described before. If

desired, the internal sweep circuit may be sny-

chronized with the modulating signal of the

transmitter by applying a small portion of the

modulator output signal to the external sync

post of the oscilloscope. The percentage of

modulation may be determined in the same

fashion as with a trapezoidal pattern. Figures

26, 27 and 28 show the modulated wave pat-

tern for various degrees of modulation.

9 -5 Receiver I -F Alignment

with an Oscilloscope

The alignment of the i -f amplifiers of a re-

ceiver consists of adjusting all the tuned cir-

cuits to resonance at the intermediate frequen-

cy and at the same time to permit passage of

a predetermined number of side bands. The

best indication of this adjustment is a reso-

nance curve representing the response of the

i -f circuit to its particular range of frequencies.

As a rule medium and low- priced receivers

use i -f transformers whose bandwidth is about

5 kc. on each side of the fundamental frequen-

cy. The response curve of these i -f transform-

ers is shown in figure 29. High fidelity re-

ceivers usually contain i -f transformers which

have a broader bandwidth which is usually 10

kc. on each side of the fundamental. The re-

sponse curve for this type transformer is shown

in figure 30.

Resonance curves such as these can be dis-

played on the screen of an oscilloscope. For

a complete understanding of the procedure it

is important to know how the resonance curve

is traced.

www.americanradiohistory.com

HANDBOOK Receiver Alignment 181

VV

EMIN E MLY.

Figure 26

(LESS THAN 100% MODULATION)

V\I"

1\1\

CARRIER WAVE PATTERN

Figure 27

(100% MODULATION)

Figure 28

(OVER MODULATION)

The Resonance To present a resonance curve

Curve on the on the screen, a frequency -

Screen modulated signal source must

be avail a b l e. This signal

source is a signal generator whose output is

the fundamental i -f frequency which is fre-

quency- modulated 5 to 10 kc. each side of the

fundamental frequency. A signal generator of

this type generally takes the form of an ordi-

nary signal generator with a rotating motor

driven tuned circuit capacitor, called a uwob-

eKC 4 K KC

Figure 29

ecc

FREQUENCY RESPONSE CURVE OF THE

I -F OF A LOW PRICED RECEIVER

ecc eKc

Figure 30

FREQUENCY RESPONSE OF

HIGH- FIDELITY I -F SYSTEM

bulator, or its electronic equivalent, a react-

ance tube.

The method of presenting a resonance curve

on the screen is to connect the vertical chan-

nel of the oscilloscope across the detector

load of the receiver as shown in the detectors

of figure 31 (between point A and ground) and

the time -base generator output to the horizon-

tal channel. In this way the d -c voltage across

the detector load varies with the frequencies

which are passed by the i -f system. Thus, if

the time -base generator is set at the frequency

of rotation of the motor driven capacitor, or

the reactance tube, a pattern resembling fig-

ure 32, a double resonance curve, appears on

the screen.

Figure 32 is explained by considering fig-

ure 33. In half a rotation of the motor driven

capacitor the frequency increases from 445

kc. to 465 kc., more than covering the range

of frequencies passed by the i -f system.

Therefore, a full resonance curve is presented

on the screen during this half cycle of rota-

tion since only half a cycle of the voltage pro-

ducing horizontal deflection has transpired.

In the second half of the rotation the motor

TRIODE DETECTOR

Figure 31

DIODE DETECTOR

CONNECTION OF THE OSCILLOSCOPE

ACROSS THE DETECTOR LOAD

www.americanradiohistory.com

182 The Oscilloscope THE RADIO

Figure 32

DOUBLE RESONANCE CURVE

445 KC 455 KC 46 KC 455 KC 445 KC

Figure 33

DOUBLE RESONANCE ACHIEVED BY

COMPLETE ROTATION OF THE MOTOR

DRIVEN CAPACITOR

Figure 34

SUPER -POSITION OF RESONANCE CURVES

driven capacitor takes the frequency of the

signal in the reverse order through the range

of frequencies passed by the i -f system. In

this interval the time -base generator sawtooth

waveform completes its cycle, drawing the

electron beam further across the screen and

then returning it to the starting point. Subse-

quent cycles of the motor driven capacitor and

the sawtooth voltage merely retrace the same

pattern. Since the signal being viewed is ap-

plied through the vertical amplifier, the sweep

can be synchronized internally.

Some signal generators, particularly those

employing a reactance tube, provide a sweep

output in the form of a sine wave which is

synchronized to the frequency with which the

reactance tube is swinging the fundamental

frequency through its limits, usually 60 cycles

per second. If such a signal is used for hori-

zontal deflection, it is already synchronized.

Since this signal is a sine wave, the response

curve is observed as it sweeps the spot across

the screen from left to right; and it is observed

again as the sine wave sweeps the spot back

again from right to left. Under these condi-

tions the two response curves are superim-

posed on each other and the high frequency

responses of both curves are at one end and

the low frequency response of both curves is

at the other end. The i -f trimmer capacitors

are adjusted to produce a response curve

which is symmetrical on each side of the fun-

damental frequency.

When using sawtooth sweep, the two re-

sponse curves can also be superimposed. If

the sawtooth signal is generated at exactly

twice the frequency of rotation of the motor

driven capacitor, the two resonance curves

will be superimposed (figure 34) if the i -f

transformers are properly tuned. If the two

curves do not coincide the i -f trimmer capaci-

tors should be adjusted. At the point of co-

incidence the tuning is correct. It should be

pointed out that rarely do the two curves agree

perfectly. As a result, optimum adjustment is

made by making the peaks coincide. This lat-

ter procedure is the one generally used in i -f

adjustment. When the two curves coincide, it

is evident that the i -f system responds equal-

ly to signals higher and lower than the funda-

mental i -f frequency.

9 -6 Single Sideband Applications

Measurement of power output and distortion

are of particular importance in SSB transmitter

adjustment. These measurements are related to

the extent that distortion rises rapidly when

the power amplifier is overloaded. The useable

power output of a SSB transmitter is often de-

fined as the maximum peak envelope power

II11IIIIIIU114ulul

m IIIIIIIIIIIIIIIIIIII

Figure 35

SINGLE TONE PRESENTATION

Oscilloscope trace of SSB signal

modulated by single tone (A).

Incomplete carrier supression or

spurious products will show

modulated envelope of (B). The

ratio of supression is:

S - 20 log A +B

A -B

www.americanradiohistory.com

HANDBOOK S.S.B. tpplications 183

R -F INPUT POWER T LIFIER

VER A TEST

INPUT

ENVELOPE

DETECTOR

OSCILLOSCOPE

Figure 36

BLOCK DIAGRAM OF

LINEARITY TRACER

obtainable with a specified signal -to- distortion

ratio. The oscilloscope is a useful instrument

for measuring and studying distortion of all

types that may be generated in single sideband

equipment.

Single Tone When aSSB transmitter is modu-

Observations laced with a single audio tone,

the r -f output should be a single

radio frequency. If the vertical plates of the

oscilloscope are coupled to the output of the

transmitter, and the horizontal amplifier sweep

is set to a slow rate, the scope presentation

will be as shown in figure 35. If unwanted dis-

tortion products or carrier are present, the top

and bottom of the pattern will develop a "rip-

ple" proportional to the degree of spurious

products.

The Linearity The linearity tracer is an aux-

Tracer iliary detector to be used with

an oscilloscope for quick ob-

servation of amplifier adjustments and para-

meter variations. This instrument consists of

two SSB envelope detectors the outputs of

which connect to the horizontal and vertical

inputs of an oscilloscope. Figure 36 shows a

block diagram of atypical linearity test set -up.

A two -tone test signal is normally employed

to supply a SSB modulation envelope, but any

modulating signal that provides an envelope

that varies from zero to full amplitude may be

used. Speech modulation gives a satisfactory

trace, so that this instrument may be used as

a visual monitor of transmitter linearity. It is

particularly useful for monitoring the signal

level and clearly shows when the amplifier

under observation is overloaded. The linearity

trace will be a straight line regardless of the

envelope shape if the amplifier has no dis-

tortion. Overloading causes a sharp break in

the linearity curve. Distortion due to too much

bias is also easily observed and the adjustment

for low distortion can easily be made.

Another feature of the linearity detector is

R -F SSB INPUT

FOM VOLTAGE

DIVIDER OR

PICMUP COIL

GERMANIUM 2.5 MM

DIODE RFC AUDIO OUTPUT

70 OSCILLOSCOPE

Figure 37

SCHEMATIC OF

ENVELOPE DETECTOR

that the distortion of each individual stage

can be observed. This is helpful in trouble-

shooting. By connecting the input envelope

detector to the output of the SSB generator,

the overall distortion of the entire r -f circuit

beyond this point is observed. The unit can

also serve as a voltage indicator which is

useful in making tuning adjustments.

The circuit of a typical envelope detector

is shown in figure 37. Two matched germainum

diodes are used as detectors. The detectors

are not linear at low signal levels, but if the

nonlinearity of the two detectors is matched,

the effect of their nonlinearity on the oscillo-

scope trace is cancelled. The effect of diode

differences is minimized by using a diode load

of 5,000 to 10,000 ohms, as shown. It is im-

portant that both detectors operate at approxi-

mately the same signal level so that their

differences will cancel more exactly. The

operating level should be 1 -volt or higher.

It is convenient to build the detector in a

small shielded enclosure such as an i -f trans-

former can fitted with coaxial input and output

connectors. Voltage dividers can be similarly

constructed so that it is easy to insert the de-

sired amount of voltage attenuation from the

various sources. In some cases it is convenient

to use a pickup loop on the end of a short

length of coaxial cable.

The phase shift of the amplifiers in the os-

cilloscope should be the same and their fre-

quency response should be flat out to at least

twenty times the frequency difference of the

two test tones. Excellent high frequency charac-

teristics are necessary because the rectified

SSB envelope contains harmonics extending

to the limit of the envelope detector's response.

Inadequate frequency response of the vertical

amplifier may cause a little "foot" to appear

on the lower end of the trace, as shown in

figure 38. If it is small, it may be safely neg-

lected.

Another spurious effect often encountered

is a double trace, as shown in figure 39. This

can usually be corrected with an R -C network

placed between one detector and the oscillo-

scope. The best method of testing the detectors

and the amplifiers is to connect the input of

www.americanradiohistory.com

184 The Oscilloscope

Figure 38

EFFECT OF INADEQUATE

RESPONSE OF VERTICAL

AMPLIFIER

Figure 39

DOUBLE TRACE

CAUSED BY PHASE

SHIFT

the envelope detectors in parallel. A perfectly

straight line trace will result when everything

is working properly. One detector is then con-

nected to the other r -f source through a voltage

divider adjusted so that no appreciable change

in the setting of the oscilloscope amplifier

controls is required. Figure 40 illustrates some

typical linearity traces. Trace A is caused by

inadequate static plate current in class A or

class B amplifiers or a mixer stage. To regain

linearity, the grid bias of the stage should be

reduced, the screen voltage should be raised,

or the signal level should be decreased. Trace

B is a result of poor grid circuit regulation

when grid current is drawn, or a result of non-

OUTPUT

SIGNAL

LEVEL

INPUT SIGNAL LEVEL

Figure 41

ORDINATES ON LINEARITY

CURVE FOR 3RD ORDER

DISTORTION EQUATION

linear plate characteristics of the amplifier

tube at large plate swings. More grid swamping

should be used, or the exciting signal should

be reduced. A combination of the effects of A

and B are shown in Trace C. Trace D illustrates

amplifier overloading. The exciting signal

should be reduced.

A means of estimating the distortion level

observed is quite useful. The first and third

order distortion components may be derived by

an equation that will give the approximate

signal -to- distortion level ratio of a two tone

test signal, operating on a given linearity curve.

Figure 41 shows a linearity curve with two

ordinates erected at half and full peak input

signal level. The length of the ordinates et

and e2 may be scaled and used in the following

equation:

Signal -to- distortion ratio in db =20 log 8 e t -e2

2 el -e2

TYPICAL LINEARITY TRACES

Figure 40

TYPICAL LINEARITY

TRACES

www.americanradiohistory.com

CHAPTER TEN

Special Vacuum Tube Circuits

A whole new concept of vacuum tube appli-

cations has been developed in recent years.

No longer are vacuum tubes chained to the

field of communication. This chapter is de-

voted to some of the more common circuits en-

countered in industrial and military applica-

tions of the vacuum tube.

10 -1 Limiting Circuits

The term limiting refers to the removal or

suppression by electronic means of the ex-

tremities of an electronic signal. Circuits

which perform this function are referred to as

limiters or clippers. Limiters are useful in

wave -shaping circuits where it is desirable to

square off the extremities of the applied sig-

nal. A sine wave may be applied to a limiter

circuit to produce a rectangular wave. A

peaked wave may be applied to a limiter cir-

cuit to eliminate either the positive or nega-

tive peaks from the output. Limiter circuits

are employed in FM receivers where it is nec-

essary to limit the amplitude of the signal ap-

plied to the detector. Limiters may be used to

reduce automobile ignition noise in short -wave

receivers, or to maintain a high average level

of modulation in a transmitter. They may also

be used as protective devices to limit input

signals to special circùits.

185

Diode Limiters The characteristics of a

diode tube are such that the

tube conducts only when the plate is at a posi-

tive potential with respect to the cathode. A

positive potential may be placed on the cath-

ode, but the tube will not conduct until the

voltage on the plate rises above an equally

positive value. As the plate becomes more

positive with respect to the cathode, the diode

conducts and passes that portion of the wave

that is more positive than the cathode voltage.

Diodes may be used as either series or paral-

lel limiters, as shown in figure 1. A diode may

be so biased that only a certain portion of the

positive or negative cycle is removed.

Audio Peak An audio peak clipper consisting

Limiting of two diode limiters may be used

to limit the amplitude of an au-

dio signal to a predetermined value to provide

a high average level of modulation without

danger of overmodulation. An effective limiter

for this service is the series -diode gate clip-

per. A circuit of this clipper is shown in fig-

ure 2. The audio signal to be clipped is cou-

pled to the clipper through C,. R, and R2 are

the clipper input and output load resistors.

The clipper plates are tied together and are

connected to the clipping level control, R.,

through the series resistor, R3. R. acts as a

voltage divider between the high voltage sup-

ply and ground. The exact point at which clip-

www.americanradiohistory.com

186 Special Vacuum Tube Circuits THE RADIO

e iN e OUT

E

IAA'

E = VOLTAGE DROP

ACROSS DIODE

e IN e OUT

E

lTV i VT

E= VOLTAGE DROP

ACROSS DIODE

ear

e OVT

E

-A-21

e OUT

Figure 1

VARIOUS DIODE LIMITING CIRCUITS

Series diodes limiting positive and negative peaks are shown in A and 8. Parallel diodes limit-

ing positive and negative peaks are shown in C and D. Parallel diodes limiting above and below

ground are shown in E and F. Parallel diode limiters which pass negative and positive peaks

are shown in G and H.

ping will occur is set by R,, which controls the

positive potential applied to the diode plates.

Under static conditions, a d -c voltage is ob-

tained from R4 and applied through R, to both

plates of the 6AL5 tube. Current flows through

R,, R and divides through the two diode

sections of the 6AL5 and the two load resis-

tors, R, and Rr. All parts of the clipper circuit

are maintained at a positive potential above

ground. The voltage drop between the plate

and cathode of each diode is very small com-

pared to the drop across the 300,000 -ohm re-

sistor (R,) in series with the diode plates.

The plate and cathode of each diode are there-

fore maintained at approximately equal poten-

tials as long as there is plate current flow.

Clipping does not occur until the peak audio

input voltage reaches a value greater than the

static voltages at the plates of the diode.

Assume that R4 has been set to a point that

will give 4 volts at the plates of the 6AL5.

When the peak audio input voltage is less than

4 volts, both halves of the tube conduct at all

times. As long as the tube conducts, its re-

sistance is very low compared with the plate

resistor R,. Whenever a voltage change occurs

across input resistor R the voltage at all of

the tube elements increases or decreases by

the same amount as the input voltage change,

and the voltage drop across R, changes by an

equal amount. As long as the peak input volt-

age is less than 4 volts, the 6AL5 acts merely

as a conductor, and the output cathode is per-

mitted to follow all voltage changes at the in-

put cathode.

If, under static conditions, 4 volts appear at

the diode plates, then twice this voltage (8

volts) will appear if one of the diode circuits

www.americanradiohistory.com

HANDBOOK Clamping Circuits 187

e IN

Ci

0.1

6AL5

R

zoom

Ra

300K

C2

0.1 e OUT

R2

200K

R 00 m

CLIPPING

LEVEL

CONTROL

Bt

Figure 2

THE SERIES -DIODE GATE CLIPPER FOR

AUDIO PEAK LIMITING

eIN E

E WNEDN GRIDO SEDR A IVENPOSITIVE

Figure 3

GRID LIMITING CIRCUIT

is opened, removing its d -c load from the cir-

cuit. As long as only one of the diodes con-

tinues to conduct, the voltage at the diode

plates cannot rise above twice the voltage se-

lected by R. In this example, the voltage can-

not rise above 8 volts. Now, if the input audio

voltage applied through C, is increased to any

peak value between zero and plus 4 volts, the

first cathode of the 6AL5 will increase in volt-

age by the same amount to the proper value be-

tween 4 and 8 volts. The other tube elements

will assume the same potential as the first

cathode. However, the 6AL5 plates cannot in-

crease more than 4 volts above their original

4 -volt static level. When the input voltage to

the first cathode of the 6AL5 increases to

more than plus 4 volts, the cathode potential

increases to more than 8 volts. Since the plate

circuit potential remains at 8 volts, the first

diode section ceases to conduct until the in-

put voltage across R, drops below 4 volts.

When the input voltage swings in a negative

direction, it will subtract from the 4 -volt drop

across R, and decrease the voltage on the in-

put cathode by an amount equal to the input

voltage. The plates and the output cathode will

follow the voltage level at the input cathode

as long as the input voltage does not swing

below minus 4 volts. If the input voltage does

not change more than 4 volts in a negative

direction, the plates of the 6AL5 will also be-

come negative. The potential at the output

cathode will follow the input cathode voltage

and decrease from its normal value of 4 volts

until it reaches zero potential. As the input

cathode voltage decreases to less than zero,

the plates will follow. however, the output

cathode, grounded through R will stop at zero

potential as the plate becomes negative. Con-

duction through the second diode is impossible

under these conditions. The output cathode

remains at zero potential until the voltage at

the input cathode swings back to zero.

The voltage developed across output resis-

tor R2 follows the input voltage variations as

long as the input voltage does not swing to a

peak value greater than the static voltage at

the diode plates, determined by R. Effective

clipping may thus be obtained at any desired

level. The square- topped audio waves generated

by this clipper are high in harmonic content,

but these higher order harmonics may be great-

ly reduced by a low -level speech filter.

Grid Limiters A triode grid limiter is shown

in figure 3. On positive peaks

of the input signal, the triode grid attempts to

swing positive, and the grid- cathode resist-

ance drops to a value on the order of 1000

ohms or so. The voltage drop across R (usu-

ally of the order of I megohm) is large com-

pared to the grid -cathode drop, and the result-

ing limiting action removes the top part of the

positive input wave.

10 -2 Clamping Circuits

A circuit which holds either amplitude ex-

treme of a waveform to a given reference level

e IN

OA POSITIVE CLAMPING CIRCUIT

e OUT

DIODE CONDUCTS

eiN eouT

Figure 4

SIMPLE POSITIVE AND NEGATIVE CLAMPING CIRCUITS

www.americanradiohistory.com

188 Special Vacuum Tube Circuits THE RADIO

e IN

r -, ¡DEFLECTION

COIL

L- J

-100Y

Figure 5

NEGATIVE CLAMPING CIRCUIT EM-

PLOYED IN ELECTROMAGNETIC SWEEP

SYSTEM

CI CHARGE PATH C2 DISCHARGE PATH

Figure 7

THE CHARGE AND DISCHARGE PATHS

IN FREE -RUNNING MULTIVIBRATOR OF

FIGURE 6

B+

Figure 6

BASIC MULTIVIBRATOR CIRCUIT

of potential is called a clamping circuit or a

d -c restorer. Clamping circuits are used after

RC cpupling circuits where the wave f o r m

swing is required to be either above or below

the reference voltage, instead of alternating

on both sides of it (figure 4). Clamping cir-

cuits are usually encountered in oscilloscope

sweep circuits. If the sweep voltage does not

always start from the same reference point,

the trace on the screen does not begin at the

same point on the screen each time the sweep

is repeated and therefore is "jittery." If a

clamping circuit is placed between the sweep

amplifier and the deflection element, the start

of the sweep can be regulated by adjusting the

d -c voltage applied to the clamping tube (fig-

ure 5).

10-3 Multivibrators

The multivibrator, or relaxation oscillator,

is used for the generation of nonsinusoidal

waveforms. The output is rich in harmonics,

but the inherent frequency stability is poor.

The multivibrator may be stabilized by the

introduction of synchronizing voltages of har-

monic or subharmonic frequency.

In its simplest form, the multivibrator is a

simple two -stage resistance -capacitance cou-

pled amplifier with the output of the second

stage coupled through a capacitor to the grid

of the first tube, as shown in figure 6. Since

the output of the second stage is of the proper

polarity to reinforce the input signal applied

to the first tube, oscillations can readily take

place, started by thermal agitation noise and

DIRECT- COUPLED CATHODE

MULTI VIBRATOR

B. B.

NIP

///

SYNCHNONIZING

SIGNAL -

B ©

ELECTRON-COUPLED

MULTIVIBRATOR MULTI VIBRATOR WITH SINE -WAVE

SYNCHRONIZING SIGNAL APPLIED

TO ONE TUBE

Figure 8

VARIOUS FORMS OF MULTIVIBRATOR CIRCUITS

www.americanradiohistory.com

Multivibrators 189

)

BASIC ECCLES-JORDAN TRIGGER

CIRCUIT

Figure 9

ONE -SHOT MULTIVIBRATOR

ECCLES -JORDAN MULTI VIBRATOR CIRCUITS

PULSE

OUTPUT

miscellaneous tube noise. Oscillation is main-

tained by the process of building up and dis-

charging the store of energy in the grid cou-

pling capacitors of the two tubes. The charg-

ing and discharging paths are shown in figure

7. Various forms of multivibrators are shown

in figure 8.

The output of a multivibrator may be used

as a source of square waves, as an electronic

switch, or as a means of obtaining frequency

division. Submultiple frequencies as low as

one -tenth of the injected synchronizing fre-

quency may easily be obtained.

The Eccles- Jordan The Eccles -Jordan trigger

Circuit circuit is shown in figure

9A. This is not a true mul-

tivibrator, but rather a circuit that possesses

two conditions of stable equilibrium. One con-

dition is when V, is conducting and V2 is cut-

off; the other when V2 is conducting and V, is

cutoff. The circuit remains in one or the other

of these two stable conditions with no change

in operating potentials until some external

action occurs which causes the nonconducting

tube to conduct. The tubes then reverse their

functions and remain in the new condition as

long as no plate current flows in the cutoff

tube. This type of circuit is known as a flip -

flop circuit.

C ouT I_ MNI3CIC04E TIMG

CUTOFF

TIME

Figure 10

SINGLE -SWING BLOCKING OSCILLATOR

Figure 9B illustrates a modified Eccles -

Jordan circuit which accomplishes a complete

cycle when triggered with a positive pulse.

Such a circuit is called a one -shot multivibra-

tor. For initial action, V, is cutoff and V2 is

conducting. A large positive pulse applied to

the grid of V, causes this tube to conduct, and

the voltage at its plate decreases by virtue of

the IR drop through R3. Capacitor C2 is charged

rapidly by this abrupt change in V, plate volt-

age, and V, becomes cutoff while V, conducts.

This condition exists until C2 discharges, al-

lowing V2 to conduct, raising the cathode bias

of V, until it is once again cutoff.

A direct, cathode -coupled multivibrator is

shown in figure 8A. RK is a common cathode

resistor for the two tubes, and coupling takes

place across this resistor. It is impossible for

a tube in this circuit to completely cutoff the

other tube, and a circuit of this type is called

a free- running multivibrator in which the con-

dition of one tube temporarily cuts off the

other.

eoUT

RF RF RF

PULSE PULSE PULSE

'1

nnl nns nnl

CUTOFF CUTOFF

TIME TIME

Figure 11

HARTLEY OSCILLATOR USED AS BLOCKING

OSCILLATOR BY PROPER CHOICE OF R, -C,

www.americanradiohistory.com

190 Special Vacuum Tube Circuits THE

eIN Pour e

POSITIVE COUNTING CIRCUIT

eouT

NEGATIVE COUNTING CIRC, POSITIVE NG CIRCUIT WITH

'.ETEA IN .. )N

Figure 12

POSITIVE AND NEGATIVE COUNTING CIRCUITS

ADIO

10 -4 The Blocking Oscillator

A blocking oscillator is any oscillator which

cuts itself off after one or more cycles caused

by the accumulation of a negative charge on

the grid capacitor. This negative charge may

gradually be drained off through the grid re-

sistor of the tube, allowing the circuit to os-

cillate once again. The process is repeated

and the tube becomes an intermittent oscilla-

tor. The rate of such an occurance is deter-

mined by the R -C time constant of the grid cir-

cuit. A single -swing blocking oscillator is

shown in figure 10, wherein the tube is cutoff

before the completion of one cycle. The tube

produces single pulses of energy, the time

between the pulses being regulated by the

discharge time of the grid R -C network. The

self-pulsing blocking oscillator is shown in

figure 11, and is used to produce pulses

of r -f energy, the number of pulses being de-

termined by the timing network in the grid cir-

cuit of the oscillator. The rate at which these

pulses occur is known as the pulse -repetition

frequency, or p.r. /.

10 -5 Counting Circuits

A counting circuit, or frequency divider is

one which receives uniform pulses, represent-

feIN

Vs

e OUTy

Figure 13

STEP -BY -STEP COUNTING CIRCUIT

ing units to be counted, and produces a- volt-

age that is proportional to the frequency of

the pulses. A counting circuit may be used in

conjunction with a blocking oscillator to pro-

duce a trigger pulse which is a submultiple of

of the frequency of the applied pulse. Either

positive or negative pulses may be counted.

A positive counting circuit is shown in figure

12A, and a negative counting circuit is shown

in figure 12B. The positive counter allows a

certain amount of current to flow through R,

each time a pulse is applied to C,.

The positive pulse charges C and makes

the plate of V, positive with respect to its

cathode. V, conducts until the exciting pulse

passes. C, is then discharged by V and the

circuit is ready to accept another pulse. The

average current flowing through R, increases

as the pulse- repetition frequency increases,

and decreases as the p.r.f. decreases.

By reversing the diode connection s, as

shown in figure 12B, the circuit is made to

respond to negative pulses. In this circuit, an

increase in the p.r.f. causes a decrease in the

average current flowing through R which is

opposite to the effect in the positive counter.

e

3

Figure 14

The step -by -step counter used to trigger a

blocking oscillator. The blocking oscillator

serves as a frequency divider.

www.americanradiohistory.com

HANDBOOK R -C Oscillators 191

CI C2 VI

4

C

RB R=LP

LP R4. 4 WATT, 110 V LAMP BULB

a1 xCI =R2 X Ca

Figure 15

THE WIEN- BRIDGE AUDIO OSCILLATOR

Figure 16

THE PHASE -SHIFT OSCILLATOR

!00'1.

A step- counter is similar to the circuits

discussed, except that a capacitor which is

large compared to C, replaces the diode load

resistor. The charge of this condenser is in-

creased during the time of each pulse, pro-

ducing a step voltage across the output (figure

13). A blocking oscillator may be connected

to a step- counter, as shown in figure 14. The

oscillator is triggered into operation when the

voltage across C, reaches a point sufficiently

positive to raise the grid of V, above cutoff.

Circuit parameters may be chosen so that a

count division up to 1/20 may be obtained

with reliability.

10 -6 Resistance -Capacity

Oscillators

In an R -C oscillator, the frequency is de-

termined by a resistance capacity network that

provides regenerative coupling between the

output and input of a feedback amplifier. No

use is made of a tank circuit consisting of in-

ductance and capacitance to control the fre-

quency of oscillation.

The Wien - Bridge oscillator employs a Wien

network in the R -C feedback circuit and is

shown in figure 15. Tube V, is the oscillator

tube, and tube V, is an amplifier and phase -

inverter tube. Since the feedback voltage

through C4 produced by V, is in phase with the

input circuit of V, at all frequencies, oscilla-

tion is maintained by voltages of any frequen-

cy that exist in the circuit. The bridge circuit

is used, then, to eliminate feedback voltages

of all frequencies except the single frequency

desired at the output of the oscillator. The

bridge allows a voltage of only one frequency

to be effective in the circuit because of the

degeneration and phase shift provided by this

circuit. The frequency at which oscillation

occurs is:

1

f- , when R,xC,=R,xC,

2n R, C,

A lamp Lp is used as the cathode resistor

of V, as a thermal stabilizer of the oscillator

amplitude. The variation of the resistance

with respect to current of the lamp bulb holds

the oscillator output voltage at a nearly con-

stant amplitude.

The phase -shi /t oscillator shown in figure

16 is a single tube oscillator using a three

section phase shift network. Each section of

the network produces a phase shift in propor-

tion to the frequency of the signal that passes

through it. For oscillations to be produced,

the signal from the plate of the tube must be

shifted 180 °. Three successive phase shifts

of 60° accomplish this, and the frequency of

oscillation is determined by this phase shift -

A high -mu triode or a pentode must be used in

this circuit. In order to increase the frequency

of oscillation, either the resistance or the

capacitance must be decreased.

Figure 17

THE BRIDGE -TYPE PHASE -SHIFT

OSCILLATOR

www.americanradiohistory.com

192 Special Vacuum Tube Circuits THE RADIO

Figure 18

THE NBS BRIDGE -T

OSCILLATOR CIRCUIT AS USED

IN THE HEATH AG -9 AUDIO

GENERATOR

A bridge -type phase shill oscillator is

shown in figure 17. The bridge is so propor-

tioned that at only one frequency is the phase

shift through the bridge 180 °. Voltages of other

frequencies are fed back to the grid of the tube

out of phase with the existing grid signal, and

are cancelled by being amplified out of phase.

The NBS Bridge -T oscillator developed by

the National Bureau of Standards consists of

a two stage amplifier having two feedback

loops, as shown in figure 18. Loop 1 consists

of a regenerative cathode -to- cathode loop, con-

sisting of Lp, and C3, The bulb regulates the

positive feedback, and tends to stabilize the

output of the oscillator, much as in the man-

ner of the Wien circuit. Loop 2 consists of a

grid -cathode degenerative circuit, containing

the bridge -T. Oscillation will occur at the

null frequency of the bridge, at which frequen-

cy the bridge allows minimum degeneration

in loop 2 (figure 19).

10 -7 Feedback

Feedback amplifiers have been discus s e d

in Chapter 6, section 15 of this Handbook. A

more general use of feedback is in automatic

control and regulating systems. Mechanical

feedback has been used for many years in such

forms as engine speed governors and steering

servo engines on ships.

A simple feedback system for temperature

control is shown in figure 20. This is a cause

-NOTCHFREQUENCY

F- I

2?RC

WHERE

C=1/Ct C2

0-FREQ. OF OSCILLATION

NEC F/B =POS F/B

NEGATIVE

FEEDBACK

(LOOP 2)

f

POSITIVE

FEEDBACK

(LOOP r)

rFREQ OF OSCILLATION

"NOTCH NETWORK

PHASE SHIFT'0

Figure 19

BRIDGE -T FEEDBACK

LOOP CIRCUITS

Oscillation will occur at the null

frequency of the bridge, at which

frequency the bridge allows

minimum degeneration in loop 2.

and effect system. The furnace (F) raises the

room temperature (T) to a predetermined value

at which point the sensing thermostat (TAI)

reduces the fuel flow to the furnace. When the

room temperature drops below the predeter-

mined value the fuel flow is increased by the

thermostat control. An interdependent control

system is created by this arrangement: the

room temperature depends upon the thermostat

action, and the thermostat action depends upon

the room temperature. This sequence of events

may be termed a closed loop feedback system.

FURNACE

(F)

ROOM

TEMPERATURE

(T)

Figure 20

SIMPLE CLOSED LOOP

FEEDBACK SYSTEM

FEEDBACK

(ERROR SIGNAL)

Room temperature (T) controls

fuel supply to furnace (F) by feed-

back loop through Thermostat

(TH) control.

www.americanradiohistory.com

HANDBOOK Feedback 193

PHASE SHIFT INPUT SIGNAL

OF SYSTEM ¡OUTPUT SIGNAL

TIME

FEEDBACK SIGNAL

NO PHASE SHIFT A

I! .. . I

OUTPUT SIGNAL

FEEDBACK

SIGNAL

WITH 180

PHASE

SHIFT

TIME

Figure 21

PHASE SHIFT OF ERROR

SIGNAL MAY CAUSE OSCILLA-

TION INCLOSED LOOP SYSTEM

To prevent oscillation, the gain

of the feedback loop must be

less than unity when the phase

shift of the system reaches 180

degrees.

Error Cancellation A feedback control system

is dependent upon a degree

of error in the output signal, since this error

component is used to bring about the correc-

tion. This component is called the error signal.

The error, or deviation from the desired signal

is passed through the feedback loop to cause

an adjustment to reduce the value of the error

signal. Care must be taken in the design of

the feedback loop to reduce over -control ten-

dencies wherein the correction signal would

carry the sytem past the point of correct oper-

ation. Under certain circumstances the new

error signal would cause the feedback control

to overcorrect in the opposite direction, result-

ing in hunting or oscillation of the closed

loop system about the correct operating point.

Negative feedback control would tend to

dampout spurious system oscillation if it were

not for the time lag or phase shift in the sys-

tem. If the overall phase shift is equal to one -

half cycle of the operating frequency of the

system the feedback will maintain a steady

state of oscillation when the circuit gain is

sufficiently high, as shown in figure 21. In

order to prevent oscillation, the gain figure of

the feedback loop must be less than unity when

the phase shift of the system reaches 180 de-

grees. In an ideal control system the gain of

the loop would be constant throughout the

operating range of the device, and would drop

rapidly outside the range to reduce the band-

width of the control system to a minimum.

The time lag in a closed loop system may

be reduced by using electronic circuits in place

of mechanical devices, or by the use of special

circuit elements having a phase -lead charac-

teristic. Such devices make use of the proper-

ties of a capacitor, wherein the current leads

the voltage applied to it.

www.americanradiohistory.com

CHAPTER ELEVEN

Electronic Computers

Mechanical computing machines were first

produced in the seventeenth century in Eu-

rope although the simple Chinese abacus (a

digital computer) had been in use for cen-

turies. Until the last decade only simple me-

chanical computers (such as adding and book-

keeping machines) were in general use.

The transformation and transmission of the

volume of information required by modern

technology requires that machines assume many

of the information processing systems former-

ly done by the human mind. Computing ma-

chines can perform routine operations more

quickly and more accurately than a human be-

ing, processing mathematical and logistical

data on a production line basis. The computer,

however, cannot create, but can only follow

instructions. If the instructions are in error,

.j

CIIP 0111

011,

THE IBM

COMPUTER

AND

"MEMORY"

The "704" Com-

puter is used with

32,000 "word"

memory storage

unit for research

programs. Heart

of this auxiliary

unit are small,

doughnut- shaped

iron ferrites which

store information

by means of mag-

netism. The unit

is the first of Ms

kind to be instal-

led with I B M's

704 computer.

Components of the

system seen in the

foreground are

(left) card punch

and (right) cord

reader. In the

center of the pic-

ture is the 704's

processing unit.

www.americanradiohistory.com

Digital Computers 195

FARN!R

NORTN SNORE

CORN N!N ^FOX

SOUTN SNORE

Figure 1

SIMPLE PUZZLES IN LOGIC MAY BE

SOLVED BY ELECTRIC COMPUTER.

THE "FARMER AND RIVER"

COMPUTER IS SHOWN HERE.

the computer will produce a wrong answer.

Computers may be divided into two classes:

the digital and the analog. The digital com-

puter counts, and its accuracy is limited only by

the number of significant figures provided for

in the instrument. The analog computer

measures, and its accuracy is limited by the

percentage errors of the devices used, multi-

plied by the range of the variables they repre-

sent.

11 -1 Digital Computers

The digital computer operates in discrete

steps. In general, the mathematical operations

are performed by combinations of additions.

Thus multiplication is performed by repeated

additions, and integration is performed by

summation. The digital computer may be

thought of as an "on -off" device operating

from signals that either exist or do not exist.

The common adding machine is a simple com-

puter of this type. The "on -off" or "yes -no"

type of situation is well suited to switches, elec-

trical relays, or to electronic tubes.

A simple electrical digital computer may be

used to solve the old "farmer and river" prob-

lem. The farmer must transport a hen, a bushel

of corn, and a fox across a river in a small

boat capable of carrying the farmer plus one

other article. If the farmer takes the fox in

the boat with him, the hen will eat the corn.

On the other hand, if he takes the corn, the

fox will eat the hen. The circuit for a simple

computer to solve this problem is shown in

figure 1. When the switches are moved from

"south shore" to "north shore" in the proper

sequence the warning buzzer will not sound.

An error of choice will sound the buzzer.

A second simple "digital computer" is shown

is figure 2. The problem is to find the three

NOTE: ALL BUTTONS NAVE ONE NORMALLY OPEN CONTACT AND

ONE NORMALLY CLOSED CONTACT.

Figure 2

A SEQUENCE COMPUTER.

Three correct buttons will sound the buzzer.

proper push buttons that will sound the buzzer.

The nine buttons are mounted on a board so

that the wiring cannot be seen.

Each switch of these simple computers ex-

ecutes an "on -off" action. When applied to a

logical problem "yes-no" may be substituted

for this term. The computer thus can act out

a logical concept concerned with a simple

choice. An electronic switch (tube) may be

substituted for the mechanical switch to in-

crease the speed of the computer. The early

computers, such as the ENIAC (Electronic Nu-

merical Integrator and Calculator) employed

over 18,000 tubes for memory and registering

circuits capable of "remembering" a 10 -digit

number.

11 -2 Binary Notation

To simply and reduce the cost of the digital

computer it was necessary to modify the system

of operation so that fewer tubes were used per

bit of information. The ENIAC -type computer

requires 50 tubes to register a 5 -digit number.

O 0 0 0 0

O O O O

O O O O O

O O O O

{S. O O 0 0

O O O O O

O O O

O O :;o- O O

Figure 3

BINARY NOTATION MAY BE USED

FOR DIGITAL DISPLAY. BINARY

BOARD ABOVE INDICATES "73092."

www.americanradiohistory.com

196 Electronic Computers THE RADIO

,

O O :O"-- -`

DIGIT TUBE(S)

I 1

a a

3 2+1

4 4

5 +1 4+2

7 +2+1

s 4

9 !+1

10 6+2

1/ 5+2+1

12 +

12 +4+1

14 +4+2

15 +4+2+1

Figure 4

BINARY DECIMAL NOTATION. ONLY

FOUR TUBES ARE REQUIRED TO

REPRESENT DIGITS FROM 1 TO 15.

THE DIGIT "12" IS INDICATED

ABOVE.

The tubes (or their indicator lamps) can be

arranged in five columns of 10 tubes each.

From right to left the columns represent units,

tens, hundreds, thousands, etc. The bottom tube

in each column represents "zero," the second

tube represents "one," the third tube "two,"

and so on. Only one tube in each column is

excited at any given instant. If the number

73092 is to be displayed, tube number seven

in the fifth column is excited, tube number

three in the fourth column, tube number zero

in the third column, etc. as shown in figure 3.

A simpler system employs the binary deci-

mal notation, wherein any number from one

to fifteen can be represented by four tubes.

Each of the four tubes has a numerical value

that is associated with its position in the tube

group. More than one tube of the group may

be excited at once, as illustrated in figure 4.

The values assigned to the tubes in this par-

ticular group are 1, 2, 4, and 8. Additional

tubes may be added to the group, doubling the

notation of the rube thus: 1, 2, 4, 8, 16, 32,

64, 128, 356, etc. Any numerical value lower

than the highest group number can be dis-

played by the correct tube combination.

A third system employs the binary notation

which makes use of a bit (binary digit) repre-

senting a single morsel of information. The

binary system has been known for over forty

centuries, and was considered a mystical revel-

ation for ages since it employed only two sym-

DECIMAL NOTATION BINARY NOTATION

0 o

1 1

2 1,0

3 1,1

1,0.0

s 1,0,1

6 1.1.0

- 7 1.1.1

0 1,0,0,0

0 1,0,0,1

10 1,0,1,0

Figure 5

BINARY NOTATION SYSTEM

REQUIRES ONLY TWO NUMBERS,

"0" AND "1."

bols for all numbers. Computer service usual-

ly employs "zero" and "one" as these symbols.

Decimal notation and binary notation for com-

mon numbers are shown in figure 5. The

binary notation represents 4 -digit numbers

(thousands) with .ten bits, and 7 -digit num-

bers (millions) with 20 bits. Only one elec-

tron tube is required to display an information

bit. The savings in components and primary

power drain of a binary -type computer over

the older ENIAC -type computer is obvious.

Figure 6 illustrates a computer board show-

ing the binary indications from one to ten.

Digital The digital computer is em-

Computer Uses ployed in a "yes -no" situa-

tion. It may be used for

routine calculations that would ordinarily re-

quire enormous man -hours of time, such as

checking stress estimates in aircraft design, or

military logistics, and problems involving the

manipulation of large masses of figures.

DECIMAL NOTATION COMPUTER NOTATION

o

1 O

a o

3 o 0

4 0

5 o o

4 O O

7 o 0 0

. o

+ 0 O

10 0 0

= OFF

Figure 6

BINARY NOTATION AS REPRESENTED

ON COMPUTER BOARD FOR NUMBERS

FROM 1 TO 10.

0 O

www.americanradiohistory.com

HANDBOOK Analog Computers 197

eour=e,+e2

e Our

R,

e, R, e2 R2 / R3

eour- R, +R2 Ri +R2 R, R2

\[R,+ R234-R3J

Figure 7

SUMMATION OF TWO VOLTAGES

BY ELECTRICAL MEANS.

11 -3 Analog Computers

The analog computer represents the use of

one physical system as a model for a second

system that is usually more difficult to con-

struct or to measure, and that obeys the equa-

tions of the same form. The term analog im-

plies similarity of relations or properties be-

tween the two systems. The common slide -rule

is a mechanical analog computer. The speedo-

meter in an automobile is a differential analog

computer, displaying information proportional

to the rate of change of speed of the vehicle.

The electronic analog computer employs cir-

cuits containing resistance, capacitance, and in-

ductance arranged to behave in accordance with

analogous equations. Variables are represented

by d -c voltages which may vary with time.

+150V.

Figure 8

SUMMATION OF TWO VOLTAGES

BY ELECTRONIC MEANS.

Thus complicated problems can be solved by

d -c amplifiers and potentiometer controls in

electronic circuits performing mathematical

functions.

Addition and

Subtraction If a linear network is ener-

gized by two voltage sources

the voltages may be summed

as shown in figure 7. Subtraction of quantities

may be accomplished by using negative and

positive voltages. A -c voltages may be em-

ployed for certain additive circuits, and more

THE

HEATHKIT

ELECTRONIC

ANALOG

DIGITAL

COMPUTER

This "electronic

slide rule" simu-

lates equations or

physical problems

electronically, sub-

stituting one phys-

ical system as a

model for a sec-

ond system that

is usually more

difficult or costly

to construct or

measure, and that

obeys equations of

the some form.

www.americanradiohistory.com

198 Electronic Computers THE RADIO

eour = R2 . e

tt

Figure 9

ELECTRONIC MULTIPLICATION

MAY BE ACCOMPLISHED BY

CALIBRATED POTENTIOMETERS,

WHEN OUTPUT VOLTAGE IS

PROPORTIONAL TO THE INPUT

VOLTAGE MULTIPLIED BY A

CONSTANT (R R1).

eour

complex circuits employ vacuum tubes, as in

figure 8. Synchronous transformers may be

used to add expressions of angular rotation,

and circuits have been developed for adding

time delays, or pulse counting.

Multiplication Electronic multiplication and

and Division division may be accomplished

with the use of potentiome-

ters where the output voltage is proportional to

the input voltage multiplied by a constant

which may be altered by changing the physical

arrangement of the potentiometer (figure 9).

Variable autotransformers may also be used to

perform multiplication.

A simple bridge may be used to obtain an

output that is the product of two inputs divided

by a third input, as shown in figure 10.

Differentiation The time derivative of a

voltage can be expressed as

a charge on a capacitor by:

i =C de

dt (1)

and is shown in figure 11A. The charging cur-

rent is converted into a voltage by the use

of a resistor, R. If the input to the RC circuit

is charging at a uniform rate so that the cur-

rent through C and R is constant, the output

voltage e is:

OUTPUT'\, RI x R3

R2

Figure 10

ELECTRONIC MULTIPLICATION BY

BRIDGE CIRCUIT PROVIDES

OUTPUT THAT IS PRODUCT OF

TWO INPUTS DIVIDED BY A

THIRD INPUT.

c

e

Figure 11

ELECTRONIC DIFFERENTIATION

The time derivative of a voltage can be

expressed as a charge on a capacitor (A).

Operational amplifier (B) employs feed

back principle for short differentiation

time.

our

RC- d (2)

For highest accuracy, a small RC product

should be used, permitting the maximum pos-

sible differentiation time. The output of the

differentiator may be amplified to any suitable

level. A more accurate differentiating device

makes use of an operational amplifier. This

unit is a high gain, negative feedback d -c

amplifier (discussed in section 11 -4) with the

resistance portion of the RC product appearing

in the feedback loop of the amplifier (figure

11B). A shorter differentiation time may be

employed if the junction point between R and

C could be held at a constant potential. The

feedback amplifier shown inverts the output

signal and applies it to the RC network, hold-

ing the junction potential constant.

Integration Integration is a process of ac-

cumulation, or summation, and

requires a device capable of storing physical

quantities. A capacitor will store an electrical

charge and will give the time integral of a

current in respect to a voltage:

eo= 1 idt (3)

In most computers, the input signal is in

the form of a voltage, and the input charging

current of the capacitor must be taken through

a series resistance as in figure 12. If the inte-

grating time is short the charging current is

approximately proportional to the input vol-

tage. The charging current may be made a

true measure of the input voltage by the use

Figure 12

SIMPLE

INTEGRATION

CIRCUIT e c

T eouT

O

Making use of

charging current

of capacitor.

www.americanradiohistory.com

HANDBOOK Operational Amplifier 199

- +

Figure 3

"MILLER FEEDBACK" INTEGRATOR

SUITABLE FOR COMPUTER USE.

eaur

Figure 14

R -L NETWORK USED FOR

INTEGRATION PURPOSES.

o

Figure 15

OPERATIONAL AMPLIFIER ( -A)

Mathematical operations may be performed

by any operational amplifier, usually a

stable, high -gain d -c amplifier, such as

shown in Figure 16.

of an operational amplifier wherein the ca-

pacitance portion of the RC product appears

in the feedback loop of the amplifier, holding

the junction point between R and C at a con-

stant potential. A simple integrator is shown

in figure 13 employing the Miller feedback

principle. Integration is also possible with an

RL network (figure 14).

11 -4 The Operational

Amplifier

Mathematical operations are performed by

using a high gain d -c amplifier, termed an

operational ampli fier. The symbol of this unit

is a triangle, with the apex pointing in the di-

rection of operation ( figure 15). The gain of

such an amplifier is -A, so:

e

eo=-Aeg,oreg= - A (4)

If -A approaches infinity, e, will be ap-

proximately zero. In practice this condition is

realized by using amplifiers having open loop

gains of 30,000 to 60,000. If ea is set at 100

volts, e, will be of the order of a few milli-

volts. Thus, considering eg equal to zero:

Rr

,oreo-

RI - Rf RI

which may be written:

eI (5)

e0 _ - Keg, where K = R I (6 )

This amounts to multiplication by a constant

coefficient, since RI and Rr may be fixed in

value. The circuit of a typical operational am-

plifier is shown in figure 16.

Amplifier

Operation Two voltages may be added by

the amplifier, as shown in figure

17. Keeping in mind that eg is

e,

u (Rv)

BIAS -4í0V

GAIN= -A

Figure 16

HIGH GAIN OPERATIONAL AMPLIFIER, SUCH AS USED IN HEATH COMPUTER.

-ze0 V

www.americanradiohistory.com

200 Electronic Computers THE RADIO

o

e

RF D ea

Figure 17

TWO VOLTAGES MAY BE

ADDED BY SUMMATION

AMPLIFIER.

o eo

essentially at zero (ground) potential:

Re Re

eo = - e + e2

R. R3 (7)

or, eo= - K. e,+K:e (8)

where K = B and K: R` =

As long as e., does not exceed the input

range of the amplifier, any number of inputs

may be used:

eo= - Re Rf RI

Rl ei + Á= ei + - - - + Ro en

F =eft/

Figure 19

"MASS- SPRING-

DAMPER" PROBLEM

MAY BE SOLVED BY

ELECTRICAL ANALOGY

WITH SIMPLE

COMPUTER.

11 -5 Solving Analog

Problems

By combining the above operations in var-

ious ways, problems of many kinds may be

solved .For example, consider the mass- spring-

damper assembly shown in figure 19. The mass

M is connected to the spring which has an

elastic constant K. The viscous damping con-

stant is C. The vertical displacement is y. The

sum of. the forces acting on mass M is:

(10) f (t) =M d$ + C d +Ky (16)

where f (t) is the applied force, or forcing

function.

The first step is to set up the analog com-

puter circuit so as to obtain an output voltage

proportional to y for a given input voltage

proportional to f (t) . Equation (16) may be

rewritten in the form:

(12) M dt' -C dt -Ky +f (t)

en

or eo = -Rr+ Ro (11)

Integration is performed by replacing the

feedback resistor Re with a capacitor Ce, as

shown in figure 18. For this circuit (with ea

approximately zero):

dc ei

t dt Ri

but g=Cr en, so d= Ce do and R = Cf deo

Thus: deo

1

and eo

(17)

If, in the analog circuit, there is a voltage

(13) equal to M d - it can be converted to -dt

by passing it through an integrator circuit hav-

ing an RC time constant equal to M. This re-

sulting voltage can be passed through a second

integrator stage with unit time constant which

will haue an output volage equal to y. The

voltages representing y, and f (t) can

1 eidt (14)

R Cr

Ri Cr ei dt (15)

RI

o AN

e o

oeo

o

Figure 18

INTEGRATION

Performed by

Summation Amplifier

by replacing feed

back resistor with a

capacitor.

then be summed to give -C do - Ky + f (t)

which is the right hand side of equation (17) ,

r

and therefore equal to M dt . Connecting the

www.americanradiohistory.com

HANDBOOK Analog Problems 201

SET VOLTAGE TO Y =YO As

TIME t'0 (INITIAL

DISPLACEMENT)

dZ

M d t 1

F(t)

-2.dr dt -Cdt -Kr+f(t

Cdt - fit)

Figure 20

ANALOG SOLUTION FOR "MASS- SPRING-

DAMPER" PROBLEM OF FIGURE 19.

TO DISPLAY

OSCILLOSCOPE

output of the summing amplifier (A3) to the

input of the first as shown in figure 20 satis-

fies the equation.

To obtain a solution to the problem, the

initial displacement and velocity must be speci-

fied. This is done by charging the integrating

capacitors to the proper voltages. Three oper-

ational amplifiers and a summing amplifier

are required.

A second problem that may be solved by

the analog computer is the example of a freely

falling body. Disregarding air resistance, the

body will fall (due to the action of gravity)

with a constant acceleration. The equation

describing this action is:

F =mg =m dt' (18)

Integration of equation (18) will give the

velocity, or dt , and integration a second time

will give displacement, or y. The block dia-

gram of a suitable computer for this problem

is shown in figure 21.

If a voltage proportional to g and hence to

d'

dY is introduced into the first amplifier, the

output of that unit will he - dt , or the

e,= G

-H ---

Ri R2

dZr -dr r

dtZ dt e o

Figure 21

ANALOG COMPUTER FOR

"FREELY FALLING BODY"

PROBLEM.

velocity. That, in turn, will become y, or dis-

tance, at the output of the second amplifier.

Before the problem can be solved on the

computer it is necessary to determine the time

of solution desirable and the output amplitude

of the solution. The time of solution is de-

termined by the RC constant of the integrating

amplifiers. If RC is set at unity, computer

time is equal to real time. The computer time

desirable is determined by the method of read-

out. When using an oscilloscope for read -out,

a short solution time is desirable. For a re-

corder, longer solution time is better.

Suppose, for example, in the problem of the

falling body, the distance of fall in 2.5 seconds

is desired. Using an RC constant of 1 would

give a solution time of 2.5 seconds. This would

be acceptable for a recorder but is slow for an

oscilloscope. A convenient time of solution for

the 'scope would be 25 milliseconds. This is

1/100 of the real time, so an RC constant of

.01 is needed. This can be obtained with C

equal to 0.1 pfd, and R equal to 100,000

ohms.

It is now necessary to choose an input volt-

age which will not overdrive the amplifiers.

The value of g is known to be approximately

32 ft /sec. /sec. A check indicates that if we

set g equal to 32 volts, the voltage represent-

ing the answer will exceed 100 volts. Since

the linear response of the amplifier is only

100 volts, this is undesirable. An input of 16

volts, however, should permit satisfactory op-

eration of the amplifiers. Output voltages near

zero should also be avoided. In general, output

voltage should be about 50 volts or so, with

amplifier gains of 20 to 60 being preferable.

Thus, for this particular problem the time -

scale factor and amplitude -scale factor have

www.americanradiohistory.com

202 Electronic Computers THE RADIO

16 V.

100 K

A RELAY A' RELAY

d2r

dt2 dr dt tr eo= Y

Figure 22

ANALOG SOLUTION FOR

"FALLING BODY PROBLEM"

OF FIGURE 21.

6

eo

VOLTS

-24

-32

-40

-46

36

64

72

TIME IN SECONDS

4 (c s)

1

)

16

DISTANCE

32 N

FEET

46

64

-80

_'98 1'00)

112

28

,44

Figure 23

READ -OUT SOLUTION OF "FREELY

FALLING BODY" PROBLEM.

been chosen. The problem now looks like

figure 22.

To solve the problem, relays A and A' are

opened. The solution should now appear on

the oscilloscope as shown in figure 23. The

solution of the problem leaves the integrating

capacitors charged. It is necessary to remove

this charge before the problem can be rerun.

This is done by closing relays A and A'.

e,

Figure 24

LIMITING CIRCUIT TO SIMULATE

NON -LINEAR FUNCTIONS SUCH AS

ENCOUNTERED IN HYSTERESIS,

BACKLASH, AND FRICTION

PROBLEMS.

eo

11 -6 Non -linear Functions

Problems are frequently encountered in

which non -linear functions must be simulated.

Non -linear potentiometers may be used to sup-

ply an unusual voltage source, or diodes may

be used as limiters in those problems in which

a function is defined differently for different

regions of the independent variable. Such a

function might be defined as follows:

e.. = - K1, e, - KI (19)

e.. = et, - KT,, e_, K: (20)

e.. = K2, el ,, K: (21)

where K, and K: are constants.

Various limiting circuits can be used, one of

which is shown in figure 24. This is a series

limiter circuit which is simple and does not

require special components. Commonly en-

countered problems requiring these or similar

limiting techniques include hysteresis, back-

lash, and certain types of friction.

C NOTE: REPLACE C WITN A I MEC. RESISTOR

FOR FUNCTION SETUP

P

SLOPE CONTROL

2 6AL5

BREAK

CONTROL

VOLTAGE

MEG

RAMP - FUNCTION

GENERATOR

620 K

1 E

620K t SIGN CHANGING

AMPLIFIER SUMMING

AMPLIFIER

X OUTPUT

Y OUTPUT

Figure 25

SIMPLIFIED DIAGRAM OF FUNCTIONAL GENERATOR TO APPROXIMATE NON- LINEAR

FUNCTIONS.

www.americanradiohistory.com

HANDBOOK Non -Linear Functions 203

o XI Xz 7X3

Figure 26

TYPICAL NON -LINEAR FUNCTION

WHICH MAY BE SET UP WITH

FUNCTION GENERATOR.

The Function A function generator may be

Generator used to approximate almost

any non -linear function. This

is done by use of straight line segments which

are combined to approximate curves such as

are found in trigonometric functions as well

as in stepped functions. In a typical generator

ten line segments are used, five in the plus -x

direction, and five in the minus -x direction.

Five 6ÁL5 double diodes are used. Each line

segment is generated by a modified bridge

circuit (figure 25). A ramp function or volt-

age is fed into one arm of the bridge while

the opposite arm is connected to a biased diode.

The other two arms of the bridge combine to

form the output. The voltage appearing across

one of these arms is fed through a sign- chang-

ing amplifier and then summed with the volt-

age appearing at the opposite arm. If the arm

of potentiometer P (the slope control) is set

in the center, the bridge will be balanced and

Figure 27

ELECTRONIC PACKAGE IN

DIGITAL COMPUTER.

Stylized diagram of tube package. Lines car-

rying negative pulses are marked by a small

circle at each end. Gates are indicated by

a semi -circle with "pins" for each input.

the output of the summing amplifier will be

zero. If, on the other hand, potentiometer P

is adjusted one way or the other from center,

the bridge will be unbalanced and the sum-

ming amplifier output will vary linearly with

respect to the input in either a positive or neg-

ative y direction, depending upon which side

of center potentiometer P is set.

The break voltage, or value of x at which a

straight line segment will begin is set by

biasing the diode to the particular voltage level

or value of x desired. The ramp function gen-

erator has either a positive or negative input

which because of the 180 degree phase shift

in the amplifier, gives a minus- or plus -x out-

put respectively. A typical function such as

shown in figure 26 may be set up with the

function generator. The initial condition volt-

IBM's new "608," the

first completely tran-

sistorized calculator

for commercial appli-

cations, operates

without the use of a

single vacuum tube.

Transistors - -tiny ger-

manium devices that

perform many of the

functions of conven-

tional vacuum tubes

-make possible 50%

reduction in compu-

ter -unit size and a

90% reduction in

power requirements

over a comparable

IBM tube -model ma-

chine. They are

mounted, along with

related circuitry, on

banks of printed wir-

ing panels in the

608.

The machine's inter-

nal storage, or "mem-

ory," is made up of

magnetic cores -mi-

nute, doughnut -shap-

ed objects that can

"remember" informa-

tion indefinitely, and

recall it for use in

calculations in a few

millionths of a

second.

www.americanradiohistory.com

204 Electronic Computers THE RADIO

age is set to the value of X. The break -voltage

control is increased until the output of the

summing amplifier increases abruptly, indi-

cating the diode is conducting. The input

voltage from the initial condition power supply

is set to the value X: The slope control (P)

is now set to value Y2. A second function gen-

erator may be used to set points X.1 and X.,

using the break -voltage control and the initial

condition voltage adjustments. Points XS and

X. are finally set with a third generator. The

x- output of the function generator system may

be read on an oscilloscope, using the x- output

of the ramp- function generator amplifier as

the horizontal sweep for the oscilloscope.

11 -7 Digital Circuitry

Digital circuits dealing with "and," "or,"

and "not" situations may be excited by electri-

cal pulses representing these logical operations.

Sorting and amplifying the pulses can be ac-

complished by the use of electronic packages,

such as shown in figure 27. Logical operations

may be accomplished by diode -resistor gates

operating into an amplifier stage. Negative and

positive output pulses from the amplifier are

obtained through diode output gates. The driv-

ing pulses may be obtained from a standard

oscillator, operating at or near 1 mc.

A circuit of a single digital package is shown

in figure 28. Other configurations, such as a

"flip-flop" may be used. Many such packages

can be connected in series to form operational

circuits. The input "and" and "or" gates are

biased to conduction by external voltages. The

"and" diode gate transmits a pulse only when

all the input terminals are pulsed positively,

and the "or" diode gate transmits a positive

pulse applied to any one of its input terminals.

The input pulses pass through the gates and

drive the amplifier stage, which delivers an

amplified pulse to the positive and negative

output gates, and to accompanying memory

circuits.

Memory Circuits A memory circuit consists

of some sort of delay line

which is capable of holding an information

pulse for a period of time. The amount of

delay is proportional to the frequency of the

input signal. A "long" transmission line may

be used as a delay line with the signal being

removed from the "far" end of the line after

being delayed an interval equal to the time

of transmission along the line. Lines of this

type are constructed in the manner of a

coaxial cable, except that the inner conductor

is a long, thin coil of wire. Other memory

circuits make use of magnetostrictive or piezo-

electric effects to retard the pulse. Information

may also be stored in electrostatic storage

tubes, upon magnetic recording tape, and in

ferro- magnetic cores capable of holding 10,000

bits of information.

"AND GATES

INPUT SI

INPUT 2

INPUTSS

INPUT *4 0

INPUT *5

INPUTS 6

INPUT* 7

INPUT

"OR"

GATE CLAMPING

DIODES 6AN5

LIMITING

DIODES

POSITIVE PULSE

NEGATIVE PULSE

Figure 28

TYPICAL DIGITAL PACKAGE SHOWING INPUT AND OUTPUT DIODE GATES AND

PULSE AMPLIFIER.

www.americanradiohistory.com

CHAPTER TWELVE

Radio Receiver Fundamentals

A conventional reproducing device such as

a loudspeaker or a pair of earphones is in-

capable of receiving directly the intelligence

carried by the carrier wave of a radio trans-

mitting station. It is necessary that an addi-

tional device, called a radio receiver, be

placed between the receiving antenna and the

loudspeaker or headphones.

Radio receivers vary widely in their com-

plexity and basic design, depending upon the

intended application and upon economic fac-

tors. A simple radio receiver for reception of

radiotelephone signals can consist of an ear-

phone, a silicon or germanium crystal as a

carrier rectifier or demodulator, and a length

of wire as an antenna. However, such a re-

ceiver is highly insensitive, and offers no

significant discrimination between two sig-

nals in the same portion of the spectrum.

On the other hand, a dual -diversity receiver

designed for single -sideband reception and

employing double or triple detection might

occupy several relay racks and would cost

many thousands of dollars. Ilowever, conven-

tional communications receivers are interme-

diate in complexity and performance between

the two extremes. This chapter is devoted to

the principles underlying the operation of

such conventional communications receivers.

205

12 -1 Detection or

Demodulation

A detector or demodulator is a device for

removing the modulation (demodulating) or

detecting the intelligence carried by an in-

coming radio wave.

Radiotelephony Figure 1 illustrates an ele-

Demodulation mentary form of radiotele-

phony receiver employing a

diode detector. Energy from a passing radio

wave will induce a voltage in the antenna and

cause a radio- frequency current to flow from

antenna to ground through coil Lt. The alter-

nating magnetic field set up around L, links

with the turns of L2 and causes an r -f current

to flow through the parallel -tuned circuit,

1..2-C1. %hen variable capacitor C, is adjusted

so that the tuned circuit is resonant at the

frequency of the applied signal, the r -f voltage

is maximum. This r -f voltage is applied to the

diode detector where it is rectified into a vary-

ing direct current and passed through the ear-

phones. The variations in this current corres-

pond to the voice modulation placed on the

signal at the transmitter. As the earphone

diaphragms vibrate back and forth in accord-

www.americanradiohistory.com

206 Radio Receiver Fundamentals THE RADIO

'Ili lquirlli

O

L+ L2

- GROUND

Figure 1

ELEMENTARY FORM OF RECEIVER

This is the basis of the "crystal set" type of re-

ceiver, although a vacuum diode may be used in

place of the crystal diode. The tank circuit L2 -C1

is tuned to the frequency it is desired to receive.

The bypass capacitor across the phones should

have a low reactance to the carrier frequency be-

ing received, but a high reactance to the modula-

tion on the received signal.

ance with the pulsating current they audibly

reproduce the modulation which was placed

upon the carrier wave.

The operation of the detector circuit is

shown graphically above the detector circuit

in figure 1. The modulated carrier is shown

at A, as it is applied to the antenna. B repre-

sents the same carrier, increased in amplitude,

as it appears across the tuned circuit. In C

the varying d -c output from the detector is

seen.

Radiotelegraphy Since a c -w telegraphy sig-

Reception nal consists of an unmodu-

lated carrier which is inter-

rupted to form dots and dashes, it is apparent

that such a signal would not be made audible

by detection alone. While the keying is a form

of modulation, it is composed of such low fre-

quency components that the keying envelope

itself is below the audible range for hand key-

ing speeds. Some means must be provided

whereby an audible tone is heard while the

unmodulated carrier is being received, the tone

stopping immediately when the carrier is in-

terrupted.

The most simple means of accomplishing

this is to feed a locally generated carrier of

a slightly different frequency into the same

detector, so that the incoming signal will mix

with it to form an audible beat note. The dif-

ference frequency, or heterodyne as the beat

note is known, will of course stop and start

in accordance with the incoming c -w radio-

telegraph signal, because the audible hetero-

dyne can exist only when both the incoming

and the locally generated carriers are present.

TRIODE

AUDIO OUTPUT

I- e +

PLATE- TICKLER REGENERATION WITH "THROTTLE'

CONDENSER REGENERATION CONTROL.

PENTODE AUDIO OUTPUT

-e +e

CATHODE-TAP REGENERATION WITH SCREEN VOLTAGE

REGENERATION CONTROL.

Figure 2

REGENERATIVE DETECTOR CIRCUITS

Regenerative detectors are seldom used at the pre-

sent time due to their poor selectivity. However,

they do illustrate the simplest type of receiver

which may be used either for radiophone or radio-

telegraph reception.

The Autodyne The local signal which is used

Detector to beat with the desired c -w

signal in the detector may be

supplied by a separate low -power oscillator

in the receiver itself, or the detector may be

made to self -oscillate, and thus serve the

dual purpose of detector and oscillator. A de-

tector which self -oscillates to provide a beat

note is known as an autodyne detector, and

the process of obtaining feedback between the

detector plate and grid is called regeneration.

An autodyne detector is most sensitive when

it is barely oscillating, and for this reason

a regeneration control is always included in

the circuit to adjust the feedback to the proper

amount. The regeneration control may be either

a variable capacitor or a variable resistor,

as shown in figure 2.

With the detector regenerative but not os-

cillating, it is also quite sensitive. When the

circuit is adjusted to operate in this manner,

modulated signals may be received with con-

siderably greater strength than with a non -

regenerative detector.

www.americanradiohistory.com

HANDBOOK Superregenerative Detectors 2J7

12 -2 Superregenerative

Receivers

At ultra -high frequencies, when it is de-

sired to keep weight and cost at a minimum,

a special form of the regenerative receiver

known as the superregenerator is often used

for radiotelephony reception. The superregen-

erator is essentially a regenerative receiver

with a means provided to throw the detector

rapidly in and out of oscillation. The frequency

at which the detector is made to go in and out

of oscillation varies with the frequency to be

received, but is usually between 20,000 and

500,000 times a second. This superregenera-

tive action considerably increases the sen-

sitivity of the oscillating detector so that the

usual "background hiss" is greatly amplified

when no signal is being received. This hiss

diminishes in proportion to the strength of the

received signal, loud signals eliminating the

hiss entirely.

Quench There are two systems in common

Methods use for causing the detector to break

in and out of oscillation rapidly. In

one, a separate interruption -frequency oscilla-

tor is arranged so as to vary the voltage rapid-

ly on one of the detector tube elements (usual-

ly the plate, sometimes the screen) at the high

rate necessary. The interruption- frequency

oscillator commonly uses a conventional tick-

ler- feedback circuit with coils appropriate for

its operating frequency.

The second, and simplest, type of super -

regenerative detector circuit is arranged so

as to produce its own interruption frequency

oscillation, without the aid of a separate tube.

The detector tube damps (or "quenches ") itself

out of signal- frequency oscillation at a high

rate by virtue of the use of a high value of

grid leak and proper size plate- blocking and

grid capacitors, in conjunction with an excess

of feedback. In this type of "self- quenched"

detector, the grid leak is quite often returned

to the positive side of the power supply (through

the coil) rather than to the cathode. A repre-

sentative self -quenched superregenerative de-

tector circuit is shown in figure 3.

Except where it is impossible to secure

sufficient regenerative feedback to permit

superregeneration, the self -quenching circuit

is to be preferred; it is simpler, is self- adjust-

ing as regards quenching amplitude, and can

have good quenching wave form. To obtain

as good results with a separately quenched

superregenerator, very careful design is re-

quired. However, separately quenched circuits

are useful when it is possible to make a cer-

tain tube oscillate on a very high frequency

but it is impossible to obtain enough regenera-

tion for self -quenching action.

TO AUDIO

AMPLIFIER

Figure 3

SUPERREGENERATIVE DETECTOR CIRCUIT

A sell -quenched superregenerative detector such

as illustrated above is capable of giving good

sensitivity in the v -h -f range. However, the circuit

has the disadvantage that its selectivity is rela-

tively poor. Also, such o circuit should be pre-

ceded by an r -I stage to suppress the radiation of

o signal by the oscillating detector.

The optimum quenching frequency is a func-

tion of the signal frequency. As the operating

frequency goes up, so does the optimum quench-

ing frequency. hen the quench frequency is

too low, maximum sensitivity is not obtained.

When it is too high, both sensitivity and selec-

tivity suffer. In fact, the optimum quench fre-

quency for an operating frequency below 15 Mc.

is in the audible range. This makes the super -

regenerator impracticable for use on the lower

frequencies.

The high background noise or hiss which

is heard on a properly designed superregener-

ator when no signal is being received is not

the quench frequency component; it is tube

and tuned circuit fluctuation noise, indicating

that the receiver is extremely sensitive.

A moderately strong signal will cause the

background noise to disappear completely,

because the superregenerator has an inherent

and instantaneous automatic volume control

characteristic. This same a -v -c characteristic

makes the receiver comparatively insensitive

to impulse noise such as ignition pulses -a

highly desirable feature. This characteristic

also results in appreciable distortion of a re-

ceived radiotelephone signal, but not enough

to affect the intelligibility.

The selectivity of a superregenerator is

rather poor as compared to a superheterodyne,

but is surprisingly good for so simple a re-

ceiver when figured on a percentage basis

rather than absolute kc. bandwidth.

FM Reception A. superregenerative receiver

will receive frequency modu-

lated signals with results comparing favorably

with amplitude modulation if the frequency

swing of the FM transmitter is sufficiently

high. For such reception, the receiver is de-

tuned slightly to either side of resonance.

www.americanradiohistory.com

208 Radio Receiver Fundamentals THE RADI O

22 MC

OUTPUT

0+100V

AUOIO

Figure 4

THE FREMODYNE SUPERREGENERATIVE

SUPERHETERODYNE DETECTOR FOR

FREQUENCY MODULATED SIGNALS

Superregenerative receivers radiate a strong,

broad, and rough signal. For this reason, it is

necessary in most applications to employ a

radio frequency amplifier stage ahead of the

detector, with thorough shielding throughout

the receiver.

The Fremodyne The Hazel tin e- Fremodyne

Detector superregenerative circuit is

expressly designed for re-

ception of FM signals. This versatile circuit

combines the action of the superregenerative

receiver with the superhetrodyne, converting

FM signals directly into audio signals in one

double triode tube (figure 4). One section of

the triode serves as a superregenerative mixer,

producing an i -f of 22 Mc., an i -f amplifier, and

a FM detector. The detector action is accom-

plished by slope detection tuning on the side

of the i -f selectivity curve.

This circuit greatly reduces the radiated

signal, characteristic of the superregenerative

detector, yet provides many of the desirable

features of the superregenerator. The pass -

band of the Fremodyne detector is about

400 kc.

12 -3 Superheterodyne

Receivers

Because of its superiority and nearly uni-

versal use in all fields of radio reception, the

RF

AMPLIFIER

I_ -

FREONCY

IOSCILLATORI

(FOR C.W.) I

AUDIO

OUTPUT

IINTERMED. I

REOUENCV "SECOND* AUDIO I

AMPLIFIER 1 DETECTOR 'AMPLIFIER

Figure 5

ESSENTIAL UNITS OF A

SUPERHETERODYNE RECEIVER

The basic portions of the receiver are shown

in solid blocks. Practicable receivers em-

ploy the dotted blocks and also usually in-

clude such additional circuits os a noise

limiter, on a -v -c circuit, and a crystal filter

in the i -f amplifier.

theory of operation of the superheterodyne

should be familiar to every radio student and

experimenter. The following discussion con-

cerns superheterodynes for amplitude- modula-

tion reception. It is, however, applicable in

part to receivers for frequency modulation.

Principle of In the superheterodyne, the in-

Operation coming signal is applied to a

mixer consisting of a non -linear

impedance such as a vacuum tube or a diode.

The signal is mixed with a steady signal gen-

erated locally in an oscillator stage, with the

result that a signal bearing all the modulation

applied to the original signal but of a fre-

quency equal to the difference between the

local oscillator and incoming signal frequen-

cies appears in the mixer output circuit. The

output from the mixer stage is fed into a fixed -

tuned intermediate -frequency amplifier, where

it is amplified and detected in the usual man-

ner, and passed on to the audio amplifier. Fig-

ure 5 shows a block diagram of the fundamen-

tal superheterodyne arrangement. The basic

components are shown in heavy lines, the

simplest superheterodyne consisting simply

of these three units. However, a good com-

munications receiver will comprise all of the

elements shown, both heavy and dotted blocks.

Superheterodyne The advantages of super -

Advantages heterodyne reception are

directly attributable to the

use of the fixed -tuned intermediate -frequency

(i -f) amplifier. Since all signals are converted

to the intermediate frequency, this section of

the receiver may be designed for optimum se-

lectivity and high amplification. High ampli-

fication is easily obtained in the intermediate -

frequency amplifier, since it operates at a

www.americanradiohistory.com

HANDBOOK The Superhetrodyne 209

INPIp VARIABLE-1/

PENTODE

1 MEG.

To A V C

BY -PASS CAPACITORS 05 TO 0.1 JIPD.

Figure 6

TYPICAL I -F AMPLIFIER STAGE

relatively low frequency, where conventional

pentode -type tubes give adequate voltage gain.

A typical i -f amplifier is shown in figure 6.

From the diagram it may be seen that both

the grid and plate circuits are tuned. The tuned

circuits used for coupling between i -f stages

are known as i-I transformers. These will be

more fully discussed later in this chapter.

Choice of Inter- The choice of a frequency

mediate Frequency for the i -f amplifier in-

volves several considera-

tions. One of these considerations concerns

selectivity; the lower the intermediate fre-

quency the greater the obtainable selectivity.

On the other hand, a rather high intermediate

frequency is desirable from the standpoint of

image elimination, and also for the reception

of signals from television and FM transmitters

and modulated self -controlled oscillators, all

of which occupy a rather wide band of frequen-

cies, making a broad selectivity characteristic

desirable. Images are a pecularity common to

all superheterodyne receivers, and for this

reason they are given a detailed discussion

later in this chapter.

While intermediate frequencies as low as

50 kc. are used where extreme selectivity is

a requirement, and frequencies of 60 Mc. and

above are used in some specialized forms of

receivers, most present -day communications

superheterodynes use intermediate frequencies

around either 455 kc. or 1600 kc.

Home -type broadcast receivers almost al-

ways use an i -f in the vicinity of 455 kc.,

while auto receivers usually use a frequency

of about 262 kc. The standard frequency for

the i -f channel of FM receivers is 10.7 Mc.

Television receivers use an i -f which covers

the band between about 21.5 and 27 Mc., al-

though a new band between 41 and 46 Mc. is

coming into more common usage.

Arithmetical Aside from allowing the use of

Selectivity fixed -tuned band -pass amplifier

stages, the superheterodyne has

an overwhelming advantage over the tuned

radio frequency (t -r -f) type of receiver because

of what is commonly known as arithmetical

selectivity.

This can best be illustrated by considering

two receivers, one of the t -r -f type and one of

the superheterodyne type, both attempting to

receive a desired signal at 10,000 kc. and

eliminate a strong interfering signal at 10,010

kc. In the t -r -f receiver, separating these two

signals in the tuning circuits is practically

impossible, since they differ in frequency by

only 0.1 per cent. However, in a superhetero-

dyne with an intermediate frequency of, for ex-

ample, 1000 kc., the desired signal will be

converted to a frequency of 1000 kc. and the

interfering signal will be converted to a fre-

quency of 1010 kc., both signals appearing at

the input of the i -f amplifier. In this case, the

two signals may be separated much more read-

ily, since they differ by 1 per cent, or 10 times

as much as in the first case.

The Converter The converter stage, or mixer,

Stage of a superheterodyne receiver

can be either one of two types:

(1) it may use a single envelope converter

tube, such as a 6K8, 6SA7, or 6BE6, or (2) it

may use two tubes, or two sets of elements in

the same envelope, in an oscillator -mixer ar-

rangement. Figure 7 shows a group of circuits

of both types to illustrate present practice

with regard to types of converter stages.

Converter tube combinations such as shown

in figures 7A and 7B are relatively simple and

inexpensive, and they do an adequate job for

most applications. With a converter tube such

as the 6SB7 -Y or the 6BA7 quite satisfactory

performance may be obtained for the reception

of relatively strong signals (as for example

FM broadcast reception) up to frequencies in

excess of 100 Mc. However, the equivalent in-

put noise resistance of such tubes is of the

order of 200,000 ohms, which is a rather high

value indeed. So such tubes are not suited for

operation without an r -f stage in the high -

frequency range if weak -signal reception is

desired.

The 6L7 mixer circuit shown in figure 7C,

and the 6BA7 circuit of figure 71), also are

characterized by an equivalent input noise re-

sistance of several hundred thousand ohms, so

that these also must be preceded by one or

more r -f stages with a fairly high gain per

stage if a low noise factor is desired of the

complete receiver.

However, the circuit arrangements shown

at figures 7F and 6F are capable of low -noise

operation within themselves, so that these

circuits may be fed directly from the antenna

without an r -f stage and still provide a good

noise factor to the complete receiver. Note

www.americanradiohistory.com

210 Radio Receiver Fundamentals THE RADIO

6SÁ7, 6SB7Y,

6BE6. 6BÁ7 roar

AMP

+250 V

2 ULF

Figure 7

TYPICAL FREQUENCY- CONVERTER (MIXER) STAGES

The relative advantages of the different circuits are discussed in the text

+ 50 V.

that both these circuits use control -grid in-

jection of both the incoming signal and the

local- oscillator voltage. Hence, paradoxically,

circuits such as these should be preceded by

an r -f stage if local- oscillator radiation is to

be held to any reasonable value of field in-

tensity.

Diode Mixers As the frequency of operation of

a superheterodyne receiver is in-

creased above a few hundred megacycles the

signal -to -noise ratio appearing in the plate

circuit of the mixer tube when triodes or pen-

todes are employed drops to a prohibitively

low value. At frequencies above the upper -fre-

quency limit for conventional mixer stages,

mixers of the diode type are most commonly

employed. The diode may be either a vacuum -

tube heater diode of a special u -h -f design

such as the 9005, or it may be a crystal diode

of the general type of the 1N21 through 1N28

series.

12 -4 Mixer Noise'

and Images

The effects of mixer noise and images are

troubles common to all superheterodynes. Since

www.americanradiohistory.com

HANDBOOK Mixer Characteristics 211

both these effects can largely be obviated by

the same remedy, they will be considered to-

gether.

Mixer Noise Mixer noise of the shot- effect

type, which is evidenced by a

hiss in the audio output of the receiver, is

caused by small irregularities in the plate cur-

rent in the mixer stage and will mask weak

signals. Noise of an identical nature is gen-

erated in an amplifier stage, but due to the

fact that the conductance in the mixer stage

is considerably lower than in an amplifier

stage using the same tube, the proportion of

inherent noise present in a mixer usually is

considerably greater than in an amplifier stage

using a comparable tube.

Although this noise cannot be eliminated,

its effects can be greatly minimized by plac-

ing sufficient signal- frequency amplification

having a high signal -to -noise ratio ahead of

the mixer. This remedy causes the signal out-

put from the mixer to be large in proportion to

the noise generated in the mixer stage. In-

creasing the gain after the mixer will be of no

advantage in eliminating mixer noise difficul-

ties; greater selectivity after the mixer will

help to a certain extent, but cannot be carried

too far, since this type of selectivity decreases

the i -f band -pass and if carried too far will

not pass the sidebands that are an essential

part of a voice -modulated signal.

Triode Mixers A triode having a high trans -

conductance is the quietest

mixer tube, exhibiting somewhat less gain but

a better signal -to -noise ratio than a compar-

able multi -grid mixer tube. However, below 30

Mc. it is possible to construct a receiver that

will get down to the atmospheric noise level

without resorting to a triode mixer. The addi-

tional difficulties experienced in avoiding

pulling, undesirable feedback, etc., when using

a triode with control -grid injection tend to make

multi -grid tubes the popular choice for this

application on the lower frequencies.

On very high frequencies, where set noise

rather than atmospheric noise limits the weak

signal response, triode mixers are more widely

used. A 6J6 miniature twin triode with grids

in push -pull and plates in parallel makes an

excellent mixer up to about 600 Mc.

Injection The amplitude of the injection volt -

Voltage age will affect the conversion trans -

conductance of the mixer, and there-

fore should be made optimum if maximum sig-

nal -to -noise ratio is desired. If fixed bias is

employed on the injection grid, the optimum

injection voltage is quite critical. If cathode

bias is used, the optimum voltage is not so

critical; and if grid leak bias is employed, the

optimum injection voltage is not at all critical

just so it is adequate. Typical optimum in-

jection voltages will run from 1 to 10 volts for

control grid injection, and 45 volts or so for

screen or suppressor grid injection.

Images There always are two signal frequen-

cies which will combine with a given

frequency to produce the same difference fre-

quency. For example: assume a superhetero-

dyne with its oscillator operating on a higher

frequency than the signal, which is common

practice in present superheterodynes, tuned to

receive a signal at 14,100 kc. Assuming an

i -f amplifier frequency of 450 kc., the mixer

input circuit will be tuned to 14,100 kc., and

the oscillator to 14,100 plus 450, or 14,550 kc.

Now, a strong signal at the oscillator frequen-

cy plus the intermediate frequency (14,550

plus 450, or 15,000 kc.) will also give a dif-

ference frequency of 450 kc. in the mixer out -

put and will be heard also. Note that the image

is always twice the intermediate frequency

away from the desired signal. Images cause

repeat points on the tuning dial.

The only way that the image could be elimi-

nated in this particular case would be to make

the selectivity of the mixer input circuit, and

any circuits preceding it, great enough so that

the 15,000 -kc. signal never reaches the mixer

grid in sufficient amplitude to produce inter-

ference.

For any particular intermediate frequency,

image interference troubles become increas-

ingly greater as the frequency to which the

signal- frequency portion of the receiver is

tuned is increased. This is due to the fact that

the percentage difference between the desired

frequency and the image frequency decreases

as the receiver is tuned to a higher frequency.

The ratio of strength between a signal at the

image frequency and a signal at the frequency

to which the receiver is tuned producing equal

output is known as the image ratio. The higher

this ratio, the better the receiver in regard to

image- interference troubles.

kith but a single tuned circuit between the

mixer grid and the antenna, and with 400 -500

kc. i -f amplifiers, image ratios of 60 db and

over are easily obtainable up to frequencies

around 2000 kc. Above this frequency, greater

selectivity in the mixer grid circuit through

the use of additional tuned circuits between

the mixer and the antenna is necessary if a

good image ratio is to be maintained.

12 -5 Z -F Stages

Since the necessLry tuned circuits between

the mixer and the antenna can be combined

with tubes to form r -f amplifier stages, the

www.americanradiohistory.com

212 Radio Receiver Fundamentals THE RADIO

INPUT C

PENTODE

Figure 8

TYPICAL PENTODE R -F AMPLIFIER STAGE

reduction of the effects of mixer noise and the

increasing of the image ratio can be accom-

plished in a single section of the receiver.

When incorporated in the receiver, this sec-

tion is known simply as an r -/ amplifier; when

it is a separate unit with a separate tuning

control it is often known as a preselector.

Either one or two stages are commonly used

in the preselector or r -f amplifier. Some pre -

selectors use regeneration to obtain still

greater amplification and selectivity. An r -f

amplifier or preselector embodying more than

two stages rarely ever is employed since two

stages will ordinarily give adequate gain to

override mixer noise.

R -F Stages in Generally speaking, atmos-

the V -H -F Range pheric noise in the frequency

range above 30 Mc. is quite

low -so low, in fact, that the noise generated

within the receiver itself is greater than the

noise received on the antenna. Hence it is of

the greatest importance that internally gener-

ated noise be held to a minimum in a receiver.

At frequencies much above 300 Mc. there is

not too much that can be done at the present

state of the art in the direction of reducing

receiver noise below that generated in the con-

verter stage. But in the v -h -f range, between

30 and 300 Mc., the receiver noise factor in a

well designed unit is determined by the char-

acteristics of the first r -f stage.

The usual v -h -f receiver, whether for com-

munications or for FM or TV reception, uses

a miniature pentode for the first r -f amplifier

stage. The 6AK5 is the best of presently avail-

able types, with the 6CB6 and the 6DC6 close-

ly approaching the 6AK5 in performance. But

when gain in the first r -f stage is not so im-

portant, and the best noise factor must be ob-

tained, the first r -f stage usually uses a triode.

Shown in figure 9 are four commonly used

types of triode r -f stages for use in the v -h -f

range. The circuit at (A) uses few components

and gives a moderate amount of gain with very

low noise. It is most satisfactory when the

first r -f stage is to be fed directly from a low-

O

GROUNDED-GRID

6AB4, 6J6,

6J4, 12AT7 --

CATHODE- COUPLED

LOW NOISE

CASCODE

6J6

70

+120v.

i

+120 V

Lo

6BK7,6B07AOR6BZ7

+120 V.

200V.

Figure 9

TYPICAL TRIODE V -H -F

R -F AMPLIFIER STAGES

Triode r -f stages contribute the least amount of

noise output for a given signal level, hence their

frequent use in the v -h -f range.

impedance coaxial transmission line. Figure

9 (B) gives somewhat more gain than (A), but

requires an input matching circuit. The effec-

tive gain of this circuit is somewhat reduced

when it is being used to amplify a broad band

of frequencies since the effective Gm of the

cathode -coupled dual tube is somewhat less

www.americanradiohistory.com

HANDBOOK The Cascode Amplifier 213

IA MC. MC. 455 KC.

TUNABLE TUNABLE FII% DULATOR

R F. MIXER I.F. MIXER AND

AMPLIFIER AMPLIFIER AMPLIFIER AUDIO

10 MC. CRYSTAL

OSCILLATOR

1A4S5KC

VARIABLE

OSCILLATOR 3545 KC.

455 KC. I I

I I

l I

II

I I

L

1

50 KC.

MIXER nx MIXER FIXED DEMODULATOR

AND

AUDIO

I.I.

AMPLIFIER AMPLIFIER

VARIABLE

OSCILLATOR

CONVENTIONAL COMMUNICATIONS

RECEIVER

11 FIXED

OSCILLATOR 505 BC.

I I

II HIGHLY SELECTIVE ACCESSORY I F.

II AMPLIFIER AND DEMODULATOR (Q5'ER)I

_JL

Figure 10

TYPICAL DOUBLE -CONVERSION SUPERHETERODYNE RECEIVERS

Illustrated at (A) is the basic circuit of a commercial double- conversion superheterodyne receiver. At (B) is

illustrated the application of on accessory sharp i -f channel for obtaining improved selectivity from a con-

ventional communications receiver through the use of the double -conversion principle.

than half the Gm of either of the two tubes

taken alone.

The Cascode The Cascode r -f amplifier, de-

Amplifier veloped at the MIT Radiation

Laboratory during World War II,

is a low noise circuit employing a grounded

cathode triode driving a grounded grid triode,

as shown in figure 9C. The stage gain of such

a circuit is about equal to that of a pentode

tube, while the noise figure remains at the low

level of a triode tube. Neutralization of the

first triode tube is usually unnecessary below

50 Mc. Above this frequency, a definite im-

provement in the noise figure may be obtained

Through the use of neutralization. The neutral-

izing coil, LN, should resonate at the operat-

ing frequency with the grid -plate capacity of

the first triode tube.

The 6B(27A and 6BZ7 tubes are designed for

use in cascode circuits, and may be used to

good advantage in the 144 Mc. and 220 Mc. am-

ateur bands (figure 9D). For operation at higher

frequencies, the 6A)4 tube is recommended.

Double Conversion As previously mentioned,

the use of a higher inter-

mediate frequency will also improve the image

ratio, at the expense of i -f selectivity, by

placing the desired signal and the image far-

ther apart. To give both good image ratio at

the higher frequencies and good selectivity in

the i -f amplifier, a system known as double

conversion is sometimes employed. In this sys-

tem, the incoming signal is first converted to

a rather high intermediate frequency, and then

amplified and again converted, this time to a

much lower frequency. The first intermediate

frequency supplies the necessary wide separa-

tion between the image and the desired sig-

nal, while the second one supplies the bulk of

the i -f selectivity.

The double -conversion system, as illus-

trated in figure 10, is receiving two general

types of application at the present time. The

first application is for the purpose of attaining

extremely good stability in a communications

receiver through the use of crystal control of

the first oscillator. In such an arrangement,

as used in several types of Collins receivers,

the first oscillator is crystal controlled and is

followed by a tunable i -f amplifier which then

is followed by a mixer stage and a fixed -tuned

i -f a m p l i f i e r on a touch lower frequency.

Through such a circuit arrangement the sta-

bility of the complete receiver is equal to the

www.americanradiohistory.com

214 Radio Receiver Fundamentals THE RADIO

stability of the oscillator which feeds the sec-

ond mixer, while the selectivity is determined

by the bandwidth of the second, fixed i -f am-

plifier.

The second common application of the

double- conversion principle is for the purpose

of obtaining a very high degree of selectivity

in the complete communications receiver. In

this type of application, as illustrated in fig-

ure 10 (B), a conventional communications re-

ceiver is modified in such a manner that its

normal i -f amplifier (which usually is in the

450 to 915 kc. range) instead of being fed to

a demodulator and then to the audio system,

is alternatively fed to a fixed -tune mixer stage

and then into a much lower intermediate fre-

quency amplifier before the signal is demodu-

lated and fed to the audio system. The acces-

sory i -f amplifier system (sometimes called a

Q5'er) normally is operated on a frequency of

175 kc., 85 kc., or 50 kc.

12 -6 Signal- Frequency

Tuned Circuits

The signal- frequency tuned circuits in high -

frequency superheterodynes and tuned radio

frequency types of receivers consist of coils

of either the solenoid or universal -wound types

shunted by variable capacitors. It is in these

tuned circuits that the causes of success or

failure of a receiver often lie. The universal -

wound type coils usually are used at frequen-

cies below 2000 kc.; above this frequency the

single -layer solenoid type of coil is more

satisfactory.

Impedance The two factors of greatest signi-

and Q ficance in determining the gain -

per -stage and selectivity, respec-

tively, of a tuned amplifier are tuned- circuit

impedance and tuned -circuit Q. Since the re-

sistance of modern capacitors is low at ordi-

nary frequencies, the resistance usually can

be considered to be concentrated in the coil.

The resistance to be considered in making Q

determinations is the r -f resistance, not the

d -c resistance of the wire in the coil. The lat-

ter ordinarily is low enough that it may be

neglected. The increase in r -f resistance over

d -c resistance primarily is due to skin effect

and is influenced by such factors as wire size

and type, and the proximity of metallic objects

or poor insulators, such as coil forms with

high losses. Higher values of Q lead to better

selectivity and increased r -f voltage across

the tuned circuit. The increase in voltage is

due to an increase in the circuit impedance

with the higher values of Q.

R F INPUT C

Figure 11

ILLUSTRATING "COMMON POINT"

BY- PASSING

To reduce the detrimental effects of cathode cir-

cuit inductance in v -h -f stages, all by -pass ca-

pacitors should be returned to the cathode terminal

at the socket. Tubes with two cathode leads can

give improved performance if the grid return is

made to one cathode terminal while the plate and

screen by -pass returns are made to the cathode

terminal which is connected to the suppressor

within the tube.

Frequently it is possible to secure an in-

crease in impedance in a resonant circuit, and

consequently an increase in gain from an am-

plifier stage, by increasing the reactance

through the use of larger coils and smaller

tuning capacitors (higher L/C ratio).

Input Resistance Another factor which in-

fluences the operation of

tuned circuits is the input resistance of the

tubes placed across these circuits. At broad-

cast frequencies, the input resistance of most

conventional r -f amplifier tubes is high enough

so that it is not bothersome. But as the fre-

quency is increased, the input resistance be-

comes lower and lower, until it ultimately

reaches a value so low that no amplification

can be obtained from the r -f stage.

The two contributing factors to the decrease

in input resistance with increasing frequency

are the transit time required by an electron

traveling between the cathode and grid, and

the inductance of the cathode lead common

to both the plate and grid circuits. As the

frequency becomes higher, the transit time

can become an appreciable portion of the time

required by an r -f cycle of the signal voltage,

and current will actually flow into the grid.

The result of this effect is similar to that

which would be obtained by placing a resis-

tance between the tube's grid and cathode.

Superheterodyne Because the oscillator in a

Tracking superheterodyne operate s

"offset" from the other front

end circuits, it is necessary to make special

provisions to allow the oscillator to track

when similar tuning capacitor sections are

www.americanradiohistory.com

HANDBOOK Tuning Circuits 215

MIXER

PADDING CAPACITOR

TUNING CAPACITOR

OSCILLATOR

SERIES TRACKING CAPACITOR

Figure 12

SERIES TRACKING EMPLOYED

IN THE H -F OSCILLATOR OF A

SUPERHETERODYNE

The series tracking capacitor permits the use of

identical gangs in a ganged capacitor, since the

tracking capacitor slows down the rate of frequen-

cy change in the oscillator so that a constant dif-

ference in frequency between the oscillator and

the r -f stage (equal to the i -f amplifier frequency)

may be maintained.

ganged. The usual method of obtaining good

tracking is to operate the oscillator on the

high- frequency side of the mixer and use a

series tracking capacitor to slow down the

tuning rate of the oscillator. The oscillator

tuning rate must be slower because it covers

a smaller range than does the mixer when both

are expressed as a percentage of frequency.

At frequencies above 7000 kc. and with ordi-

nary intermediate frequencies, the difference

in percentage between the two tuning ranges

is so small that it may be disregarded in re-

ceivers designed to cover only a small range,

such as an amateur band.

A mixer and oscillator tuning arrangement

in which a series tracking capacitor is provid-

ed is shown in figure 12. The value of the

tracking capacitor varies considerably with

different intermediate frequencies and tuning

ranges, capacitances as low as .0001 pfd.

being used at the lower tuning -range frequen-

cies, and values up to .01 µfd. being used at

the higher frequencies.

Superheterodyne receivers designed to cover

only a single frequency range, such as the

standard broadcast band, sometimes obtain

tracking between the oscillator and the r -f cir-

cuits by cutting the variable plates of the os-

cillator tuning section to a different shape

from those used to tune the r -f stages.

Frequency Range The frequency to which a

Selection receiver responds may be

varied by changing the size

of either the coils or the capacitors in the tun-

ing circuits, or both. In short -wave receivers

Figure 13

BANDSPREAD CIRCUITS

Parallel bandspread is illustrated at (A) and (B),

series bandspread at (C), and tq,ped.coil band-

spread at (D),

a combination of both methods is usually em-

ployed, the coils being changed from one band

to another, and variable capacitors being used

to tune the receiver across each band. In prac-

tical receivers, coils may be changed by one

of two methods: a switch, controllable from

the panel, may be used to switch coils of dif-

ferent sizes into the tuning circuits or, alter-

natively, coils of different sizes may be

plugged manually into the receiver, the con-

nection into the tuning circuits being made by

suitable plugs on the coils. Where there are

several plug -in cods for each band, they are

sometimes arranged to a single mounting strip,

allowing them all to be plugged in simultan-

eously.

Bandspread In receivers using large tuning

Tuning capacitors to cover the short-

wave spectrum with a minimum

of coils, tuning is likely to be quite difficult,

owing to the large frequency range covered by

a small rotation of the variable capacitors.

To alleviate this condition, some method of

slowing down the tuning rate, or bandspread -

ing, must be used.

Quantitatively, bandspread is usually desig-

nated as being inversely proportional to the

range covered. Thus, a large amount of band-

spread indicates that a small frequency range

is covered by the bandspread control. Con-

versely, a small amount of bandspread is taken

to mean that a large frequency range is covered

by the bandspread dial.

Types of Bandspreading systems are of

Bandspread two general types: electrical and

mechanical. Mechanical systems

are exemplified by high -ratio dials in which

the tuning capacitors rotate much more slowly

www.americanradiohistory.com

216 Radio Receiver Fundamentals THE RADIO

than the dial knob. In this system, there is

often a separate scale or pointer either con-

nected or geared to the dial knob to facilitate

accurate dial readings. However, there is a

practical limit to the amount of mechanical

bandspread which can be obtained in a dial

and capacitor before the speed- reduction unit

and capacitor bearings become prohibitively

expensive. Hence, most receivers employ a

combination of electrical and mechanical band -

spread. In such a system, a moderate reduc-

tion in the tuning rate is obtained in the dial,

and the rest of the reduction obtained by elec-

trical bandspreading.

Stray Circuit In this book and in other radio

Capacitance literature, mention is sometimes

made of stray or circuit capaci-

tance. This capacitance is in the usual sense

defined as the capacitance remaining across

a coil when all the tuning, bandspread, and

padding capacitors across the circuit are at

their minimum capacitance setting.

Circuit capacitance can be attributed to two

general sources. One source is that due to the

input and output capacitance of the tube when

its cathode is heated. The input capacitance

varies somewhat from the static value when

the tube is in actual operation. Such factors

as plate load impedance, grid bias, and fre-

quency will cause a change in input capaci-

tance. However, in all except the extremely

high -transconductance tubes, the published

measured input capacitance is reasonably close

to the effective value when the tube is used

within its recommended frequency range. But

in the high -transconductance types the effec-

tive capacitance will vary considerably from

the published figures as operating conditions

are changed.

The second source of circuit capacitance,

and that which is more easily controllable, is

that contributed by the minimum capacitance

of the variable capacitors across the circuit

and that due to capacitance between the wir-

ing and ground. In well -designed high -fre-

quency receivers, every effort is made to keep

this portion of the circuit capacitance at a

minimum since a large capacitance reduces

the tuning range available with a given coil

and prevents a good L/C ratio, and conse-

quently a high- impedance tuned circuit, from

being obtained.

A good percentage of stray circuit capaci-

tance is due also to distributed capacitance

of the coil and capacitance between wiring

points and chassis.

Typical values of circuit capacitance may

run from 10 to 75 µpfd. in high- frequency re-

ceivers, the first figure representing concen-

tric -line receivers with acorn or miniature

tubes and extremely small tuning capacitors,

and the latter representing all -wave sets with

bandswitching, large tuning capacitors, and

conventional tubes.

12 -7 I -F Tuned Circuits

I -f amplifiers usually employ bandpass cir-

cuits of some sort. A bandpass circuit is ex-

actly what the name implies -a circuit for pass-

ing a band of frequencies. Bandpass arrange-

ments can be designed for almost any degree

of selectivity, the type used in any particular

case depending upon the ultimate application

of the amplifier.

I.F Intermediate frequency trans -

Transformers formers ordinarily consist of

two or more tuned circuits and

some method of coupling the tuned circuits

together. Some representative arrangements

are shown in figure 14. The circuit shown at

A is the conventional i -f transformer, with the

coupling, M, between the tuned circuits being

provided by inductive coupling from one coil

to the other. As the coupling is increased, the

selectivity curve becomes less peaked, and

when a condition known as critical coupling

is reached, the top of the curve begins to flat-

ten out. When the coupling is increased still

more, a dip occurs in the top of the curve.

The windings for this type of i -f transformer,

as well as most others, nearly always consist

of small, flat universal -wound pies mounted

either on a piece of dowel to provide an air

core or on powdered -iron for iron core i -f trans-

formers. The iron -core transformers generally

have somewhat more gain and better selectivity

than equivalent air -core units.

The circuits shown at figure 14 -B and C are

quite similar. Their only difference is the type

of mutual coupling used, an inductance being

used at B and a capacitance at C. The opera-

tion of both circuits is similar. Three reson-

ant circuits are formed by the components. In

B, for example, one resonant circuit is formed

by L C1, C, and L2 all in series. The fre-

quency of this resonant circuit is just the same

as that of a single one of the coils and capaci-

tors, since the coils and capacitors are simi-

lar in both sides of the circuit, and the reson-

ant frequency of the two capacitors and the

two coils all in series is the same as that of

a single coil and capacitor. The second reson-

ant frequency of the complete circuit is deter-

mined by the characteristics of each half of

the circuit containing the mutual coupling de-

vice. In B, this second frequency will be lower

than the first, since the resonant frequency of

L C, and the inductance, M, or L2, C, and M

is lower than that of a single coil and capaci-

www.americanradiohistory.com

HANDBOOK I -F Amplifiers 217

tor, due to the inductance of M being added to

the circuit.

The opposite effect takes place at figure

14 -C, where the common coupling impedance

is a capacitor. Thus, at C the second reson-

ant frequency is higher than the first. In either

case, however, the circuit has two resonant fre-

quencies, resulting in a flat- topped selectivity

curve. The width of the top of the curve is

controlled by the reactance of the mutual

coupling component. As this reactance is in-

creased (inductance made greater, capacitance

made smaller), the two resonant frequencies

become further apart and the curve is broad-

ened.

In the circuit of figure 14 -D, there is induc-

tive coupling between the center coil and each

of the outer coils. The result of this arrange-

ment is that the center coil acts as a sharply

tuned coupler between the other two. A signal

somewhat off the resonant frequency of the

transformer will not induce as much current

in the center coil as will a signal of the cor-

rect frequency. When a smaller current is in-

duced in the center coil, it in turn transfers

a still smaller current to the output coil. The

effective coupling between the outer coils in-

creases as the resonant frequency is ap-

proached, and remains nearly constant over a

small range and then decreases again as the

resonant band is passed.

Another very satisfactory bandpass arrange-

ment, which gives a very straight- sided, flat -

topped curve, is the negative- mutual arrange-

ment shown at figure 14 -E. Energy is trans-

ferred between the input and output circuits in

this arrangement by both the negative- mutual

coils, M, and the common capacitive reactance,

C. The negative- mutual coils are interwound

on the same form, and connected backward.

Transformers usually are made tunable over

a small range to permit accurate alignment in

the circuit in which they are employed. This

is accomplished either by means of a variable

capacitor across a fixed inductance, or by

means of a fixed capacitor across a variable

inductance. The former usually employ either

a mica -compression capacitor (designated

"mica tuned "), or a small air dielectric vari-

able capacitor (designated "air tuned "). Those

which use a fixed capacitor usually employ a

powdered iron core on a threaded rod to vary

the inductance, and are known as "permea-

bility tuned."

Shape Factor It is obvious that to pass modu-

lation sidebands and to allow

for slight drifting of the transmitter carrier fre-

quency and the receiver local oscillator, the

i -f amplifier must pass not a single frequency

but a band of frequencies. The width of this

pass band, usually 5 to 8 kc. at maximum

M M

E

Figure 14

I -F AMPLIFIER COUPLING

ARRANGEMENTS

The interstoge coupling arrangements illustrated

above give a better shape factor (more straight

sided selectivity curve) than would the some num-

ber of tuned circuits coupled by means of tubes.

width in a good communications receiver, is

known as the pass band, and is arbitrarily

taken as the width between the two frequen-

cies at which the response is attenuated 6 db,

or is "6 db down." However, it is apparent

that to discriminate against an interfering sig-

nal which is stronger than the desired signal,

much more than 6 db attenuation is required.

The attenuation arbitrarily taken to indicate

adequate discrimination against an interfering

signal is 60 db.

www.americanradiohistory.com

218 Radio Receiver Fundamentals THE RADIO

Figure 15

I -F PASS BAND OF TYPICAL

COMMUNICATIONS RECEIVER

It is apparent that it is desirable to have

the bandwidth at 60 db down as narrow as

possible, but it must be done without making

the pass band (6 db points) too narrow for satis-

factory reception of the desired signal. The

figure of merit used to show the ratio of band-

width at 6 db down to that at 60 db down is

designated shape factor. The ideal i -f curve,

a rectangle, would have a shape factor of 1.0.

The i -f shape factor in typical communications

receivers runs from 3.0 to 5.5.

The most practicable method of obtaining a

low shape factor for a given number of tuned

circuits is to employ them in pairs, as in fig-

ure 14 -A, adjusted to critical coupling (the

value at which two resonance points just be-

gin to become apparent). If this gives too

sharp a "nose" or pass band, then coils of

lower Q should be employed, with the coupling

maintained at the critical value. As the Q is

lowered, closer coupling will be required for

critical coupling.

Conversely if the pass band is too broad,

coils of higher Q should be employed, the

coupling being maintained at critical. If the

pass band is made more narrow by using looser

coupling instead of raising the Q and main -

taninig critical coupling, the shape factor will

not be as good.

The pass band will not be much narrower

for several pairs of identical, critically coupled

tuned circuits than for a single pair. However,

the shape factor will be greatly improved as

each additional pair is added, up to about 5

pairs, beyond which the improvement for each

additional pair is not significant. Commer-

cially available communications receivers of

L C R

Figure 16

ELECTRICAL EQUIVALENT OF

QUARTZ FILTER CRYSTAL

The crystal is equivalent to o very large value of

inductance in series with small values of capaci-

tance and resistance, with a larger though still

small value of capacitance across the whole cir-

cuit Yrepresenting holder capacitance plus stray

capacitances).

good quality normally employ 3 or 4 double

tuned transformers with coupling adjusted to

critical or slightly less.

The pass band of a typical communication

receiver having a 455 kc. i -f amplifier is shown

in figure 15.

"Miller As mentioned previously, the dyna-

Effect" mic input capacitance of a tube varies

slightly with bias. As a -v -c voltage

normally is applied to i -f tubes for radiotele-

phony reception, the effective grid -cathode

capacitance varies as the signal strength

varies, which produces the same effect as

slight detuning of the i -f transformer. This

effect is known as "Miller effect," and can

be minimized to the extent that it is not

troublesome either by using a fairly low L/C

ratio in the transformers or by incorporating

a small amount of degenerative feedback, the

latter being most easily accomplished by leav-

ing part of the cathode resistor unbypassed

for r.f.

Crystal Filters The pass band of an inter-

mediate frequency amplifier

may be made very narrow through the use of a

piezoelectric filter crystal employed as a

series resonant circuit in a bridge arrange-

ment known as a crystal filter. The shape fac-

tor is quite poor, as would be expected when

the selectivity is obtained from the equivalent

of a single tuned circuit, but the very narrow

pass band obtainable as a result of the ex-

tremely high Q of the crystal makes the crys-

tal filter useful for c -w telegraphy reception.

The pass band of a 455 kc. crystal filter may

be made as narrow as 50 cycles, while the

narrowest pass band that can be obtained with

a 455 kc. tuned circuit of practicable dimen-

sions is about 5 kc.

The electrical equivalent of a filter crystal

is shown in figure 16. For a given frequency,

L is very high, C very low, and R (assuming

www.americanradiohistory.com

HANDBOOK Crystal Filters 219

Figure 17

EQUIVALENT OF CRYSTAL

FILTER CIRCUIT

For a given voltage out of the generator, the volt-

age developed across Z1 depends upon the ratio

of the impedance of X to the sum of the impedances

of Z and Z1. Because of the high Q of the crystal,

its impedance changes rapidly with changes in

frequency.

a good crystal of high Q) is very low. Capaci-

tance C, represents the shunt capacitance of

the electrodes, plus the crystal holder and

wiring, and is many times the capacitance of

C. This makes the crystal act as a parallel

resonant circuit with a frequency only slightly

higher than that of its frequency of series

resonance. For crystal filter use it is the

series resonant characteristic that we are pri-

marily interested in.

The electrical equivalent of the basic crys-

tal filter circuit is shown in figure 17. If the

impedance of Z plus Z, is low compared to the

impedance of the crystal X at resonance, then

the current flowing through Z and the voltage

developed across it, will be almost in inverse

proportion to the impedance of X, which has

a very sharp resonance curve.

If the impedance of Z plus Z, is made high

compared to the resonant impedance of X, then

there will be no appreciable drop in voltage

across Z, as the frequency departs from the

resonant frequency of X until the point is

reached where the impedance of X approaches

that of Z plus Z,. This has the effect of broad-

ening out the curve of frequency versus voltage

developed across Z which is another way of

saying that the selectivity of the crystal filter

(but not the crystal proper) has been reduced.

In practicable filter circuits the impedances

Z and Z, usually are represented by some form

of tuned circuit, but the basic principle of

operation is the same.

Fractical Filters It is necessary to balance

out the capacitance across

the crystal holder (C in figure 16) to prevent

bypassing around the crystal undesired signals

off the crystal resonant frequency. The bal-

ancing is done by a phasing circuit which

takes out -of -phase voltage from a balanced in-

CRYSTAL

SELECTIVITY

E CONTROL

PHASING

CONTROL

Figure 18

TYPICAL CRYSTAL FILTER CIRCUIT

put circuit and passes it to the output side of

the crystal in proper phase to neutralize that

passed through the holder capacitance. A rep-

resentative practical filter arrangement is

shown in figure 18. The balanced input circuit

may be obtained either through the use of a

split- stator capacitor as shown, or by the use

of a center -tapped input coil.

Variable- Selec- In the circuit of figure 18, the

tivity Filters selectivity is minimum with

the crystal input circuit tuned

to resonance, since at resonance the imped-

ance of the tuned circuit is maximum. As the

input circuit is detuned from resonance, how-

ever, the impedance decreases, and the selec-

tivity becomes greater. In this circuit, the out-

put from the crystal filter is tapped down on

the i -f stage grid winding to provide a low

value of series impedance in the output cir-

cuit. It will be recalled that for maximum selec-

tivity, the total impedance in series with the

crystal (both input and output circuits) must

be low. If one is made low and the other is

made variable, then the selectivity may be

varied at will from sharp to broad.

The circuit shown in figure 19 also achieves

variable selectivity by adding a variable im-

pedance in series with the crystal circuit. In

this case, the variable impedance is in series

with the crystal output circuit. The impedance

of the output circuit is varied by varying the

Q. As the Q is reduced (by adding resistance

in series with the coil), the impedance de-

creases and the selectivity becomes greater.

The input circuit impedance is made low by

using a non -resonant secondary on the input

transformer.

A variation of the circuit shown at figure

19 consists of placing the variable resistance

across the coil and capacitor, rather than in

series with them. The result of adding the re-

sistor is a reduction of the output impedance,

and an increase in selectivity. The circuit be-

haves oppositely to that of figure 19, however;

as the resistance is lowered the selectivity

becomes greater. Still another variation of fig-

ure 19 is to use the tuning capacitor across

the output coil to vary the output impedance.

www.americanradiohistory.com

220 Radio Receiver Fundamentals THE RADIO

CRYSTwL

SELECTIVITY

CONTROL

Figure 19

VARIABLE SELECTIVITY

CRYSTAL FILTER

This circuit permits of a greater control of selec-

tivity than does the circuit of figure 16, and does

not require a split- stator variable capacitor.

As the output circuit is detuned from reso-

nance, its impedance is lowered, and the

selectivity increases. Sometimes a set of

fixed capacitors and a multipoint switch are

used to give step -by -step variation of the out-

put circuit tuning, and thus of the crystal

filter selectivity.

Rejection As previously discussed, a filter

Notch crystal has both a resonant(series

resonant) and an anti -resonant

(parallel resonant) frequency, the impedance

of the crystal being quite low at the former

frequency, and quite high at the latter fre-

quency. The anti- resonant frequency is just

slightly higher than the resonant frequency,

the difference depending upon the effective

shunt capacitance of the filter crystal and

holder. As adjustment of the phasing capacitor

controls the effective shunt capacitance of the

crystal, it is possible to vary the anti -reso-

nant frequency of the crystal slightly without

unbalancing the circuit sufficiently to let un-

desired signals leak through the shunt capac-

itance in appreciable amplitude. At the exact

anti -resonant frequency of the crystal the at-

tenuation is exceedingly high, because of the

high impedance of the crystal at this frequen-

cy. This is called the rejection notch, and

can be utilized virtually to eliminate the

heterodyne image or repeat tuning of c -w sig-

nals. The beat frequency oscillator can be

so adjusted and the phasing capacitor so ad-

justed that the desired beat note is of such

a pitch that the image (the same audio note

on the other side of zero beat) falls in the re-

jection notch and is inaudible. The receiver

then is said to be adjusted for single -signal

operation.

The rejection notch sometimes can be em-

ployed to reduce interference from an un-

desired phone signal which is very close in

frequency to a desired phone signal. The filter

is adjusted to "broad" so as to permit tele-

r CRYSTAL NOTCH

z 3 3S

O

0 40

J

m

V

CI so

-3 -2 -I 455 +I +2 +3 +

KC

Figure 20

I -F PASS BAND OF TYPICAL

CRYSTAL FILTER

COMMUNICATIONS RECEIVER

phony reception, and the receiver tuned so

that the carrier frequency of the undesired

signal falls in the rejection notch. The modu-

lation sidebands of the undesired signal still

will come through, but the carrier heterodyne

will be effectively eliminated and interference

greatly reduced.

A typical crystal selectivity curve for a

communications receiver is shown in figure 20.

Crystal Filter A crystal filter, especially

Considerations when adjusted for single sig-

nal reception, greatly reduces

interference and background noise, the latter

feature permitting signals to be copied that

would ordinarily be too weak to be heard above

the background hiss. However, when the filter

is adjusted for maximum selectivity, the pass

band is so narrow that the received signal

must have a high order of stability in order to

stay within the pass band. Likewise, the local

oscillator in the receiver must be highly stable,

or constant retuning will be required. Another

effect that will be noticed with the filter ad-

justed too "sharp" is a tendency for code

characters to produce a ringing sound, and

have a hangover or "tails." This effect limits

the code speed that can be copied satisfac-

torily when the filter is adjusted for extreme

selectivity.

The Mechanical The Collins Mechanical Fil -

Filter ter (figure 21) is a new con-

cept in the field of selec-

tivity. It is an electro- mechanical bandpass

filter about half the size of a cigarette pack-

age. As shown in figure 22, it consists of an

input transducer, a resonant mechanical sec-

www.americanradiohistory.com

HANDBOOK Collins Mechanical Filter 221

tion comprised of a number of metal discs, and

an output transducer.

The frequency characteristics of the reso-

nant mechanical section provide the almost

rectangular selectivity curves shown in figure

23. The input and output transducers serve

only as electrical to mechanical coupling de-

vices and do not affect the selectivity charac-

teristics which are determined by the metal

discs. An electrical signal applied to the in-

put terminals is converted into a mechanical

vibration at the input transducer by means of

magnetostriction. This mechanical vibration

travels through the resonant mechanical sec-

tion to the output transducer, where it is con-

verted by magnetostriction to an electrical

signal which appears at the output terminals.

In order to provide the most efficient electro-

mechanical coupling, a small magnet in the

mounting above each transducer applies a mag-

netic bias to the nickel transducer core. The

electrical impulses then add to or subtract

from this magnetic bias, causing vibration of

the filter elements that corresponds to the

exciting signal. There is no mechanical motion

except for the imperceptible vibration of the

metal discs.

Magnetostrictively -driven mechanical filters

have several advantages over electrical equi-

valents. In the region from 100 kc. to 500 kc.,

the mechanical elements are extremely small,

and a mechanical filter having better selectivi-

ty than the best of conventional i -f systems

may be enclosed in a package smaller than one

i -f transformer.

Since mechanical elements with Q's of 5000

or more are readily obtainable, mechanical fil-

ters may be designed in accordance with the

theory for lossless elements. This permits fil-

ter characteristics that are unobtainable with

electrical circuits because of the relatively

high losses in electrical elements as compared

with the mechanical elements used in the

filters.

ONE SUPPORTING

DISC AT

EACH END COUPLING RODS

RESONANT MECHANICAL SECTION

ill iì%U

(0 RESONANT DISCS) DIAS MAGNET

MAGNETOSTRICTIVE RANSDUCER

DRIVING ROD COIL

ELECTRICAL SIGNAL

(INPUT OR OUTPUT)

Figure 22

MECHANICAL FILTER

FUNCTIONAL DIAGRAM

ELECTRICAL SIGNAL

(INPUT OR OUTPUT)

c.

Figure 21

COLLINS MECHANICAL FILTERS

The Collins Mechanical Filter is an

electro- mechanical bandpass filter which

surpasses, in one small unit, the se-

lectivity of conventional, space-consuming

filters. At the left is the miniaturized

filter, less than 2!4' long. Type H is

next, and two horizontal mounting types

are at right. For exploded view of Collins

Mechanical Filter, see figure 46.

The frequency characteristics of the mechan-

ical filter are permanent, and no adjustment is

required or is possible. The filter is enclosed

in a hermetically sealed case.

In order to realize full benefit from the me-

chanical filter's selectivity characteristics,

it is necessary to provide shielding between

the external input and output circuits, capable

of reducing transfer of energy external to the

o

Figure 23

Selectivity curves of 4554c. mechanical filters

with nominal 0.8 -%c. (dotted line) and 3.1 -kc.

(solid line) bandwidth at -6 db.

www.americanradiohistory.com

222 Radio Receiver Fundamentals THE RADIO

65J7 I VERY SMALL

//1

I-rl p 2Np

Por I DETECTOR

J

Figure 24

VARIABLE -OUTPUT B -F -O CIRCUIT

A beat- frequency oscillator whose output is con-

trollable is of considerable assistance in copying

c -w signals over a wide ronge of levels, and such

a control is almost a necessity for satisfactory

copying of single -sideband radiophone signals.

filter by a minimum value of 100 db. If the in-

put circuit is allowed to couple energy into

the output circuit external to the filter, the

excellent skirt selectivity will deteriorate and

the passband characteristics will be distorted.

As with almost any mechanically resonant

circuit, elements of the mechanical filter have

multiple resonances. These result in spurious

modes of transmission through the filter and

produce minor passbands at frequencies on

other sides of the primary passband. Design

of the filter reduces these sub -bands to a low

level and removes them from the immediate

area of the major passband. Two conventional

i -f transformers supply increased attenuation

to these spurious responses, and are sufficient

to reduce them to an insignificant level.

Beat - Frequency The beat -frequency oscillator,

Oscillators usually called the b.J.o., is a

necessary adjunct for recep-

tion of c -w telegraph signals on superhetero-

dynes which have no other provision for ob-

taining modulation of an incoming c -w tele-

graphy signal. The oscillator is coupled into

or just ahead of second detector circuit and

supplies a signal of nearly the same frequency

as that of the desired signal from the i -f am-

plifier. If the i -f amplifier is tuned to 455 kc.,

for example, the b.f.o. is tuned to approxi-

mately 454 or 456 kc. to produce an audible

(1000 cycle) beat note in the output of the

second detector of the receiver. The carrier

signal itself is, of course, inaudible. The b.f.o.

is not used for voice reception, except as an

aid in searching for weak stations.

The b -f -o input to the second detector need

only be sufficient to give a good beat note on

an average signal. Too much coupling into the

second detector will give an excessively high

hiss level, masking weak signals by the high

noise background.

Figure 24 shows a method of manually ad-

I.F. STAGE DET.

BY

pA GRID LEAK DETECTOR

I.F STAGE DET.

OB DIODE DETECTOR

I.F. STAGE DET.

AUDIO

OD INFINITE IMPEDANCE DETECTOR

Figure 25

TYPICAL CIRCUITS FOR GRID -LEAK,

DIODE, PLATE AND INFINITE IMPE-

DANCE DETECTOR STAGES

justing the b -f -o output to correspond with the

strength of received signals. This type of vari-

able b -f -o output control is a useful adjunct

to any superheterodyne, since it allows suffi-

cient b-f -o output to be obtained to beat with

strong signals or to allow single - sideband

reception and at the same time permits the

b -f -o output, and consequently the hiss, to be

reduced when attempting to receive weak sig-

nals. The circuit shown is somewhat better

than those in which one of the electrode volt-

www.americanradiohistory.com

HANDBOOK Detector Circuits 223

Figure 26

TYPICAL A -V -C CIRCUIT USING A DOUBLE DIODE

Any of the small dual-diode tubes may be used in this circuit. Or, if desired, a duo- diode-

triode may be used, with the triode acting as the first audio stage. The left-hand diode serves

as the detector, while the right -hand side acts as the a -v -c rectifier. The use of separate

diodes for detector and o-v -c reduces distortion when receiving an AM signal with a high

modulation percentage.

ages on the b -f -o tube is changed, as the latter

circuits usually change the frequency of the

b.f.o. at the same time they change the strength,

making it necessary to reset the trimmer each

time the output is adjusted.

The b.f.o. usually is provided with a small

trimmer which is adjustable from the front

panel to permit adjustment over a range of 5

or 10 kc. For single -signal reception the b.f.o.

always is adjusted to the high- frequency side,

in order to permit placing the heterodyne image

in the rejection notch.

In order to reduce the b -f -o signal output

voltage to a reasonable level which will pre-

vent blocking the second detector, the signal

voltage is delivered through a low- capacitance

(high -reactance) capacitor having a value of

1 to 2 fgifd.

Care must be taken with the b.f.o. to pre-

vent harmonics of the oscillator from being

picked up at multiples of the b -f -o frequency.

The complete b.f.o. together with the coupling

circuits to the second detector, should be thor-

oughly shielded to prevent pickup of the har-

monics by the input end of the receiver.

If b-f -o harmonics still have a tendency to

give trouble after complete shielding and iso-

lation of the b -f -o circuit has been accom-

plished, the passage of these harmonics from

the b-f -o circuit to the rest of the receiver can

be stopped through the use of a low -pass filter

in the lead between the output of the b -f -o cir-

cuit and the point on the receiver where the

b-f -o signal is to be injected.

12 -8 Detector, Audio, and

Control Circuits

Detectors Second detectors for use in super -

heterodynes are usually of the

diode, plate, or infinite -impedance types. Oc-

casionally, grid -leak detectors are used in re-

ceivers using one i -f stage or none at all, in

which case the second detector usually is

made regenerative.

Diodes are the most popular second detect-

ors because they allow a simple method of

obtaining automatic volume control to be used.

Diodes load the tuned circuit to which they

are connected, however, and thus reduce the

selectivity slightly. Special i -f transformers

are used for the purpose of providing a low -

impedance input circuit to the diode detector.

Typical circuits for grid -leak, diode, plate

and infinite -impedance detectors are shown

in figure 25.

Automatic Vol- The elements of an automatic

urne Control volume control (a.v.c.) sys-

tem are shown in figure 26.

A dual -diode tube is used as a combination

diode detector and a -v -c rectifier. The left -

hand diode operates as a simple rectifier in

the manner described earlier in this chapter.

Audio voltage, superimposed on a d -c voltage,

appears across the 500,000 -ohm potentiometer

(the volume control) and the .0001 -µfd. capa-

citor, and is passed on to the audio amplifier.

The right -hand diode receives signal voltage

directly from the primary of the last i -f ampli-

fier, and acts as the a -v -c rectifier. The pul-

sating d -c voltage across the 1- megohm a.v.c.-

diode load resistor is filtered by a 500,000 -ohm

resistor and a .05-pfd. capacitor, and applied

as bias to the grids of the r -f and i -f amplifier

tubes; an increase or decrease in signal

strength will cause a corresponding increase

or decrease in a -v -c bias voltage, and thus the

gain of the receiver is automatically adjusted

to compensate for changes in signal strength.

www.americanradiohistory.com

224 Radio Receiver Fundamentals THE RADIO

A -C Loading of By disassociating the a.v.c.

Second Detector and detecting functions

through using separate

diodes, as shown, most of the ill effects of

a -c shunt loading on the detector diode are

avoided. This type of loading causes serious

distortion, and the additional components re-

quired to eliminate it are well worth their cost.

Even with the circuit shown, a -c loading can

occur unless a very high (5 megohms, or more)

value of grid resistor is used in the following

audio amplifier stage.

A.V.C. in In receivers having a beat -

B-F-O- Equipped frequency oscillator for the

Receivers reception of radiotelegraph

signals, the use of a.v.c.

can result in a great loss in sensitivity when

the b.f.o. is switched on. This is because the

beat oscillator output acts exactly like a

strong received signal, and causes the a -v -c

circuit to put high bias on the r -f and i -f stages,

thus greatly reducing the receiver's sensitivi-

ty. Due to the above effect, it is necessary to

provide a method of making the a -v -c circuit

inoperative when the b.f.o. is being used. The

simplest method of eliminating the a -v -c ac-

tion is to short the a -v -c line to ground when

the b.f.o. is turned on. A two -circuit switch

may be used for the dual purpose of turning on

the beat oscillator and shorting out the a.v.c.

if desired.

Signal Strength Visual means for determining

Indicators whether or not the receiver is

properly tuned, as well as an

indication of the relative signal strength, are

both provided by means of tuning indicators

(S meters) of the meter or vacuum -tube type.

A d -c milliammeter can be connected in the

plate supply circuit of one or more r -f or i -f

amplifiers, as shown in figure 27A, so that the

change in plate current, due to the action of

the a -v -c voltage, will be indicated on the in-

strument. The d -c instrument MA should have

a full -scale reading approximately equal to the

total plate current taken by the stage or stages

whose plate current passes through the instru-

ment. The value of this current can be esti-

mated by assuming a plate current on each

stage (with no signal input to the receiver) of

about 6 ma. However, it will be found to be

more satisfactory to measure the actual plate

current on the stages with a milliammeter of

perhaps 0 -100 ma. full scale before purchasing

an instrument for use as an S meter. The

50 -ohm potentiometer shown in the drawing is

used to adjust the meter reading to full scale

with no signal input to the receiver.

When an ordinary meter is used in the plate

circuit of a stage, for the purpose of indicat-

ing signal strength, the meter reads backwards

RF IF IF

RFoR IF

R Fon I F

6U5/6G5

OR 6E5

TO A V C

t250v

+70

Figure 27

SIGNAL -STRENGTH -METER CIRCUITS

Shown above are four circuits for obtaining a sig-

nal- strength reading which is a function of incom-

ing carrier amplitude. The circuits are discussed

lin the accompanying text.

with respect to strength. This is because in-

creased a -v -c bias on stronger signals causes

lower plate current through the meter. For this

reason, special meters which indicate zero at

the right -hand end of the scale are often used

for signal strength indicators in commercial

receivers using this type of circuit. Alter-

natively, the meter may be mounted upside

down, so that the needle moves toward the

right with increased strength.

The circuit of figure 27B can frequently be

used to advantage in a receiver where the cath-

ode of one of the r -f or i -f amplifier stages

runs directly to ground through the cathode

bias resistor instead of running through a cath-

www.americanradiohistory.com

HANDBOOK Noise Suppression 225

ode -voltage gain control. In this case a 0 -1

d -c milliammeter in conjunction with a resistor

from 1000 to 3000 ohms can be used as shown

as a signal- strength meter. With this circuit

the meter will read backwards with increasing

signal strength as in the circuit previously

discussed.

Figure 27C is the circuit of a forward -read-

ingS meter as is often used in communications

receivers. The instrument is used in an un-

balanced bridge circuit with the d -c plate re-

sistance of one i -f tube as one leg of the

bridge and with resistors for the other three

legs. The value of the resistor R must be de-

termined by trial and error and will be some-

where in the vicinity of 50,000 ohms. Some-

times the screen circuits of the r -f and i -f

stages are taken from this point along with

the screen -circuit voltage divider.

Electron -ray tubes (sometimes called "magic

eyes ") can also be used as indicators of rela-

tive signal strength in a circuit similar to that

shown in figure 27D. A 6U5/6G5 tube should

be used where the a -v -c voltage will be from

5 to 20 volts and a type 6E5 tube should be

used when the a -v -c voltage will run from 2

to 8 volts.

Audio Ampl if iers Audio amplifiers are em-

ployed in nearly all radio

receivers. The audio amplifier stage or stages

are usually of the Class A type, although Class

AB push -pull stages are used in some re-

ceivers. The purpose of the audio amplifier is

to bring the relatively weak signal from the

detector up to a strength sufficient to operate

a pair of headphones or a loud speaker. Either

triodes, pentodes, or beam tetrodes may be

used, the pentodes and beam tetrodes usually

giving greater output. In some receivers, par-

ticularly those employing grid leak detection,

it is possible to operate the headphones di-

rectly from the detector, without audio ampli-

fication. In such receivers, a single audio

stage with a beam tetrode or pentode tube is

ordinarily used to drive the loud speaker.

Most communications receivers, either home -

constructed or factory -made, have a single -

ended beam tetrode (such as a 6L6 or 6V6) or

pentode (6F6 or 6K6 -GT) in the audio output

stage feeding the loudspeaker. If precautions

are not taken such a stage will actually bring

about a decrease in the effective signal -to-

noise ratio of the receiver due to the rising

high- frequency characteristic of such a stage

when feeding a loud -speaker. One way of im-

proving this condition is to place a mica or

paper capacitor of approximately 0.003 pfd.

capacitance across the primary of the output

transformer. The use of a capacitor in this

manner tends to make the load impedance seen

by the plate of the output tube more constant

over the audio -frequency range. The speaker

and transformer will tend to present a rising

impedance to the tube as the frequency in-

creases, and the parallel capacitor will tend

to make the total impedance more constant

since it will tend to present a decreasing im-

pedance with increasing audio frequency.

A still better way of improving the frequency

characteristic of the output stage, and at the

same time reducing the harmonic distortion,

is to use shunt feedback from the plate of the

output tube to the plate of a tube such as a

6SJ7 acting as an audio amplifier stage ahead

of the output stage.

12 -9 Noise Suppression

The problem of noise suppression confronts

the listener who is located in places where

interference from power lines, electrical ap-

pliances, and automobile ignition systems is

troublesome. This noise is often of such in-

tensity as to swamp out signals from desired

stations.

There are two principal methods for reduc-

ing this noise:

(1) A -c line filters at the source of inter-

ference, if the noise is created by an

electrical appliance.

(2) Noise -limiting circuits for the reduc-

tion, in the receiver itself, of interfer-

ence of the type caused by automobile

ignition systems.

Power Line Many household appliances, such

Filters as electric mixers, heating pads,

vacuum sweepers, refrigerators,

oil burners, sewing machines, doorbells, etc.,

create an interference of an intermittent na-

ture. The insertion of a line filter near the

source of interference often will effect a com-

plete cure. Filters for small appliances can

consist of a 0.1 -µfd. capacitor connected a-

cross the 110 -volt a -c line. Two capacitors

in series across the line, with the midpoint

connected to ground, can be used in conjunc-

tion with ultraviolet ray machines, refriger-

ators, oil burner furnaces, and other more stub-

born offenders. In severe cases of interfer-

ence, additional filters in the form of heavy -

duty r -f choke coils must be connected in

series with the 110 -volt a -c line on both sides

of the line right at the interfering appliance.

Peak Noise Numerous noise -limiting circuits

Limiters which are beneficial in overcom-

ing key clicks, automobile ig-

nition interference, and similar noise impulses

have become popular. They operate on the

principle that each individual noise pulse is

www.americanradiohistory.com

226 Radio Receiver Fundamentals THE RADIO

of very short duration, yet of very high ampli-

tude. The popping or clicking type of noise

from electrical ignition systems may produce

a signal having a peak value ten to twenty

times as great as the incoming radio signal,

but an average power much less than the sig-

nal. As the duration of this type of noise peak

is short, the receiver can be made inoperative

during the noise pulse without the human ear

detecting the total loss of signal. Some noise

limiters actually punch a bole in the signal,

while others merely limit the maximum peak

signal which reaches the headphones or loud-

speaker.

The noise peak is of such short duration

that it would not be objectionable except for

the fact that it produces an over -loading effect

on the receiver, which increases its time con-

stant. A sharp voltage peak will give a kick

to the diaphragm of the headphones or speak-

er, and the momentum or inertia keeps the

diaphragm in motion until the dampening of

the diaphragm stops it. This movement pro-

duces a popping sound which may completely

obliterate the desired signal. If the noise pulse

can be limited to a peak amplitude equal to

that of the desired signal, the resulting inter-

ference is practically negligible for moder-

ately low repetition rates, such as ignition

noise.

In addition, the i -f amplifier of the receiver

will also tend to lengthen the duration of the

noise pulses because the relatively high -Q i -f

tuned circuits will ring or oscillate when ex-

cited by a sharp pulse, such as produced by

ignition noise. The most effective noise limiter

would be placed before the high -Q i -f tuned

circuits. At this point the noise pulse is the

sharpest and has not been degraded by pass-

age through the i -f transformers. In addition,

the pulse is eliminated before it can produce

ringing effects in the i -f chain.

The Lomb An i -f noise limiter is shown in

Noise Limiter figure 28. This is an adapta-

tion of the Lamb noise silenc-

er circuit. The i -f signal is fed into a double

grid tube, such as a 6L7, and thence into the

i -f chain. A 6AB7 high gain pentode is capa-

city coupled to the input of the i -f system.

This auxiliary tube amplifies both signal and

noise that is fed to it. It has a minimum of

selectivity ahead of it so that it receives the

true noise pulse before it is degraded by the

i -f strip. A broadly tuned i -f transformer is used

to couple the noise amplifier to a 6H6 noise

rectifier. The gain of the noise amplifier is

controlled by a potentiometer in the cathode

of the 6AB7 noise amplifier. This potenti-

ometer controls the gain of the noise amplifier

IST DET ISTI.F.

617

2ND I.F.

Figure 28

THE LAMB I -F NOISE SILENCER

stage and in addition sets the bias level on

the 6H6 diode so that the incoming signal will

not be rectified. Only noise peaks louder than

the signal can overcome the resting bias of

the 6H6 and cause it to conduct. A noise pulse

rectified by the 6H6 is applied as a negative

voltage to the control grid of the 6L7 i -f tube,

disabling the tube, and punching a hole in the

signal at the instant of the noise pulse. By

varying the bias control of the noise limiter,

the negative control voltage applied to the

6L7 may be adjusted until it is barely suffi-

cient to overcome the noise impulses applied

to the al control grid without allowing the

modulation peaks of the carrier to become

badly distorted.

The Bishop Another effective i -f noise

Noise Limiter limiter is the Bishop limiter.

This is a full -wave shunt type

diode limiter applied to the primary of the last

i -f transformer of a receiver. The limiter is

self- biased and automatically adjusts itself

to the degree of modulation of the received

signal. The schematic of this limiter is shown

in figure 29. The bias circuit time constant is

determined by C, and the shunt resistance,

which consists of R, and R2 in series. The

plate resistance of the last i -f tube and the

capacity of C, determine the charging rate of

the circuit. The limiter is disabled by opening

S which allows the bias to rise to the value

of the i -f signal.

www.americanradiohistory.com

HANDBOOK Noise Limiters 227

Figure 29

THE BISHOP I -F NOISE LIMITER

Audio Noise Some of the simplest and most

Limiters practical peak limiters for radio-

telephone reception employ one

or two diodes either as shunt or series limiters

in the audio system of the receiver. when a

noise pulse exceeds a certain predetermined

threshold value, the limiter diode acts either

as a short or open circuit, depending upon

whether it is used in a shunt or series circuit.

The threshold is made to occur at a level high

enough that it will not clip modulation peaks

enough to impair voice intelligibility, but low

enough to limit the noise peaks effectively.

Because the action of the peak limiter is

needed most on very weak signals, and these

usually are not strong enough to produce proper

a -v -c action, a threshold setting that is cor-

rect for a strong phone signal is not correct

for optimum limiting on very weak signals. For

this reason the threshold control often is tied

in with the a -v -c system so as to make the

optimum threshold adjustment automatic in-

stead of manual.

Suppression of impulse noise by means of

an audio peak limiter is best accomplished at

the very front end of the audio system, and

for this reason the function of superheterodyne

second detector and limiter often are combined

in a composite circuit.

The amount of limiting that can be obtained

is a function of the audio distortion that can

be tolerated. Because excessive distortion

will reduce the intelligibility as much as will

background noise, the degree of limiting for

which the circuit is designed has to be a com-

promise.

Peak noise limiters working at the second

detector are much more effective when the i -f

bandwidth of the receiver is broad, because a

sharp i -f amplifier will lengthen the pulses by

the time they reach the second detector, mak-

ing the limiter less effective. V -h -f super -

heterodynes have an i -f bandwidth consider-

ably wider than the minimum necessary for

voice sidebands (to take care of drift and in-

stability). Therefore, they are capable of bet-

ter peak noise suppression than a standard

communications receiver having an i -f band-

width of perhaps 8 kc. Likewise, when a crys-

tal filter is used on the "sharp" position an

a -f peak' limiter is of little benefit.

Practical Noise limiters range all the

Peak Noise way from an audio stage

Limiter Circuits running at very low screen

or plate voltage, to elabor-

ate affairs employing 5 or more tubes. Rather

than attempt to show the numerous types, many

of which are quite complex considering the

results obtained, only two very similar types

will be described. Either is just about as ef-

fective as the most elaborate limiter that can

be constructed, yet requires the addition of

but a single diode and a few resistors and

capacitors over what would be employed in a

good superheterodyne without a limiter. Both

circuits, with but minor modifications in re-

sistance and capacitance values, are incor-

porated in one form or another in different

types of factory -built communications receiv-

ers. Referring to figure 30, the first circuit shows

a conventional superheterodyne second de-

tector, a.v.c., and first audio stage with the

addition of one tube element, D which may

be either a separate diode or part of a twin -

diode as illustrated. Diode D, acts as a series

gate, allowing audio to get to the grid of the

a -f tube only so long as the diode is conduct-

ing. The diode is biased by a d -c voltage ob-

tained in the same manner as a -v -c control volt-

age, the bias being such that pulses of short

duration no longer conduct when the pulse

voltage exceeds the carrier by approximately

60 per cent. This also clips voice modulation

peaks, but not enough to impair intelligibility.

It is apparent that the series diode clips

only positive modulation peaks, by limiting

upward modulation to about 60 per cent. Neg-

ative or downward peaks are limited automati-

cally to 100 per cent in the detector, because

obviously the rectified voltage out of the

diode detector cannot be less than zero. Limit-

ing the downward peaks to 60 per cent or so

instead of 100 per cent would result in but

little improvement in noise reduction, and the

results do not justify the additional compon-

ents required.

It is important that the exact resistance

values shown be used, for best results, and

that 10 per cent tolerance resistors be used

for R, and R,. Also, the rectified carrier volt-

age developed across C, should be at least 5

volts for good limiting.

The limiter will work well on c -w telegraphy

if the amplitude of beat frequency oscillator

injection is not too high. Variable injection is

to be preferred, adjustable from the front panel.

www.americanradiohistory.com

228 Radio Receiver Fundamentals THE RADIO

VI

LAST I. F.TUDC I FT.

V2

AUDIO C1 -0.1 -µfd. paper

C, -50 -4µtd. mice

C_100 -44fd. mica

C4, Cs -0.01 -µfd. paper

Rr, R2 -1 megohm,

15 watt

R,, R4- 220,000 ohms,

Vt watt

R6, R6-1 megohm,

4T watt

Rr -2- megohm poten-

tiometer

Figure 30

NOISE LIMITER CIRCUIT, WITH ASSOCIATED A -V -C

This limiter is of the series type, and is self -adjusting to carrier strength for phone reception. For proper

operation several volts should be developed across the secondary of the last i -f transformer (IFT) under

carrier conditions.

If this feature is not provided, the b -f -o injec-

tion should be reduced to the lowest value that

will give a satisfactory beat. When this is

done, effective limiting and a good beat can

be obtained by proper adjustment of the r -f

and a -f gain controls. It is assumed, of course,

that the a.v.c. is cut out of the circuit for

c -w telegraphy reception.

Alternative The circuit of figure 31 is

Limiter Circuit more effective than that

shown in figure 30 under cer-

tain conditions and requires the addition of

only one more resistor and one more capaci-

tor than the other circuit. Also, this circuit

involves a smaller loss in output level than

the circuit of figure 30. This circuit can be

used with equal effectiveness with a com-

bined diode - triode or diode -pentode tube (6R7,

6SR7, 6Q7, 6SQ7 or similar diode -triodes, or

6B8, 6SF7, or similar diode -pentodes) as diode

detector and first audio stage. However, a

separate diode must be used for the noise

limiter, D,. This diode may be one -half of a

6H6, 6AL5, 7A6, etc., or it may be a triode

connected 6J5, 6C4 or similar type.

Note that the return for the volume control

must be made to the cathode of the detector

diode (and not to ground) when a dual tube is

used as combined second- detector first- audio.

This means that in the circuit shown in figure

31 a connection will exist across the points

where the "X" is shown on the diagram since

a common cathode lead is brought out of the

tube for Dr and V,. If desired, of course, a

single dual diode may be used for Dr and D,

in this circuit as well as in the circuit of fig-

ure 30. Switching the limiter in and out with

the switch S brings about no change in volume.

In any diode limiter circuit such as the ones

shown in these two figures it is important that

the mid -point of the heater potential for the

noise -limiter diode be as close to ground

potential as possible. This means that the

center -tap of the heater supply for the tubes

should be grounded wherever possible rather

than grounding one side of the heater supply

as is often done. Difficulty with hum pickup

in the limiter circuit may be encountered when

one side of the heater is grounded due to the

high values of resistance necessary in the

limiter circuit.

The circuit of figure 31 has been used with

excellent success in several home -constructed

receivers, and in the BC- 312/BC -342 and BC-

348 series of surplus communications receiv-

ers. It is also used in certain manufactured

receivers.

An excellent check on the operation of the

noise limiter in any communications receiver

can be obtained by listening to the Loran sig-

nals in the 160 -meter band. With the limiter

out a sharp rasping buzz will be obtained

when one of these stations is tuned in. With

the noise limiter switched into the circuit the

buzz should be greatly reduced and a low -

pitched hum should be heard.

The Full -Wave The most satisafctory diode

Limiter noise limiter is the series full -

wave limiter, shown in figure

32. The positive noise peaks are clipped by

diode A, the clipping level of which may be

adjusted to clip at any modulation level be-

tween 25 per cent and 100 per cent. The nega-

ative noise peaks are clipped by diode B at

a fixed level.

The TNS Limiter The Twin Noise Squelch.

popularized by CQ maga-

zine, is a combination of a diode noise clipper

and an audio squelch tube. The squelch cit.-

www.americanradiohistory.com

HANDBOOK U -H -F Circuits 229

This circuit is of the self -

adjusting type and gives less

distortion for a given degree

of modulation than the more

common limiter circuits.

R1, R2 -470K, 14 won

R3 -100K, V2 watt

R., RS -1 megohm, VI watt

R6 -2- megohm potentiometer

C1- 0.00025 mica (approx.)

c2- 0.01 -µtd. paper

C3- 0.01 -4fd. paper

C4- 0.01 -12fd. paper

D3, 02 -6116, 6AL5, 7A6, or

diode sections of a 6S8 -GT

Figure 31

ALTERNATIVE NOISE LIMITER CIRCUIT

cuit is useful in eliminating the grinding back-

ground noise that is the residual left by the

diode clipper. In figure 33, the setting of the

470K potentiometer determines the operating

level of the squelch action and should be set

to eliminate the residual background noise.

Because of the low inherent distortion of the

TNS, it may be left in the circuit at all times.

As with other limiters, the TNS requires a

high signal level at the second detector for

maximum limiting effect.

12 -10 Special Considerations

in U -H -F Receiver Design

Transmission At increasingly higher frequen-

Line Circuits cies, it becomes progressively

more difficult to obtain a satis-

factory amount of selectivity and impedance

from an ordinary coil and capacitor used as a

resonant circuit. On the other hand, quarter

F T 2ND DET

Figure 32

THE FULL -WAVE SERIES AUDIO

NOISE LIMITER

AuD10

wavelength sections of parallel conductors or

concentric transmission line are not only more

efficient but also become of practical dimen-

sions.

Tuning Tubes and tuning capacitors con -

Short Lines netted to the open end of a trans-

mission line provide a capaci-

tance that makes the resonant length less than

a quarter wave -length. The amount of shorten-

ing for a specified capacitive reactance is

determined by the surge impedance of the line

F STAGE 2ND DEI. -AUDIO

Figure 33

THE TNS AUDIO NOISE LIMITER

S n. DIO

www.americanradiohistory.com

230 Radio Receiver Fundamentals THE RADIO

Figure 34

COUPLING AN ANTENNA TO A

COAXIALRESONANT CIRCUIT

(A) shows the recommended method for coupling

a coaxial line to a coaxial resonant circuit. (B)

shows an alternative method for use with an open -

wire type of antenna feed line.

CAVITY

CONCENTRIC

LINE

CAVITY

HOLE

© O

CAVITY

GVITY

GRIDS ELECTRON

BEAM

0LOOV

LINE

CAVITY

Figure 35

METHODS OF EXCITING A RESONANT

CAVITY

section. It is given by the equation for reso-

nance:

1

2rr /C

in which 77 = 3.1416, / is the frequency, C the

capacitance, Z. the surge impedance of the

line, and tan / is the tangent of the electrical

length in degrees.

The capacitive reactance of the capacitance

across the end is 1 /(277 / C) ohms. For reson-

ance, this must equal the surge impedance of

the line times the tangent of its electrical

length (in degrees, where 90° equals a quar-

ter wave). It will be seen that twice the capac-

itance will resonate a line if its surge imped-

ance is halved; also that a given capacitance

has twice the loading effect when the frequen-

cy is doubled.

Coupling Into It is possible to couple into

Lines and a parallel -rod line by tapp-

Coaxial Circuits ing directly on one or both

rods, preferably through

blocking capacitors if any d.c. is present.

More commonly, however, a hairpin is induc-

tively coupled at the shorting bar end, either

to the bar or to the two rods, or both. This

normally will result in a balanced load. Should

a loop unbalanced to ground be coupled in,

any resulting unbalance reflected into the rods

can be reduced with a simple Faraday screen,

made of a few parallel wires placed between

the hairpin loop and the rods. These should

be soldered at only one end and grounded.

An unbalanced tap on a coaxial resonant

= Z. tan 1

circuit can be made directly on the inner con-

ductor at the point where it is properly matched

(figure 34). For low impedances, such as a con-

centric line feeder, a small one -half turn loop

can be inserted through a hole in the outer

conductor of the coaxial circuit, being in ef-

fect a half of the hairpin type recommended

for coupling balanced feeders to coaxial re-

sonant lines. The size of the loop and close-

ness to the inner conductor determines the

impedance matching and loading. Such loops

coupled in near the shorting disc do not alter

the tuning appreciably, if not overcoupled.

Resonant A cavity is a closed resonant

Cavities chamber made of metal. It is

known also as a rhumbatron. The

cavity, having both inductance and capaci-

tance, supersedes coil -capacitor and capaci-

tance- loaded transmission -line tuned circuits

at extremely high frequencies where conven-

tional L and C components, of even the most

refined design, prove impractical because of

the tiny electrical and physical dimensions

they must have. Microwave cavities have high

Q factors and are superior to conventional

tuned circuits. They may be employed in the

manner of an absorption wavemeter or as the

tuned circuit in other r -f test instruments, and

in microwave transmitters and receivers.

Resonant cavities usually are closed on all

sides and all of their walls are made of elec-

trical conductor. However, in some forms,

small openings are present for the purpose of

excitation. Cavities have been produced in

several shapes including the plain sphere,

www.americanradiohistory.com

HANDBOOK U -H -F Circuits 231

TUNING

SLUGS MOVEABLE

DISC

Figure 36

TUNING METHODS FOR CYLINDRICAL

RESONANT CAVITIES

dimpled sphere, sphere with reentrant cones

of various sorts, cylinder, prism (including

cube), ellipsoid, ellipsoid -hyperboloid, dough-

nut- shape, and various reentrant types. In ap-

pearance, they resemble in their simpler forms

metal boxes or cans.

The cavity actually is a linear circuit, but

one which is superior to a conventional co-

axial resonator in the s -h -f range. The cavity

resonates in much the same manner as does a

barrel or a closed room with reflecting walls.

Because electromagnetic energy, and the

associated electrostatic energy, oscillates to

and fro inside them in one mode or another,

resonant cavities resemble wave guides. The

mode of operation in a cavity is affected by

the manner in which micro -wave energy is in-

jected. A cavity will resonate to a large num-

ber of frequencies, each being associated with

a particular mode or standing -wave pattern.

The lowest mode (lowest frequency of opera-

tion) of a cavity resonator normally is the one

used. The resonant frequency of a cavity may be

varied, if desired, by means of movable plung-

ers or plugs, as shown in figure 36A, or a

movable metal disc (see figure 36B). A cavity

that is too small for a given wavelength will

not oscillate.

The resonant frequencies of simple spheri-

cal, cylindrical, and cubical cavities may be

calculated simply for one particular mode.

Wavelength and cavity dimensions (in centi-

meters) are related by the following simple

resonance formulae:

For Cylinder A, = 2.6 x radius

" Cube Ar = 2.83 x half of 1 side

" Sphere A, = 2.28x radius

Butterfly Unlike the cavity resonator, which

Circuit in its conventional form is a device

which can tune over a relatively

narrow band, the butterfly circuit is a tunable

resonator which permits coverage of a fairly

Figure 37

THE BUTTERFLY RESONANT CIRCUIT

Shown at (A) is the physical appearance of the

butterfly circuit as used in the v -h -f and lower

u-h-f ronge. (B) shows an electrical representa-

tion of the circuit.

wide u -h -f band. The butterfly circuit is very

similar to a conventional coil -variable capaci-

tor combination, except that both inductance

and capacitance are provided by what appears

to be a variable capacitor alone. The Q of this

device is somewhat less than that of a con-

centric -line tuned circuit but is entirely ade-

quate for numerous applications.

Figure 37A shows construction of a single

butterfly section. The butterfly- shaped rotor,

from which the device derives its name, turns

in relation to the unconventional stator. The

two groups of stator "fins" or sectors are in

effect joined together by a semi -circular metal

band, integral with the sectors, which provides

the circuit inductance. When the rotor is set

to fill the loop opening (the position in which

it is shown in figure 37A), the circuit induct-

ance and capacitance are reduced to minimum.

When the rotor occupies the position indicated

by the dotted lines, the inductance and capaci-

tance are at maximum. The tuning range of

practical butterfly circuits is in the ratio of

1.5:1 to 3.5:1.

Direct circuit connections may be made to

points A and B. If balanced operation is de-

sired, either point C or D will provide the

electrical mid -point. Coupling may be effected

by means of a small singleturn loop placed

near point E or F. The butterfly thus permits

continuous variation of both capacitance and

inductance, as indicated by the equivalent

circuit in figure 37B, while at the same time

eliminating all pigtails and wiping contacts.

Several butterfly sections may be stacked

in parallel in the same way that variable ca-

pacitors are built up. In stacking these sec-

tions, the effect of adding inductances in paral-

lel is to lower the total circuit inductance,

while the addition of stators and rotors raises

the total capacitance, as well as the ratio of

maximum to minimum capacitance.

www.americanradiohistory.com

232 Radio Receiver Fundamentals THE RADIO

Butterfly circuits have been applied specifi-

cally to oscillators for transmitters, super-

heterodyne receivers, and heterodyne frequen-

cy meters in the 100 - 1000 -Mc. frequency range.

Receiver The types of resonant circuits de-

Circuits scribed in the previous paragraphs

have largely replaced conventional

coil -capacitor circuits in the range above 100

Mc. Tuned short lines and butterfly circuits

are used in the range from about 100 Mc. to

perhaps 3500 Mc., and above about 3500 Mc.

resonant cavities are used almost exclusively.

The resonant cavity is also quite generally

employed in the 2000 -Mc. to 3500 -Mc. range.

In a properly designed receiver, thermal

agitation in the first tuned circuit is amplified

by subsequent tubes and predominates in the

output. For good signal-to- set -noise ratio,

therefore, one must strive for a high -gain low -

noise r -f stage. Hiss can be held down by giv-

ing careful attention to this point. A mixer

has about 0.3 of the gain of an r -f tube of the

same type; so it is advisable to precede a

mixer by an efficient r -f stage. It is also of

some value to have good r -f selectivity before

the first detector in order to reduce noises

produced by beating noise at one frequency

against noise at another, to produce noise at

the intermediate frequency in a superheter-

odyne.

The frequency limit of a tube is reached

when the shortest possible external connec-

tions are used as the tuned circuit, except for

abnormal types of oscillation. Wires or size-

able components are often best considered as

sections of transmission lines rather than as

simple resistances, capacitances, or induct-

ances.

So long as small triodes and pentodes will

operate normally, they are generally preferred

as v -h -f tubes over other receiving methods

that have been devised. However, the input

capacitance, input conductance, and transit

time of these tubes limit the upper frequency

at which they may be operated. The input re-

sistance, which drops to a low value at very

short wave -lengths, limits the stage gain and

broadens the tuning.

V -H -F The first tube in a v -h -f receiver is

Tubes most important in raising the signal

above the noise generated in succes-

sive stages, for which reason small v -h -f types

are definitely preferred.

Tubes employing the conventional grid -con-

trolled and diode rectifier principles have been

modernized, through various expedients, for

operation at frequencies as high, in some new

types, as 4000 Mc. Beyond that frequency,

electron transit time becomes the limiting fac-

tor and new principles must be enlisted. In

general, the improvements embodied in exist-

ing tubes have consisted of (1) reducing elec-

trode spacing to cut down electron transit

time, (2) reducing electrode areas to decrease

interelectrode capacitances, and (3) shorten-

ing of electrode leads either by mounting the

electrode assembly close to the tube base or

by bringing the leads out directly through the

glass envelope at nearby points. Through re-

duction of lead inductance and interelectrode

capacitances, input and output resonant fre-

quencies due to tube construction have bee':

increased substantially.

Tubes embracing one or more of the fea-

tures just outlined include the later !octal

types, high- frequency acorns, button -base

types, and the lighthouse types. Type 6J4

button -base triode will reach 500 Mc. Type

6F4 acorn triode is recommended for use up to

1200 Mc. Type 1A3 button -base diode has a

resonant frequency of 1000 Mc., while type

9005 acorn diode resonates at 1500 Mc. Light-

house type 2C40 can be used at frequencies

up to 3500 Mc. as an oscillator.

Crystal More than two de c a d e s have

Rectifiers passed since the crystal (mineral)

rectifier enjoyed widespread use

in radio receivers. Low -priced tubes complete-

ly supplanted the fragile and relatively insen-

sitive crystal detector, although it did con-

tinue for a few years as a simple meter recti-

fier in absorption wavemeters after its demise

as a receiver component.

Today, the crystal detector is of new im-

portance in microwave communication. It is

being employed as a detector and as a mixer

in receivers and test instruments used at ex-

tremely high radio frequencies. At some of

the frequencies employed in microwave opera-

tions, the crystal rectifier is the only satis-

factory detector or mixer. The chief advan-

tages of the crystal rectifier are very low ca-

pacitance, relative freedom from transit -time

difficulties, and its two -terminal nature. No

batteries or a -c power supply are required for

its operation.

The crystal detector consists essentially of

a small piece of silicon or germanium mount-

ed in a base of low- melting -point alloy and

contacted by means of a thin, springy feeler

wire known as the cat whisker. This arrange-

ment is shown in figure 38A.

The complex physics of crystal rectification

is beyond the scope of this discussion. It is

sufficient to state that current flows from sev-

eral hundred to several thousand times more

readily in one direction through the contact of

cat whisker and crystal than in the opposite

direction. Consequently, an alternating current

(including one of microwave frequency) will

www.americanradiohistory.com

HANDBOOK Receiver Adjustment 233

SYMBOL f \'i BRASS BASE CONNECTOR

-CERAMIC SLEEVE

BRASS CAP

BRASS CONNECTOR PIN

Figure 38

1N23 MICROWAVE -TYPE

CRYSTAL DIODE

A small silicon crystal is attached to

the base connector and o fine "cat -

whisker" wire is set to the most sensi-

tive spot on the crystal. After adjust-

ment the ceramic shell is filled with

compound to hold the contact wire in

position. Crystals of this type are used

to over 30,000 mc.

be rectified by the crystal detector. The load,

through which the rectified currents flow, may

be connected in series or shunt with the crys-

tal, although the former connection is most

generally employed.

The basic arrangement of a modern fixed

crystal detector developed during World War II

for microwave work, particularly radar, is

shown in figure 38B. Once the cat whisker of

this unit is set at the factory to the most sen-

sitive spot on the surface of the silicon crys-

tal and its pressure is adjusted, a filler com-

pound is injected through the filling hole to

hold the cat whisker permanently in position.

12 -11 Receiver Adjustment

A simple regenerative receiver requires

little adjustment other than that necessary

to insure correct tuning and smooth regenera-

tion over some desired range. Receivers of

the tuned radio- frequency type and superheter-

odynes require precise alignment to obtain the

highest possible degree of selectivity and

sensitivity.

Good results can be obtained from a receiv-

er only when it is properly aligned and ad-

justed. The most practical technique for mak-

ing these adjustments is given below.

Instruments A very small number of instru-

ments will suffice to check and

align a communications receiver, the most im-

portant of these testing units being a modu-

lated oscillator and a d -c and a -c voltmeter.

The meters are essential in checking the volt-

age applied at each circuit point from the pow-

er supply. If the a -c voltmeter is of the oxide -

rectifier type, it can be used, in addition, as

an output meter when connected across the

receiver output when tuning to a modulated

signal. If the signal is a steady tone, such as

from a test oscillator, the output meter will

indicate the value of the detected signal. In

this manner, alignment results may be visually

noted on the meter.

T -R -F Receiver Alignment procedure in a mul-

Alignment tistage t -r -f receiver is exact-

ly the same as aligning a

single stage. If the detector is regenerative,

each preceding stage is successively aligned

while keeping the detector circuit tuned to the

test signal, the latter being a station signal

or one locally generated by a test oscillator

loosely coupled to the antenna lead. During

these adjustments, the r -f amplifier gain con-

trol is adjusted for maximum sensitivity, as-

suming that the r -f amplifier is stable and

does not oscillate. Often a sensitive receiver

can be roughly aligned by tuning for maximum

noise pickup.

Superheterodyne Aligning a superhet is a de-

Alignment tailed task requiring a great

amount of care and patience.

It should never be undertaken without a thor-

ough understanding of the involved job to be

done and then only when there is abundant

time to devote to the operation. There are no

short cuts; every circuit must be adjusted in-

dividually and accurately if the receiver is to

give peak performance. The precision of each

adjustment is dependent upon the accuracy

with which the preceding one was made.

Superhet alignment requires (1) a good sig-

nal generator (modulated oscillator) covering

the radio and intermediate frequencies and

equipped with an attenuator; (2) the necessary

socket wrenches, screwdrivers, or "neutral-

izing tools" to adjust the various i -f and r -f

trimmer capacitors; and (3) some convenient

type of tuning indicator, such as a copper -

oxide or electronic voltmeter.

Throughout the alignment process, unless

specifically stated otherwise, the r -f gain con-

trol must be set for maximum output, the beat

oscillator switched off, and the a.v.c. turned

off or shorted out. When the signal output of

the receiver is excessive, either the attenuator

or the a -f gain control may be turned down, but

never the r -f gain control.

I -F Alignment After the receiver has been

given a rigid electrical and

mechanical inspection, and any faults which

may have been found in wiring or the selec-

tion and assembly of parts corrected, the i -f

amplifier may be aligned as the first step in

the checking operations.

Vl ith the signal generator set to give a modu-

lated signal on the frequency at which the i -f

www.americanradiohistory.com

234 Radio Receiver Fundamentals THE RADIO

amplifier is to operate, clip the "hot" output

lead from the generator to the last i -f stage

through a small fixed capacitor to the control

grid. Adjust both trimmer capacitor's in the

last i -f transformer (the one between the last

i -f amplifier and the second detector) to reson-

ance as indicated by maximum deflection of

the output meter.

Each i -f stage is adjusted in the same man-

ner, moving the hot lead, stage by stage, back

toward the front end of the receiver and back-

ing off the attenuator as the signal strength

increases in each new position. The last ad-

justment will be made to the first i -f trans-

former, with the hot signal generator lead con-

nected to the control grid of the mixer. Oc-

casionally it is necessary to disconnect the

mixer grid lead from the coil, grounding it

through a 1,000- or 5,000 -ohm resistor, and

coupling the signal generator through a small

capacitor to the grid.

When the last i -f adjustment has been com-

pleted, it is good practice to go back through

the i -f channel, re- peaking all of the trans-

formers. It is imperative that this recheck be

made in sets which do not include a crystal

filter, and where the simple alignment of the

i -f amplifier to the generator is final.

I -F with There are several ways of align -

Crystal Filter ing an i -f channel which con-

tains a crystal -filter circuit.

However, the following method is one which

has been found to give satisfactory results in

every case: An unmodulated signal generator

capable of tuning to the frequency of the filter

crystal in the receiver is coupled to the grid of

the stage which precedes the crystal filter in

the receiver. Then, with the crystal filter

switched in, the signal generator is tuned

slowly to find the frequency where the crystal

peaks. The receiver "S" meter may be used

as the indicator, and the sound heard from the

loudspeaker will be of assistance in finding

the point. When the frequency at which the

crystal peaks has been found, all the i -f trans-

formers in the receiver should be touched up

to peak at that frequency.

B -F -Q Adjustment Adjusting the beat oscil-

lator on a receiver that has

no front panel adjustment is relatively simple.

It is only necessary to tune the receiver to

resonance with any signal, as indicated by

the tuning indicator, and then turn on the b.f.o.

and set its trimmer (or trimmers) to produce

the desired beat note. Setting the beat oscil-

lator in this way will result in the beat note

being stronger on one "side" of the signal

than on the other, which is what is desired

for c -w reception. The b.f.o. should not be set

to zero beat when the receiver is tuned to

I -F SIGNAL IN

ALONE

t ` F PLUS O MULTIPLIER

w

f

W f

55 CC

FREQUENCY

Figure 39

THE Q- MULTIPLIER

The loss resistance of a high -Q circuit

is neutralized by regeneration in a

simple feedback amplifier. A highly

selective passband is produced which

is coupled to the i -f circuit of the

receiver.

resonance with the signal, as this will cause

an equally strong beat to be obtained on both

sides of resonance.

Front -End Alignment of the front end of a

Alignment home -constructed receiver is a

relatively simple process, con-

sisting of first getting the oscillator to cover

the desired frequency range and then of peak-

ing the various r -f circuits for maximum gain.

However, if the frequency range covered by

the receiver is very wide a fair amount of cut

and try will be required to obtain satisfactory

tracking between the r -f circuits and the oscil-

lator. Manufactured communications receivers

should always be tuned in accordance with

the instructions given in the maintenance man-

ual for the receiver.

12 -12 Receiving Accessories

The Q- Multiplier The selectivity of a receiver

may be increased by raising

the Q of the tuned circuits of the i -f strip. A

simple way to accomplish this is to add a con-

trolled amount of positive feedback to a tuned

circuit, thus increasing its Q. This is done in

the 0- multiplier, whose basic circuit is shown

in figure 39. The circuit L -C1 -C2 is tuned to

www.americanradiohistory.com

HANDßCOK Receiving Accessories 235

I.F SIGNAL IN I -F SIGNAL OJT

O MULTIPLIER "NULL"

455 KC

FREQUENCY

Figure 40

Q- MULTIPLIER NULL CIRCUIT

The addition of a second triode permits

the 0- Multiplier to be used for nulling

out an unwanted hetrodyne.

the intermediate frequency, and the loss re-

sistance of the circuit is neutralized by the

positive feedback circuit composed of C3 and

the vacuum tube. Too great a degree of positive

feedback will cause the circuit to break into

oscillation.

At the resonant frequency, the impedance of

the tuned circuit is very high, and when shunted

across an i -f stage will have little effect upon

the signal. At frequencies removed from reso-

nance, the impedance of the circuit is low, re-

sulting in high attenuation of the i -f signal.

The resonant frequency of the Q- multiplier may

by varied by changing the value of one of the

components in the tuned circuit.

The Q- multiplier may also be used to "null"

a signal by employing negative feedback to

control the plate resistance of an auxiliary

amplifier stage as shown in figure 40. Since the

grid- cathode phase shift through the Q- multiplier

is zero, the plate resistance of a second tube

may be readily controlled by placing it across

the Q- multiplier. At resonance, the high nega-

tive feedback drops the plate resistance of

V2, shunting the i -f circuit. Off resonance, the

feedback is reduced and the plate resistance

of V2 rises, reducing the amount of signal at-

tenuation in the i -f strip. A circuit combining

both the "peak" and "null" features is shown

in figure 41.

The Product Detector A version of the common

mixer or converter stage

12

TO PLATE TERMINAL 10X7

OF FIRST I-F TUBE TNRU _ 6

2. OF COAXIAL LINE

.005 Lt

2 12Ax7

L2

.005

NULL PEAR

SELECTIfIrY

CONTROLS

3

1.SK 32 MEG 10K A.

I 1 n T

- B +200-300 V. - 6.3 V.

LI. GRAY6'URNE V6 CNOKE (0.6-6.OAIN)

L22GRAY6URNE "LOOPSTICK" COIL

Figure 41

SCHEMATIC OF A 455KC

0- MULTIPLIER

Coil LI is required to tune out the

reactance of the coaxial line. It is ad-

justed for maximum signal response.

LI may be omitted if the Q- multiplier

is connected to the receiver with o

short length of wire, and the i -f trans-

former within the receiver is retuned.

may be used as a second detector in a receiver

in place of the usual diode detector. The diode

is an envelope detector (section 12 -1) and de-

velops a d -c output voltage from a single r -f

signal, and audio "beats" from two or more

input signals. A product detector (figure 42)

requires that a local carrier voltage be present

in order to produce an audio output signal.

R -F SIGNAL PRODUCT

DETECTOR HAUDIO OUTPUT

LOCAL

OSCILLATOR

Figure 42

THE PRODUCT

DETECTOR

Audio output signal is

developed only when

local oscillator is on.

www.americanradiohistory.com

236 Radio Receiver Fundamentals

I -F SIGNAL

Figure 43

PENTAGRID MIXER

USED AS PRODUCT

DETECTOR

Such a detector is useful for single sideband

work, since the inter -modulation distortion is

extremely low.

A pentagrid product detector is shown in

figure 43. The incoming signal is applied to

grid 3 of the mixer tube, and the local oscillator

is injected on grid 1. Grid bias is adjusted for

operation over the linear portion of the tube

characteristic curve. When grid 1 injection

is removed, the audio output from an unmodu-

lated signal applied to grid 3 should be reduced

approximately 30 to 40 db below normal de-

tection level. When the frequency of the local

oscillator is synchronized with the incoming

carrier, amplitude modulated signals may be

received by exalted carrier reception, wherein

the local carrier substitutes for the transmitted

carrier of the a -m signal.

Three triodes may be used as a product

detector (figure 44). Triodes V1 and V2 act

as cathode followers, delivering the sideband

signal and the local oscillator signal to a

grounded grid triode (V3) which functions as

the mixer stage. A third version of the product

detector is illustrated in figure 45. A twin

triode tube is used. Section V1 functions as

a cathode follower amplifier. Section V2 is a

. ì ..

I-F

SIC.

V,

12AU7 Va VS

12AU7 0, r+AU01O OUT

SO

SEAT OSC.

SIGNAL

Figure 44

TRIPLE -TRIODE

PRODUCT DETECTOR

4711

VI and V2 act as cathode follow-

ers, delivering sideband signal

and local oscillator signal to

grounded grid triode mixer (V3).

Figure 45

DOUBLE -TRIODE

PRODUCT DETECTOR

"plate" detector, the cathode of which is

common with the cathode follower amplifier.

The local oscillator signal is injected into the

grid circuit of tube V2.

?tl II14611 IIIIII vIIl°1i 'If#I!IIItIYIIiIIIIvIIIII'I+II

i.h°1 .1. 1 1

Figure 46

EXPLODED VIEW OF COLLINS

MECHANICAL FILTER

www.americanradiohistory.com

CHAPTER THIRTEEN

Generation of

Radio Frequency Energy

A radio communication or broadcast trans-

mitter consists of a source of radio frequency

power, or carrier; a system for modulating the

carrier whereby voice or telegraph keying or

other modulation is superimposed upon it; and

an antenna system, including feed line, for

radiating the intelligence- carrying radio fre-

quency power. The power supply employed to

convert primary power to the various voltages

required by the r -f and modulator portions of

the transmitter may also be considered part

of the transmitter.

Voice modulation usually is accomplished

by varying either the amplitude or the frequen-

cy of the radio frequency carrier in accordance

with the components of intelligence to be

transmitted.

Radiotelegraph modulation (keying) normal-

ly is accomplished either by interrupting,

shifting the frequency of, or superimposing an

audio tone on the radio -frequency carrier in

accordance with the dots and dashes to be

transmitted.

The complexity of the radio- frequency gen-

erating portion of the transmitter is dependent

upon the power, order of stability, and frequen-

cy desired. An oscillator feeding an antenna

directly is the simplest form of radio- frequency

generator. A modern high -frequency transmitter,

on the other hand, is a very complex generator.

Such an equipment usually comprises a very

237

stable crystal -controlled or self -controlled

oscillator to stabilize the output frequency, a

series of frequency multipliers, one or more

amplifier stages to increase the power up to

the level which is desired for feeding the an-

tenna system, and a filter system for keeping

the harmonic energy generated in the trans-

mitter from being fed to the antenna system.

13 -1 Self- Controlled

Oscillators

In Chapter Four, it was explained that the

amplifying properties of a tube having three

or more elements give it the ability to gener-

ate an alternating current of a frequency de-

termined by the components associated with

it. A vacuum tube operated in such a circuit

is called an oscillator, and its function is

essentially to convert direct current into radio -

frequency alternating current of a predeter-

mined frequency.

Oscillators for controlling the frequency of

conventional radio transmitters can be divided

into two general classes: self -controlled and

crystal- controlled.

There are a great many types of self -con-

trolled oscillators, each of which is best suited

www.americanradiohistory.com

238 Generation of R -F Energy THE RADIO

OA SHUNT -FED HARTLEY

R

250

GRID

COIL

OB SHUNT -FED COLPITTS

R

L+ 250 Li

OD TUNED -PLATE UNTUNED GRID EO ELECTRON COUPLED

L

OG CLAPP HO CLAPP ELECTRON COUPLED

FO COLPITTS ELECTRON COUPLED

Figure 1

COMMON TYPES OF SELF -EXCITED OSCILLATORS

Fixed capacitor values are typical, but will vary somewhat with the application.

In the Clapp oscillator circuits (G) and (H), capacitors Cr and C2 should have a

reactance of 50 to 100 ohms at the operating frequency of the oscillator. Tuning of

these two oscillators is accomplished by capacitor C. In the circuits of (E), (F), and

(H), tuning of the tank circuit in the plate of the oscillator tube will have relatively

small effect on the frequency of oscillation. The plate tank circuit also may, if de-

sired, be tuned to a harmonic of the oscillation frequency, or a broadly resonant

circuit may be used in this circuit position.

to a particular application. They can further

be subdivided into the classifications of: neg-

ative -grid oscillators, electron -orbit oscilla-

tors, negative -resistance oscillators, velocity

modulation oscillators, and magnetron oscil-

lators.

Negative -Grid A negative -grid oscillator is

Oscillators essentially a vacuum -tube am-

plifier with a sufficient por-

tion of the output energy coupled back into the

input circuit to sustain oscillation. The con-

trol grid is biased negatively with respect to

the cathode. Common types of negative -grid

oscillators are diagrammed in figure 1.

The Hartley Illustrated in figure 1 (A) is the

oscillator circuit which finds the

most general application at the present time;

this circuit is commonly called the Hartley.

The operation of this oscillator will be de-

scribed as an index to the operation of all

negative -grid oscillators; the only real differ-

www.americanradiohistory.com

HANDBOOK Oscillators 239

ence between the various circuits is the man-

ner in which energy for excitation is coupled

from the plate to the grid circuit.

When plate voltage is applied to the Hartley

oscillator shown at (A), the sudden flow of

plate current accompanying the application of

plate voltage will cause an electro- magnetic

field to be set up in the vicinity of the coil.

The building -up of this field will cause a po-

tential drdp to appear from turn -to -turn along

the coil. Due to the inductive coupling be-

tween the portion of the coil in which the plate

current is flowing and the grid portion, a po-

tential will be induced in the grid portion.

Since the cathode tap is between the grid

and plate ends of the coil, the induced grid

voltage acts in such a manner as to increase

further the plate current to the tube. This ac-

tion will continue for a short period of time

determined by the inductance and capacitance

of the tuned circuit, until the flywheel effect

of the tuned circuit causes this action to come

to a maximum and then to reverse itself. The

plate current then decreases, the magnetic

field around the coil also decreasing, until a

minimum is reached, when the action starts

again in the original direction and at a greater

amplitude than before. The amplitude of these

oscillations, the frequency of which is de-

termined by the coil -capacitor circuit, will in-

crease in a very short period of time to a limit

determined by the plate voltage of the oscil-

lator tube.

The Colpitts Figure 1 (8) shows a version of

the Colpitts oscillator. It can

be seen that this is essentially the same cir-

cuit as the Hartley except that the ratio of a

pair of capacitances in series determines the

effective cathode tap, instead of actually us-

ing a tap on the tank coil. Also, the net ca-

pacitance of these two capacitors comprises

the tank capacitance of the tuned circuit. This

oscillator circuit is somewhat less suscep-

tible to parasitic (spurious) oscillations than

the Hartley.

For best operation of the Hartley and Col-

pitts oscillators, the voltage from grid to cath-

ode, determined by the tap on the coil or the

setting of the two capacitors, normally should

be from 1/3 to 1/5 that appearing between

plate and cathode.

The T.P.T.G. The tuned -plate tuned -grid os-

cillator illustrated at (C) has

a tank circuit in both the plate and grid cir-

cuits. The feedback of energy from the plate

to the grid circuits is accomplished by the

plate -to -grid inter -electrode capacitance with-

in the tube. The necessary phase reversal in

feedback voltage is provided by tuning the

grid tank capacitor to the low side of the de-

sired frequency and the plate capacitor to the

high side. A broadly resonant coil may be sub-

stituted for the grid tank to form the T.N. T.

oscillator shown at (D).

Electron -Coupled In any of the oscillator cir-

Oscillators cuits just described it is

possible to take energy from

the oscillator circuit by coupling an external

load to the tank circuit. Since the tank circuit

determines the frequency of oscillation of the

tube, any variations in the conditions of the

external circuit will be coupled back into the

frequency determining portion of the oscillator.

These variations will result in frequency in-

stability.

The frequency determining portion of an

oscillator may be coupled to the load circuit

only by an electron stream, as illustrated in

(E) and (F) of figure 1. When it is considered

that the screen of the tube acts as the plate

to the oscillator circuit, the plate merely act-

ing as a coupler to the load, then the sim-

ilarity between the cathode -grid- screen circuit

of these oscillators and the cathode -grid -plate

circuits of the corresponding prototype can be

seen.

The electron- coupled oscillator has good

stability with respect to load and voltage var-

iation. Load variations have a relatively small

effect on the frequency, since the only cou-

pling between the oscillating circuit and the

load is through the electron stream flowing

through the other elements to the plate. The

plate is electrostatically shielded from the

oscillating portion by the bypassed screen.

The stability of the e.c.o. with respect to

variations in supply voltages is explained as

follows: The frequency will shift in one direc-

tion with an increase in screen voltage, while

an increase in plate voltage will cause it to

shift in the other direction. By a proper pro-

portioning of the resistors that comprise the

voltage divider supplying screen voltage, it is

possible to make the frequency of the oscil-

lator substantially independent of supply volt-

age variations.

The Clapp A relatively new type of oscillator

Oscillator circuit which is capable of giving

excellent frequency stability is

illustrated in figure 1G. Comparison between

the more standard circuits of figure IA through

IF and the Clapp oscillator circuits of figures

1G and 1H will immediately show one marked

difference: the tuned circuit which controls

the operating frequency in the Clapp oscillator

is series resonant, while in all the more stand-

ard oscillator circuits the frequency control-

ling circuit is parallel resonant. Also, the

capacitors C, and C, are relatively large in

terms of the usual values for a Colpitts oscil-

www.americanradiohistory.com

240 Generation of R -F Energy THE RADIO

lator. In fact, the value of capacitors C, and

C, will be in the vicinity of 0.001 µfd. to

0.0025 µfd. for an oscillator which is to be

operated in the 1.8 -Mc. band.

The Clapp oscillator operates in the follow-

ing manner: at the resonant frequency of the

oscillator tuned circuit (L, C) the impedance

of this circuit is at minimum (since it oper-

ates in series resonance) and maximum cur-

rent flows through it. Note however, that C,

and C, also are included within the current

path for the series resonant circuit, so that at

the frequency of resonance an appreciable

voltage drop appears across these capacitors.

The voltage drop appearing across C, is ap-

plied to the grid of the oscillator tube as ex-

citation, while the amplified output of the

oscillator tube appears across C, as the driv-

ing power to keep the circuit in oscillation.

Capacitors C, and C, should be made as

large in value as possible, while still permit-

ting the circuit to oscillate over the full tun-

ing range of C. The larger these capacitors

are made, the smaller will be the coupling be-

tween the oscillating circuit and the tube, and

consequently the better will be oscillator sta-

bility with respect to tube variations. High Gm

tubes such as the 6AC7, 6ÁG7, and 6CB6 will

permit the use of larger values of capacitance

at C, and C, than will more conventional tubes

such as the 6SJ7, 6V6, and such types. In gen-

eral it may be said that the reactance of ca-

pacitors C, and C, should be on the order of

40 to 120 ohms at the operating frequency of

the oscillator -with the lower values of re-

actance going with high -Gm tubes and the

higher values being necessary to permit oscil-

lation with tubes having Gm in the range of

2000 micromhos such as the 6SJ7.

It will be found that the Clapp oscillator

will have a tendency to vary in power output

over the frequency range of tuning capacitor

C. The output will be greatest where C is at

its largest setting, and will tend to fall off

with C at minimum capacitance. In fact, if

capacitors C, and C, have too large a value

the circuit will stop oscillation near the mini-

mum capacitance setting of C. Hence it will

be necessary to use a slightly smaller value

of capacitance at C, and C, (to provide an in-

crease in the capacitive reactance at this

point), or else the frequency range of the oscil-

lator must be restricted by paralleling a fixed

capacitor across C so that its effective capaci-

tance at minimum setting will be increased to

a value which will sustain oscillation.

In the triode Clapp oscillator, such as shown

at figure 1G, output voltage for excitation of

an amplifier, doubler, or isolation stage nor-

mally is taken from the cathode of the oscil-

lator tube by capacitive coupling to the grid

of the next tube. However, where greater iso-

lation of succeeding stages from the oscillat-

ing circuit is desired, the electron- coupled

Clapp oscillator diagrammed in figure 1H may

be used. Output then may be taken from the

plate circuit of the tube by capacitive coupling

with either a tuned circuit, as shown, or with

an r -f choke or a broadly resonant circuit in

the plate return. Alternatively, energy may be

coupled from the output circuit L,-C, by link

coupling. The considerations with regard to

C C and the grid tuned circuit are the same

as for the triode oscillator arrangement of

figure 1G.

Negative Resist- Negative- resistance oscil-

ance Oscillators lators often are used when

unusually high frequency

stability is desired, as in a frequency meter.

The dynatron of a few years ago and the newer

transitron are examples of oscillator circuits

which make use of the negative resistance

characteristic between different elements in

some multi -grid tubes.

In the dynatron, the negative resistance is a

consequence of secondary emission of elec-

trons from the plate of a tetrode tube. By a

proper proportioning of the electrode voltage,

an increase in screen voltage will cause a

decrease in screen current, since the increased

screen voltage will cause the screen to attract

a larger number of the secondary electrons

emitted by the plate. Since the net screen cur-

rent flowing from the screen supply will be

decreased by an increase in screen voltage,

it is said that the screen circuit presents a

negative resistance.

If any type of tuned circuit, or even a re-

sistance- capacitance circuit, is connected in

series with the screen, the arrangement will

oscillate -provided, of course, that the external

circuit impedance is greater than the negative

resistance. A negative resistance effect simi-

lar to the dynatron is obtained in the transitron

circuit, which uses a pentode with the suppres-

sor coupled to the screen. The negative re-

sistance in this case is obtained from a com-

bination of secondary emission and inter -elec-

trode coupling, and is considerably more stable

than that obtained from uncontrolled secondary

emission alone in the dynatron. A representa-

tive transitron oscillator circuit is shown in

figure 2.

The chief distinction between a conven-

tional negative grid oscillator and a negative

resistance oscillator is that in the former the

tank circuit must act as a phase inverter in

order to permit the amplification of the tube

to act as a negative resistance, while in the

latter the tube acts as its own phase inverter.

Thus a negative resistance oscillator requires

only an untapped coil and a single capacitor

www.americanradiohistory.com

HANDBOOK Oscillators 241

6SK7

O TRANSITRON OSCILLATOR

6SN7 OR 6J6

O CATHODE COUPLED OSCILLATOR

Figure 2

TWO -TERMINAL OSCILLATOR CIRCUITS

Both circuits may be used for an audio

oscillator or for frequencies into the

v -h -f range simply by placing a tank cir-

cuit tuned to the proper frequency where

indicated on the drawing. Recommended

values for the components are given be-

low for both oscillators.

TRANSITION OSCILLATOR

C,- 0.01 -µfd. mica for r.f. 10 -µfd. elect. for c.f.

C2- 0.00005 -µfd. mica for r.f. 0.1 -4fd. paper for

a.f.

C3- 0.003 -µfd. mico for r.f. 03-4fd. paper for

a.f.

C4- 001 -µfd. mica for r.f. 8 -4fd. elect. for a.f.

R3-220K S5 -watt carbon

R2 1800 ohms 55 -watt carbon

R3 -22K 2 -watt carbon

R4 -22K 2 -watt carbon

CATHODE -COUPLED OSCILLATOR

C3- 0.00005 -µfd. mica for r.f. 0.1 -µfd. paper

for audio

C2- 0.003 -µfd. mica for rd. 8 -pfd. lect. for

audio

R1-47K S5 -watt carbon

R2 -1K 1 -watt carbon

as the frequency determining tank circuit, and

is classed as a two terminal oscillator. In fact,

the time constant of an R/C circuit may be

used as the frequency determining element and

such an oscillator is rather widely used as a

tunable audio frequency oscillator.

The Fronklin The Franklin oscillator makes

Oscillator use of two cascaded tubes to

obtain the negative- resistance

effect (figure 3). The tubes may be either a

pair of triodes, tetrodes, or pentodes, a dual

triode, or a combination of a triode and a multi -

grid tube. The chief advantage of this oscil-

lator circuit is that the frequency determining

tank only has two terminals, and one side of

the circuit is grounded.

The second tube acts as a phase inverter to

give an effect similar to that obtained with the

dynatron or transitron, except that the effective

transconductance is much higher. If the tuned

circuit is omitted or is replaced by a resistor,

the circuit becomes a relaxation oscillator or

a multivibrator.

Oscillator The Clapp oscillator has proved

Stability to be inherently the most stable

of all the oscillator circuits dis-

cussed above, since minimum coupling be-

tween the oscillator tube and its associated

tuned circuit is possible. However, this in-

herently good stability is with respect to tube

variations; instability of the tuned circuit with

respect to vibration or temperature will of

course have as much effect on the frequency of

oscillation as with any other type of oscillator

circuit. Solid mechanical construction of the

components of the oscillating circuit, along

with a small negative -coefficient compensating

capacitor included as an element of the tuned

circuit, usually will afford an adequate degree

of oscillator stability.

Figure 3

THE FRANKLIN OSCILLATOR CIRCUIT

A separate phase inverter tube is used in

this oscillator to feed a portion of the output

back to the input in the proper phase to sus-

tain oscillation. The values of Cr and C2

should be as small as will permit oscillations

to be sustained over the desired frequency

range.

www.americanradiohistory.com

242 Generation of R -F Energy THE RADIO

V. F.O. Transmit - When used to control the fre-

ter Controls quency of a transmitter in

which there are stringent

limitations on frequency tolerance, several pre-

cautions are taken to ensure that a variable

frequency oscillator will stay on frequency.

The oscillator is fed from a voltage regulated

power supply, uses a well designed and tem-

perature compensated tank circuit, is of rugged

mechanical construction to avoid the effects

of shock and vibration, is protected against

excessive changes in ambient room tempera-

ture, and is isolated from feedback or stray

coupling from other portions of the transmitter

by shielding, filtering of voltage supply leads,

and incorporation of one or more buffer- ampli-

fier stages. In a high power transmitter a small

amount of stray coupling from the final ampli-

fier to the oscillator can produce appreciable

degradation of the oscillator stability if both

are on the same frequency. Therefore, the os-

cillator usually is operated on a subharmonic

of the transmitter output frequency, with one

or more frequency multipliers between the os-

cillator and final amplifier.

13 -2 Quartz Crystal

Oscillators

Quartz is a naturally occuring crystal hav-

ing a structure such that when plates are cut

in certain definite relationships to the crystal-

lographic axes, these plates will show the

piezoelectric effect -the plates will be de-

formed in the influence of an electric field,

and, conversely, when such a plate is com-

pressed or deformed in any way a potential

difference will appear upon its opposite sides.

The crystal has mechanical resonance, and

will vibrate at a very high frequency because

of its stiffness, the natural period of vibration

depending upon the dimensions, the method of

electrical excitation, and crystallographic

orientation. Because of the piezoelectric pro-

perties, it is possible to cut a quartz plate

which, when provided with suitable electrodes,

will have the characteristics of a series reso-

nant circuit with a very high L/C ratio and

very high Q. The Q is several times as high

as can be obtained with an inductor -capacitor

combination in conventional physical sizes.

The equivalent electrical circuit is shown in

figure 4A, the resistance component simply

being an acknowledgment of the fact that the

Q, while high, does not have an infinite value.

The shunt capacitance of the electrodes and

associated wiring (crystal holder and socket,

plus circuit wiring) is represented by the dot-

ted portion of figure 4B. In a high frequency

CI

(SMALL)

Lt

(LARGE)

RI

(SMALL)

4: C2

I (sraAr

SHUNT)

- J

L

Figure 4

EQUIVALENT ELECTRICAL CIRCUIT OF

QUARTZ PLATE IN A HOLDER

At (A) is shown the equivalent series -reso-

nant circuit of the crystal itself, at (B) is

shown how the shunt capacitance of the

holder electrodes and associated wiring af-

fects the circuit to the combination circuit

of (C) which exhibits both series resonance

and parallel resonance (anti -resonance), the

separation in frequency between the two

modes being very small and determined by

the ratio of C1 to C,.

crystal this will be considerably greater than

the capacitance component of an equivalent

series L/C circuit, and unless the shunt ca-

pacitance is balanced out in a bridge circuit,

the crystal will exhibit both resonant (series

resonant) and anti- resonant (parallel resonant)

frequencies, the latter being slightly higher

than the series resonant frequency and ap-

proaching it as C, is increased.

The series resonance characteristic is em-

ployed in crystal filter circuits in receivers

and also in certain oscillator circuits wherein

the crystal is used as a selective feedback

element in such a manner that the phase of the

feedback is correct and the amplitude ade-

quate only at or very close to the series reso-

nant frequency of the crystal.

While quartz, tourmaline, Rochelle salts,

ADP, and EDT crystals all exhibit the piezo-

electric effect, quartz is the material widely

employed for frequency control.

As the cutting and grinding of quartz plates

has progressed to a high state of development

and these plates may be purchased at prices

which discourage the cutting and grinding by

simple hand methods for one's own use, the

procedure will be only lightly touched upon

here. The crystal blank is cut from the raw quartz

at a predetermined orientation with respect to

the optical and electrical axes, the orientation

determining the activity, temperature coeffi-

cient, thickness coefficient, and other charac-

teristics. Various orientations or "cuts" hav-

ing useful characteristics are illustrated in

figure 5.

www.americanradiohistory.com

HANDBOOK Crystal Oscillators 243

Figure 5

ORIENTATION OF THE

COMMON CRYSTAL CUTS

neo T[.neror[ co rmus..T

OSCILLATORS eD fllT[eS

MK.. fe[0U[CT A7.11

LOW feCWCMCT CT.DT,[T,fT

AT SSiS'

ST

CT 34.

¡ T .N` er -

IT -ST. b

y

27e0 Tóir[eTUet

Xff: ó 1 L

a` OSCILLATORS -'

1 >

veD.tl D

' xo.o ^p.¢

Ta

L

,

2[1.10

S e S,

URUC

`«e. s.ocs

a o nA ZERO

.T ..[ee , .

r OXillTOe3

.r.D filTCeS p.c...., _

cDC CCnTUT f lTtes rr - [.

The crystal blank is then rough -ground al-

most to frequency, the frequency increasing

in inverse ratio to the oscillating dimension

(usually the thickness). It is then finished to

exact frequency either by careful lapping, by

etching, or plating. The latter process con-

sists of finishing it to a frequency slightly

higher than that desired and then silver plating

the electrodes right on the crystal, the fre-

quency decreasing as the deposit of silver is

increased. If the crystal is not etched, it must

be carefully scrubbed and "baked" several

times to stabilize it, or otherwise the frequen-

cy and activity of the crystal will change with

time. Irradiation by X -rays recently has been

used in crystal finishing.

Unplated crystals usually are mounted in

pressure holders, in which two electrodes are

held against the crystal faces under slight

pressure. Unplated crystals also are some-

times mounted in an air -gap holder, in which

there is a very small gap between the crystal

and one or both electrodes. By making this

gap variable, the frequency of the crystal may

be altered over narrow limits (about 0.3% for

certain types).

The temperature coefficient of frequency for

various crystal cuts of the " -T" rotated fam-

ily is indicated in figure 5. These angles are

typical, but crystals of a certain cut will vary

slightly. By controlling the orientation and di-

mensioning, the turning point (point of zero

temperature coefficient) for a BT cut plate may

be made either lower or higher than the 75 de-

grees shown. Also, by careful control of axes

and dimensions, it is possible to get AT cut

crystals with a very flat temperature- frequency

characteristic.

The first quartz plates used were either Y

cut or X cut. The former had a very high tem-

perature coefficient which was discontinuous,

causing the frequency to jump at certain criti-

cal temperatures. The X cut had a moderately

bad coefficient, but it was more continuous,

and by keeping the crystal in a temperature

controlled oven, a high order of stability could

be obtained. However, the X cut crystal was

considerably less active than the Y cut, es-

pecially in the case of poorly ground plates.

For frequencies between 500 kc. and about

6 Mc., the AT cut crystal now is the most

widely used. It is active, can be made free

from spurious responses, and has an excellent

temperature characteristic. However, above

about 6 Mc. it becomes quite thin, and a diffi-

cult production job. Between 6 Mc. and about

12 Mc., the BT cut plate is widely used. It

also works well between 500 kc. and 6 Mc.,

but the AT cut is more desirable when a high

order of stability is desired and no crystal

oven is employed.

For low frequency operation on the order of

100 kc., such as is required in a frequency

standard, the GT cut crystal is recommended,

though CT and DT cuts also are widely used

for applications between 50 and 500 kc. The

CT, DT, and GT cut plates are known as con-

tour cuts, as these plates oscillate along the

long dimension of the plate or bar, and are

much smaller physically than would be the

case for a regular AT or BT cut crystal for

the same frequency.

Crystal Holders Crystals normally are pur-

chased ready mounted. The

www.americanradiohistory.com

244 Generation of R -F Energy THE RADIO

best type mount is determined by the type crys-

tal and its application, and usually an opti-

mum mounting is furnished with the crystal.

However, certain features are desirable in all

holders. One of these is exclusion of moisture

and prevention of electrode oxidization. The

best means of accomplishing this is a metal

holder, hermetically sealed, with glass insula-

tion and a metal -to -glass bond. However, such

holders are more expensive, and a ceramic or

phenolic holder with rubber gasket will serve

where requirements are not too exacting.

Temperature Control; Where the frequency tol-

Crystal Ovens erance requirements are

not too stringent and the

ambient temperature does not include extremes,

an AT -cut plate, or a BT -cut plate with opti-

mum (mean temperature) turning point, will

often provide adequate stability without re-

sorting to a temperature controlled oven. How-

ever, for broadcast stations and other applica-

tions where very close tolerances must be

maintained, a thermostatically controlled oven,

adjusted for a temperature slightly higher than

the highest ambient likely to be encountered,

must of necessity be employed.

Harmonic Cut Just as a vibrating string can

Crystals be made to vibrate on its har-

monics, a quartz crystal will

exhibit mechanical résonance (and therefore

electrical resonance) at harmonics of its funda-

mental frequency. When employed in the usual

holder, it is possible to excite the crystal

only on its odd harmonics (overtones).

By grinding the crystal especially for har-

monic operation, it is possible to enhance its

operation as a harmonic resonator. BT and AT

cut crystals designed for optimum operation

on the 3d, 5th and even the 7th harmonic are

available. The 5th and 7th harmonic types,

especially the latter, require special holder

and oscillator circuit precautions for satis-

factory operation, but the 3d harmonic type

needs little more consideration than a regular

fundamental type. A crystal ground for optimum

operation on a particular harmonic may or may

not be a good oscillator on a different har-

monic or on the fundamental. One interesting

characteristic of a harmonic cut crystal is that

its harmonic frequency is not quite an exact

multiple of its fundamental, though the dis-

parity is very small.

The harmonic frequency for which the crys-

tal was designed is the working frequency. It

is not the fundamental since the crystal itself

actually oscillates on this working frequency

when it is functioning in the proper manner.

When a harmonic -cut crystal is employed, a

selective tuned circuit must be employed some-

where in the oscillator in order to discrimi-

6J5 ETC EXCITATION

EXCITATION

+5

loo -isov.

BASIC PIERCE" OSCILLATOR HOT -CATHODE -PIERCE"

OSCILLATOR

Figure 6

THE PIERCE CRYSTAL OSCILLATOR

CIRCUIT

Shown at (A) is the basic Pierce crystal os-

cillator circuit. A capacitance of 10 to 75

µtd. normally will be required at C1 for

optimum operation. If a plate supply voltage

higher thon indicated is to be used, RFC'

may be replaced by a 22,000 -ohm 2 -watt re-

sistor. Shown at (B) is an alternative ar-

rangement with the r -f ground moved to the

plate, and with the cathode floating. This

alternative circuit has the advantage that

the full r -f voltage developed across the

crystal may be used os excitation to the next

stage, since one side of the crystal is

grounded.

nate against the fundamental frequency or un-

desired harmonics. Otherwise the crystal might

not always oscillate on the intended frequency.

For this reason the Pierce oscillator, later

described in this chapter, is not suitable for

use with harmonic -cut crystals, because the

only tuned element in this oscillator circuit

is the crystal itself.

Crystal Current; For a given crystal op-

Heating and Fracture erating as an anti -reso-

nant tank in a given os-

cillator at fixed load impedance and plate and

screen voltages, the r -f current through the

crystal will increase as the shunt capacitance

C2 of figure 4 is increased, because this effec-

tively increases the step -up ratio of C, to Cr.

For a given shunt capacitance, C the crystal

current for a given crystal is directly propor-

tional to the r -f voltage across C,. This volt-

age may be measured by means of a vacuum

tube voltmeter having a low input capacitance,

and such a measurement is a more pertinent

one than a reading of r -f current by means of

a thermogalvanometer inserted in series with

one of the leads to the crystal holder.

The function of a crystal is to provide accu-

rate frequency control, and unless it is used

in such a manner as to take advantage of its

inherent high stability, there is no point in

using a crystal oscillator. For this reason a

www.americanradiohistory.com

HANDBOOK Crystal Oscillators 245

crystal oscillator should not be run at high

plate input in an attempt to obtain consider-

able power directly out of the oscillator, as

such operation will cause the crystal to heat,

with resultant frequency drift and possible

fracture.

13 -3 Crystal Oscillator

Circuits

Considerable confusion exists as to nomen-

clature of crystal oscillator circuits, due to a

tendency to name a circuit after its discoverer.

Nearly all the basic crystal oscillator circuits

were either first used or else developed inde-

pendently by G. W. Pierce, but he has not been

so credited in all the literature.

Use of the crystal oscillator in master os-

cillator circuits in radio transmitters dates

back to about 1924 when the first application

articles appeared.

The Pierce The circuit of figure6A is the sim-

Oscillator plest crystal oscillator circuit. It

is one of those developed by

Pierce, and is generally known among ama-

teurs as the Pierce oscillator. The crystal

simply replaces the tank circuit in a Colpitts

or ultra -audion oscillator. The r -f excitation

voltage available to the next stage is low, be-

ing somewhat less than that developed across

the crystal. Capacitor C, will make more of

the voltage across the crystal available for

excitation, and sometimes will be found nec-

essary to ensure oscillation. Its value is small,

usually approximately equal to or slightly

greater than the stray capacitance from the

plate circuit to ground (including the grid of

the stage being driven).

If the r -f choke has adequate inductance, a

crystal (even a harmonic cut crystal) will al-

most invariably oscillate on its fundamental.

The Pierce oscillator therefore cannot be used

with harmonic cut crystals.

The circuit at (B) is the same as that of

(A) except that the plate instead of the cath-

ode is operated at ground r -f potential. All of

the r -f voltage developed across the crystal

is available for excitation to the next stage,

but still is low for reasonable values of crys-

tal current. For best operation a tube with low

heater - cathode capacitance is required. Exci-

tation for the next stage may also be taken

from the cathode when using this circuit.

Tuned -Plate The circuit shown in fig -

Crystal Oscillator ure 7A is also one used by

Pierce, but is more widely

referred to as the " \filler "oscillator. To avoid

confusion, we shall refer to it as the tuned -

plate crystal oscillator. It is essentially an

Armstrong or tuned plate -tuned grid oscillator

with the crystal replacing the usual L -C grid

tank. The plate tank must be tuned to a fre-

quency slightly higher than the anti -resonant

(parallel resonant) frequency of the crystal.

Whereas the Pierce circuits of figure 6 will

oscillate at (or very close to) the anti -reso-

nant frequency of the crystal, the circuits of

figure 7 will oscillate at a frequency a little

above the anti -resonant frequency of the

crystal.

The diagram shown in figure 7A is the basic

circuit. The most popular version of the tuned -

plate oscillator employs a pentode or beam

tetrode with cathode bias to prevent excessive

plate dissipation when the circuit is not oscil-

lating. The cathode resistor is optional. Its

omission will reduce both crystal current and

oscillator efficiency, resulting in somewhat

more output for a given crystal current. The

tube usually is an audio or video beam pentode

or tetrode, the plate -grid capacitance of such

tubes being sufficient to ensure stable oscilla-

tion but not so high as to offer excessive feed-

back with resulting high crystal current. The

6AG7 makes an excellent all- around tube for

this type circuit.

Pentode The usual type of crystal -

Harmonic Crystal controlled h -f transmitter

Oscillator Circuits operates, at least part of

the time, on a frequency

which is an integral multiple of the operating

frequency of the controlling crystal. Hence,

oscillator circuits which are capable of pro-

viding output on the crystal frequency if de-

sired, but which also can deliver output energy

on harmonics of the crystal frequency have

come into wide use. Four such circuits which

have found wide application are illustrated in

figures 7C, 7D, 7E, and 7F.

The circuit shown in figure 7C is recom-

mended for use with harmonic -cut crystals

when output is desired on a multiple of the

oscillating frequency of the crystal. As an

example, a 25-Mc. harmonic -cut crystal may

be used in this circuit to obtain output on 50

Mc., or a 48 -Mc. harmonic -cut crystal may be

used to obtain output on the 144 -Mc. amateur

band. The circuit is not recommended for use

with the normal type of fundamental- frequency

crystal since more output with fewer variable

elements can be obtained with the circuits of

7D and 7F.

The Pierce -harmonic circuit shown in fig-

ure 7D is satisfactory for many applications

which require very low crystal current, but

has the disadvantage that both sides of the

crystal are above ground potential. The Tri -tet

circuit of figure 7E is widely used and can

www.americanradiohistory.com

246 Generation of R -F Energy THE RADIO

6J5,Erc F 6V6. 6AQ5, ETC 6 AG7, 6AQ5. 5763 3F

002

150V.

BASIC TUNED -PLATE OSCILLATOR

6AG7 F, 2F, 3F

+250 V.

RECOMMENDED TUNED -PLATE

OSCILLATOR

6AG7 F, 2F, 3F, 4F

+250 V.

SPECIAL C RCUIT FOR USE WITH

HARMONIC CUT CRYSTAL.

10111JF

6AG7 F. 2F, 3F, 4F

22 K

PIERCE HARMONIC CIRCUIT "TRITET" CIRCUIT

150 ULF

+250 V.

COLPITTS HARMONIC OSCILLATOR

Figure 7

COMMONLY USED CRYSTAL OSCILLATOR CIRCUITS

Shown at (A) is the basic tuned -plate crystal oscillator with a triode oscillator tube.

The plate tank must be tuned on the low -capacitance side of resonance to sustain

oscillation. (8) shows the tuned -plat oscillator as it is normally used with an a -f

power pentode to permit high output with relatively low crystal current.

Schematics (C), (D), (E), and (F) illustrate crystal oscillator circuits which can de-

liver moderate output energy on harmonics of the oscillating frequency of the crys-

tal. (C) shows a special circuit which will permit use of a harmonic -cut crystal to

obtain output energy well into the v -h -f range. (D) is valuable when extremely low

crystal current is a requirement, but delivers relatively low output. (E) is commonly

used, but is subject to crystal damage if the cathode circuit is mistuned. (F) is

recommended as the most generally satisfactory from the standpoints of: low crys-

tal current regardless of mis- adjustment, good output on harmonic frequencies, one

side of crystal is grounded, will oscillate with crystals from 1.5 to 70 Mc. without

adjustment, output tank may be tuned to the crystal frequency for fundamental out-

put without stopping oscillation or changing frequency.

give excellent output with low crystal current.

However, the circuit has the disadvantages of

requiring a cathode coil, of requiring careful

setting of the variable cathode capacitor to

avoid damaging the crystal when changing fre-

quency ranges, and of having both sides of

the crystal above ground potential.

The Colpitts harmonic oscillator of figure

7F is recommended as being the most gener-

ally satisfactory harmonic crystal oscillator

circuit since it has the following advantages:

(1) the circuit will oscillate with crystals

over a very wide frequency range with no

change other than plugging in or switching in

the desired crystal; (2) crystal current is ex-

tremely low; (3) one side of the crystal is

grounded, which facilitates crystal- switching

circuits; (4) the circuit will operate straight

through without frequency pulling, or it may

be operated with output on the second, third,

or fourth harmonic of the crystal frequency.

Crystal Oscillator The tunable circuits of all

Tuning oscillators illustrated

should be tuned for maxi-

mum output as indicated by maximum excita-

tion to the following stage, except that the

oscillator tank of tuned -plate oscillators (fig-

ure 7A and figure 7B) should be backed off

slightly towards the low capacitance side from

www.americanradiohistory.com

HANDBOOK All -band Crystal Oscillator 247

6AG7

77. a 305

eaW M /N /JUC TOR

(2 0 vN) +300 V.

NOTES

I. Li'/sUN (2 f" OF eew 0301s)

2. L2 = /.gLN (/" OF 80W .3003)

3. FOR 160 METER OPERATION ADD s 4/1/F. CONDENSER

BETWEEN PINS 418 OF 6A67. PLATE COIL= 35 2/P+.

(21-.0; Dew R awe)

4. X' 7 MC. CRYSTAL FOR HARMON /C OPERATION

Figure 8

ALL -3AND 6AG7 CRYSTAL OSCILLATOR

CAPABLE OF DRIVING

BEAM PENTODE TUBE

maximum output, as the oscillator then is in

a more stable condition and sure to start im-

mediately when power is applied. This is es-

pecially important when the oscillator is keyed,

as for break -in c -w operation.

Crystal Switching It is desirable to keep stray

shunt capacitances in the

crystal circuit as low as possible, regardless

of the oscillator circuit. If a selector switch

is used, this means that both switch and crys-

tal sockets must be placed close to the oscil-

lator tube socket. This is especially true of

harmonic -cut crystals operating on a compara-

tively high frequency. In fact, on the highest

frequency crystals it is preferable to use a

turret arrangement for switching, as the stray

capacitances can be kept lower.

Crystol Oscillator When the crystal oscillator

Keying is keyed, it is necessary

that crystal activity and os-

cillator -tube transconductance be moderately

high, and that oscillator loading and crystal

shunt capacitance be low. Below 2500 kc. and

above 6 Mc. these considerations become es-

pecially important. Keying of the plate voltage

(in the negative lead) of a crystal oscillator,

with the screen voltage regulated at about

150 volts, has been found to give satisfactory

results.

A Versatile 6AG7 The 6AG7 tube may be

Crystal Oscillator used in a modified Tri -tet

crystal oscillator, capable

of delivering sufficient power on all bands

from 160 meters through 10 meters to fully

drive a pentode tube, such as the 807, 2E26

or 6146. Such an oscillator is extremely use-

ful for portable or mobile work, since it com-

bines all essential exciter functions in one

tube. The circuit of this oscillator is shown

in figure 8. For 160, 80 and 40 meter opera-

tion the 6AG7 functions as a tuned -plate os-

cillator. Fundamental frequency crystals are

used on these three bands. For 20, 15 and 10

meter operation the 6AG7 functions as a Tri-

tet oscillator with a fixed -tuned cathode cir-

cuit. The impedance of this cathode circuit

does not affect operation of the 6AG7 on the

lower frequency bands so it is left in the cir-

cuit at all times. A 7 -Mc. crystal is used for

fundamental output on 40 meters, and for har-

monic output on 20, 15 and 10 meters. Crystal

current is extremely low regardless of the out-

put frequency of the oscillator. The plate cir-

cuit of the 6AG7 is capable of tuning a fre-

quency range of 2:1, requiring only two output

coils: one for 80 -40 meter operation, and one

for 20, 15 and 10 meter operation. In some

cases it may be necessary to add 5 micromicro-

farads of external feedback capacity between

the plate and control grid of the 6AG7 tube to

sustain oscillation with sluggish 160 meter

crystals.

Triode Overtone The recent development of

Oscillators reliable overtone crystals

capable of operation on the

third, fifth, seventh (or higher) overtones has

made possible v -h -f output from a low frequen-

cy crystal by the use of a double triode regen-

erative oscillator circuit. Some of the new

twin triode tubes such as the 12AU7, 12AV7

and 6J6 are especially satisfactory when used

in this type of circuit. Crystals that are ground

for overtone service may be made to oscillate

on odd overtone frequencies other than the

one marked on the crystal holder. A 24 -Mc.

overtone crystal, for example, is a specially

ground 8 -Mc. crystal operating on its third

overtone. In the proper circuit it may be made

to oscillate on 40 Mc. (fifth overtone), 56 Mc.

(seventh overtone), or 72 Mc. (ninth overtone).

Even the ordinary 8 -Mc. crystals not designed

for overtone operation may be made to oscil-

late readily on 24 Mc. (third overtone) in these

circuits.

A variety of overtone oscillator circuits is

shown in figure 9. The oscillator of figure 9A

is attributed to Frank Jones, W6AJF. The first

section of the 6J6 dual triode comprises a re-

generative oscillator, with output on either the

third or fifth overtone of the crystal frequency.

The regenerative loop of this oscillator con-

sists of a condenser bridge made up of C, and

C2, with the ratio C2 /C, determining the amount

of regenerative feedback in the circuit. With

www.americanradiohistory.com

248 Generation of R -F Energy THE RADIO

+300 V.

AO JONES HARMONIC OSCILLATOR

12AÚ7

RFC

3F

6,9,10 o915 F

+300 v.

6, 9F

FOR MC. CRYSTAL

L I= 9 7. a 5003 9iW M /N /DUCTOR

L2. 4 T 13003 817 MIN /DOCTOR

THESE COLS MADE FROM SINGLE SECT /ON

NE TRN OF O /V /OE I INDUCTOR INTO TWO COILS S

BROKEN TO

12AT7

+300 V.

6,9F

E© CATHODE FOLLOWER OVERTONE OSCILLATOR

2F 6J6

+300 V

4F

FOR 7 MC. CRYSTAL

LI =28T e 24 ON NATIONAL XPSO

FORM

L 2. $ T Il /! ON NATIONAL XRSO

FORM

6J6

3F /s0 50

+300V

6,9F

FOR 8MC. CRYSTAL

L I = /07 a 30 /I 64W M NIDUC TOR,

TAP AT 37. FROM GRID ENO

OD REGENERATIVE HARMONIC OSCILLATOR

- 12AT7 (or 6AB4)

_ _ !00

F=IA4MC. 1 L2

_ +200 V.

LI=STe/E, f- O.SPACED

L2=/ T. NOOXUPWIRE, O.

OF V.H.F. OVERTONE OSCILLATOR

Figure 9

VARIOUS TYPES OF OVERTONE OSCILLATORS USING MINIATURE DOUBLE -TRIODE

TUBES

an 8 -plc. crystal, output from the first section

of the 6J6 tube may be obtained on either 24

Mc. or 40 Mc., depending upon the resonant

frequency of the plate circuit inductor, L,. The

second half of the 636 acts as a frequency

multiplier, its plate circuit, L2, tuned to the

sixth or ninth harmonic frequency when L, is

tuned to the third overtone, or to the tenth

harmonic frequency when L, is tuned to the

fifth overtone.

Figure 9B illustrates a Colpitts overtone

oscillator employing a 636 tube. This is an

outgrowth of the Colpitts harmonic oscillator

of figure 7F. The regenerative loop in this

case consists of C C2 and RFC between the

grid, cathode and ground of the first section

of the 6J6. The plate circuit of the first sec-

tion is tuned to the second overtone of the

crystal, and the second section of the 636

doubles to the fourth harmonic of the crystal.

This circuit is useful in obtaining 28 -Mc. out-

put from a 7 -Mc. crystal and is highly popular

in mobile work.

The circuit of figure 9C shows a typical re-

generative overtone oscillator employing a

12AÚ7 double triode tube. Feedback is con-

trolled by the number of turns in L2, and the

coupling between L2 and L,. Only enough feed-

www.americanradiohistory.com

HANDBOOK R -F Amplifiers 249

back should be employed to maintain proper

oscillation of the crystal. Excessive feedback

will cause the first section of the 12ÁU7 to

oscillate as a self- excited TNT oscillator,

independent of the crystal. A variety of this

circuit is shown in figure 9D, wherein a tapped

coil, L is used in place of the two separate

coils. Operation of the circuit is the same in

either case, regeneration now being controlled

by the placement of the tap on L,.

A cathode follower overtone oscillator is

shown in figure 9E. The cathode coil, L is

chosen so as to resonate with the crystal and

tube capacities just below the third overtone

frequency of the crystal. For example, with an

8 -Mc. crystal, L3 is tuned to 24 Mc.. L, reson-

ates with the circuit capacities to 23.5 Mc.,

and the harmonic tank circuit of the second

section of the 12AT7 is tuned either to 48 Mc.

or 72 11c. If a 24 -Mc. overtone crystal is used

in this circuit, L, may be tuned to 72 Mc., L,

resonates with the circuit capacities to 70

Mc., and the harmonic tank circuit, L is tuned

to 144 Mc. If there is any tendency towards

self -oscillation in the circuit, it may be elimi-

nated by a small amount of inductive coupling

between L2 and L3. Placing these coils near

each other, with the winding of L, correctly

polarized with respect to L3 will prevent self -

oscillation of the circuit.

The use of a 144 -Mc. overtone crystal is

illustrated in figure 9F. A 6AB4 or one -half

of a 12AT7 tube may be used, with output

directly in the 2 -meter amateur band. A slight

amount of regeneration is provided by the one

turn link, L which is loosely coupled to the

144 -Mc. tuned tank circuit, L, in the plate cir-

cuit of the oscillator tube. If a 12AT7 tube

and a 110 -Mc. crystal are employed, direct out-

put in the 220 -Mc. amateur band may be ob-

tained from the second half of the 12AT7.

13 -4 Radio Frequency

Amplifiers

The output of the oscillator stage in a trans-

mitter (whether it be self -controlled or crystal

controlled) must be kept down to a fairly low

level to maintain stability and to maintain a

factor of safety from fracture of the crystal

when one is used. The low power output of

the oscillator is brought up to the desired

power level by means of radio -frequency am-

plifiers. The two classes of r -f amplifiers that

find widest application in radio transmitters

are the Class B and Class C types.

The Class B Class B amplifiers are used in a

Amplifier radio -telegraph transmitter when

maximum power gain and mini-

mum harmonic output is desired in a particu-

lar stage. A Class B amplifier operates with

cutoff bias and a comparatively small amount

of excitation. Power gains of 20 to 200 or so

are obtainable in a well- designed Class B

amplifier. The plate efficiency of a Class B

c -w amplifier will run around 65 per cent.

The Class B Another type of Class B ampli -

Linear fier is the Class B linear stage

as employed in radiophone work.

This type of amplifier is used to increase the

level of a modulated carrier wave, and de-

pends for its operation upon the linear rela-

tion between excitation voltage and output

voltage. Or, to state the fact in another man-

ner, the power output of a Class B linear stage

varies linearly with the square of the excita-

tion voltage.

The Class B linear amplifier is operated

with cutoff bias and a small value of excita-

tion, the actual value of exciting power being

such that the power output under carrier con-

ditions is one -fourth of the peak power capa-

bilities of the stage. Class B linears are very

widely employed in broadcast and commercial

installations, but are comparatively uncommon

in amateur application, since tubes with high

plate dissipation are required for moderate

output. The carrier efficiency of such an am-

plifier will vary from approximately 30 per

cent to 35 per cent.

The Class C Class C amplifiers are very wide -

Amplifier ly used in all types of trans-

mitters. Good power gain may be

obtained (values of gain from 3 to 20 are com-

mon) and the plate circuit efficiency may be,

under certain conditions, as high as 85 per

cent. Class C amplifiers operate with consider-

ably more than cutoff bias and ordinarily with

a large amount of excitation as compared to a

Class B amplifier. The bias for a normal Class

C amplifier is such that plate current on the

stage flows for approximately 120° of the 360°

excitation cycle. Class C amplifiers are used

in transmitters where a fairly large amount of

excitation power is available and good plate

circuit efficiency is desired.

Plate Modulated The characteristic of a Class

Class C C amplifier which makes it

linear with respect to

changes in plate voltage is that which allows

such an amplifier to be plate modulated for

radiotelephony. Through the use of higher bias

than is required for a c -w Class C amplifier

and greater excitation, the linearity of such

an amplifier may be extended from zero plate

voltage to twice the normal value. The output

power of a Class C amplifier, adjusted for

plate modulation, varies with the square of the

www.americanradiohistory.com

250 Generation of R -F Energy THE RADIO

plate voltage. This is the same condition that

would take place if a resistor equal to the

voltage on the amplifier, divided by its plate

current, were substituted for the amplifier.

Therefore, the stage presents a resistive load

to the modulator.

Grid Modulated If the grid current to a Class

Class C C amplifier is reduced to a

low value, and the plate load-

ing is increased to the point where the plate

dissipation approaches the rated value, such

an amplifier may be grid modulated for radio-

telephony. If the plate voltage is raised to

quite a high value and the stage is adjusted

carefully, efficiencies as high as 40 to 43 per

cent with good modulation capability and com-

paratively low distortion may be obtained.

Fixed bias is required. This type of operation

is termed Class C grid -bias modulation.

Grid Excitation Adequate grid excitation

must be available for Class

B or Class C service. The excitation for a

plate- modulated Class C stage must be suffi-

cient to produce a normal value of d -c grid cur-

rent with rated bias voltage. The bias voltage

preferably should be obtained from a combina-

tion of grid leak and fixed C -bias supply.

Cutoff bias can be calculated by dividing

the amplification factor of the tube into the

d -c plate voltage. This is the value normally

used for Class B amplifiers (fixed bias, no

grid resistor). Class C amplifiers use from 1

to 5 times this value, depending upon the avail-

able grid drive, or excitation, and the desired

plate efficiency. Less grid excitation is need-

ed for c -w operation, and the values of fixed

bias (if greater than cutoff) may be reduced, or

the value of the grid leak resistor can be low-

ered until normal rated d -c grid current flows.

The values of grid excitation listed for each

type of tube may be reduced by as much as

50 per cent if only moderate power output and

plate efficiency are desired. When consulting

the tube tables, it is well to remember that

the power lost in the tuned circuits must be

taken into consideration when calculating the

available grid drive. At very high frequencies,

the r -f circuit losses may even exceed the

power required for actual grid excitation.

Link coupling between stages, particularly

to the final amplifier grid circuit, normally will

provide more grid drive than can be obtained

from other coupling systems. The number of

turns in the coupling link, and the location of

the turns on the coil, can be varied with res-

pect to the tuned circuits to obtain the great-

est grid drive for allowable values of buffer

or doubler plate current. Slight readjustments

sometimes can be made after plate voltage

has been applied to the driver tube.

Excessive grid current damages tubes by

overheating the grid structure; beyond a cer-

tain point of grid drive, no increase in power

output can be obtained for a given plate volt-

age.

13 -5 Neutralization of

R.F. Amplifiers

The plate -to -grid feedback capacitance of

triodes makes it necessary that they be neu-

tralized for operation as r -f amplifiers at fre-

quencies above about 500 kc. Those screen -

grid tubes, pentodes, and beam tetrodes which

have a plate -to -grid capacitance of 0.1 µµEd.

or less may be operated as an amplifier with-

out neutralization in a well -designed amplifier

up to 30 Mc.

Neutralizing The object of neutralization is

Circuits to cancel or neutralize the ca-

pacitive feedback of energy from

plate to grid. There are two general methods

by which this energy feedback may be elimi-

nated: the first, and the most common method,

is through the use of a capacitance bridge,

and the second method is through the use of a

parallel reactance of equal and opposite po-

larity to the grid -to -plate capacitance, to nul-

lify the effect of this capacitance.

Examples of the first method are shown in

figure 10. Figure l0A shows a capacity neu-

tralized stage employing a balanced tank cir-

cuit. Phase reversal in the tank circuit is ob-

tained by grounding the center of the tank coil

to radio frequency energy by condenser C.

Points A and B are 180 degrees out of phase

with each other, and the correct amount of out

of phase energy is coupled through the neu-

tralizing condenser NC to the grid circuit of

the tube. The equivalent bridge circuit of this

is shown in figure 11A. It is seen that the

bridge is not in balance, since the plate -fila-

ment capacity of the tube forms one leg of the

bridge, and there is no corresponding capacity

from the neutralizing condenser (point B) to

ground to obtain a complete balance. In addi-

tion, it is mechanically difficult to obtain a

perfect electrical balance in the tank coil, and

the potential between point A and ground and

point B and ground in most cases is unequal.

This circuit, therefore, holds neutralization

over a very small operating range and unless

tubes of low interelectrode capacity are used

the inherent unbalance of the circuit will per-

mit only approximate neutralization.

Split-Stator

Plate Neutrali-

zation

Figure 10B shows the neu-

tralization circuit which is

most widely used in single -

ended r -f stages. The use of

www.americanradiohistory.com

HANDBOOK Neutralization 251

Figure 10

COMMON NEUTRALIZING CIRCUITS FOR SINGLE -ENDED AMPLIFIERS

a split -stator plate capacitor makes the electri-

cal balance of the circuit substantially inde-

pendent of the mutual coupling within the coil

and also makes the balance independent of the

place where the coil is tapped. With conven-

tional tubes this circuit will allow one neutral-

ization adjustment to be made on, say, 28 Mc.,

and this adjustment usually will hold suffi-

ciently close for operation on all lower fre-

quency bands.

Condenser C, is used to balance out the

plate -filament capacity of the tube to allow a

perfect neutralizing balance at all frequencies.

The equivalent bridge circuit is shown in fig-

ure 11B. If the plate- filament capacity of the

tube is extremely low (100TH triode, for ex-

ample), condenser C, may be omitted, or may

merely consist of the residual capacity of NC

to ground.

Grid Neutralization A split grid tank circuit

may also be used for neu-

tralization of a triode tube as shown in figure

IOC. Out of phase voltage is developed across

a balanced grid circuit, and coupled through

NC to the single -ended plate circuit of the

tube. The equivalent bridge circuit is shown

in figure 11C. This circuit is in balance until

the stage is in operation when the loading ef-

fect of the tube upon one -half of the grid cir-

cuit throws the bridge circuit out of balance.

The amount of unbalance depends upon the

grid -plate capacity of the tube, and the amount

of mutual inductance between the two halves

of the grid coil. If an r -f voltmeter is placed

between point A and ground, and a second

voltmeter placed between point B and ground

the loading effect of the tube will be notice-

able. When the tube is supplied excitation

with no plate voltage, NC may be adjusted

until the circuit is in balance. When plate

voltage is applied to the stage, the voltage

from point A to ground will decrease, and the

voltage from point B to ground will increase,

both in direct proportion to the amount of cir-

cuit unbalance. The use of this circuit is not

recommended above 7 Mc., and it should be

used below that frequency only with low in-

ternal capacity tubes.

Push -Pull Two tubes of the same type

Neutralization can be connected for push -pull

operation so as to obtain twice

as much output as that of a single tube. A

push -pull amplifier, such as that shown in fig-

ure 12 also has an advantage in that the cir-

cuit can more easily be balanced than a single -

tube r -f amplifier. The various inter -electrode

capacitances and the neutralizing capacitors

are connected in such a manner that the re-

actances on one side of the tuned circuits are

exactly equal to those on the opposite side.

For this reason, push -pull r -f amplifiers can

be more easily neutralized in very- high -fre-

quency transmitters; also, they usually remain

in perfect neutralization when tuning the am-

plifier to different bands.

The circuit shown in figure 12 is perhaps

www.americanradiohistory.com

252 Generation of R -F Energy THE RADIO

F.AC

OA BRIDGE EQUIVALENT OF FIGURE IO -A

C

OB BRIDGE EQUIVALENT OF FIGURE 10 -B

C

(RES.DUAL

CAPACITY)`

CG-r

(s1IALL) RFC

Figure 11

EQUIVALENT NEUTRALIZING CIRCUITS

the most commonly used arrangement for a

push -pull r -f amplifier stage. The rotor of the

grid capacitor is grounded, and the rotor of the

plate tank capacitor is by- passed to ground.

Shunt or Coil The feedback of energy from

Neutralization grid to plate in an unneutral-

ized r -f amplifier is a result of

the grid -to -plate capacitance of the amplifier

tube. A neutralization circuit is merely an

electrical arrangement for nullifying the effect

of this capacitance. All the previous neutrali-

zation circuits have made use of a bridge cir-

cuit for balancing out the grid -to -plate energy

feedback by feeding hack an equal amount of

energy of opposite phase.

Another method of eliminating the feedback

effect of this capacitance, and hence of neu-

tralizing the amplifier stage, is shown in fig-

ure 13. The grid -to -plate capacitance in the

triode amplifier tube acts as a capacitive re-

Figure 12

STANDARD CROSS- NEUTRALIZED

PUSH -PULL TRIODE AMPLIFIER

actance, coupling energy back from the plate

to the grid circuit. If this capacitance is par-

alleled with an inductance having the same

value of reactance of opposite sign, the re-

actance of one will cancel the reactance of

the other and a high -impedance tuned circuit

from grid to plate will result.

This neutralization circuit can be used on

ultra -high frequencies where other neutraliza-

tion circuits are unsatisfactory. This is true

because the lead length in the neutralization

circuit is practically negligible. The circuit

can also be used with push -pull r -f amplifiers.

In this case, each tube will have its own neu-

tralizing inductor connected from grid to plate.

The main advantage of this arrangement is

that it allows the use of single -ended tank

circuits with a single -ended amplifier.

The chief disadvantage of the shunt neutral-

ized arrangement is that the stage must be re-

neutralized each time the stage is retuned to

a new frequency sufficiently removed that the

grid and plate tank circuits must be retuned to

resonance. However, by the use of plug -in

coils it is possible to change to a different

band of operation by changing the neutral-

izing coil at the same time that the grid and

plate coils are changed.

The 0.0001-pfd. capacitor in series with

the neutralizing coil is merely a blocking ca-

pacitor to isolate the plate voltage from the

grid circuit. The coil L will have to have a

very large number of turns for the band of oper-

ation in order to be resonant with the compara-

tively small grid -to -plate capacitance. But

since, in all ordinary cases with tubes operat-

ing on frequencies for which they were de-

signed, the L/C ratio of the tuned circuit will

be very high, the coil can use comparatively

small wire, although it must be wound on air

or very low loss dielectric and must be insu-

lated for the sum of the plate r -f voltage and

the grid r -f voltage.

www.americanradiohistory.com

HANDBOOK Neutralizing Procedure 253

Figure 13

COIL NEUTRALIZED AMPLIFIER

This neutralization circuit is very effective

with triode tubes on any frequency, but is

particularly effective in the v -h -f range. The

coil L is adjusted so that it resonates at the

operating frequency with the grid -to -plate

capacitance of the tube. Capacitor C may be

a very small unit of the low- capacitance

neutralizing type and is used to trim the cir-

cuit to resonance at the operating frequency.

If some means of varying the inductance of

the coil a small amount is available, the

trimmer capacitor is not needed.

13 -6 Neutralizing

Procedure

An r -f amplifier is neutralized to prevent

self -oscillation or regeneration. A neon bulb,

a flashlight lamp and loop of wire, or an r -f

galvanometer can be used as a null indicator

for neutralizing low -power stages. The plate

voltage lead is disconnected from the r -f am-

plifier stage while it is being neutralized.

Normal grid drive then is applied to the r -f

stage, the neutralizing indicator is coupled

to the plate coil, and the plate tuning capac-

itor is tuned to resonance. The neutralizing

capacitor (or capacitors) then can be adjusted

until minimum r.f. is indicated for resonant

settings of both grid and plate tuning capac-

itors. Both neutralizing capacitors are ad-

justed simultaneously and to approximately the

same value of capacitance when a physically

symmetrical push -pull stage is being neu-

tralized.

A final check for neutralization should be

made with a d -c milliammeter connected in the

grid leak or grid -bias circuit. There will be

no movement of the meter reading as the plate

circuit is tuned through resonance (without

plate voltage being applied) when the stage

is completely neutralized.

Plate voltage should be completely removed

by actually opening the d -c plate circuit. If

there is a d -c return through the plate supply,

a small amount of plate current will flow when

grid excitation is applied, even though no pri-

mary a -c voltage is being fed to the plate trans-

former.

A further check on the neutralization of any

r -f amplifier can be made by noting whether

maximum grid current on the stage comes at

the same point of tuning on the plate tuning

capacitor as minimum plate current. This check

is made with plate voltage on the amplifier

and with normal antenna coupling. As the plate

tuning capacitor is detuned slightly from reso-

nance on either side the grid current on the

stage should decrease the same amount and

without any sudden jumps on either side of

resonance. This will be found to be a very

precise indication of accurate neutralization

in either a triode or beam -tetrode r -f amplifier

stage, so long as the stage is feeding a load

which presents a resistive impedance at the

operating frequency.

Push -pull circuits usually can be more com-

pletely neutralized than single -ended circuits

at very high frequencies. In the intermediate

range of from 3 to 15 Mc., single -ended cir-

cuits will give satisfactory results.

Neutralization of Radio -frequency amplifiers

Screen -Grid R -F using screen -grid tubes can

Amplifiers be operated without any ad-

ditional provision for neu-

tralization at frequencies up to about 15 Mc.,

provided adequate shielding has been provided

between the input and output circuits. Special

v -h -f screen -grid and beam tetrode tubes such

as the 2E26, 807W, and 5516 in the low -power

category and HK -257B, 4E27/8001, 4 -125A,

and 4 -250A in the medium -power category can

frequently be operated at frequencies as high

as 100 Mc. without any additional provision

for neutralization. Tubes such as the 807,

2E22, HY -69, and 813 can be operated with

good circuit design at frequencies up to 30

Mc. without any additional provision for neu-

tralization. The 815 tube has been found to

require neutralization in many cases above

20 Mc., although the 829B tube will operate

quite stably at 100 Mc. without neutralization.

None of these tubes, however, has perfect

shielding between the grid and the plate, a

condition brought about by the inherent in-

ductance of the screen leads within the tube

itself. In addition, unless "watertight" shield-

ing is used between the grid and plate circuits

of the tube a certain amount of external leak-

age between the two circuits is present. These

difficulties may not be serious enough to re-

quire neutralization of the stage to prevent

oscillation, but in many instances they show

up in terms of key -clicks when the stage in

question is keyed, or as parasitics when the

stage is modulated. Unless the designer of the

equipment can carefully check the tetrode

www.americanradiohistory.com

254 Generation of R -F Energy THE RADIO

Figure 14

NEUTRALIZING CIRCUITS FOR

BEAM TETRODES

A conventional cross neutralized circuit for use with push -pull beam tetrodes is

shown at (A). The neutralizing capacitors (NC) usually consist of small plates or

rods mounted alongside the plate elements of the tubes. (B) and (C) show "grid

neutralized" circuits for use with a single -ended tetrode stage having either link

coupling or capacitive coupling into the grid tank. (D) shows a method of tuning the

screen -lead inductance to accomplish neutralization in a single -frequency v -h-f

tetrode amplifier, while (E) shows a method of neutralization by increasing the grid -

to -plate capacitance on a tetrode when the operating frequency is higher than that

frequency where the tetrode is "self- neutralized" as a result of series resonance

in the screen lead. Methods (D) and (E) normally are not practicable at frequencies

below about 50 Mc. with the usual types of beam tetrode tubes.

stage for miscellaneous feedback between the

grid and plate circuits, and make the neces-

sary circuit revisions to reduce this feedback

to an absolute minimum, it is wise to neutral-

ize the tetrode just as if it were a triode tube.

In most push -pull tetrode amplifiers the sim-

plest method of accomplishing neutralization

is to use the cross -neutralized capacitance

bridge arrangement as normally employed with

triode tubes. The neutralizing capacitances,

however, must be very much smaller than used

with triode tubes, values of the order of 0.2

wifd. normally being required with beam tetrode

tubes. This order of capacitance is far less

than can be obtained with a conventional neu-

tralizing capacitor at minimum setting, so the

neutralizing arrangement is most commonly

made especially for the case at hand. Most

common procedure is to bring a conductor (con-

nected to the opposite grid) in the vicinity of

the plate itself or of the plate tuning capacitor

of one of the tubes. Either one or two such

capacitors may be used, two being normally

used on a higher frequency amplifier in order

to maintain balance within the stage.

An example of this is shown in figure 14A.

www.americanradiohistory.com

H A N D B O O K Tetrode Neutralization 255

Neutralizing A single -ended tetrode r -f am-

Single -Ended plifier stage may be neutral -

Tetrode Stages ized in the same manner as

illustrated for a push -pull

stage in figure 14A, provided a split- stator

tank capacitor is in use in the plate circuit.

However, in the majority of single -ended tet-

rode r -f amplifier stages a single- section ca-

pacitor is used in the plate tank. Hence, other

neutralization procedures must be employed

when neutralization is found necessary.

The circuit shown in figure 14B is not a

true neutralizing circuit, in that the plate -to-

grid capacitance is not balanced out. However,

the circuit can afford the equivalent effect by

isolating the high resonant impedance of the

grid tank circuit from the energy fed back from

plate to grid. When NC and C are adjusted to

bear the following ratio to the grid -to -plate

capacitance and the total capacitance from

grid -to- ground in the output tube:

NC CsP

C Cas

both ends of the grid tank circuit will be at the

same voltage with respect to ground as a result

of r -f energy fed back to the grid circuit. This

means that the impedance from grid to ground

will be effectively equal to the reactance of

the grid -to- cathode capacitance in parallel

with the stray grid -to- ground capacitance, since

the high resonant impedance of the tuned cir-

cuit in the grid has been effectively isolated

from the feedback path. It is important to note

that the effective grid -to- ground capacitance

of the tube being neutralized includes the

rated grid -to- cathode or input capacitance of

the tube, the capacitance of the socket, wiring

capacitances and other strays, but it does not

include the capacitances associated with the

grid tuning capacitor. Also, if the tube is be-

ing excited by capacitive coupling from a pre-

ceding stage (as in figure 14C), the effective

grid -to- ground capacitance includes the out-

put capacitance of the preceding stage and

its associated socket and wiring capacitances.

Cancellation of The provisions discussed in

Screen -Lead the previous paragraphs are

Inductance for neutralization of the small,

though still important at the

higher frequencies, grid -to -plate capacitance

of beam -tetrode tubes. However, in the vicinity

of the upper- frequency limit of each tube type

the inductance of the screen lead of the tube

becomes of considerable importance. With a

tube operating at a frequency where the in-

ductance of the screen lead is appreciable,

the screen will allow a considerable amount

of energy leak- through from plate to grid even

though the socket terminal on the tube is care-

fully by- passed to ground. This condition takes

place even though the socket pin is bypassed

since the reactance of the screen lead

will allow a moderate amount of r -f potential

to appear on the screen itself inside the elec-

trode assembly in the tube. This effect has

been reduced to a very low amount in such

tubes as the Hytron 5516, and the Eimac 4X150A

and 4X500A but it is still quite appreciable in

most beam -tetrode tubes.

The effect of screen -lead inductance on the

stability of a stage can be eliminated at any

particular frequency by one of two methods.

These methods are: (1) Tuning out the screen -

lead inductance by series resonating the screen

lead inductance with a capacitor to ground.

This method is illustrated in figure 14D and is

commonly employed in commercially -built equip-

ment for operation on a narrow frequency band

in the range above about 75 Mc. The other

method (2) is illustrated in figure 14E and

consists in feeding back additional energy

from plate to grid by means of a small capac-

itor connected between these two elements.

Note that this capacitor is connected in such

a manner as to increase the effective grid -to-

plate capacitance of the tube. This method

has been found to be effective with 807 tubes

in the range above 50 Mc. and with tubes such

as the 4 -125A and 4 -250A in the vicinity of

their upper frequency limits.

Note that both these methods of stabilizing

a beam -tetrode v -h -f amplifier stage by can-

cellation of screen -lead inductance are suit-

able only for operation over a relatively narrow -

band of frequencies in the v -h -f range. At low-

er frequencies both these expedients for re-

ducing the effects of screen -lead inductance

will tend to increase the tendency toward os-

cillation of the amplifier stage.

Neutralizing When a stage cannot be com-

Problems pletely neutralized, the difficulty

usually can be traced to one or

more of the following causes: (1) Filament

leads not by- passed to the common ground of

that particular stage. (2) Ground lead from the

rotor connection of the split -stator tuning ca-

pacitor to filament open or too long. (3) Neu-

tralizing capacitors in a field of excessive

r.f. from one of the tuning coils. (4) Electro-

magnetic coupling between grid and plate

coils, or between plate and preceding buffer

or oscillator circuits. (5) Insufficient shielding

or spacing between stages, or between grid

and plate circuits in compact transmitters.

(6) Shielding placed too close to plate circuit

coils, causing induced currents in the shields.

(7) Parasitic oscillations when plate voltage

is applied. The cure for the latter is mainly a

matter of cut and try -rearrange the parts,

www.americanradiohistory.com

256 Generation of R -F Energy THE RADIO

GRID

LEAK

INTERWOUND COILS

(UNITY COUPLING)

Figure 15

GROUNDED -GRID AMPLIFIER

This type of triode amplifier requires no

neutralization, but can be used only with

tubes having o relatively low plate -to- cathode

capacitance

change the length of grid or plate or neutraliz-

ing leads, insert a parasitic choke in the grid

lead or leads, or eliminate the grid r -f chokes

which may be the cause of a low- frequency

parasitic(in conjunction with plate r -f chokes).

13- 7 Grounded Grid

Amplifiers

Certain triodes have a grid configuration

and lead arrangement which results in very low

plate to filament capacitance when the control

grid is grounded, the grid acting as an effec-

tive shield much in the manner of the screen

in a screen -grid tube.

By connecting such a triode in the circuit of

figure 15, taking the usual precautions against

stray capacitive and inductive coupling be-

tween input and output leads and components,

a stable power amplifier is realized which re-

quires no neutralization.

At ultra -high frequencies, where it is diffi-

cult to obtain satisfactory neutralization with

conventional triode circuits (particularly when

a wide band of frequencies is to be covered),

the grounded -grid arrangement is about the only

practicable means of employing a triode am-

plifier.

Because of the large amount of degeneration

inherent in the circuit, considerably more ex-

citation is required than if the same tube were

employed in a conventional grounded- cathode

circuit. The additional power required to drive

a triode in a grounded -grid amplifier is not

lost, however, as it shows up in the output cir-

cuit and adds to the power delivered to the

load. But nevertheless it means that a larger

driver stage is required for an amplifier of

OUT

Figure 16

CONVENTIONAL TRIODE FREQUENCY

MULTIPLIER

Small triodes such as the 604 operate satis-

factorily as frequency multipliers, and can

deliver output well into the v -h -t ronge. Re-

sistor R normally will have a value in the

vicinity of 100,000 ohms.

given output, because a moderate amount of

power is delivered to the amplifier load by the

driver stage of a grounded -grid amplifier.

13 -8 Frequency Multipliers

Quartz crystals and variable- frequency os-

cillators are not ordinarily used for direct con-

trol of the output of high- frequency transmit-

ters. Frequency multipliers are usually em-

ployed to multiply the frequency to the desired

value. These multipliers operate on exact mul-

tiples of the excitation frequency; a 3.6 -Mc.

crystal oscillator can be made to control the

output of a transmitter on 7.2 or 14.4 Mc., or

on 28.8 Mc., by means of one or more frequency

multipliers. Chen used at twice frequency,

they are often termed frequency doublers. A

simple doubler circuit is shown in figure 16.

It consists of a vacuum tube with its plate cir-

cuit tuned to twice the frequency of the grid

driving circuit. This doubler can be excited

from a crystal oscillator or another multiplier

or amplifier stage.

Doubling is best accomplished by operating

the tube with high grid bias. The grid circuit

is driven approximately to the normal value of

d -c grid current through the r -f choke and grid -

leak resistor, shown in figure 16. The resist-

ance value generally is from two to five times

as high as that used with the same tube for

straight amplification. Consequently, the grid

bias is several times as high for the same

value of grid current.

Neutralization is seldom necessary in a

doubler circuit, since the plate is tuned to

twice the frequency of the grid circuit. The

impedance of the grid driving circuit is very

low at the doubling frequency, and thus there

is little tendency for self -excited oscillation.

www.americanradiohistory.com

HANDBOOK Frequency Multipliers 257

Figure 17

FREQUENCY MULTIPLIER CIRCUITS

The output of a triode v -h -f frequency multi-

plier often maybe increased by neutralization

of the grid -to -plate capacitance as shown at

(A) above. Such o stage also may be oper-

ated as a straight amplifier when the occa-

sion demands. A pentode frequency multi-

plier is shown at (B). Conventional power

tetrodes operate satisfactorily as multipliers

so long as the output frequency is below

about 100 Mc. Above this frequency special

v -h -f tetrodes must be used to obtain satis-

factory output.

Frequency doublers require bias of several

times cutoff; high -ft tubes therefore are desir-

able for this type of service. Tubes which

have amplification factors from 20 to 200 are

suitable for doubler circuits. Tetrodes and

pentodes make excellent doublers. Low -ft

triodes, having amplification constants of from

3 to 10, are not applicable for doubler service.

In extreme cases the grid voltage must be as

high as the plate voltage for efficient doubling

action.

Angle of Flow The angle of plate current flow

in Frequency in a frequency multiplier is a

Multipliers very important factor in deter-

mining the efficiency. As the

angle of flow is decreased for a given value

of grid current, the efficiency increases. To

reduce the angle of flow, higher grid bias is

required so that the grid excitation voltage

will exceed the cutoff value for a shorter por-

tion of the exciting -voltage cycle. For a high

order of efficiency, frequency doublers should

have an angle of flow of 90 degrees or less,

tripiers 60 degrees or less, and quadruplers

I (CUTOrr)

TANN CIRCUIT OUTPUT VOLTAGE

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\ EXCITATION

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Figure 18

ILLUSTRATING THE ACTION OF A

FREQUENCY DOUBLER

45 degrees or less. Under these conditions

the efficiency will be on the same order as

the reciprocal of the harmonic on which the

stage operates. In other words the efficiency

of a doubler will be approximately % or 50

per cent, the efficiency of a tripler will be

approximately / or 33 per cent and that of a

quadrupler will be about 25 per cent. With good

stage design the efficiency can be somewhat

greater than these values, but as the angle

of flow is made greater than these limiting

values, the efficiency falls off rapidly. The

reason is apparent from a study of figure 18.

The pulses ABC, EFG, JKL illustrate 180 -

degree excitation pulses under Class B opera-

tion, the solid straight line indicating cutoff

bias. If the bias is increased by N times, to

the value indicated by the dotted straight line,

and the excitation increased until the peak

r -f voltage with respect to ground is the same

as before, then the excitation frequency can

be cut in half and the effective excitation

pulses will have almost the same shape as

before. The only difference is that every other

pulse is missing; MNO simply shows where

the missing pulse would go. However, if the

Q of the plate tank circuit is high, it will have

sufficient flywheel effect to carry over through

the missing pulse, and the only effect will be

that the plate input and r -f output at optimum

loading drop to approximately half. As the in-

put frequency is half the output frequency, an

efficient frequency doubler is the result.

By the same token, a tripler or quadrupler

can be analyzed, the tripler skipping two ex-

citation pulses and the quadrupler three. In

each case the excitation pulse ideally should

be short enough that it does not exceed 180

degrees at the output frequency; otherwise the

excitation actually is bucking the output over

a portion of the cycle.

In actual practice, it is found uneconomical

to provide sufficient excitation to run a tripler

or quadrupler in this fashion. Usually the ex-

www.americanradiohistory.com

258 Generation of R -F Energy THE RADIO

Figure 19

PUSH -PUSH FREQUENCY DOUBLER

The output of o doubler stage may be materi-

ally increased through the use of a push -push

circuit such as illustrated above.

citation pulses will be at least 90 degrees at

the exciting frequency, with correspondingly

low efficiency, but it is more practicable to

accept the low efficiency and build up the out-

put in succeeding amplifier stages. The effi-

ciency can become quite low before the power

gain becomes less than unity.

Push -Push Two tubes can be connected in

Multipliers parallel to give twice the output

of a single -tube doubler. If the

grids are driven out of phase instead of in

phase, the tubes then no longer work simul-

taneously, but rather one at a time. The effect

is to fill in the missing pulses (figure 18).

Not only is the output doubled, but several

advantages accrue which cannot be obtained

by straight parallel operation.

Chief among these is the effective neutral-

ization of the fundamental and all odd harmon-

ics, an advantage when spurious emissions

must be minimized. Another advantage is that

when the available excitation is low and ex-

citation pulses exceed 90 degrees, the output

and efficiency will be greater than for the

same tubes connected in parallel.

The same arrangement may be used as a

quadrupler, with considerably better efficiency

than for straight parallel operation, because

seldom is it practicable to supply sufficient

excitation to permit 45 degree excitation

pulses. As pointed out above, the push -push

arrangement exhibits better efficiency than a

single ended multiplier when excitation is in-

adequate for ideal multiplier operation.

A typical push -push doubler is illustrated

in figure 19. When high transconductance tubes

are employed, it is necessary to employ a

split- stator grid tank capacitor to prevent self

oscillation; with well screened tetrodes or

pentodes having medium values of transcon-

ductance, a split -coil arrangement with a sin-

gle- section capacitor may be employed (the

Figure 20

PUSH -PULL FREQUENCY TRIPLER

The push -pull tripler is advantageous in the

v -h -f ronge since circuit balance is main-

tained both in the input and output circuits.

If the circuit is neutralized it may be used

either as a straight amplifier or as a tripler.

Either triodes or tetrodes may be used; dual -

unit tetrodes such as the 815, 832A, and

8298 are particularly effective in the v -h -f

range.

center tap of the grid coil being by- passed to

ground).

Push -Pull Frequency It is frequently desirable

Tripiers in the case of u -h -f and

v -h -f transmitters that

frequency multiplication stages be balanced

with respect to ground. Further it is just as

easy in most cases to multiply the crystal or

v -f -o frequency by powers of three rather than

multiplying by powers of two as is frequently

done on lower frequency transmitters. Hence

the use of push -pull tripiers has become quite

prevalent in both commercial and amateur

v -h -f and u -h -f transmitter designs. Such stages

are balanced with respect to ground and appear

in construction and on paper essentially the

same as a push -pull r -f amplifier stage with

the exception that the output tank circuit is

tuned to three times the frequency of the grid

tank circuit. A circuit for a push -pull tripler

stage is shown in figure 20.

A push -pull tripler stage has the further

advantage in amateur work that it can also be

used as a conventional push -pull r -f amplifier

merely by changing the grid and plate coils

so that they tune to the same frequency. This

is of some advantage in the case of operating

the 50 -Mc. band with 50 -bic. excitation, and

then changing the plate coil to tune to 144

Mc. for operation of the stage as a tripler from

excitation on 48 Mc. This circuit arrangement

is excellent for operation with push -pull beam

tetrodes such as the 6360 and 829B, although

a pair of tubes such as the 2E26, or 5763 could

just as well be used if proper attention were

given to the matter of screen -lead inductance.

www.americanradiohistory.com

HANDBOOK Tank Circuits 259

13 -9 Tank Circuit

Capacitances

It is necessary that the proper value of Q

be used in the plate tank circuit of any r -f

amplifier. The following section has been de-

voted to a treatment of the subject, and charts

are given to assist the reader in the determina-

tion of the proper L/C ratio to be used in a

radio -frequency amplifier stage.

A Class C amplifier draws plate current in

the form of very distorted pulses of short dura-

tion. Such an amplifier is always operated in-

to a tuned inductance- capacitance or tank cir-

cuit which tends to smooth out these pulses,

by its storage or tank action, into a sine wave

of radio -frequency output. Any wave -form dis-

tortion of the carrier frequency results in har-

monic interference in higher- frequency chan-

nels. A Class A r -f amplifier would produce a sine

wave of radio -frequency output if its exciting

waveform were also a sine wave. However, a

Class A amplifier stage converts its d -c input

to r -f output by acting as a variable resistance,

and therefore heats considerably. A Class C

amplifier when driven hard with short pulses

at the peak of the exciting waveform acts more

as an electronic switch, and therefore can con-

vert its d -c input to r -f output with relatively

good efficiency. Values of plate circuit effi-

ciency from 65 to 85 per cent are common in

Class C amplifiers operating under optimum

conditions of excitation, grid bias, and load-

ing.

Tank Circuit Q As stated before, the tank cir-

cuit of a Class C amplifier

receives energy in the form of short pulses of

plate current which flow in the amplifier tube.

But the tank circuit must be able to store

enough energy so that it can deliver a current

essentially sine wave in form to the load. The

ability of a tank to store energy in this man-

ner may be designated as the effective Q of

the tank circuit. The effective circuit Q may

be stated in any of several ways, but essen-

tially the Q of a tank circuit is the ratio of the

energy stored to 2e times the energy lost per

cycle. Further, the energy lost per cycle must,

by definition, be equal to the energy delivered

to the tank circuit by the Class C amplifier

tube or tubes.

The Q of a tank circuit at resonance is equal

to its parallel resonant impedance (the reso-

nant impedance is resistive at resonance) di-

vided by the reactance of either the capaci-

tor or the inductor which go to make up the

tank. The inductive reactance is equal to the

capacitive reactance, by definition, at reso-

nance. Hence we may state:

DYNAMIC

CHARACTERISTIC (0\ A_

GRID SWING

Figure 21

CLASS C AMPLIFIER OPERATION

Plate current pulses are shown at (A), (e),

and (C). The dip in the top of the plate cur-

rent waveform will occur when the excitation

voltage is such that the minimum plate volt-

age dips below the maximum grid voltage.

A detailed discussion of the operation of

Class C amplifiers is given in Chapter Seven.

RL RL

Q = -=- Xc XL

where RL is the resonant impedance of the

tank and Xc is the reactance of the tank ca-

pacitor and XL is the reactance of the tank

coil. This value of resonant impedance, RL,

is the load which is presented to the Class C

amplifier tube in a single -ended circuit such

as shown in figure 21.

The value of load impedance, RL, which the

Class C amplifier tube sees may be obtained,

looking in the other direction from the tank

coil, from a knowledge of the operating con-

ditions on the Class C tube. This load imped-

ance may be obtained from the following ex-

pression, which is true in the general case of

any Class C amplifier:

Epm=

RL

2 Np lb Ebb

where the values in the equation have the char-

acteristics listed in the beginningof Chapter 6.

The expression above is academic, since

the peak value of the fundamental component

of plate voltage swing, Epm, is not ordinarily

known unless a high -voltage peak a -c voltmeter

is available for checking. Also, the decimal

value of plate circuit efficiency is not ordinari-

ly known with any degree of accuracy. How-

ever, in a normally operated Class C amplifier

www.americanradiohistory.com

260 Generation of R -F Energy THE RADIO

> s

w >

!

54

V- 8 u

3.

x

100 10 ,S 20

TANK CIRCUIT Q

25

Figure 22

RELATIVE HARMONIC OUTPUT

PLOTTED AGAINST TANK CIRCUIT Q

30

the plate voltage swing will be approximately

equal to 0.85 to 0.9 times the d -c plate voltage

on the stage, and the plate circuit efficiency

will be from 70 to 80 per cent (Np of 0.7 to

0.8), the higher values of efficiency normally

being associated with the higher values of

plate voltage swing. With these two assump-

tions as to the normal Class C amplifier, the

expression for the plate load impedance can

be greatly simplified to the following approxi-

mate but useful expression:

Rd. c.

RL .v

2

which means simply that the resistance pre-

sented by the tank circuit to the Class C tube

is approximately equal to one -half the d -c load

resistance which the Class C stage presents

to the power supply (and also to the modulator

in case high -level modulation of the stage is

to be used).

Combining the above simplified expression

for the r -f impedance presented by the tank to

the tube, with the expression for tank Q given

in a previous paragraph we have the following

expression which relates the reactance of the

tank capacitor or coil to the d -c input to the

Class C stage: , Rd.c.

XC = XL

2Q

The above expression is the basis of the

usual charts giving tank capacitance for the

various bands in terms of the d -c plate voltage

and current to the Class C stage, including

the charts of figure 23, figure 24 and figure 25.

Harmonic Rodio- The problem of harmonic

tion vs. Q radiation from transmitters

has long been present, but

it has become critical only relatively recently

along with the extensive occupation of the

v -h -f range. Television signals are particularly

susceptible to interference from other signals

falling within the pass band of the receiver,

so that the TVI problem has received the major

emphasis of all the services in the v -h -f range

which are susceptible to interference from

harmonics of signals in the h -f or lower v -h -f

range.

w i

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20000

15

10

\11111III1N1 Q=12

111111!\IIIIIII

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1 11 . IIII

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1110M1121111101111111111

\11'I \\111Ii

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3 10 20 30 100 200 500 1000

TOTAL CAPACITANCE ACROSS LC C RCUIT (CO 2000

NEUTRALIZING

COIL

-e

RFC

re

O

Figure 23

PLATE -TANK CIRCUIT ARRANGEMENTS

Shown above in the case of each of the tank circuit types is the recommended tank circuit ca-

pacitance. (A) is a conventional tetrode amplifier, (B) is a coil -neutralized triode amplifier,

(C) is a grounded -grid triode amplifier, (D) is a grid -neutralized triode amplifier.

www.americanradiohistory.com

HANDBOOK Tank Circuits 261

10

1.3

o >

á

II

¢

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1MINE111011; I111111MII1,vIM11111MMnII

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2 3 5 7 10 lo 30 50 100 lao 500 1000

CORRECT VALUES OF TANK CIRCUIT CAPACITANCE (C FOR

OPERATING Q OF 12 WITH SINGLE-ENDED SPLIT TANK COILS

Figure 24

PLATE -TANK CIRCUIT ARRANGEMENTS

Shown above for each of the tank circuit types is the recommended tank circuit capacitance of

the operating frequency for an operating Q of 12. (A) is a split -stator tank, each section of which

is twice the capacity value read on the graph. (8) is circuit using tapped coil for phase reversal.

et

Inspection of figure 22 will show quickly

that the tank circuit of a Class C amplifier

should have an operating Q of 12 or greater

to afford satisfactory rejection of second har-

monic energy. The curve begins to straighten

out above a Q of about 15, so that a consider-

able increase in Q must be made before an ap-

preciable reduction in second -harmonic energy

is obtained. Above a circuit Q of about 10 any

increase will not afford appreciable reduction

in the third -harmonic energy, so that additional

harmonic filtering circuits external to the am-

plifier proper must be used if increased atten-

uation of higher order harmonics is desired.

The curves also show that push -pull amplifiers

may be operated at Q values of 6 or so, since

the second harmonic is cancelled to a large

extent if there is no unbalanced coupling be-

tween the output tank circuit and the antenna

system.

Capacity Charts for Figures 23, 24 and 25 il-

Correct Tank Q lustrate the correct value

of tank capacity for vari-

ous circuit configurations. A Q value of 12

has been chosen as optimum for single ended

circuits, and a value of 6 has been chosen for

push -pull circuits. Figure 23 is used when a

single ended stage is employed, and the ca-

pacitance values given are for the total ca-

pacitance across the tank coil. This value in-

cludes the tube interelectrode capacitance

(plate to ground), coil distributed capacitance,

wiring capacities, and the value of any low-

inductance plate -to- ground by -pass capacitor

as used for reducing harmonic generation, in

addition to the actual "in -use" capacitance

of the plate tuning capacitor. Total circuit

stray capacitance may vary from perhaps 5

micromicrofarads for a v -h -f stage to 30 micro -

microfarads for a medium power tetrode h -f

stage.

When a split plate tank coil is employed in

the stage in question, the graph of figure 24

should be used. The capacity read from the

graph is the total capacity across the tank

coil. If the split- stator tuning capacitor is

used, each section of the capacitor should

have a value of capacity equal to twice the

value indicated by the graph. As in the case

of figure 23, the values of capacity read on

the graph of figure 24 include all residual cir-

cuit capacities.

For push -pull operation, the correct values

of tank circuit capacity may be determined

with the aid of figure 25. The capacity values

obtained from figure 25 are the effective values

across the tank circuit, and if a split- stator

tuning capacitor is used, each section of the

capacitor should have a value of capacity e-

qual to twice the value indicated by the graph.

As in the case of figures 23 and 24, the values

of capacity read on the graph of figure 25 in-

clude all residual circuit capacities.

The tank circuit operates in the same man-

ner whether the tube feeding it is a pentode,

beam tetrode, neutralized triode, grounded -

grid triode, whether it is single ended or push-

www.americanradiohistory.com

262 Generation of R -F Energy THE RADIO

uz á 0:

>ú w

« J

O. a.

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20000

16000

8000

6000

Q°6

1000 2 3 ! 7 10 20 30 30 00 200 300 1000

CORRECT VALUES OF TANK CIRCUIT CAPACITANCE C ) FOR

OPERAT NG Q OF 6 WITH PUSH -PULL TANK CIRCUITS

0

Figure 25

PLATE -TANK CIRCUIT ARRANGEMENTS FOR PUSH -PULL STAGES

Shown above is recommended tank circuit capacity at operating frequency for a Q of 6. (A) is

split -stator tank, each section of which is twice the capacity value read on the graph. (B) is

circuit using topped coil for phase reversal.

pull, or whether it is shunt fed or series fed.

The important thing in establishing the oper-

ating Q of the tank circuit is the ratio of the

loaded resonant impedance across its termi-

nals to the reactance of the L and the C which

make up the tank.

Due to the unknowns involved in determin-

ing circuit stray capacitances it is sometimes

more convenient to determine the value of L

required for the proper circuit Q (by the method

discussed earlier in this Section) and then to

vary the tuned circuit capacitance until reso-

nance is reached. This method is most fre-

quently used in obtaining proper circuit Q in

commercial transmitters.

The values of Rp for using the charts are

easily calculated by dividing the d -c plate sup-

ply voltage by the total d -c plate current (ex-

pressed in amperes). Correct values of total

tuning capacitance are shown in the chart for

the different amateur bands. The shunt stray

capacitance can be estimated closely enough

for all practical purposes. The coil inductance

should then be chosen which will produce

resonance at the desired frequency with the

total calculated tuning capacitance.

Effect of Load- The Q of a circuit depends

ing on Q upon the resistance in series

with the capacitance and in-

ductance. This series resistance is very low

for a low -loss coil not loaded by an antenna

circuit. The value of Q may be from 100 to 600

under these conditions. Coupling an antenna

circuit has the effect of increasing the series

resistance, though in this case the power is

consumed as useful radiation by the antenna.

Mathematically, the antenna increases the

value of R in the expression Q = oiL /R where

L is the coil inductance in microhenrys and

is the term 2nf, f being in megacycles.

The coupling from the final tank circuit to

the antenna or antenna transmission line can

be varied to obtain values of Q from perhaps

3 at maximum coupling to a value of Q equal

to the unloaded Q of the circuit at zero an-

tenna coupling. This value of unloaded Q can

be as high as 500 or 600, as mentioned in the

preceding paragraph. However, the value of

Q = 12 will not be obtained at values of nor-

mal d -c plate current in the Class C amplifier

stage unless the C -to -L ratio in the tank cir-

cuit is correct for that frequency of operation.

Tuning Capacitor To determine the required

Air Gap tuning capacitor air gap for

a particular amplifier cir-

cuit it is first necessary to estimate the peak

r -f voltage which will appear between the

plates of the tuning capacitor. Then, using

figure 26, it is possible to estimate the plate

spacing which will be required.

The instantaneous r -f voltage in the plate

circuit of a Class C amplifier tube varies from

nearly zero to nearly twice the d -c plate volt-

age. If the d -c voltage is being 100 per cent

modulated by an audio voltage, the r -f peaks

will reach nearly four times the d -c voltage.

www.americanradiohistory.com

HANDBOOK L and Pi Networks 263

FIGURE 26

USUAL BREAKDOWN

COMMON PLATE

Air -gap in

inches

RATINGS OF

SPACINGS

Peak voltage

breakdown

.030 1,000

.050 2,000

.070 3,000

.100 4,000

.125 4,500

.150 5,200

.170 6,000

.200 7,500

.250 9,000

.350 11,000

.500 15,000

.700 20,000

Recommended air -gap for use when no d -c

voltage appears across plate tank condenser

(when plate circuit is shunt fed, or when the

plate tank condenser is insulated from

ground).

D.C. PLATE

VOLTAGE C.W. PLATE

MOD.

400 .030 .050

600 .050 .070

750 .050 .084

1000 .070 .100

1250 .070 .144

1500 .078 .200

2000 .100 .250

2500 .175 .375

3000 .200 .500

3500 .250 .600

Spacings should be multiplied by 1.5 for

some safety factor when d -c voltage appears

across plate tank condenser.

These rules apply to a loaded amplifier or

buffer stage. If either is operated without an

r -f load, the peak voltages will be greater and

can exceed the d -c plate supply voltage. For

this reason no amplifier should be operated

without load when anywhere near normal d -c

plate voltage is applied.

If a plate blocking condenser is used, it

must be rated to withstand the d -c plate volt-

age plus any audio voltage. This capacitor

should be rated at a d -c working voltage of at

least twice the d -c plate supply in a plate mod-

ulated amplifier, and at least equal to the d -c

supply in any other type of r -f amplifier.

13 -10 L and Pi Matching

Networks

The L and pi networks often can be put to

advantageous use in accomplishing an imped-

ance match between two differing impedances.

Common applications are the matching between

a transmission line and an antenna, or between

the plate circuit of a single -ended amplifier

stage and an antenna transmission line. Such

networks may be used to accomplish a match

RF

+e = FOR OPERATING CIRCUIT

Q OF 15

RP RA(Q2+1)(txACT)

RP = Q2 RA (APPROX.)

Q=xs__X.. XC

-BL-B.e

RA RA XL

XL =Xc

RP= APPROX. LATE VOLTAG4

'PLATE CURRENT

RP= 225 RA

XC 19

XL= *

Figure 27

THE L NETWORK IMPEDANCE

TRANSFORMER

The L network is useful with a moderate

operating Q for high values of impedance

transformation, and it may be used for appli-

cations other than in the plate circuit of a

tube with relatively low values of operating

Q for moderate impedance transformations.

Exact and approximate design equations ore

given.

between the plate tank circuit of an amplifier

and a transmission line, or they may be used

to match directly from the plate circuit of an

amplifier to the line without the requirement

for a tank circuit -provided the network is de-

signed in such a manner that it has sufficient

operating Q for accomplishing harmonic atten-

uation.

The L Matching The L network is of limited

Network utility in impedance match-

ing since its ratio of imped-

ance transformation is fixed at a value equal

to (Q2 +1). The operating Q may be relatively

low (perhaps 3 to 6) in a matching network be-

tween the plate tank circuit of an amplifier

and a transmission line; hence impedance

transformation ratios of 10 to 1 and even lower

may be attained. But when the network also

acts as the plate tank circuit of the amplifier

stage, as in figure 27, the operating Q should

be at least 12 and preferably 15. An operating

Q of 15 represents an impedance transforma-

tion of 225; this value normally will be too

high even for transforming from the 2000 to

10,000 ohm plate impedance of a Class C am-

plifier stage down to a 50 -ohm transmission

line. However, the L network is interesting since

it forms the basis of design for the pi network.

Inspection of figure 27 will show that the L

network in reality must be considered as a

parallel- resonant tank circuit in which RA

represents the coupled -in load resistance;

only in this case the load resistance is di-

rectly coupled into the tank circuit rather than

being inductively coupled as in the conven-

www.americanradiohistory.com

264 Generation of R -F Energy THE RADIO

Roc = Ebb

I

Rp Roc.

z

Rp

XCn = Q

XL, Á

Rp

XCZ -RA RA(Q2+1)-Rp

XLZ' RAZ XCi

RI12tXG22

ALTOT. XL1+XL2

Figure 28

THE PI NETWORK

The pi network is valuable for use as an im-

pedance transformer over a wide ratio of

transformation values. The operating Q should

be at least 12 and preferably 15 to 20 when

the circuit is to be used in the plate circuit

of a Class C amplifier. Design equations

are given above. The inductor Ltot repre-

sents a single inductance, usually variable,

with a value equal to the sum of Lt and L2.

tional arrangement where the load circuit is

coupled to the tank circuit by means of a link.

When RA is shorted, L and C comprise a con-

ventional parallel- resonant tank circuit, since

for proper operation L and C must be resonant

in order for the network to present a resistive

load to the Class C amplifier.

The Pi Network The pi impedance matching

network, illustrated in figure

28, is much more general in its application

than the L network since it offers greater har-

monic attenuation, and since it can be used

to match a relatively wide range of impedances

while still maintaining any desired operating

Q. The values of C, and L, in the pi network

of figure 28 can be thought of as having the

same values of the L network in figure 27 for

the same operating Q, but what is more impor-

tant from the comparison standpoint these val-

ues will be the same as in a conventional tank

circuit.

The value of the capacitance may be deter-

mined by calculation, with the operating Q and

the load impedance which should be reflected

to the plate of the Class C amplifier as the

two knowns -or the actual values of the ca-

pacitance may be obtained for an operating Q

of 12 by reference to figures 23, 24 and 25.

The inductive arm in the pi network can be

thought of as consisting of two inductances

in series, as illustrated in figure 28. The first

portion of this inductance, L is that value of

inductance which would resonate with C, at

the operating frequency -the same as in a con-

ventional tank circuit. However, the actual

value of inductance in this arm of the pi net-

work, L10, will be greater than L, for normal

values of impedance transformation. For high

transformation ratios Lot will be only slightly

greater than Li; for a transformation ratio of

1.0, L10t will be twice as great as L. The

amount of inductance which must be added to

L, to restore resonance and maintain circuit

Q is obtained through use of the expression

for X12 in figure 28.

The peak voltage rating of the main tuning

capacitor C, should be the normal value for a

Class C amplifier operating at the plate volt-

age to be employed. The inductor L101 may be

a plug -in coil which is changed for each band

of operation, or some sort of variable inductor

may be used. A continuously variable slider -

type of variable inductor, such as used in cer-

tain items of surplus military equipment, may

be used to good advantage if available, or a

tapped inductor such as used in the ART -13

may be employed. However, to maintain good

circuit Q on the higher frequencies when a

variable or tapped coil is used on the lower

frequencies, the tapped or variable coil should

be removed from the circuit and replaced by

a smaller coil which has been especially de-

signed for the higher frequency ranges.

The peak voltage rating of the output or

loading capacitor, C2i is determined by the

power level and the impedance to be fed. If a

50 -ohm coaxial line is to be fed from the pi

network, receiving -type capacitors will be

satisfactory even up to the power level of a

plate -modulated kilowatt amplifier. In any

event, the peak voltage which will be im-

pressed across the output capacitor is ex-

pressed by: Epk2 = 2 R. Wo, where Epk is the

peak voltage across the capacitor, R. is the

value of resistive load which the network is

feeding, and W. is the maximum value of the

average power output of the stage. The har-

monic attenuation of the pi network is quite

good, although an external low -pass filter will

be required to obtain harmonic attenuation

value upward of 100 db such as normally re-

quired. The attenuation to second harmonic

energy will be approximately 40 db for an oper-

ating Q of 15 for the pi network; the value

increases to about 45 db for a 1:1 transforma-

tion and falls to about 38 db for an impedance

step -down of 80:1, assuming that the oper-

ating Q is maintained at 15.

www.americanradiohistory.com

HANDBOOK Grid Bias 265

RFC CB

-B

PLATE LOAD (OHMS) E.

Zx[B

COAX

OUTPUT

WHERE ES IS PLATE VOLTAGE

AND I B IS PLATE CURRENT

IN 4MPE II C S

Ce- .000252/F MICA CAPACITOR RATED AT TWICE THE D.C.

PLATE VOLTAGE

RFC i -Ne 28 ENAMELED. CLOSE -WOUND ON A CERAMIC INSULATOR

I'OIA., 4 -LONG OR NATIONAL R-1754

RFC2- 2,1 MN, NATIONAL R -100

Est mated Plate

load (ohms) 1,000 1,500 2,000 2,500 3,000 3,500 4,000 4,500 5,000 6,000 NOTES

C, n µµf, 3.5 Mc 520 360 280 210 180 155 135 120 110 90 The actuel capacitance setting

7 260 180 140 105 90 76 68 60 56 45 for C, equals the value in this

14 130 90 70 52 45 38 34 30 28 23 table minus the published tube

21 85 60 47 35 31 25 23 20 19 15 output capacitance. Air gap

28 65 45 35 26 33 19 17 15 14 11 approx. 10 mils 100 v E,,.

I. in ph, 3.5 Mc 4.5 6.5 8.5 10.5 12.5 14 15.5 18 20 25 Inductance values are for a

7 2.2 3.2 4.2 5.2 6.2 7 7.8 9 10 12.5 50 -ohm load. For a 70 -ohm

14 1.1 1.6 2.1 2.6 3.1 3.5 3.9 4.5 5 6.2 load, values ore approx. 3%

21 0.73 1.08 1.38 1.7 2.05 2.3 2.6 3 3.3 4.1 higher.

28 0.55 0.8 1.05 1.28 1.55 1.7 1.95 2.25 2.5 3.1

C1 in puf, 3.5 Mc 2,400 2,100 1,800 1,550 1,400 1,250 1,100 1,000 900 700 For 50 -ohm transmission line.

7 1,200 1,060 900 760 700 630 560 500 460 350 Air gap for Cr is approx. 1

14 600 530 450 380 350 320 280 250 230 175 mil 100 v E,.

21 400 350 300 250 230 210 185 165 155 120

28 300 265 225 190 175 160 140 125 115 90

Cr in gal, 3.5 Mc 1.800 1,500 1,300 1,100 1,000 900 800 720 640 500 For 70 -ohm transmission line.

7 900 750 650 560 500 450 400 360 320 250

14 450 370 320 280 250 220 200 180 160 125

21 300 250 215 190 170 145 130 120 110 85

28 225 185 160 140 125 110 100 90 80 65

Values given are approximations. All components shown in Table 1 are for a Q of 12. for other values of Q, use

Q. C. Q. L,.

-- and - _ When the estimated plate load is higher than 5,000 ohms, it is recommended that the

Q,, C,, Q L.

components be selected for a circuit Q between 20 and 30.

Table 1 Components for Pi- Coupled Final Amplifiers

Component Chart To simplify design pro -

for Pi- Networks cedure, a pi- network chart,

compiled by M. Seybold,

W2RYI (reproduced by courtesy of R.C.A. Tube

Division, Harrison, N.J.) is shown in table 1.

This chart summarizes the calculations of fig-

ure 28 for various values of plate load.

13 -11 Grid Bias

Radio -frequency amplifiers require some

form of grid bias for proper operation. Prac-

tically all r -f amplifiers operate in such a man-

ner that plate current flows in the forni of

short pulses which have a duration of only a

fraction of an r -f cycle. To accomplish this

with a sinusoidal excitation voltage, the oper-

ating grid bias must be at least sufficient to

cut off the plate current. In very high efficien-

cy Class C amplifiers the operating bias may

be many times the cutoff value. Cutoff bias,

it will be recalled, is that value of grid volt-

age which will reduce the plate current to

zero at the plate voltage employed. The method

for calculating it has been indicated previous-

ly. This theoretical value of cutoff will not

reduce the plate current completely to zero,

due to the variable-ft tendency or "knee"

which is characteristic of all tubes as the

cutoff point is approached.

Class C Bias Amplitude modulated Class C

amplifiers should be operated

with the grid bias adjusted to a value greater

than twice cutoff at the operating plate volt-

www.americanradiohistory.com

266 Generation of R- F Energy THE RADIO

FOU DRIVER

Figure 29

GRID -LEAK BIAS

The grid leak on an amplifier or multiplier

stage may also be used as the shunt feed

impedance to the grid of the tube when o

high value of grid leak (greater than perhaps

20,000 ohms) is used. When a lower value of

grid leak is to be employed, an r -f choke

should be used between the grid of the tube

and the grid leak to reduce r -f losses in the

grid leak resistance.

age. This procedure will insure that the tube

is operating at a bias greater than cutoff when

the plate voltage is doubled on positive modu-

lation peaks. C -w telegraph and FM trans-

mitters can be operated with bias as low as

cutoff, if only limited excitation is available

and moderate plate efficiency is satisfactory.

In a c -w transmitter, the bias supply or re-

sistor should be adjusted to the point which

will allow normal grid current to flow for the

particular amount of grid driving r -f power

available. This form of adjustment will allow

more output from the under -excited r -f ampli-

fier than when higher bias is used with corre-

sponding lower values of grid current. In any

event, the operating bias should be set at as

low a value as will give satisfactory opera-

tion, since harmonic generation in a stage in-

creases rapidly as the bias is increased.

Grid -Leak Bias A resistor can be connected

in the grid circuit of a Class

C amplifier to provide grid -leak bias. This re-

sistor, R, in figure 29, is part of the d -c path

in the grid circuit.

The r -f excitation applied to the grid cir-

cuit of the tube causes a pulsating direct cur-

rent to flow through the bias supply lead, due

to the rectifying action of the grid, and any

current flowing through R, produces a voltage

drop across that resistor. The grid of the tube

is positive for a short duration of each r -f

cycle, and draws electrons from the filament

or cathode of the tube during that time. These

electrons complete the circuit through the d -c

grid return. The voltage drop across the re-

sistance in the grid return provides a nega-

tive bias for the grid.

Grid -leak bias automatically adjusts itself

over fairly wide variations of r -f excitation.

The value of grid -leak resistance should be

such that normal values of grid current will

flow at the maximum available amount of r -f

FROLIC/RIVER

Figure 30

COMBINATION GRID -LEAK AND

FIXED BIAS

Grid -leak bias often is used in conjunction

with a fixed minimum value of power supply

bias. This arrangement permits the operating

bias to be established by the excitation ener-

gy, but in the absence of excitation the elec-

trode currents to the tube will be held to safe

values by the fixed- minimum power supply

bias. If a relatively low value of grid leak

is to be used, an r -f choke should be con-

nected between the grid of the tube and the

grid leak as discussed in figure 29.

excitation. Grid -leak bias cannot be used for

grid -modulated or linear amplifiers in which

the average d -c grid current is constantly

varying with modulation.

Safety Bias Grid -leak bias alone provides no

protection against e x c e s s i v e

plate current in case of failure of the source

of r -f grid excitation. A C- battery or C -bias

supply can be connected in series with the

grid leak, as shown in figure 30. This fixed

protective bias will protect the tube in the

event of failure of grid excitation. "Zero- bias"

tubes do not require this bias source in addi-

tion to the grid leak, since their plate current

will drop to a safe value when the excitation

is removed.

Cathode Bias A resistor can be connected in

series with the cathode or cen-

ter- tapped filament lead of an amplifier to se-

cure automatic bias. The plate current flows

through this resistor, then back to the cathode

or filament, and the voltage drop across the

resistor can be applied to the grid circuit by

connecting the grid bias lead to the grounded

or power supply end of the resistor R, as shown

in figure 31.

The grounded (B- minus) end of the cathode

resistor is negative relative to the cathode

by an amount equal to the voltage drop across

the resistor. The value of resistance must be

so chosen that the sum of the desired grid

and plate current flowing through the resistor

will bias the tube for proper operation.

This type of bias is used more extensively

in audio -frequency than in radio -frequency am-

plifiers. The voltage drop across the resistor

www.americanradiohistory.com

HANDBOOK Protective Circuits 267

Figure 31

RF STAGE WITH CATHODE BIAS

Cathode bias sometimes is advantageous for

use it on r -f stage that operates with a rela-

tively small amount of r -f excitation.

PROM

DRIVER

Figure 32

R -F STAGE WITH BATTERY BIAS

Battery bias is seldom used, due to deteriora-

tion of the cells by the reverse grid current.

However, it may be used in certain special

applications, or the fixed bias voltage may

be supplied by a bias power supply.

must be subtracted from the total plate supply

voltage when calculating the power input to

the amplifier, and this loss of plate voltage

in an r -f amplifier may be excessive. A Class

A audio amplifier is biased only to approxi-

mately one -half cutoff, whereas an r -f amplifier

may be biased to twice cutoff, or more, and

thus the plate supply voltage loss may be a

large percentage of the total available voltage

when using low or medium It tubes.

Oftentimes just enough cathode bias is em-

ployed in an r -f amplifier to act as safety bias

to protect the tubes in case of excitation fail-

ure, with the rest of the bias coming from a

grid leak.

Separate Bias An external supply often is

Supply used for grid bias, as shown in

figure 32. Battery bias gives

very good voltage regulation and is satisfac-

tory for grid- modulated or linear amplifiers,

which operate at low grid current. In the case

of Class C amplifiers which operate with high

grid current, battery bias is not satisfactory.

This direct current has a charging effect on

the dry batteries; after a few months of service

the cells will become unstable, bloated, and

noisy.

A separate a -c operated power supply is

commonly used for grid bias. The bleeder re-

sistance across the output of the filter can be

made sufficiently low in value that the grid

current of the amplifier will not appreciably

change the amount of negative grid -bias volt-

age. Alternately, a voltage regulated grid -bias

supply can be used. This type of bias supply

is used in Class B audio and Class B r -f lin-

ear amplifier service where the voltage regu-

lation in the C -bias supply is important. For

a Class C amplifier, regulation is not so im-

portant, and an economical design of compo-

nents in the power supply, therefore, can be

utilized. In this case, the bias voltage must

be adjusted with normal grid current flowing,

as the grid current will raise the bias con-

siderably when it is flowing through the bias -

supply bleeder resistance.

13 -12 Protective Circuits for

Tetrode Transmitting Tubes

The tetrode transmitting tube requires three

operating voltages: grid bias, screen voltage,

and plate voltage. The current requirements of

these three operating voltages are somewhat

interdependent, and a change in potential of

one voltage will affect the current drain of the

tetrode in respect to the other two voltages.

In particular, if the grid excitation voltage is

interrupted as by keying action, or if the plate

supply is momentarily interrupted, the resulting

voltage or current surges in the screen circuit

are apt to permanently damage the tube.

The Series Screen A simple method of obtain -

Supply ing screen voltage is by

means of a dropping resis-

tor from the high voltage plate supply, as shown

in figure 33. Since the current drawn by the

screen is a function of the exciting voltage

applied to the tetrode, the screen voltage will

rise to equal the plate voltage under condi-

tions of no exciting voltage. If the control grid

is overdriven, on the other hand, the screen

current may become excessive. In either case,

damage to the screen and its associated com-

ponents may result. In addition, fluctuations

in the plate loading of the tetrode stage will

cause changes in the screen current of the

tube. This will result in screen voltage fluc-

tuations due to the inherently poor voltage

regulation of the screen series dropping resis-

tor. These effects become dangerous to tube

life if the plate voltage is greater than the

screen voltage by a factor of 2 or so.

www.americanradiohistory.com

268 Generation of R -F Energy THE RADIO

Figure 33

DROPPING- RESISTOR SCREEN SUPPLY

The Clomp Tube A clamp tube may be added

to the series screen supply,

as shown in figure 34. The clamp tube is nor-

mally cut off by virtue of the d -c grid bias drop

developed across the grid resistor of the tet-

rode tube. When excitation is removed from

the tetrode, no bias appears across the grid

resistor, and the clamp tube conducts heavily,

dropping the screen voltage to a safe value.

When excitation is applied to the tetrode the

clamp tube is inoperative, and fluctuations of

the plate loading of the tetrode tube could

allow the screen voltage to rise to a damaging

value. Because of this factor, the clamp tube

does not offer complete protection to the tet-

rode.

The Separate A low voltage screen supply

Screen Supply may be used instead of the

series screen dropping resis-

tor. This will protect the screen circuit from

excessive voltages when the other tetrode

operating parameters shift. However, the screen

can be easily damaged if plate or bias volt-

age is removed from the tetrode, as the screen

current will reach high values and the screen

dissipation will be exceeded. If the screen

supply is capable of providing slightly more

screen voltage than the tetrode requires for

proper operation, a series wattage -limiting re-

sistor may be added to the circuit as shown

in figure 35. With this resistor in the circuit

it is possible to apply excitation to the tet-

rode tube with screen voltage present (but in

the absence of plate voltage) and still not dam-

age the screen of the tube. The value of the

resistor should be chosen so that the product

of the voltage applied to the screen of the

tetrode times the screen current never exceeds

the maximum rated screen dissipation of the

tube.

13 -13 Interstage Coupling

Energy is usually coupled from one circuit

of a transmitter into another either by capaci-

tive coupling, inductive coupling, or link cou-

RFC

NEGATIVE

OPERATING r CLAMP

8/AS CUTS TUBE

OFF CLAMP{

rUBE

4B

Figure 34

CLAMP -TUBE SCREEN SUPPLY

piing. The latter is a special form of induc-

tive coupling. The choice of a coupling method

depends upon the purpose for which it is to

be used.

Capacitive Capacitive coupling between an

Coupling amplifier or doubler circuit and a

preceding driver stage is shown

in figure 36. The coupling capacitor, C, iso-

lates the d -c plate supply from the next grid

and provides a low impedance path for the r -f

energy between the tube being driven and the

driver tube. This method of coupling is simple

and economical for low power. amplifier or ex-

citer stages, but has certain disadvantages,

particularly for high frequency stages. The

grid leads in an amplifier should be as short

as possible, but this is difficult to attain in

the physical arrangement of a high power am-

plifier with respect to a capacitively- coupled

driver stage.

Disadvantages of One significant disadvan-

Capacitive tage of capacitive coupling

Coupling is the difficulty of adjusting

the load on the driver stage.

Impedance adjustment can be accomplished

by tapping the coupling lead a part of the way

down on the plate coil of the tuned stage of

the driver circuit; but often when this is done

SERIES RESISTOR

LOW VOLTAGE

SCREEN SUPPLY +B

Figure 35

A PROTECTIVE WATTAGE -LIMITING RE-

SISTOR FOR USE WITH LOW- VOLTAGE

SCREEN SUPPLY

www.americanradiohistory.com

HANDBOOK Interstage Coupling 269

Figure 36

CAPACITIVE INTERSTAGE COUPLING

a parasitic oscillation will take place in the

stage being driven.

One main disadvantage of capacitive coupl-

ing lies in the fact that the grid -to- filament

capacitance of the driven tube is placed di-

rectly across the driver tuned circuit. This

condition sometimes makes the r -f amplifier

difficult to neutralize, and the increased mini-

mum circuit capacitance makes it difficult to

use a reasonable size coil in the v -h -f range.

Difficulties from this source can be partially

eliminated by using a center -tapped or split -

stator tank circuit in the plate of the driver

stage, and coupling capacitively to the oppo-

site end from the plate. This method places

the plate -to- filament capacitance of the driver

across one -half of the tank and the grid -to-

filament capacitance of the following stage

across the other half. This type of coupling is

shown in figure 37.

Capacitive coupling can be used to advan-

tage in reducing the total number of tuned cir-

cuits in a transmitter so as to conserve space

and cost. It also can be used to advantage be-

tween stages for driving beam tetrode or pen-

tode amplifier or doubler stages.

Inductive Inductive coupling (figure 38) re-

Coupling sults when two coils are electro-

magnetically coupled to one an-

other. The degree of coupling is controlled by

varying the mutual inductance of the two coils,

which is accomplished by changing the spac-

ing or the relationship between the axes of

the coils.

Figure 38

INDUCTIVE INTERSTAGE COUPLING

Figure 37

BALANCED CAPACITIVE COUPLING

Balanced capacitive coupling sometimes is

useful when it is desirable to use o relatively

large inductance in the interstage tank cir-

cuit, or where the exciting stage is neutral-

ized as shown above.

Inductive coupling is used extensively for

coupling r -f amplifiers in radio receivers. How-

ever, the mechanical problems involved in ad-

justing the degree of coupling limit the use-

fulness of direct inductive coupling in trans-

mitters. Either the primary or the secondary

or both coils may be tuned.

Unity Coupling If the grid tuning capacitor of

figure 38 is removed and the

coupling increased to the maximum practicable

value by interwinding the turns of the two coils,

the circuit insofar as r.f. is concerned acts

like that of figure 36, in which one tank serves

both as plate tank for the driver and grid tank

for the driven stage. The inter -wound grid

winding serves simply to isolate the d -c plate

voltage of the driver from the grid of the driven

stage, and to provide a return for d -c grid cur-

rent. This type of coupling, illustrated in fig-

ure 39, is commonly known as unity coupling.

Because of the high mutual inductance, both

primary and secondary are resonated by the

one tuning capacitor.

INTERWOUND

Figure 39

"UNITY" INDUCTIVE COUPLING

Due to the high value of coupling between

the two coils, one tuning capacitor tunes

both circuits. This arrangement often is usa

ful in coupling from a single -ended to a push -

pull stage.

www.americanradiohistory.com

270 Generation of R -F Energy THE RADIO

LINK COUPLING

AT ..COLD. ENDS.

UPPER ENDS "MOT"

Figure 40

INTERSTAGE COUPLING BY MEANS

OF A "LINK"

Link interstoge coupling is very commonly

used since the two stages may be separated

by a considerable distance, since the amount

of a coupling between the two stages may he

easily varied, and since the capacitances of

the two stages may be isolated to permit use

of larger inductances in the v -h -f range.

Link Coupling A special form of inductive

coupling which is widely em-

ployed in radio transmitter circuits is known

as link coupling. A low impedance r -f trans-

mission line couples the two tuned circuits

together. Each end of the line is terminated

in one or more turns of wire, or links, wound

around the coils which are being coupled to-

gether. These links should be coupled to each

tuned circuit at the point of zero r -f potential,

or nodal point. A ground connection to one

side of the link usually is used to reduce har-

monic coupling, or where capacitive coupling

between two circuits must be minimized. Co-

axial line is commonly used to transfer energy

between the two coupling links, although Twin -

Lead may be used where harmonic attenuation

is not so important.

Typical link coupled circuits are shown in

figures 40 and 41. Some of the advantages of

link coupling are the following:

(1) It eliminates coupling taps on tuned cir-

cuits.

(2) It permits the use of series power supply

connections in both tuned grid and tuned

plate circuits, and thereby eliminates the

need of shunt -feed r -f chokes.

(3) It allows considerable separation between

transmitter stages without appreciable

r -f losses or stray chassis currents.

(4) It reduces capacitive coupling and there-

by makes neutralization more easily at-

tainable in r -f amplifiers.

(5) It provides semi- automatic impedance

matching between plate and grid tuned

circuits, with the result that greater grid

drive can be obtained in comparison to

capacitive coupling.

(6) It effectively reduces the coupling of har-

monic energy.

LINK COUPLING

AT COLO CENTER

ENDS "HOT*

Figure 41

PUSH -PULL LINK COUPLING

The link -coupling line and links can be

made of no. 18 push -back wire for coupling

between low -power stages. For coupling be-

tween higher powered stages the 150 -ohm

Twin -Lead transmission line is quite effective

and has very low loss. Coaxial transmission is

most satisfactory between high powered am-

plifier stages, and should always be used

where harmonic attenuation is important.

13 -14 Radio- Frequency

Chokes

Radio -frequency chokes are connected in

circuits for the purpose of stopping the pas-

sage of r -f energy while still permitting a di-

rect current or audio -frequency current to pass.

They consist of inductances wound with a

large number of turns, either in the form of a

solenoid, a series of solenoids, a single uni-

versal pie winding, or a series of pie wind-

ings. These inductors are designed to have as

much inductance and as little distributed or

shunt capacitance as possible. The unavoid-

able small amount of distributed capacitance

resonates the inductance, and this frequency

normally should be much lower than the fre-

quency at which the transmitter or receiver

circuit is operating. R -f chokes for operation

on several bands must be designed carefully

so that the impedance of the choke will be ex-

tremely high (several hundred thousand ohms)

in each of the bands.

The direct current which flows through the

r -f choke largely determines the size of wire

to be used in the winding. The inductance of

r -f chokes for the v -h -f range is much less

than for chokes designed for broadcast and

ordinary short -wave operation. A very high

inductance r -f choke has more distributed ca-

pacitance than a smaller one, with the result

www.americanradiohistory.com

HANDBOOK Shunt and Series Feed 271

+5G +11V

PARALLEL PLATE FEED

+SG +Nv

SERIES PLATE FEED

Figure 42

ILLUSTRATING PARALLEL AND

SERIES PLATE FEED

Parallel plate feed is desirable from a safety

standpoint since the tank circuit is at ground

potential with respect to d.c. However, a

high- impedance r -f choke is required, and

the r -t choke must be able to withstand the

peak r -f voltage output of the tube. Series

plate feed eliminates the requirement for a

high -performance r -f choke, but requires the

use of a relatively large value of by-pass

capacitance at the bottom end of the tank

circuit, as contrasted to the moderate value

of coupling capacitance which may be used

at the top of the tank circuit for parallel

plate feed.

that it will actually offer less impedance at

very high frequencies.

Another consideration, just as important as

the amount of d.c. the winding will carry, is

the r -f voltage which may be placed across

the choke without its breaking down. This is

a function of insulation, turn spacing, frequen-

cy, number and spacing of pies and other fac-

tors. Some chokes which are designed to have a

high impedance over a very wide range of fre-

quency are, in effect, really two chokes: a

u -h -f choke in series with a high -frequency

choke. A choke of this type is polarized; that

is, it is important that the correct end of the

combination choke be connected to the "hot"

side of the circuit.

Shunt and Direct-current grid and plate

Series Feed connections are made either by

series or parallel leed systems.

Simplified forms of each are shown in figures

42 and 43.

Series feed can be defined as that in which

the d -c connection is made to the grid or plate

circuits at a point of very low r -f potential.

Shunt feed always is made to a point of high

r -f voltage and always requires a high imped-

ance r -f choke or a relatively high resistance

to prevent waste of r -f power.

-BIAS

PARALLEL BIAS FEED

-BIAS

SERIES BIAS FEED

Figure 43

ILLUSTRATING SERIES AND

PARALLEL BIAS FEED

13 -15 Parallel and

Push -Pull Tube Circuits

The comparative r -f power output from paral-

lel or push -pull operated amplifiers is the same

if proper impedance matching is accomplished,

if sufficient grid excitation is available in

both cases, and if the frequency of measure-

ment is considerably lower than the frequency

limit of the tubes.

Parallel Operating tubes in parallel has

Operation some advantages in transmitters

designed for operation below 10

NIc., particularly when tetrode or pentode tubes

are to be used. Only one neutralizing capacitor

is required for parallel operation of triode

tubes, as against two for push -pull. Above

about 10 etc., depending upon the tube type,

parallel tube operation is not ordinarily recom-

mended with triode tubes. However, parallel

operation of grounded -grid stages and stages

using low -C beam tetrodes often will give ex-

cellent results well into the v -h -f range.

Push -Pull The push -pull connection provides

Operation a well -balanced circuit insofar as

miscellaneous capacitances are

concerned; in addition, the circuit can be neu-

tralized more completely, especially in high -

frequency amplifiers. The L/C ratio in a push -

pull amplifier can be made higher than in a

plate- neutralized parallel -tube operated am-

plifier. Push -pull amplifiers, when perfectly

balanced, have less second -harmonic output

than parallel or single -tube amplifiers, but in

practice undesired capacitive coupling and

circuit unbalance more or less offset the theo-

retical harmonic -reducing advantages of push -

pull r -f circuits.

www.americanradiohistory.com

CHAPTER FOURTEEN

R -F Feedback

Comparatively high gain is required in sin-

gle sideband equipment because the signal is

usually generated at levels of one watt or less.

To get from this level to a kilowatt requires

about 30 db of gain. High gain tetrodes may

be used to obtain this increase with a minimum

number of stages and circuits. Each stage con-

tributes some distortion; therefore, it is good

practice to keep the number of stages to a

minimum. It is generally considered good prac-

tice to operate the low level amplifiers below

their maximum power capability in order to

confine most of the distortion to the last two

amplifier stages. R -f feedback can then be

utilized to reduce the distortion in the last

two stages. This type of feedback is no dif-

ferent from the common audio feedback used

in high fidelity sound systems. A sample of

the output waveform is applied to the ampli-

fier input to correct the distortion developed

in the amplifier. The same advantages can be

obtained at radio frequencies that are obtained

at audio frequencies when feedback is used.

14 -1 R -F Feedback

Circuits

R -f feedback circuits have been developed

by the Collins Radio Co. for use with linear

amplifiers. Tests with large receiving and small

transmitting tubes showed that amplifiers us-

ing these tubes without feedback developed

signal -to- distortion ratios no better than 30 db

or so. Tests were run employing cathode fol-

lower circuits, such as shown in figure 1A.

Lower distortion was achieved, but at the cost

of low gain per stage. Since the voltage gain

through the tube is less than unity, all gain

has to be achieved by voltage step -up in the

tank circuits. This gain is limited by the dis-

sipation of the tank coils, since the circuit

capacitance across the coils in a typical trans-

mitter is quite high. In addition, the tuning

of such a stage is sharp because of the high

Q circuits.

The cathode follower performance of the

tube can be retained by moving the r -f ground

Br

B

Bi>5

`J

Figure 1

SIMILAR CATHODE FOLLOWER CIRCUITS HAVING DIFFERENT R -F GROUND POINTS.

272

www.americanradiohistory.com

R -F Feedback Circuits 273

B*

Figure 2

SINGLE STAGE AMPLIFIER WITH

R -F FEEDBACK CIRCUIT

BIAS = B}

R F OUT

Figure 3

SINGLE STAGE FEEDBACK

AMPLIFIER WITH GROUND

RETURN POINT MODIFIED FOR

UNBALANCED INPUT AND

OUTPUT CONNECTIONS.

point of the circuit from the plate to the cath-

ode as shown in figure 1B. Both ends of the

input circuit are at high r -f potential so in-

ductive coupling to this type of amplifier is

necessary.

Inspection of figure 1B shows that by mov-

ing the top end of the input tank down on a

voltage divider tap across the plate tank cir-

cuit, the feedback can be reduced from 100 %,

as in the case of the cathode follower circuit,

down to any desired value. A typical feedback

circuit is illustrated in figure 2. This circuit

is more practical than those of figure 1, since

the losses in the input tank are greatly reduced.

A feedback level of 12 db may be achieved

as a good compromise between distortion and

stage gain. The voltage developed across C=

will be three times the grid- cathode voltage.

B

BIAS B4

Figure 4

R -F AMPLIFIER WITH FEEDBACK

AND IMPEDANCE MATCHING

OUTPUT NETWORK.

Tuning and loading are accomplished by Cr

and C,. C, and L, are tuned in unison to

establish the correct degree of feedback.

Inductive coupling is required for this cir-

cuit, as shown in the illustration.

The circuit of figure 3 eliminates the need

for inductive coupling by moving the r -f

ground to the point common to both tank

circuits. The advantages of direct coupling be-

tween stages far outweigh the disadvantages of

having the r -f feedback voltage appear on the

cathode of the amplifier tube.

In order to match the amplifier to a load,

the circuit of figure 4 may be used. The ratio

of XL, to XC, determines the degree of feed-

back, so it is necessary to tune them in unison

when the frequency of operation is changed.

Tuning and loading functions are accomplished

by varying C2 and G. L5 may also be varied to

adjust the loading.

Feedback Around o The maximum phase

Two -Stage Amplifier shift obtainable over

two simple tuned cir-

cuits does not exceed 180 degrees, and feed-

back around a two stage amplifier is possible.

The basic circuit of a two stage feedback

amplifier is shown in figure 5. This circuit

is a conventional two -stage tetrode amplifier

except that r -f is fed back from the plate

circuit of the PA tube to the cathode of the

driver tube. This will reduce the distortion

E 1

Figure 5

BASIC CIRCUIT OF TWO -STAGE AMPLIFIER WITH R -F FEEDBACK

Feedback voltage is obtained from a voltage divider across the output circuit and

applied directly to the cathode of the first tube. The input tank circuit is thus

outside the feedback loop.

www.americanradiohistory.com

274 R -F Feedback THE RADIO

of both tubes as effectively as using individual

feedback loops around each stage, yet will

allow a higher level of overall gain. With

only two tuned circuits in the feedback loop,

it is possible to use 12 to 15 db of feedback

and still leave a wide margin for stability. It

is possible to reduce the distortion by nearly

as many db as are used in feedback. This cir-

cuit has two advantages that are lacking in the

single stage feedback amplifier. First, the fila-

ment of the output stage can now be operated

at r -f ground potential. Second, any conven-

tional pi output network may be used.

R -f feedback will correct several types of

distortion. It will help correct distortion caused

by poor power supply regulation, too low grid

bias, and limiting on peaks when the plate

voltage swing becomes too high.

Neutralization The purpose of neutraliza-

ond R -F Feedback tion of an r -f amplifier

stage is to balance out ef-

fects of the grid -plate capacitance coupling in

the amplifier. In a conventional amplifier us-

ing a tetrode tube, the effective input capacity

is given by:

Input Capacitance = Cis + Cy. (1 + A cos e )

where: Ci,, = tube input capacitance

C.. = grid -plate capacitance

A = voltage amplification from grid

to plate

e = phase angle of load

In a typical unneutralized tetrode amplifier

having a stage gain of 33, the input capaci-

tance of the tube with the plate circuit in

resonance is increased by 8 µµfd. due to the

unneutralized grid -plate capacitance. This is

unimportant in amplifiers where the gain (A )

remains constant but if the tube gain varies,

serious detuning and r -f phase shift may result.

A grid or screen modulated r -f amplifier is an

example of the case where the stage gain var-

ies from a maximum down to zero. The gain

of a tetrode r -f amplifier operating below plate

current saturation varies with loading so that

if it drives a following stage into grid current

the loading increases and the gain falls off.

The input of the grid circuit is also affected

by the grid -plate capacitance, as shown in this

equation:

Input Resistance - 27rf X C.N ( Asine )

This resistance is in shunt with the grid

current loading, grid tank circuit losses, and

driving source impedance. When the plate cir-

cuit is inductive there is energy transferred

from the plate to the grid circuit (positive

feedback ) which will introduce negative resist-

ance in the grid circuit. When this shunt

negative resistance across the grid circuit is

lower than the equivalent positive resistance

of the grid loading, circuit losses, and driving

source impedance, the amplifier will oscillate.

When the plate circuit is in resonance

( phase angle equal to zero) the input resist-

ance due to the grid -plate capacitance becomes

infinite. As the plate circuit is tuned to the

capacitive side of resonance, the input resist-

ance becomes positive and power is actually

transferred from the grid to the plate circuit.

This is the reason that the grid current in an

unneutralized tetrode r -f amplifier varies from

a low value with the plate circuit tuned on the

low frequency side of resonance to a high value

on the high frequency side of resonance The

grid current is proportional to the r -f voltage

on the grid which is varying under these con-

ditions. In a tetrode class All amplifier, the

effect of grid -plate feedback can be observed

by placing a r -f voltmeter across the grid cir-

cuit and observing the voltage change as the

plate circuit is tuned through resonance.

If the amplifier is over -neutralized, the ef-

fects reverse so that with the plate circuit

tuned to the low frequency side of resonance

the grid voltage is high, and on the high fre-

quency side of resonance, it is low.

Amplifier A useful "rule of

Neutralization Check thumb" method of

checking neutraliza-

tion of an amplifier stage (assuming that it

is nearly correct to start with) is to tune both

grid and plate circuits to resonance. Then, ob-

serving the r -f grid current, tune the plate cir-

cuit to the high frequency side of resonance.

If the grid current rises, more neutralization

capacitance is required. Conversely, if the grid

current decreases, less capacitance is needed.

This indication is very sensitive in a neutral-

ized triode amplifier, and correct neutraliza-

tion exists when the grid current peaks at the

point of plate current dip. In tetrode power

amplifiers this indication is less pronounced.

Sometimes in a supposedly neutralized tetrode

amplifier, there is practically no change in

grid voltage as the plate circuit is tuned

through resonance, and in some amplifiers it

is unchanged on one side of resonance and

drops slightly on the other side. Another ob-

servation sometimes made is a small dip in

the center of a broad peak of grid current.

These various effects are probably caused by

www.americanradiohistory.com

HANDBOOK R -F Feedback Circuits 275

R -F nur

Figure 6

SINGLE STAGE R -F AMPLIFIER

WITH FEEDBACK RATIO OF

C C to C C - DETERMINES

STAGE NEUTRALIZATION

coupling from the plate to the grid circuit

through other paths which are not balanced

out by the particular neutralizing circuit used.

Feedback and Figure 6 shows an r -f am-

Neutralization plifier with negative feed -

of a One -Stage back. The voltage developed

R -F Amplifier across G due to the voltage

divider action of G and C,

is introduced in series with the voltage devel-

oped across the grid tank circuit and is in

phase- opposition to it. The feedback can be

made any value from zero to 100% by proper-

ly choosing the values of C:. and G.

For reasons stated previously, it is necessary

to neutralize this amplifier, and the relation-

ship for neutralization is:

G == GP

G G.

It is often necessary to add capacitance from

plate to grid to satisfy this relationship

Figure 7 is identical to figure 6 except that

it is redrawn to show the feedback inherent in

this neutralization circuit more clearly. G and

C replace G and C., and the main plate tank

tuning capacitance is G. The circuit of figure

7 presents a problem in coupling to the grid

circuit. Inductive coupling is ideal, but the

extra tank circuits complicate the tuning of a

transmitter which uses several cascaded am-

plifiers with feedback around each one. The

grid could be coupled to a high source imped-

ance such as a tetrode plate, but the driver

then cannot use feedback because this would

cause the source impedance to be low. A pos-

sible solution is to move the circuit ground

point from the cathode to the bottom end of

the grid tank circuit. The feedback voltage then

appears between the cathode and ground

( figure 8 ) . The input can be capacitively

coupled, and the plate of the amplifier can

be capacitively coupled to the next stage. Also,

cathode type transmitting tubes are available

that allow the heater to remain at ground po-

Figure 7

NEUTRALIZED AMPLIFIER AND

INHERENT FEEDBACK CIRCUIT.

Neutralization is achieved by varying

the capacity of Cn.

tential when r -f is impressed upon the cathode.

The output voltage available with capacity

coupling, of course, is less than the plate -

cathode r -f voltage developed by the amount

of feedback voltage across G.

14 -2 Feedback and

Neutralization of a

Two -Stage R -F Amplifier

Feedback around two r -f stages has the ad-

vantage that more of the rube gain can be

realized and nearly as much distortion reduc-

tion can be obtained using 12 db around two

stages as is realized using 12 db around each

of two stages separately. Figure 9 shows a

basic circuit of a two stage feedback ampli-

fier. Inductive output coupling is used, al-

though a pi- network configuration will also

work well. The small feedback voltage required

is obtained from the voltage divider C. - G

and is applied to the cathode of the driver

tube. C. is only a few gcfd., so this feedback

voltage divider may be left fixed for a wide

frequency range. If the combined tube gain is

160, and 12 db of feedback is desired, the ratio

of G to C. is about 40 to 1. This ratio in

practice may be 400 µµtd. to 2.5 µµfd., for

example.

A complication is introduced into this sim-

plified circuit by the cathode -grid capacitance

R-F.N -i

R F our

Figure 8

UNBALANCED INPUT AND OUTPUT

CIRCUITS FOR SINGLE -STAGE

R -F AMPLIFIER WITH FEEDBACK

www.americanradiohistory.com

276 R -F Feedback THE RADIO

Figure 9

TWO -STAGE AMPLIFIER WITH FEEDBACK.

Included is a capacitor ,C) for neutralizing the cathode -grid capacity of the first tube. V. is neutralized

by capacitor C , and V; is neutralized by the correct ratio of C C-.

of the first tube which causes an undersired

coupling to the input grid circuit. It is neces-

sary to neutralize out this capacitance coupling,

as illustrated in figure 9. The relationship for

neutralization is:

G Cgt

G G

The input circuit may be made unbalanced

by making C. five times the capacity of G.

This will tend to reduce the voltage across

the coil and to minimize the power dissipated

by the coil. For proper balance in this case,

G must be five times the grid -filament capaci-

tance of the tube.

Except for tubes having extremely small

grid -plate capacitance, it is still necessary to

properly neutralize both tubes. If the ratio of

G to G is chosen to be equal to the ratio of

the grid -plate capacitance to the grid -filament

capacitance in the second tube (Vg), this tube

will be neutralized. Tubes such as a 4X -150A

have very low grid -plate capacitance and prob-

ably will not need to be neutralized when used

in the first (V.) stage. If neutralization is

necessary, capacitor G is added for this pur-

pose and the proper value is given by the

following relationship:

CRO _ Cet G

G G C.

If neither tube requires neutralization, the

bottom end of the interstage tank circuit may

be returned to r -f ground. The screen and

suppressor of the first tube should then be

grounded to keep the tank output capaci-

tance directly across this interstage circuit and

to avoid common coupling between the feed-

back on the cathode and the interstage circuit.

A slight amount of degeneration occurs in the

first stage since the tube also acts as a grounded

grid amplifier with the screen as the grounded

grid. The p. of the screen is much lower than

that of the control grid so that this effect may

be unnoticed and would only require slightly

more feedback from the output stage to over-

come.

Tests For

Neutralization Neutralizing the circuit of

figure 9 balances out cou-

pling between the input

tank circuit and the output tank circuit, but it

does not remove all coupling from the plate

circuit to the grid -cathode tube input. This

latter coupling is degenerative, so applying a

signal to the plate circuit will cause a signal

to appear between grid and cathode, even

though the stage is neutralized. A bench test

for neutralization is to apply a signal to the

plate of the tube and detect the presence of a

signal in the grid coil by inductive coupling

to it. No signal will be present when the stage

is neutralized. Of course, a signal could be in-

ductively coupled to the input and neutraliza-

tion accomplished by adjusting one branch of

the neutralizing circuit bridge (G for ex-

ample) for minimum signal on the plate cir-

cuit.

Neutralizing the cathode -grid capacitance of

the first stage of figure 9 may be accomplished

by applying a signal to the cathode of the tube

and adjusting the bridge balance for minimum

signal on a detector inductively coupled to the

input coil.

Tuning o Two -Stage Tuning the two -stage

Feedback Amplifier feedback amplifier of

figure 9 is accom-

plished in an unconventional way because the

output circuit cannot be tuned for maximum

output signal. This is because the output cir-

cuit must be tuned so the feedback voltage

applied to the cathode is in -phase with the

input signal applied to the first grid. When

the feedback voltage is not in- phase, the result-

ant grid- cathode voltage increases as shown

in figure 10. When the output circuit is

properly tuned, the resultant grid -cathode volt-

age on the first tube will be at a minimum, and

the voltage on the interstage tuned circuit will

also be at a minimum.

www.americanradiohistory.com

HANDBOOK Neutralization 277

VOLTAGE -

INPUT GRID

TO

GROUND

1 VOLTAGE - GRID TO CATHODE

VOLTAGE - CATHODE TO GROUND

(PEEDBACM)

(A.

Figure 10

VECTOR RELATIONSHIP OF

FEEDBACK VOLTAGE

A Output Circuit Properly Tuned

B Output Circuit Mis -Tuned

The two -stage amplifier may be tuned by

placing a r -f voltmeter across the interstage

tank circuit ( "hot" side to ground) and tuning

the input and interstage circuits for maximum

meter reading, and tuning the output circuit

for minimum meter reading. If the second tube

is driven into the grid current region, the grid

current meter may be used in place of the r -f

voltmeter. On high powered stages where oper-

ation is well into the Class AB region, the

plate current dip of the output tube indicates

correct output circuit tuning, as in the usual

amplifier.

Parasitic Oscillations in Quite often low fre-

the Feedback Amplifier q u e n c y parasitics

may be found in

the interstage circuit of the two -stage feedback

amplifier. Oscillation occurs in the first stage

due to low frequency feedback in the cathode

circuit. R -f chokes, coupling capacitors, and

bypass capacitors provide the low frequency

tank circuits. When the feedback and second

stage neutralizing circuits are combined, it is

necessary to use the configuration of figure 11.

This circuit has the advantage that only one

capacitor (G) is required from the plate of

the output tube, thus keeping the added ca-

pacitance across the output tank at a minimum.

+ BAS

Figure 11

INTERSTAGE CIRCUIT COMBINING

NEUTRALIZATION AND

FEEDBACK NETWORKS.

Figure 12

INTERSTAGE CIRCUIT WITH

SEPARATE NEUTRALIZING

AND FEEDBACK CIRCUITS.

It is convenient, however, to separate these cir-

cuits so neutralization and feedback can be

adjusted independently. Also, it may be de-

sirable to be able to switch the feedback out

of the circuit. For these reasons, the circuit

shown in figure 12 is often used. Switch S1

removes the feedback loop when it is closed.

A slight tendency for low frequency para-

sitic oscillations still exists with this circuit.

Li should have as little inductance as possible

without upsetting the feedback. If the value of

Li is too low, it cancels out part of the re-

actance of feedback capacitor G and causes

the feedback to increase at low values of radio

frequency. In some cases, a swamping resistor

may be necessary across L. The value of this

resistor should be high compared to the re-

actance of G to avoid phase -shift of the r -f

feedback.

14 -3 Neutralization

Procedure in

Feedback -Type Amplifiers

Experience with feedback amplifiers has

brought out several different methods of neu-

tralizing. An important observation is that

when all three neutralizing adjustments are

correctly made the peaks and dips of various

tuning meters all coincide at the point of cir-

cuit resonance. For example, the coincident in-

dications when the various tank circuits are

tuned through resonance with feedback oper-

ating are:

A -When the PA plate circuit is tuned

through resonance:

1 -PA plate current dip

2 -Power output peak

3 -PA r -f grid voltage dip

4 -PA grid current dip

(Note: The PA grid current peaks

when feedback circuit is disabled

and the tube is heavily driven)

www.americanradiohistory.com

278 R -F Feedback THE RADIO

R-F IN- c

Li o pi.

O ° Tc.

cll

TY

11(-

F

ÌIr gifIIN 1 c

ct0 C:7T your RFC o T

HI, 1- 1- 1

BIAS l-

GF

6+ BIAS

Figure 13

TWO -STAGE AMPLIFIER WITH FEED

p FOUT

° T

BACK CIRCUIT.

B -When the PA grid circuit is tuned

through resonance:

1- Driver plate current dip

2 -PA r -f grid voltage peak

3 -PA grid current peak

4 -PA power output peak

C -When the driver grid circuit is tuned

through resonance:

1- Driver r -f grid voltage peak

2- Driver plate current peak

3 -PA r -f grid current peak

4 -PA plate current peak

5 -PA power output peak

Four meters may be employed to measure

the most important of these parameters. The

meters should be arranged so that the follow-

ing pairs of readings are displayed on meters

located close together for ease of observation

of coincident peaks and dips:

1 -PA plate current and power output

2 -PA r -f grid current and PA plate

current

3 -PA r -f grid voltage and power out-

put

4- Driver plate current and PA r -f

grid voltage

The third pair listed above may not be

necessary if the PA plate current dip is pro-

nounced. When this instrumentation is pro-

vided, the neutralizing procedure is as follows:

1- Remove the r -f feedback

Figure 14

FEEDBACK SHORTING DEVICE.

2- Neutralize the grid -plate capaci-

tance of the driver stage

3- Neutralize the grid -plate capaci-

tance of the power amplifier (PA)

stage

4 -Apply r -f feedback

5- Neutralize driver grid- cathode ca-

pacitance

These steps will be explained in more detail

in the following paragraphs:

Step 1. The removal of r -f feedback through

the feedback circuit must be complete. The

switch ( ) shown in the feedback circuit

( figure 13 ) is one satisfactory method. Since

C. is effectively across the PA plate tank cir-

cuit it is desirable to keep it across the circuit

when feedback is removed to avoid appreciable

detuning of the plate tank circuit. Another

method that can be used if properly done is

to ground the junction of C and C. Ground-

ing this common point through a switch or

relay is not good enough because of common

coupling through the length of the grounding

lead. The grounding method shown in figure

14 is satisfactory.

Step 2. Plate power and excitation are applied.

The driver grid tank is resonated by tuning

for a peak in driver r -f grid voltage or driver

plate current. The power amplifier grid tank

circuit is then resonated and adjusted for a

dip in driver plate current. Driver neutraliza-

tion is now adjusted until the PA r -f grid

voltage (or PA grid current) peaks at exactly

the point of driver plate current dip. A handy

rule for adjusting grid -plate neutralization of

a tube without feedback: with all circuits in

resonance, detune the plate circuit to the high

frequency side of resonance: If grid current

to next stage (or power output of the stage

under test) increases, more neutralizing capaci-

tance is required and vice versa.

If the driver tube operates class A so that

a plate current dip cannot be observed, a dif-

www.americanradiohistory.com

HANDBOOK Neutralization 279

T"

Figure 15

FEEDBACK NEUTRALIZING

CIRCUIT USING

AUXILIARY RECEIVER.

ferent neutralizing procedure is necessary This

will be discussed in a subsequent section.

Step 3. This is the same as step 2 except it

is applied to the power amplifier stage. Ad-

just the neutralization of this stage for a peak

in power output at the plate current dip.

Step 4. Reverse step 1 and apply the r -f feed-

back.

Step 5. Apply plate power and an exciting sig-

nal to drive the amplifier to nearly full out-

put. Adjust the feedback neutralization for a

peak in amplifier power output at the exact

point of minimum amplifier plate current.

Decrease the feedback neutralization capaci-

tance if the power output rises when the tank

circuit is tuned to the high frequency side of

resonance.

The above sequence applies when the neu-

tralizing adjustments are approximately cor-

rect to start with. If they are far off, some "cut -

and -try" adjustment may be necessary. Also,

the driver stage may break into oscillation if

the feedback neutralizing capacitance is not

near the correct setting.

It is assumed that a single tone test signal

is used for amplifier excitation during the

above steps, and that all tank circuits are at

resonance except the one being detuned to

make the observation. There is some interaction

between the driver neutralization and the feed-

back neutralization so if an appreciable change

is made in any adjustment the others should

be rechecked. It is important that the grid -plate

neutralization be accomplished first when using

the above procedure, otherwise the feedback

neutralization will be off a little, since it par-

tially compensates for that error.

Neutralization

Techniques The method of neutralization

employing a sensitive r -f de-

tector inductively coupled to

a tank coil is difficult to apply in some cases

because of mechanical construction of the

equipment, or because of undesired coupling.

Another method for observing neutralization

can be used, which appears to be more ac-

curate in actual practice. A sensitive r -f detec-

tor such as a receiver is loosely coupled to the

grid of the stage being neutralized, as shown

in figure 15. The coupling capacitance is of

the order of one or two µµfd. It must be small

enough to avoid upsetting the neutralization

when it is removed because the total grid -

ground capacitance is one leg of the neutraliz-

ing bridge. A signal generator is connected at

point S and the receiver at point R. If Coo is

not properly adjusted the S -meter on the re-

ceiver will either kick up or down as the grid

tank circuit is tuned through resonance. Go

may be adjusted for minimum deflection of the

S -meter as the grid circuit is tuned through

resonance.

The grid -plate capacitance of the tube is

then neutralized by connecting the signal gen-

erator to the plate of the tube and adjusting

Cti of figure 13 for minimum deflection again

as the grid tank is tuned through resonance.

The power amplifier stage is neutralized in

the same manner by connecting a receiver

loosely to the grid circuit, and attaching a

signal generator to the plate of the tube. The

r -f signal can be fed into the amplifier output

terminal if desired.

Some precautions are necessary when using

this neutralization method. First, some driver

tubes (the 6CL6, for example) have appre-

ciably more effective input capacitance when

in operation and conducting plate current than

when in standby condition. This increase in

input capacitance may be as great as three or

four µµfd, and since this is part of the neu-

tralizing bridge circuit it must be taken into

consideration. The result of this change in

input capacitance is that the neutralizing ad-

justment of such tubes must be made when

they are conducting normal plate current. Stray

coupling must be avoided, and it may prove

helpful to remove filament power from the

preceding stage or disable its input circuit in

some manner.

It should be noted that in each of the above

adjustments that minimum reaction on the

grid is desired, not minimum voltage. Some

residual voltage is inherent on the grid when

this neutralizing circuit is used.

www.americanradiohistory.com

CHAPTER FIFTEEN

Amplitude Modulation

If the output of a c -w transmitter is varied

in amplitude at an audio frequency rate in-

stead of interrupted in accordance with code

characters, a tone will be heard on a receiver

tuned to the signal. If the audio signal con-

sists of a band of audio frequencies com-

prising voice or music intelligence, then the

voice or music which is superimposed on the

radio frequency carrier will be heard on the

receiver.

When voice, music, video, or other intelli-

gence is superimposed on a radio frequency

carrier by means of a corresponding variation

in the amplitude of the radio frequency output

of a transmitter, amplitude modulation is the

result. Telegraph keying of a c -w transmitter

is the simplest form of amplitude modulation,

while video modulation in a television trans-

mitter represents a highly complex form. Sys-

tems for modulating the amplitude of a carrier

envelope in accordance with voice, music, or

similar types of complicated audio waveforms

are many and varied, and will be discussed

later on in this chapter.

15-1 Sidebands

Modulation is essentially a form of mixing

or combining already covered in a previous

chapter. To transmit voice at radio frequencies

by means of amplitude modulation, the voice

frequencies are mixed with a radio frequency

carrier so that the voice frequencies are con-

verted to radio frequency sidebands. Though

it may be difficult to visualize, the amplitude

of the radio frequency carrier does not vary

during conventional amplitude modulation.

Even though the amplitude of radio fre-

quency voltage representing the composite

signal ( resultant of the carrier and sidebands,

called the envelope) will vary from zero to

twice the unmodulated signal value during

full modulation, the amplitude of the carrier

component does not vary. Also, so long as

the amplitude of the modulating voltage does

not vary, the amplitude of the sidebands will

remain constant. For this to be apparent, how-

ever, it is necessary to measure the amplitude

of each component with a highly selective

filter. Otherwise, the measured power or volt-

age will be a resultant of two or more of the

components, and the amplitude of the resultant

will vary at the modulation rate.

If a carrier frequency of 5000 kc. is modu-

lated by a pure tone of 1000 cycles, or 1 kc.,

two sidebands are formed: one at 5001 kc.

(the sum frequency) and one at 4999 kc. (the

difference frequency). The frequency of each

sideband is independent of the amplitude of

the modulating tone, or modulation percent-

age; the frequency of each sideband is deter-

mined only by the frequency of the modulat-

ing tone. This assumes, of course, that the

transmitter is not modulated in excess of its

linear capability.

280

www.americanradiohistory.com

Modulation 281

When the modulating signal consists of

multiple frequencies, as is the case with

voice or music modulation, two sidebands will

be formed by each modulating frequency (one

on each side of the carrier), and the radiated

signal will consist of a band of frequencies.

The band width, or channel taken up in the

frequency spectrum by a conventional double -

sideband amplitude- modulated signal, is equal

to twice the highest modulating frequency.

For example, if the highest modulating fre-

quency is 5000 cycles, then the signal (as-

suming modulation of complex and varying

waveform) will occupy a band extending from

5000 cycles below the carrier to 5000 cycles

above the carrier.

Frequencies up to at least 2500 cycles, and

preferably 3500 cycles, are necessary for good

speech intelligibility. If a filter is incorpo-

rated in the audio system to cut out all fre-

quencies above approximately 3000 cycles,

the band width of a radio- telephone signal can

be limited to 6 kc. without a significant loss

in intelligibility. However, if harmonic distor-

tion is introduced subsequent to the filter, as

would happen in the case of an overloaded

modulator or overmodulation of the carrier,

new frequencies will be generated and the

signal will occupy a band wider than 6 kc.

15-2 Mechanics of

Modulation

A c -w or unmodulated r -f carrier wave is

represented in figure 1A. An audio frequency

sine wave is represented by the curve of

figure 113. When the two are combined or

"mixed," the carrier is said to be amplitude

modulated, and a resultant similar to 1C or

1D is obtained. It should be noted that under

modulation, each half cycle of r -f voltage

differs slightly from the preceding one and

the following one; therefore at no time during

modulation is the r -f waveform a pure sine

wave. This is simply another way of saying

that during modulation, the transmitted r -f

energy no longer is confined to a single radio

frequency.

It will be noted that the average amplitude

of the peak r -f voltage, or modulation enve-

lope, is the same with or without modulation.

This simply means that the modulation is

symmetrical (assuming a symmetrical modu-

lating wave) and that for distortionless modu-

lation the upward modulation is limited to a

value of twice the unmodulated carrier wave

amplitude because the amplitude cannot go

below zero on downward portions of the mod-

ulation cycle. Figure 1D illustrates the maxi-

fl f

C.W. OR UNMODULATED CARRIER

SINE WAVE

AUDIO SIGNAL FROM MODULATOR

A 2

ItiÌ% Ì 1TjTI1 1 ZIUIÌ

1111111111111111111111111

lA/2 t

III 111 ÌI, _A /2

50 % MODULATED CARRIER

A

00% MODULATED CARRIER

Figure 1

AMPLITUDE MODULATED WAVE

Top drawing (Al represents an unmodulated

carrier wave; (B) shows the audio output of

the modulator. Drawing (C) shows the audio

signal impressed on the carrier wave to the

extent of 50 per cent modulation; (D) shows

the carrier with 100 per cent amplitude modu-

lation.

A

mum obtainable distortionless modulation with

a sine modulating wave, the r -f voltage at the

peak of the r -f cycle varying from zero to

twice the unmodulated value, and the r -f power

varying from zero to four times the unmodu-

lated value ( the power varies as the square

of the voltage).

While the average r -f voltage of the modu-

lated wave over a modulation cycle is the

same as for the unmodulated carrier, the aver-

age power increases with modulation. If the

radio frequency power is integrated over the

audio cycle, it will be found with 100 per cent

sine wave modulation the average r -f power

has increased 50 per cent. This additional

power is represented by the sidebands, be-

cause as previously mentioned, the carrier

power does not vary under modulation. Thus,

when a 100 -watt carrier is modulated 100 per

cent by a sine wave, the total r -f power is 150

watts; 100 watts in the carrier and 25 watts

in each of the two sidebands.

www.americanradiohistory.com

282 Amplitude Modulation THE RADIO

Modulation So long as the relative propor-

Percentage Lion of the various sidebands

making up voice modulation is

maintained, the signal may be received and

detected without distortion. However, the

higher the average amplitude of the sidebands,

the greater the audio signal produced at the

receiver. For this reason it is desirable to

increase the modulation percentage, or degree

of modulation, to the point where maximum

peaks just hit 100 per cent. If the modulation

percentage is increased so that the peaks ex-

ceed this value, distortion is introduced, and

if carried very far, bad interference to signals

on nearby channels will result.

Modulation The amount by which a carrier

Measurement is being modulated may be ex-

pressed either as a modulation

factor, varying from zero to 1.0 at maximum

modulation, or as a percentage. The percent-

age of modulation is equal to 100 times the

modulation factor. Figure 2A shows a carrier

wave modulated by a sine -wave audio tone.

A picture such as this might be seen on the

screen of a cathode -ray oscilloscope with

sawtooth sweep on the horizontal plates and

the modulated carrier impressed on the verti-

cal plates. The same carrier without modulation

would appear on the oscilloscope screen as

figure 2B.

The percentage of modulation of the posi-

tive peaks and the percentage of modulation

of the negative peaks can be determined sepa-

rately from two oscilloscope pictures such

as shown.

The modulation factor of the positive peaks

may be determined by the formula:

Emax - Ecar

M =

Ecar

The factor for negative peaks may be de-

termined from this formula:

Ecar - Emin

M -

Ecar

In the above two formulas Ern ax is the max-

imum carrier amplitude with modulation and

Ellin is the minimum amplitude; Ecar is the

steady -state amplitude of the carrier with-

out modulation. Since the deflection of the

spot on a cathode -ray tube is linear with re-

spect to voltage, the relative voltages of

these various amplitudes may be determined

by measuring the deflections, as viewed on

the screen, with a rule calibrated in inches

or centimeters. The percentage of modulation

of the carrier may be had by multiplying the

modulation factor thus obtained by 100. The

above procedure assumes that there is no

ECAR

Figure 2

GRAPHICAL DETERMINATION OF MODU-

LATION PERCENTAGE

The procedure for determining modulation

percentage from the peak voltage points in-

dicated is discussed in the text.

carrier shift, or change in average amplitude,

with modulation.

If the modulating voltage is symmetrical,

such as a sine wave, and modulation is ac-

complished without the introduction of dis-

tortion, then the percentage modulation will

be the same for both negative and positive

peaks. However, the distribution and phase

relationships of harmonics in voice and music

waveforms are such that the percentage modu-

lation of the negative modulation peaks may

exceed the percentage modulation of the posi-

tive peaks, and vice versa. The percentage

modulation when referred to without regard

to polarity is an indication of the average of

the negative and positive peaks.

Modulation The modulation capability of a

Capability transmitter is the maximum per-

centage to which that transmitter

may be modulated before spurious sidebands

are generated in the output or before the dis-

tortion of the modulating waveform becomes

objectionable. The highest modulation cap-

ability which any transmitter may have on the

negative peaks is 100 per cent. The maximum

permissible modulation of many transmitters

is less than 100 per cent, especially on posi-

tive peaks. The modulation capability of a

transmitter may be limited by tubes with in-

sufficient filament emission, by insufficient

excitation or grid bias to a plate- modulated

stage, too light loading of any type of ampli-

fier carrying modulated r.f., insufficient power

output capability in the modulator, or too much

excitation to a grid -modulated stage or a

Class B linear amplifier. In any case, the

FCC regulations specify that no transmitter

be modulated in excess of its modulation

capability. Hence, it is desirable to make the

modulation capability of a transmitter as near

as possible to 100 per cent so that the carrier

power may be used most effectively.

www.americanradiohistory.com

HANDBOOK Modulation Systems 283

Speech Waveform The manner in which the

Dissymmetry human voice is produced

by the vocal cords gives

rise to a certain dissymmetry in the waveform

of voice sounds when they are picked up by

a good -quality microphone. This is especially

pronounced in the male voice, and more so

on certain voiced sounds than on others. The

result of this dissymmetry in the waveform is

that the voltage peaks on one side of the

average value of the wave will be consider-

ably greater, often two or three times as great,

as the voltage excursions on the other side

of the zero axis. The average value of volt-

age on both sides of the wave is, of course,

the same.

As a result of this dissymmetry in the male

voice waveform, there is an optimum polarity

of the modulating voltage that must be ob-

served if maximum sideband energy is to be

obtained without negative peak clipping and

generation of "splatter" on adjacent channels.

A double -pole double -throw "phase revers-

ing" switch in the input or output leads of any

transformer in the speech amplifier system will

permit poling the extended peaks in the direc-

tion of maximum modulation capability. The

optimum polarity may be determined easily by

listening on a selective receiver tuned to a

frequency 30 to 50 kc. removed from the de-

sired signal and adjusting the phase reversing

switch to the position which gives the least

"splatter" when the transmitter is modulated

rather heavily. If desired, the switch then may

be replaced with permanent wiring, so long as

the microphone and speech system are not to

be changed.

A more conclusive illustration of the lop-

sidedness of a speech waveform may be ob-

tained by observing the modulated waveform

of a radiotelephone transmitter on an oscillo-

scope. A portion of the carrier energy of the

transmitter should be coupled by means of a

link directly to the vertical plates of the

'scope, and the horizontal sweep should be a

sawtooth or similar wave occurring at a rate

of approximately 30 to 70 sweeps per second.

With the speech signal from the speech am-

plifier connected to the transmitter in one po-

larity it will be noticed that negative -peak

clipping -as indicated by bright "spots" in

the center of the 'scope pattern whenever the

carrier amplitude goes to zero -will occur at

a considerably lower level of average modula-

tion than with the speech signal being fed to

the transmitter in the other polarity. When the

input signal to the transmitter is polarized in

such a manner that the "fingers" of the

speech wave extend in the direction of posi-

tive modulation these fingers usually will be

clipped in the plate circuit of the modulator

at an acceptable peak modulation level.

The use of the proper polarity of the incom-

ing speech wave in modulating a transmitter

can afford an increase of approximately two

to one in the amount of speech audio power

which may be placed upon the carrier for an

amplitude -modulated transmitter for the same

amount of sideband splatter. More effective

methods for increasing the amount of audio

power on the carrier of an AM phone trans-

mitter are discussed later in this chapter.

Single- Sideband Because the same intelli-

Transmission gibility is contained in each

of the sidebands associated

with a modulated carrier, it is not necessary

to transmit sidebands on both sides of the

carrier. Also, because the carrier is simply a

single radio frequency wave of unvarying am-

plitude, i t is no t necessary to transmit the

carrier if some means is provided for inserting

a locally generated carrier at the receiver.

When the carrier is suppressed but both

upper and lower sidebands are transmitted, it

is necessary to insert a locally generated

carrier at the receiver of exactly the same

frequency and phase as the carrier which was

suppressed. For this reason, suppressed -

carrier double -sideband systems have little

practical application.

When the carrier is suppressed and only the

upper or the lower sideband is transmitted, a

highly intelligible signal may be obtained at

the receiver even though the locally generated

carrier differs a few cycles from the frequency

of the carrier which was suppressed at the

transmitter. A communications system utiliz-

ing but one group of sidebands with carrier

suppressed is known as a single sideband

system. Such systems are widely used for

commercial point to point work, and are being

used to an increasing extent in amateur com-

munication. The two chief advantages of the

system are: (1) an effective power gain of

about 9 db results from putting all the radiat-

ed power in intelligence carrying sideband

frequencies instead of mostly into radiated

carrier, and (2) elimination of the selective

fading and distortion that normally occurs in

a conventional double - sideband system when

the carrier fades and the sidebands do not, or

the sidebands fade differently.

15 -3 Systems of Amplitude

Modulation

There are many different systems and meth-

ods for amplitude modulating a carrier, but

most may be grouped under three general clas-

sifications: (1) variable efficiency systems

in which the average input to the stage re-

www.americanradiohistory.com

284 Amplitude Modulation THE RADIO

mains constant with and without modulation

and the variations in the efficiency of the

stage in accordance with the modulating sig-

nal accomplish the modulation; (2) constant

efficiency systems in which the input to the

stage is varied by an external source of modu-

lating energy to accomplish the modulation;

and (3) so- called high -efficiency systems in

which circuit complexity is increased to ob-

high plate circuit efficiency in the modulated

stage without the requirement of an external

high -level modulator. The various systems

under each classification have individual

characteristics which make certain ones best

suited to particular applications.

Variable Efficiency Since the average input

Modulation remains constant in a

stage employing variable

efficiency modulation, and since the average

power output of the stage increases with modu-

lation, the additional average power output

from the stage with modulation must come from

the plate dissipation of the tubes in the stage.

Hence, for the best relation between tube cost

and power output the tubes employed should

have as high a plate dissipation rating per

dollar as possible.

The plate efficiency in such an amplifier is

doubled when going from the unmodulated

condition to the peak of the modulation cycle.

Hence, the unmodulated efficiency of such an

amplifier must always be less than 45 per

cent, since the maximum peak efficiency ob-

tainable in a conventional amplifier is in the

vicinity of 90 per cent. Since the peak effi-

ciency in certain types of amplifiers will be

as low as 60 per cent, the unmodulated effi-

ciency in such amplifiers will be in the vici-

nity of 30 per cent.

Assuming a typical amplifier having a peak

efficiency of 70 per cent, the following fig-

ures give an idea of the operation of an ideal-

ized efficiency -modulated stage adjusted for

100 per cent sine -wave modulation. It should

be kept in mind that the plate voltage is con-

stant at all times, even over the audio cycles.

Plate input without modulation 100 watts

Output without modulation 35 watts

Efficiency without modulation 35%

Input on 100% positive modulation

peak (plate current doubles) 200 watts

Efficiency on 100% positive peak 70%

Output on 100% positive modula-

tion peak 140 watts

Input on 100% negative peak

Efficiency on 100% negative peak

Output on 100% negative peak

0 watts

0%

0 watts

Average input with 100%

modulation 100 watts

Average output with 100% modula-

tion (35 watts carrier plus 17.5

watts sideband) 52.5 watts

Average efficiency with 100%

modulation 52.5%

Systems of Efficiency There are many sys-

Modulation tems of efficiency mod-

ulation, but they all

have the general limitation discussed in the

previous paragraph -so long as the carrier

amplitude is to remain constant with and

without modulation, the efficiency at carrier

level must be not greater than one -half the

peak modulation efficiency if the stage is to

be capable of 100 per cent modulation.

The classic example of efficiency modula-

tion is the Class B linear r -f amplifier, to be

discussed below. The other three common

forms of efficiency modulation are control -

grid modulation, screen -grid modulation, and

suppressor -grid modulation. In each case,

including that of the Class B linear amplifier

note that the modulation, or the modulates

signal, is impressed on a control electrode

of the stage.

The Class B This is the simplest practi-

Linear Amplifier cable type amplifier for an

amplitude -modulated wave

or a single -sideband signal. The system pos-

sesses the disadvantage that excitation, grid

bias, and loading must be carefully controlled

to preserve the linearity of the stage. Also,

the grid circuit of the tube, in the usual appli-

cation where grid current is drawn on peaks,

presents a widely varying value of load im-

pedance to the source of excitation. Hence it

is necessary to include some sort of swamping

resistor to reduce the effect of grid- imped-

ance variations with modulation. If such a

swamping resistance across the grid tank is

not included, or is too high in value, the posi-

tive modulation peaks of the incoming modu-

lated signal will tend to be flattened with

resultant distortion of the wave being amplified.

The Class B linear amplifier has long been

used in broadcast transmitters, but recently

has received much more general usage in the

h -f range for two significant reasons: (a) the

Class B linear is an excellent way of increas-

ing the power output of a single -sideband

transmitter, since the plate efficiency with

full signal will be in the vicinity of 70 per

cent, while with no modulation the input to

the stage drops to a relatively low value; and

(b) the Class B linear amplifier operates with

relatively low harmonic output since the grid

bias on the stage normally is slightly less

www.americanradiohistory.com

HANDBOOK Class B Linear Amplifier 285

than the value which will cut off plate current

to the stage in the absence of excitation.

Since s Class B linear amplifier is biased

to extended cutoff with no excitation ( the

grid bias at extended cutoff will be approxi-

mately equal to the plate voltage divided by

the amplification factor for a triode, and will

be approximately equal to the screen voltage

divided by the grid- screen mu factor for a

tetrode or pentode) the plate current will flow

essentially in 180 -degree pulses. Due to the

relatively large operating angle of plate cur-

rent flow the theoretical peak plate efficiency

is limited to 78.5 per cent, with 65 to 70 per

cent representing a range of efficiency nor-

mally attainable, and the harmonic output

will be low.

The carrier power output from a Class B

linear amplifier of a normal 100 per cent mod-

ulated AM signal will be about one -half the

rated plate dissipation of the stage, with opti-

mum operating conditions. The peak output

from a Class B linear, which represents the

maximum- signal output as a single -sideband

amplifier, or peak output with a 100 per cent

AM signal, will be about twice the plate dis-

sipation of the tubes in the stage. Thus the

carrier -level input power to a Class B linear

should be about 1.5 times the rated plate dis-

sipation of the stage.

The schematic circuit of a Class B linear

amplifier is the same as a conventional single -

ended or push -pull stage, whether triodes or

beam tetrodes are used. However, a swamping

resistor, as mentioned before, must be placed

across the grid tank of the stage if the oper-

ating conditions of the tube are such that

appreciable gridcurrent will be drawn on modu-

lation peaks. Also, a fixed source of grid bias

must be provided for the stage. A regulated

grid -bias power supply is the usual source of

negative bias voltage.

Adjustment of a Class With grid bias adjusted

8 Linear Amplifier to the correct value,

and with provision for

varying the excitation voltage to the stage

and the loading of the plate circuit, a fully

modulated signal is applied to the grid circuit

of the stage. Then with an oscilloscope cou-

pled to the output of the stage, excitation and

loading are varied until the stage is drawing

the normal plate input and the output wave -

shape is a good replica of the input signal.

The adjustment procedure normally will re-

quire a succession of approximations, until

the optimum set of adjustments is attained.

Then the modulation being applied to the in-

put signal should be removed to check the

linearity. With modulation removed, in the

case of a 100 per cent AM signal, the input

to the stage should remain constant, and the

peak output of the r -f envelope should fall to

half the value obtained on positive modula-

tion peaks.

Class C . One widely used system of

Grid Modulation efficiency modulation for

communications work is

Class C control -grid bias modulation. The dis-

tortion is slightly higher than for a properly

operated Class B linear amplifier, but the effi-

ciency is also higher, and the distortion can

be kept within tolerable limits for communi-

cations work.

Class C grid modulation requires high plate

voltage on the modulated stage, if maximum

output is desired. The plate voltage is nor-

mally run about 50 per cent higher than for

maximum output with plate modulation.

The driving power required for operation of

a grid -modulated amplifier under these condi-

tions is somewhat more than is required for

operation at lower bias and plate voltage, but

the increased power output obtainable over-

balances the additional excitation require-

ment. Actually, almost half as much excitation

is required as would be needed if the same

stage were to be operated as a Class C plate -

modulated amplifier. The resistor R across

the grid tank of the stage serves as swamping

to stabilize the r -f driving voltage. At least

50 per cent of the output of the driving stage

should be dissipated in this swamping resistor

under carrier conditions.

A comparatively small amount of audio power

will be required to modulate the amplifier stage

100 per cent. An audio amplifier having 20

watts output will be sufficient to modulate an

amplifier with one kilowatt input. Proportion-

ately smaller amounts of audio will be re-

quired for lower powered stages. However, the

audio amplifier that is being used as the grid

modulator should, in any case, either employ

low plate resistance tubes such as 2A3's,

employ degenerative feedback from the output

stage to one of the preceding stages of the

speech amplifier, or be resistance loaded with

a resistor across the secondary of the modu-

lation transformer. This provision of low drive

ing impedance in the grid modulator is to insure

good regulation in the audio driver for the grid

modulated stage. Good regulation of both the

audio and the r -f drivers of a grid -modulated

stage is quite important if distortion -free

modulation approaching 100 per cent is desired,

because the grid impedance of the modulated

stage varies widely over the audio cycle.

A practical circuit for obtaining grid -bias

modulation is shown in figure 3. The modula-

tor and bias regulator tube have been com-

bined in a single 6B4G tube.

The regulator -modulator tube operates as

a cathode - follower. The average d -c voltage

www.americanradiohistory.com

286 Amplitude Modulation THE RADIO

R.F. AMPLIFIER

R

AUDIO INPUT

FROM 65J7

ETC

115 V A C

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w.w.

25K IOW

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MIDGET CHOKE

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70 K

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B C TRANSFORMER

Figure 3

GRID -BIAS MODULATOR CIRCUIT

on the control grid is controlled by the 70, 000 -

ohm wire -wound potentiometer and this poten-

tiometer adjusts the average grid bias on the

modulated stage. However, a -c signal voltage

is also impressed on the control -grid of the

tube and since the cathode follows this a -c

wave the incoming speech wave is superim-

posed on the average grid bias, thus effecting

grid -bias modulation of the r -f amplifier stage.

An audio voltage swing is required on the grid

of the 6B4G of approximately the same peak

value as will be required as bias -voltage

swing on the grid -bias modulated stage. This

voltage swing will normally be in the region

from 50 to 200 peak volts. Up to about 100

volts peak swing can be obtained from a 6SJ7

tube as a conventional speech amplifier stage.

The higher voltages may be obtained from a

tube such as a 6J5 through an audio trans-

former of 2:1 or 21/3:1 ratio.

With the normal amount of comparatively

tight antenna coupling to the modulated stage,

a non -modulated carrier efficiency of 40 per

cent can be obtained with substantially dis-

tortion -free modulation up to practically 100

per cent. If the antenna coupling is decreased

slightly from the condition just described, and

the excitation is increased to the point where

the amplifier draws the same input, carrier

efficiency of 50 per cent is obtainable with

tolerable distortion at 90 per cent modulation.

Tuning the The most satisfactory pro -

Grid -Bias cedure for tuning a stage

Modulated Stage for grid -bias modulation of

the Class C type is as

follows. The amplifier should first be neutra-

lized, and any possible tendency toward para-

sitics under any condition of operation should

be eliminated. Then the antenna should be

coupled to the plate circuit, the grid bias

should be run up to the maximum available

value, and the plate voltage and excitation

should be applied. The grid bias voltage

should then be reduced until the amplifier

draws the approximate amount of plate cur-

rent it is desired to run, and modulation corre-

sponding to about 80 per cent is then applied.

If the plate current kicks up when modulation

is applied, the grid bias should be reduced;

if the plate meter kicks down, increase the

grid bias.

When the amount of bias voltage has been

found (by adjusting the fine control, R2, on

the bias supply) where the plate meter re-

mains constant with modulation, it is more

than probable that the stage will be drawing

either too much or too little input. The an-

tenna coupling should then be either increased

or decreased (depending on whether the in-

put was too little or too much, respectively)

until the input is more nearly the correct value.

The bias should then be readjusted until the

plate meter remains constant with modulation

as before. By slight jockeying back and forth

of antenna coupling and grid bias, a point can

be reached where the tubes are running at

rated plate dissipation, and where the plate

milliammeter on the modulated stage remains

substantially constant with modulation.

The linearity of the stage should then be

checked by any of the conventional methods;

the trapezoidal pattern method employing a

cathode -ray oscilloscope is probably the most

satisfactory. The check with the trapezoidal

pattern will allow the determination of the

proper amount of gain to employ on the speech

amplifier. Too much audio power on the grid

of the modulated stage should not be used in

the tuning -up process, as the plate meter will

kick erratically and it will be impossible to

make a satisfactory adjustment.

Screen -Grid Amplitude modulation may be

Modulation accomplished by varying the

screen -grid voltage in a Class

C amplifier which employs a pentode, beam

www.americanradiohistory.com

H A N D B O O K Screen Grid Modulation 287

tetrode, or other type of screen -grid tube. The

modulation obtained in this way is not es-

pecially linear, but screen -grid modulation

does offer other advantages and the linearity

is quite adequate for communications work.

There are two significant and worthwhile

advantages of screen -grid modulation for com-

munications work: (1) The excitation require-

ments for an amplifier which is to be modu-

lated in the screen are not at all critical, and

good regulation of the excitation voltage is

not required. The normal rated grid- circuit

operating conditions specified for Class C

c -w operation are quite adequate for screen -

grid modulation. (2) The audio modulating

power requirements for screen -grid modulation

are relatively low.

A screen -grid modulated r -f amplifier oper-

ates as an efficiency -modulated amplifier, the

same as does a Class B linear amplifier and

a grid -modulated stage. Hence, plate circuit

loading is relatively critical as in any effi-

ciency- modulated stage, and must be adjusted

to the correct value if normal power output

with full modulation capability is to be ob-

tained. As in the case of any efficiency -modu-

lated stage, the operating efficiency at the

peak of the modulation cycle will be between

70 and 80 per cent, with efficiency at the car-

rier level (if the stage is operating in the nor-

mal manner with full carrier) about half of the

peak- modulation value.

There are two main disadvantages of screen -

grid modulation, and several factors which

must be considered if satisfactory operation

of the screen -grid modulated stage is to be

obtained. The disadvantages are: (I) As men-

tioned before, the linearity of modulation with

respect to screen -grid voltage of such a stage

is satisfactory only for communications work,

unless carrier- rectified degenerative feed -back

is employed around the modulated stage to

straighten the linearity of modulation. (2) The

impedance of the screen grid to the modulating

signal is non -linear. This means that the mod-

ulating signal must be obtained from a source

of quite low impedance if audio distortion of

the signal appearing at the screen grid is to

be avoided.

Screen -Grid Instead of being linear with re-

Impedance spect to modulating voltage, as

is the plate circuit of a plate -

modulated Class C amplifier, the screen grid

presents approximately a square -law imped-

ance to the modulating signal over the region

of signal excursion where the screen is posi-

tive with respect to ground. This non -linearity

may be explained in the following manner: At

the carrier level of a conventional screen -

modulated stage the plate -voltage swing of

the modulated tube is one -half the voltage

swing at peak- modulation level. This condition

must exist in any type of conventional effi-

ciency- modulated stage if 100 per cent posi-

tive modulation is to be attainable. Since the

plate -voltage swing is at half amplitude, and

since the screen voltage is at half its full -

modulation value, the screen current is rela-

tively low. But at the positive modulation peak

the screen voltage is approximately doubled,

and the plate -voltage swing also is at twice

the carrier amplitude. Due to the increase in

plate -voltage swing with increasing screen

voltage, the screen current increases more than

linearly with increasing screen voltage.

In a test made on an amplifier with an 813

tube, the screen current at carrier level was

about 6 ma. with screen potential of 190 volts;

but under conditions which represented a posi-

tive modulation peak the screen current meas-

ured 25 ma. at a potential of 400 volts. Thus

instead of screen current doubling with twice

screen voltage as would be the case if the

screen presented a resistive impedance, the

screen current became about four times as

great with twice the screen voltage.

Another factor which must be considered

in the design of a screen -modulated stage, if

full modulation is to be obtained, is that the

power output of a screen -grid stage with zero

screen voltage is still relatively large. Hence,

if anything approaching full modulation on

negative peaks is to be obtained, the screen

potential must be made negative with respect

to ground on negative modulation peaks. In

the usual types of beam tetrode tubes the

screen potential must be 20 to 50 volts nega-

tive with respect to ground before cut -off of

output is obtained. This condition further com-

plicates the problem of obtaining good linearity

in the audio modulating voltage for the screen -

modulated stage, since the screen voltage

must be driven negatively with respect to

ground over a portion of the cycle. Hence the

screen draws no current over a portion of the

modulating cycle, and over the major portion

of the cycle when the screen does draw cur-

rent, it presents approximately a square -law

impedance.

Circuits for Laboratory analysis of a large

ScreenGrid number of circuits for accom-

Modulation plishing screen modulation has

led to the conclusion that the

audio modulating voltage must be obtained

from a low- impedance source if low- distor-

tion modulation is to be obtained. Figure 4

shows a group of sketches of the modulation

envelope obtained with various types of modu-

lators and also with insufficient antenna coup-

ling. The result of this laboratory work led

to the conclusion that the cathode -follower

modulator of the basic circuit shown in figure

www.americanradiohistory.com

288 Amplitude Modulation THE RADIO

+MOO. +5.0. -50 V. APPROX.

O

ENVELOPE OBTAINED WITH

INSUFFICIENT ANTENNA

COUPLING

Figure 4

SCREEN -MODULATION CIRCUITS

Three common screen modulation circuits are illustrated above. All three circuits

are capable of giving intelligible voice modulation although the waveform distortion

in the circuits of (A) and (B) is likely to be rather severe. The arrangement at (A)

is often called "clamp tube" screen modulation; by returning the grid leak on the

clomp tube to ground the circuit will give controlled- carrier screen modulation. This

circuit has the advantage that it is simple and is well suited to use in mobile trans-

mitters. (B) is an arrangement using a transformer coupled modulator, and offers no

particular advantages. The arrangement at (C) is capable of giving good modulation

linearity due to the low impedance of the cathode -follower modulator. However, due

to the relatively low heater -cathode ratings on tubes suited for use as the modula-

tor, a separate heater supply for the modulator tube normally is required. This limi-

tation makes application of the circuit to the mobile transmitter a special problem,

since an isolated heater supply normally is not available. Shown at (D) as an assist-

ance in the tuning of a screen -modulated transmitter (or any efficiency -modulated

transmitter for that matter) is the type of modulation envelope which results when

loading to the modulated stage is insufficient.

5 is capable of giving good -quality screen -

grid modulation, and in addition the circuit

provides convenient adjustments for the car-

rier level and the output level on negative

modulation peaks. This latter control, P2 in

figure 5, allows the amplifier to be adjusted

in such a manner that negative -peak clipping

cannot take place, yet the negative modulation

peaks may be adjusted to a level just above

that at which sideband splatter will occur.

The Cathode -

Follower Modulator The cathode follower is

ideally suited for use as

the modulator for a screen-

grid stage since it acts as a relatively low -

impedance source of modulating voltage for

the screen -grid circuit. In addition the cathode -

follower modulator allows the supply voltage

both for the modulator and for the screen grid

of the modulated tube to be obtained from the

high -voltage supply for the plate of the screen -

grid tube or beam tetrode. In the usual case

the plate supply for the cathode follower, and

hence for the screen grid of the modulated

tube, may be taken from the bleeder on the

high- voltage power supply. A tap on the bleeder

may be used, or two resistors may be connect-

ed in series to make up the bleeder, with ap-

www.americanradiohistory.com

HANDBOOK Modulation Systems 289

propriate values such that the voltage applied

to the plate of the cathode follower is appro-

priate for the tube to be modulated. It is im-

portant that a bypass capacitor be used from

the plate of the cathode - follower modulator

to ground.

The voltage applied to the plate of the

cathode follower should be about 100 volts

greater than the rated screen voltage for the

tetrode tube as a c -w Class C amplifier. Hence

the cathode -follower plate voltage should be

about 350 volts for an 815, 2E26, or 829B,

about 400 volts for an 807 or 4 -125A, about

500 volts for an 813, and about 600 volts for

a 4 -250A or a 4E27. Then potentiometer P1

in figure 5 should be adjusted until the carrier -

level screen voltage on the modulated stage

is about one -half the rated screen voltage

specified for the tube as a Class C c -w ampli-

fier. The current taken by the screen of the

modulated tube under carrier conditions will

be about one - fourth the normal screen current

for c -w operation.

The only current taken by the cathode

follower itself will be that which will flow

through the 100,000 -ohm resistor between the

cathode of the 6L6 modulator and the nega-

tive supply. The current taken from the bleeder

on the high -voltage supply will be the carrier -

level screen current of the tube being modu-

lated (which current passes of course through

the cathode follower) plus that current which

will pass through the 100,000 -ohm resistor.

The loading of the modulated stage should

be adjusted until the input to the tube is about

50 per cent greater than the rated plate dissi-

pation of the tube or tubes in the stage. If the

carrier -level screen voltage value is correct

for linear modulation of the stage, the loading

will have to be somewhat greater than that

amount of loading which gives maximum output

from the stage. The stage may then be modu-

lated by applying an audio signal to the grid

of the cathode -follower modulator, while ob-

serving the modulated envelope on an oscillo-

scope.

If good output is being obtained, and the

modulation envelope appears as shown in fig-

ure 4C, all is well, except that P2 in figure 5

should be adjusted until negative modulation

peaks, even with excessive modulating signal,

do not cause carrier cutoff with its attendant

sideband splatter. If the envelope appears as

at figure 4D, antenna coupling should be in-

creased while the carrier level is backed down

by potentiometer PI in figure 5 until a set of

adjustments is obtained which will give a satis-

factory modulation envelope as shown in

figure 4C.

Changing Bands After a satisfactory set of ad-

justments has been obtained,

Figure 5

CATHODE -FOLLOWER

SCREEN -MODULATION CIRCUIT

A detailed discussion of this circuit, which

also is represented in figure 4C, is given in

the accompanying text.

it is not difficult to readjust the amplifier for

operation on different bands. Potentiometers

P1 (carrier level), and P2 ( negative peak level)

may be left fixed after a satisfactory adjust-

ment, with the aid of the scope, has once been

found. Then when changing bands it is only

necessary to adjust excitation until the correct

value of grid current is obtained, and then to

adjust antenna coupling until correct plate

current is obtained. Note that the correct plate

current for an efficiency -modulated amplifier

is only slightly less than the out -of- resonance

plate current of the stage. Hence carrier -level

screen voltage must be low so that the out -of-

resonance plate current will not be too high,

and relatively heavy antenna coupling must be

used so that the operating plate current will

be near the out -of- resonance value, and so that

the operating input will be slightly greater

than 1.5 times the rated plate dissipation of

the tube or tubes in the stage. Since the carrier

efficiency of the stage will be only 35 to 40

per cent, the tubes will be operating with plate

dissipation of approximately the rated value

without modulation.

Speech Clipping in The maximum r -f output

the Modulated Stage of an efficiency -modu-

lated stage is limited

by the maximum possible plate voltage swing

on positive modulation peaks. In the modula -

lation circuit of figure 5 the minimum output

is limited by the minimum voltage which the

screen will reach on a negative modulation

peak, as set by potentiometer P2 Hence the

screen -grid- modulated stage, when using the

modulator of figure 5, acts effectively as a

speech clipper, provided the modulating signal

amplitude is not too much more than that value

www.americanradiohistory.com

290 Amplitude Modulation THE RADIO

which will accomplish full modulation. With

correct adjustments of the operating conditions

of the stage it can be made to clip positive

and negative modulation peaks symmetrically.

However, the inherent peak clipping ability of

the stage should not be relied upon as a means

of obtaining a large amount of speech com-

pression, since excessive audio distortion and

excessive screen current on the modulated

stage will result.

Characteristics of a An important character -

Typical Screen istic of the screen -modu-

Modulated Stage lated stage, when using

the cathode -follower mod-

ulator, is that excessive plate voltage on the

modulated stage is not required. In fact, full

output usually may be obtained with the larger

tubes at an operating plate voltage from one -

half to two- thirds the maximum rated plate

voltage for c -w operation. This desirable con-

dition is the natural result of using a low -

impedance source of modulating signal for

the stage.

As an example of a typical screen -modu-

lated stage, full output of 75 watts of carrier

may be obtained from an 813 tube operating

with a plate potential of only 1250 volts. No

increase in output from the 813 may be ob-

tained by increasing the plate voltage, since

the tube may be operated with full rated plate

dissipation of 125 watts, with normal plate

efficiency for a screen -modulated stage, 37.5

per cent, at the 1250 -volt potential.

The operating conditions of a screen -modu-

lated 813 stage are as follows:

Plate voltage -1250 volts

Plate current -160 ma.

Plate input -200 watts

Grid current -11 ma.

Grid bias -I 10 volts

Carrier screen voltage -190 volts

Carrier screen current -6 ma.

Power output -approx. 75 watts

With full 100 per cent modulation the plate

current decreases about 2 ma. and the screen

current increases about 1 ma.; hence plate,

screen, and grid current remain essentially

constant with modulation. Referring to figure

5, which was the circuit used as modulator

for the 813, (El) measured plus 155 volts, (E2)

measured -50 volts, (E3) measured plus 190

volts, (Et) measured plus 500 volts, and the

r.m.s. swing at (E5) for full modulation meas-

ured 210 volts, which represents a peak swing

of about 296 volts. Due to the high positive

voltage, and the large audio swing, on the

cathode of the 6L6 (triode connected) modu-

lator tube, it is important that the heater of

of this tube be fed from a separate filament

transformer or filament winding. Note also that

the operating plate -to- cathode voltage on the

6L6 modulator tube does not exceed the 360 -

volt rating of the tube, since the operating

potential of the cathode is considerably above

ground potential.

Suppressor -Grid Still another form of effi-

Modulation ciency modulation may be

obtained by applying the

audio modulating signal to the suppressor grid

of a pentode Class C r -f amplifier. Basically,

suppressor -grid modulation operates in the

same general manner as other forms of effi-

ciency modulation; carrier plate circuit effi-

ciency is about 35 per cent, and antenna coup-

ling must be rather tight. However, suppressor -

grid modulation has one sizeable disadvantage,

in addition to the fact that pentode tubes are

not nearly so widely used as beam tetrodes

which of course do not have the suppressor

element. This disadvantage is that the screen -

grid current to a suppressor -grid modulated

amplifier is rather high. The high screen cur-

rent is a natural consequence of the rather high

negative bias on the suppressor grid, which

reduces the plate- voltage swing and plate cur-

rent with a resulting increase in the screen

current.

In tuning a suppressor -grid modulated am-

plifier, the grid bias, grid current, screen volt-

age, and plate voltage are about the same as

for Class C c -w operation of the stage. But

the suppressor grid is biased negatively to a

value which reduces the plate- circuit effici-

ency to about one -half the maximum obtainable

from the particular amplifier, with antenna

coupling adjusted until the plate input is about

1.5 times the rated plate dissipation of the

stage. It is important that the input to the

screen grid be measured to make sure that the

rated screen dissipation of the tube is not

being exceeded. Then the audio signal is ap-

plied to the suppressor grid. In the normal

application the audio voltage swing on the

suppressor will be somewhat greater than the

negative bias on the element. Hence sup-

pressor -grid current will flow on modulation

peaks, so that the source of audio signal volt-

age must have good regulation. Tubes suitable

for suppressor -grid modulation are: 2E22,

837, 4E27/8001, 5 -125, 804 and 803. A typi-

cal suppressor -grid modulated amplifier is

illustrated in figure 6.

15 -4 Input Modulation

Systems

Constant efficiency variable -input modula-

tion systems operate by virtue of the addition

www.americanradiohistory.com

HANDBOOK Plate Modulation 291

R.F INPUT

A.F INPUT

4E27 CARRIER

OUTPUT

'33w

IG= AIA

ISG'

44 M

-130 V. -

6J5 2.1 STEPUP

IP=)OMA.

+1500 V.

PEAK SWING FOR FULL

MODULATION = 210 V.

+300 V -210 V.

Figure 6

AMPLIFIER WITH SUPPRESSOR -GRID

MODULATION

Recommended operating conditions for lin-

ear suppressor -grid modulation of a 4E27/

2578/8001 stage are given on the drawing.

of external power to the modulated stage to

effect the modulation. There are two general

classifications that come under this heading;

those systems in which the additional power

is supplied as audio frequency energy from a

modulator, usually called plate modulation

systems, and those systems in which the addi-

tional power to effect modulation is supplied

as direct current from the plate supply.

Under the former classification comes Heis-

ing modulation (probably the oldest type of

modulation to be applied to a continuous car-

rier), Class B plate modulation, and series

modulation. These types of plate modulation

are by far the easiest to get into operation,

and they give a very good ratio of power input

to the modulated stage to power output; 65 to

80 per cent efficiency is the general rule. It

is for these two important reasons that these

modulation systems, particularly Class B plate

modulation, are at present the most popular

for communications work.

Modulation systems coming under the sec-

ond classification are of comparatively recent

development but have been widely applied to

broadcast work. There are quite a few systems

in this class. Two of the more widely used

are the Doherty linear amplifier, and the Ter -

man- Woodyard high- efficiency grid- modulated

amplifier. Both systems operate by virtue of

a carrier amplifier and a peak amplifier con-

nected together by electrical quarter -wave

lines. They will be described later in this

section.

Plate Modulation Plate modulation is the ap-

plication of the audio power

to the plate circuit of an r -f amplifier. The r -f

amplifier must be operated Class C for this

type of modulation in order to obtain a radio -

frequency output which changes in exact ac-

cordance with the variation in plate voltage.

The r -f ampli fier is 100 per cent modulated

when the peak a -c voltage from the modulator

is equal to the d.c. voltage applied to the r -f

tube. The positive peaks of audio voltage in-

crease the instantaneous plate voltage on the

r -f tube to twice the .1c value, and the nega-

tive peaks reduce the voltage to zero.

The instantaneous plate current to the r -f

stage also varies in accordance with the modu-

lating voltage. The peak alternating current

in the output of a modulator must be equal to

the d -c plate current of the Class C r -f stage

at the point of 100 per cent modulation. This

combination of change in audio voltage and

current can be most easily referred to in terms

of audio power in watts.

In a sinusoidally modulated wave, the an-

tenna current increases approximately 22 per

cent for 100 per cent modulation with a pure

tone input; an r -f meter in the antenna circuit

indicates this increase in antenna current.

The average power of the r -f wave increases

50 per cent for 100 per cent modulation, the

efficiency remaining constant.

This indicates that in a plate- modulated

radiotelephone transmitter, the audio- frequency

channel must supply this additional 50 per

cent increase in average power for sine -wave

modulation. If the power input to the modu-

lated stage is 100 watts, for example, the

average power will increase to 150 watts at

100 per cent modulation, and this additional

50 watts of power must be supplied by the

modulator when plate modulation is used. The

actual antenna power is a constant percentage

of the total value of input power.

One of the advantages of plate (or power)

modulation is the ease with which proper ad-

justments can be made in the transmitter. Also.

there is less plate loss in the r -f amplifier for

a given value of carrier power than with other

forms of modulation because the plate effi-

ciency is higher.

By properly matching the plate impedance

of the r -f tube to the output of the modulator,

the ratio of voltage and current swing to d -c

voltage and current is automatically obtained.

The modulator should have a peak voltage

output equal to the average d -c plate voltage

on the modulated stage. The modulator should

also have a peak power output equal to the

d -c plate input power to the modulated stage.

The average power output of the modulator will

depend upon the type of waveform. If the am-

plifier is being Heising modulated by a Class

A stage, the modulator must have an average

www.americanradiohistory.com

292 Amplitude Modulation THE RADIO

MODULATED CLASS C

R. f. AMPLIFIER

+9

Figure 7

HEISING PLATE MODULATION

This type of modulation was the first form

of plate modulation. It is sometimes known

as "constant current" modulation. Because

of the effective 1:1 ratio of the coupling

choke, it is impossible to obtain 100 per cent

modulation unless the plate voltage to the

modulated stage is dropped slightly by re-

sistor R. The capacitor C merely byp

the audio around R, so that the full a-f out-

put voltage of the modulator is impressed

on the Class C stage.

CLASS C

AMPLIFIER

CLASS IS

MODULATOR

MOD. +5 R F. .13

Figure 8

CLASS B PLATE MODULATION

This type of modulation is the most flexible

in that the loading adjustment can be made

in a short period of time and without elabo-

rate test equipment after a change in oper-

ating frequency of the Class C amplifier has

been made.

power output capability of one -half the input

to the Class C stage. If the modulator is a

Class B audio amplifier, the average power

required of it may vary from one -quarter to more

than one -half the Class C input depending

upon the waveform. However, the peak power

output of any modulator must be equal to the

Class C input to be modulated.

Heising Heising modulation is the oldest

Modulation system of plate modulation, and

usually consists of a Class A

audio amplifier coupled to the r -f amplifier by

means of a modulation choke coil, as shown

in figure 7.

The d.c. plate voltage and plate current in

the r -f amplifier must be adjusted to a value

which will cause the plate impedance to match

the output of the modulator, since the modula-

tion choke gives a 1 -to -1 coupling ratio. A

series resistor, by- passed for audio frequen-

cies by means of a capacitor, must be connect-

ed in series with the plate of the r -f amplifier

to obtain modulation up to 100 per cent. The

peak output voltage of a Class A amplifier

does not reach a value equal to the d -c voltage

applied to the amplifier and, consequently,

the d -c plate voltage impressed across the

r -f tube must be reduced to a value equal to

the maximum available a -c peak voltage if

100% modulation is to be obtained.

A higher degree of distortion can be toler-

ated in low -power emergency phone transmitters

which use a pentode modulator tube, and the

series resistor and by -pass capacitor are

usually omitted in such transmitters.

Class B High -level Class B plate

Plata Modulation modulation is the least ex-

pensive method of plate

modulation. Figure 8 shows a conventional

Class B plate -modulated Class C amplifier.

The statement that the modulator output

power must be one -half the Class C input for

100 per cent modulation is correct only if the

waveform of the modulating power is a sine

wove. Where the modulator waveform is un-

clipped speech, the average modulator power

for 100 per cent modulation is considerably

less than one -half the Class C input.

Power Relations in It has been determined ex-

Speech Waveforms perimentally that the ratio

of peak to average power

in a speech waveform is approximately 4 to 1

as contrasted to a ratio of 2 to 1 in a sine

wave. This is due to the high harmonic con-

tent of such a waveform, and to the fact that

www.americanradiohistory.com

HANDBOOK Plate Modulation 293

this high harmonic content manifests itself by

making the wave unsymmetrical and causing

sharp peaks or "fingers" of high energy con-

tent to appear. Thus for unclipped speech, the

average modulator plate current, plate dissi-

pation, and power output are approximately

one -half the sine wave values for a given peak

output power.

Both peak power and average power are

necessarily associated with waveform. Peak

power is just what the name implies; the power

at the peak of a wave. Peak power, although

of the utmost importance in modulation, is of

no great significance in a -c power work, ex-

cept insofar as the average power may be de-

termined from the peak value of a known wave

form.

There is no time element implied in the

definition of peak power; peak power may be

instantaneous -and for this reason average

power, which is definitely associated with

time, is the important factor in plate dissipa-

tion. It is possible that the peak power of a

given waveform be several times the average

value; for a sine wave, the peak power is twice

the average value, and for unclipped speech

the peak power is approximately four times

the average value. For 100 per cent modula-

tion, the peak (instantaneous) audio power

must equal the Class C input, although the

average power for this value of peak varies

widely depending upon the modulator wave-

form, being greater than 50 per cent for speech

that has been clipped and filtered, 50 per cent

for a sine wave, and about 25 per cent for typ-

ical unclipped speech tones.

Modulation The modulation transformer is

Transformer a device for matching the load

Calculations impedance of the Class C am-

plifier to the recommended load

impedance of the Class B modulator tubes.

Modulation transformers intended for com-

munications work are usually designed to

carry the Class C plate current through their

secondary windings, as shown in figure 8.

The manufacturer's ratings should be con-

sulted to insure that the d -c plate current

passed through the secondary winding does

not exceed the maximum rating.

A detailed discussion of the method of

making modulation transformer calculations

has been given in Chapter Six. However, to

emphasize the method of making the calcula-

tion, an additional example will be given.

Suppose we take the case of a Class C am-

plifier operating at a plate voltage of 2000

with 225 ma. of plate current. This amplifier

would present a load resistance of 2000 divi-

ded by 0.225 amperes or 8888 ohms. The plate

power input would be 2000 times 0.225 or 450

watts. By reference to Chapter Six we see that

a pair of 811 tubes operating at 1500 plate

volts will deliver 225 watts of audio output.

The plate -to -plate load resistance for these

tubes under the specified operating conditions

is 18,000 ohms. Hence our problem is to match

the Class C amplifier load resistance of 8888

ohms to the 18,000 -ohm load resistance re-

quired by the modulator tubes.

A 200 -to -300 watt modulation transformer

will be required for the job. If the taps on the

transformer are given in terms of impedances

it will only be necessary to connect the sec-

ondary for 8888 ohms (or a value approximately

equal to this such as 9000 ohms) and the pri-

mary for 18,000 ohms. If it is necessary to

determine the proper turns ratio required of the

transformer it can be determined in the follow-

ing manner. The square root of the impedance

ratio is equal to the turns ratio, hence:

8888 = V 0.494 = 0.703

18000

The transformer must have a turns ratio of

approximately 1- to -0.7 step down, total pri-

mary to total secondary. The greater number

of turns always goes with the higher imped-

ance, and vice versa.

Plate- andScreen

Modulation When only the plate of a

screen -grid tube is modu-

lated, it is impossible to ob-

tain high -percentage linear modulation under

ordinary conditions. The plate current of such

a stage is not linear with plate voltage. How-

ever, if the screen is modulated simultaneously

with the plate, the instantaneous screen volt-

age drops in proportion to the drop in the plate

voltage, and linear modulation can then be ob-

tained. Four satisfactory circuits for accom-

plishing combined plate and screen modula-

tion are shown in figure 9.

The screen r -f by -pass capacitor C2 should

not have a greater value than 0.005 µfd., pref-

erably not larger than 0.001 tad. It should be

large enough to bypass effectively all r -f volt-

age without short- circuiting high- frequency

audio voltages. The plate by -pass capacitor

can be of any value from 0.002 µfd. to 0.005

µfd. The screen -dropping resistor, 111. should

reduce the applied high voltage to the value

specified for operating the particular tube in

the circuit. Capacitor C1 is seldom required

yet some tubes may require this capacitor in

order to keep C2 from attenuating the high fre-

quencies. Different values between .0002 and

.002 µfd. should be tried for best results.

Figure 9C shows another method which uses

a third winding on the modulation transformer,

through which the screen -grid is connected to

www.americanradiohistory.com

294 Amplitude Modulation THE RADIO

E 3

3

B+ S.G. B+

Figure 9

PLATE MODULATION OF A BEAM TETRODE OR SCREEN -GRID TUBE

These alternative arrangements for plate modulation of tetrodes or pentodes are dis-

cussed in detail in the text. The arrangements shown at (B) or (D) are recommended

for most applications.

a low- voltage power supply. The ratio of turns

between the two output windings depends upon

the type of screen -grid tube which is being

modulated. Normally it will be such that the

screen voltage is being modulated 60 per cent

when the plate voltage is receiving 100 per

cent modulation.

If the screen voltage is derived from a drop-

ping resistor ( not a divider) that is bypassed

for r.f. but not a.f., it is possible to secure

quite good modulation by applying modulation

only to the plate. Under these conditions, the

screen tends to modulate itself, the screen

voltage varying over the audio cycle as a re-

sult of the screen impedance increasing with

plate voltage, and decreasing with a decrease

in plate voltage. This circuit arrangement is

illustrated in figure 9B.

A similar application of this principle is

shown in figure 9D. In this case the screen

voltage is fed directly from a low- voltage sup-

ply of the proper potential through a choke L.

A conventional filter choke having an induc-

tance from 10 to 20 henries will be satisfac-

tory for L.

To afford protection of the tube when plate

voltage is not applied but screen voltage is

supplied from the exciter power supply, when

using the arrangement of figure 9D, a resistor

of 3000 to 10,000 ohms can be connected in

series with the choke L. In this case the screen

supply voltage should be at least 1%Z times as

www.americanradiohistory.com

HANDBOOK Cathode Modulation 295

much as is required tor actual screen voltage,

and the value of resistor is chosen such that

with normal screen current the drop through

the resistor and choke will be such that nor-

mal screen voltage will be applied to the tube.

When the plate voltage is removed the screen

current will increase greatly and the drop

through resistor R will increase to such a

value that the screen voltage will be lowered

to the point where the screen dissipation on

the tube will not be exceeded. However, the

supply voltage and value of resistor R must

be chosen carefully so that the maximum rated

screen dissipation cannot be exceeded. The

maximum possible screen dissipation using

this arrangement is equal to: W = E' /4R where

E is the screen supply voltage and R is the

combined resistance of the resistor in figure

9D and the d -c resistance of the choke L. It

is wise, when using this arrangement to check,

using the above formula, to see that the value

of W' obtained is less than the maximum rated

screen dissipation of the tube or tubes used

in the modulated stage. This same system can

of course also be used in figuring the screen

supply circuit of a pentode or tetrode ampli-

fier stage where modulation is not to be

applied.

The modulation transformer for plate -and-

screen- modulation, when utilizing a dropping

resistor as shown in figure 9A, is similar to

the type of transformer used for any plate

modulated phone. The combined screen and

plate current is divided into the plate voltage

in order to obtain the Class C amplifier load

impedance. The peak audio power required to

obtain 100 per cent modulation is equal to the

d -c power input to the screen, screen resistor,

and plate of the modulated r -f stage.

15 -5 Cathode Modulation

Cathode modulation offers a workable com-

promise between the good plate efficiency but

expensive modulator of high -level plate modu-

lation, and the poor plate efficiency but in-

expensive modulator of grid modulation. Cathode

modulation consists essentially of an ad-

mixture of the two.

The efficiency of the average well- designed

plate -modulated transmitter is in the vicinity

of 75 to 80 per cent, with a compromise per-

haps at 77.5 per cent. On the other hand, the

efficiency of a good grid -modulated transmitter

may run from 28 to maybe 40 per cent, with

the average falling at about 34 per cent. Now

since cathode modulation consists of simul-

taneous grid and plate modulation, in phase

with each other, we can theoretically obtain

any efficiency from about 34 to 77.5 per cent

from our cathode -modulated stage, depending

upon the relative percentages of grid and

plate modulation.

Since the system is a compromise between

the two fundamental modulation arrangements,

a value of efficiency approximately half way

between the two would seem to be the best

compromise. Experience has proved this to be

the case. A compromise efficiency of about

56.5 per cent, roughly half way between the

two limits, has proved to be optimum. Cal-

culation has shown that this value of effi-

ciency can be obtained from a cathode -modu-

lated amplifier when the audio- frequency modu-

lating power is approximately 20 per cent of

the d -c input to the cathode -modulated stage.

An Economical Series cathode modulation is

Series Cathode ideally suited as an economi-

Modulator cal modulating arrangement

for a high -power triode c -w

transmitter. The modulator can be constructed

quite compactly and for a minimum component

cost since no power supply is required for it.

When it is desired to change over from c -w to

'phone, it is only necessary to cut the series

modulator into the cathode return circuit of the

c -w amplifier stage. The plate voltage for the

modulator tubes and for the speech amplifier

is taken from the cathode voltage drop of the

modulated stage across the modulator unit.

Figure 10 shows the circuit of such a modu-

lator, designed to cathode modulate a Class C

amplifier using push -pull 810 tubes, running

at a supply voltage of 2500, and with a plate

input of 660 watts. The modulated stage runs

at about 50% efficiency, giving a power output

of nearly 350 watts, fully modulated. The volt-

age drop across the cathode modulator is 400

volts, allowing a net plate to cathode voltage

of 2100 volts on the final amplifier. The plate

current of the 810's should be about 330 ma.,

and the grid current should be approximately

40 ma., making the total cathode current of the

modulated stage 370 ma. Four parallel 6L6

modulator tubes can pass this amount of plate

current without difficulty. It must be remem-

bered that the voltage drop across the cathode

modulator is also the cathode bias of the modu-

lated stage. In most cases, no extra grid bias

is necessary. If a bias supply is used for c -w

operation, it may be removed for cathode modu-

lation, as shown in figure 11. With low -mu

triodes, some extra grid bias (over and above

that amount supplied by the cathode modulator)

may be needed to achieve proper linearity of

the modulated stage. In any case, proper oper-

ation of a cathode modulated stage should be

determined by examining the modulated output

waveform of the stage on an oscilloscope.

Excitation The r -f driver for a cathode -mod-

ulated stage should have about

www.americanradiohistory.com

296 Amplitude Modulation THE RADIO

6AU6

.002

6AU6 6L6

500K 6L6 6 L6

TO CATHODE

MODULATED

STAGE

6 L6

10 K l W

T °T

ALL RESISTORS 0.5 wArr (INCEST

OTHERWISE NOTED

ALL CAPACITORS IN LIP UNLESS

OTHERWISE NOTED.

CAUTION - FILAMENTS OF OL° rueES MUST SE Al OPERATING

TEMPERATURE BEFORE PLATE VOLTAGE IS APPLIED

TO MODULATED AMPLIFIER.

Figure 10

SERIES CATHODE MODULATOR FOR A HIGH -POWERED TRIODE R -F

AMPLIFIER

the same power output capabilities as would

be required to drive a c -w amplifier to the same

input as it is desired to drive the cathode -

modulated stage. However, some form of exci-

tation control should be available since the

amount of excitation power has a direct bearing

on the linearity of a cathode -modulated am-

plifier stage. If link coupling is used between

the driver and the modulated stage, variation

in the amount of link coupling will afford

apple excitation variation. If much less than

40% plate modulation is employed, the stage

begins to resemble a grid -bias modulated

stage, and the necessity for good r -f regula-

tion will apply.

Cathode Modulation Cathode modulation has

of Tetrodes not proved too satisfac-

tory for use with beam

tetrode tubes. This is a result of the small

excitation and grid swing requirements for

such tubes, plus the fact that some means for

holding the screen voltage at the potential of

the cathode as far as audio is concerned is

usually necessary. Because of these factors,

cathode modulation is not recommended for

use with tetrode r -f amplifiers.

15 -6 The Doherty and the

Terman- Woodyard

Modulated Amplifiers

These two amplifiers will be described to-

gether since they operate upon very similar

principles. Figure 12 shows a greatly simpli-

fied schematic diagram of the operation of both

types. Both systems operate by virtue of a car-

rier tube (V, in both figures 12 and 13) which

supplies the unmodulated carrier, and whose

output is reduced to supply negative peaks,

and a peak tube (V2) whose function is to

supply approximately half the positive peak

of the modulation cycle and whose additional

function is to lower the load impedance on the

carrier tube so that it will be able to supply

the other half of the positive peak of the modu-

lation cycle.

The peak tube is enabled to increase the

output of the carrier tube by virtue of an im-

pedance inverting line between the plate cir-

cuits of the two tubes. This line is designed

to have a characteristic impedance of one -half

the value of load into which the carrier tube

operates under the carrier conditions. Then a

load of one -half the characteristic impedance

of the quarter -wave line is coupled into the

output. By experience with quarter -wave lines

in antenna -matching circuits we know that

such a line will vary the impedance at one

end of the line in such a manner that the geo-

metric mean between the two terminal imped-

ances will be equal to the characteristic im-

pedance of the line. Thus, if we have a value

of load of one -half the characteristic imped-

ance of the line at one end, the other end of

the line will present a value of Juice the char-

acteristic impedance of the lines to the car-

rier tube V,.

This is the situation that exists under the

carrier conditions when the peak tube merely

floats across the load end of the line and con-

tributes no power. Then as a positive peak of

modulation comes along, the peak tube starts

to contribute power to the load until at the

peak of the modulation cycle it is contributing

enough power so that the impedance at the

load end of the line is equal to R, instead of

www.americanradiohistory.com

HANDBOOK Doherty Amplifier 297

R F. AMPLIFIER

BIAS SUPPLY

FOR C. W. PHONE PHONE

MIC BAIA! BAUE -ELE'S

CATHODE

MODULATOR

Figure 11

CATHODE MODULATOR INSTALLATION

SHOWING PHONE -C.W. TRANSFER SWITCH

the R/2 that is presented under the carrier

conditions. This is true because at a positive

modulation peak (since it is delivering full

power) the peak tube subtracts a negative

resistance of R/2 from the load end of the

line. Now, since under the peak condition of modu-

lation the load end of the line is terminated

in R ohms instead of R /2, the impedance at

the carrier -tube will be reduced from 2R ohms

to R ohms. This again is due to the impedance

inverting action of the line. Since the load re-

sistance on the carrier tube has been reduced

to half the carrier value, its output at the peak

of the modulation cycle will be doubled. Thus

we have the necessary condition for a 100

per cent modulation peak; the amplifier will

deliver four times as much power as it does

under the carrier conditions.

On negative modulation peaks the peak tube

does not contribute; the output of the carrier

tube is reduced until on a 100 per cent nega-

tive peak its output is zero.

The Electrical While an electrical quarter -

Quorter -Wave wave line (consisting of a pi

Line network with the inductance

and capacitance units having

a reactance equal to the characteristic imped-

ance of the line) does have the desired im-

pedance- inverting effect, it also has the un-

desirable effect of introducing a 90° phase

shift across such a line. If the shunt elements

are capacitances, the phase shift across the

line lags by 90 °; if they are inductances, the

phase shift leads by 90 °. Since there is an un-

ELECTRICAL 5/4

vi (LINE ZO'R

040

Figure 12

DIAGRAMMATIC REPRESENTATION OF

THE DOHERTY LINEAR

LOAD

desirable phase shift of 90° between the plate

circuits of the carrier and peak tubes, an equal

and opposite phase shift must be introduced in

the exciting voltage to the grid circuits of the

two tubes so that the resultant output in the

plate circuit will be in phase. This additional

phase shift has been indicated in figure 12 and

a method of obtaining it has been shown in

figure 13.

Comparison Between The difference between

Linear and the Doherty linear am-

Grid Modulator plifier and the Terman-

Woodyard grid -modulated

amplifier is the same as the difference between

any linear and grid -modulated stages.Modulated

r.f.is applied to the grid circuit of the Doherty

linear amplifier with the carrier tube biased to

cutoff and the peak tube biased to the point

where it draws substantially zero plate current

at the carrier condition.

In the Terman -Woodyard grid -modulated am-

plifier the carrier tube runs Class C with com-

paratively high bias and high plate efficiency,

while the peak tube again is biased so that it

draws almost no plate current. Unmodulated

r.f. is applied to the grid circuits of the two

tubes and the modulating voltage is inserted

in series with the fixed bias voltages. From

one -half to two -thirds as much audio voltage

is required at the grid of the peak tube as is

required at the grid of the carrier tube.

Operating The resting carrier efficiency of

Efficiencies the grid- modulated amplifier may

run as high as is obtainable in

any Class C stage, 80 per cent or better. The

resting carrier efficiency of the linear will be

about as good as is obtainable in any Class

13 amplifier, 60 to 70 per cent. The overall

efficiency of the bias -modulated amplifier at

100 per cent modulation will run about 75 per

cent; of the linear, about 60 per cent.

In figure 13 the plate tank circuits are de-

tuned enough to give an effect equivalent to

the shunt elements of the quarter -wave "line"

of figure 12. At resonance, the coils L, and

L2 in the grid circuits of the two tubes have

www.americanradiohistory.com

298 Amplitude Modulation THE RADIO

EXCITA-

TION

Q ó r

BIAS V.I-1_

Ci

2

NC

LI

O

T Tc3

a

L30 d

TO

ANT.

Figure 13

SIMPLIFIED SCHEMATIC OF A

"HIGH EFFICIENCY" AMPLIFIER

The basic system, comprising a "carrier"

tube and a "peak" tube interconnected by

lumped -constant quarter -wave lines, is the

some for either grid -bias modulation or for

use as a linear amplifier of a modulated

wave.

each an inductive reactance equal to the ca-

pacitive reactance of the capacitor C1, Thus

we have the effect of a pi network consisting

of shunt inductances and series capacitance.

In the plate circuit we want a phase shift of

the same magnitude but in the opposite direc-

tion; so our series element is the inductance

L3 whose reactance is equal to the character-

istic impedance desired of the network. Then

the plate tank capacitors of the two tubes C2

and C3 are increased an amount past reson-

ance, so that they have a capacitive reactance

equal to the inductive reactance of the coil L3.

It is quite important that there be no coupling

between the inductors.

Although both these types of amplifiers are

highly efficient and require no high -level audio

equipment, they are difficult to adjust- parti-

cularly so on the higher frequencies -and it

would be an extremely difficult problem to de-

sign a multiband transmitter employing the

circuit. However, the grid -bias modulation sys-

tem has advantages for the high -power trans-

mitter which will be operated on a single fre-

quency band.

Other High- Efficiency Many other high- efficien-

Modulation Systems cy modulation systems

have been described since

about 1936. The majority of these, however

have received little application either by com-

mercial interests or by amateurs. In most cases

the circuits are difficult to adjust, or they have

other undesirable features which make their

use impracticable alongside the more conven-

tional modulation systems. Nearly all these

circuits have been published in the 1.R.E.

Proceedings and the interested reader can re-

fer to them in back copies of that journal.

15 -7 Speech Clipping

Speech waveforms are characterized by fre-

quently recurring high -intensity peaks of very

short duration. These peaks will cause over -

modulation if the average level of modulation

on loud syllables exceeds approximately 30

per cent. Careful checking into the nature of

speech sounds has revealed that these high -

intensity peaks are due primarily to the vowel

sounds. Further research has revealed that the

vowel sounds add little to intelligibility, the

major contribution to intelligibility coming

from the consonant sounds such as v, b, k, s,

t, and 1. Measurements have shown that the

power contained in these consonant sounds

may be down 30 db or more from the energy in

the vowel sounds in the same speech passage.

Obviously, then, if we can increase the rel-

ative energy content of the consonant sounds

with respect to the vowel sounds it will be

possible to understand a signal modulated with

such a waveform in the presence of a much

higher level of background noise and inter-

ference. Experiment has shown that it is pos-

sible to accomplish this desirable result sim-

ply by cutting off or clipping the high- intensity

peaks and thus building up in a relative man-

ner the effective level of the weaker sounds.

Such clipping theotetically can be accom-

plished simply by increasing the gain of the

speech amplifier until the average level of

modulation on loud syllables approaches 90

per cent. This is equivalent to increasing the

speech power of the consonant sounds by about

10 times or, conversely, we can say that 10 db

of clipping has been applied to the voice wave.

However, the clipping when accomplished in

this manner will produce higher order side -

bands known as "splatter," and the transmitted

signal would occupy a relatively tremendous

slice of spectrum. So another method of accom-

plishing the desirable effects of clipping must

be employed.

A considerable reduction in the amount of

splatter caused by a moderate increase in the

gain of the speech amplifier can be obtained

by poling the signal from the speech amplifier

to the transmitter such that the high- intensity

peaks occur on upward or positive modulation.

Overloading on positive modulation peaks pro-

duces less splatter than the negative -peak

clipping which occurs with overloading on the

www.americanradiohistory.com

HANDBOOK Speech Clipping 299

Figure 14

SPEECH -WAVEFORM AMPLITUDE

MODULATION

Showing the effect of using the prop-

er polarity of a speech wave for

modulating a transmitter. (A) shows

the effect of proper speech polarity

on a transmitter having an upward

modulation capability of greater

than 100 per cent. (B) shows the

effect of using proper speech polar-

ity on a transmitter having an up-

ward modulation capability of only

100 per cent. Both these conditions

will give a clean signal without

objectionable splatter. (C) shows

the effect of the use of improper

speech polarity. This condition will

cause serious splatter due to nega-

tive -peak clipping in the modulated -

amplifier stage.

1001b NEG MODULATION

_100 %b POS. MODULAT I Q

AVERAGE LEVEL

NEGATIVE

PEAR CLIPPING

100 % NEG. MODULATION

100 % POS. MODULATION

AVERAGE LEVEL

r 100 %b NEG. MODULATION

1

negative peaks of modulation. This aspect of

the problem has been discussed in more detail

in the section on Speech Waveform Dissymmetry

earlier in this chapter. The effect of feeding

the proper speech polarity from the speech am-

plifier is shown in figure 14.

A much more desirable and effective method

of obtaining speech clipping is actually to em-

ploy a clipper circuit in the earlier stages of

the speech amplifier, and then to filter out the

objectionable distortion components by means

of a sharp low -pass filter having a cut -off fre-

quency of approximately 3000 cycles. Tests on

clipper -filter speech systems have shown that

6 db of clipping on voice is just noticeable,

12 db of clipping is quite acceptable, and

values of clipping from 20 to 25 db are toler-

able under such conditions that a high degree

of clipping is necessary to get through heavy

QRM or QRN. A signal with 12 db of clipping

doesn't sound quite natural but it is not un-

pleasant to listen to and is much more read-

able than an unclipped signal in the presence

of strong interference.

The use of a clipper- filter in the speech am-

plifier, to be completely effective, requires

that phase shift between the clipper- filter

stage and the final modulated amplifier be kept

to a minimum. However, if there is phase shift

after the clipper- filter the system does not

completely break down. The presence of phase

shift merely requires that the audio gain fol-

lowing the clipper- filter be reduced to the point

where the cant applied to the clipped speech

waves still cannot cause overmodulation. This

effect is illustrated in figures 15 and 16.

The cant appearing on the tops of the square

waves leaving the clipper -filter centers about

the clipping level. Hence, as the frequency

being passed through the system is lowered,

the amount by which the peak of the canted

wave exceeds the clipping level is increased.

Phase Shift In a normal transmitter having a

Correction moderate amount of phase shift

the cant applied to the tops of

the waves will cause overmodulation on fre-

quencies below those for which the gain fol-

lowing the clipper -filter has been adjusted un-

less remedial steps have been taken. The fol-

lowing steps are advised:

(1) Introduce bass suppression into the speech

amplifier ahead of the clipper- filter.

(2) improve the low- frequency response char-

acteristic insofar as it is possible in the

www.americanradiohistory.com

300 Amplitude Modulation THE RADIO

INCOMING SPEECH WAVE

POSITIVE CLIPPING LEVEL

AVERAGE LEVEL

IttGATIVE ÇLIPPMGJ-41F,L

POSITIVE CLIPPING LEVEL

AVERAGE LEVEL

NEGATIVE CLIPPING LEVEL

CLIPPED AND FILTERED SPEECH WAVE

_100% POSITIVE MODULATION

70% POSITIVEMOOUÇATIQN

AVERAGE LEVEL

70 % NEGATIVE MODULATION

100% NEGATIVE MODULATION

MODULATED WAVE AFTER UNDERGOING PHASE SHIFT

Figure 15

ACTION OF A CLIPPER -FILTER

ON A SPEECH WAVE

The drawing (A) shows the incom-

ing speech wave before it reaches

the clipper stage. (B) shows the

output of the clipper- filter, illus-

trating the manner in which the

peaks are clipped and then the

sharp edges of the clipped wave

removed by the filter. (C) shows

the effect of p hase shift In the

stages following the clipper- filter.

(C) also shows the manner in which

the transmitter may be adjusted for

100 per cent modulation of' the

"canted" peaks of the wave, the

sloping top of the wave reaching

about 70 per cent modulation.

stages following the clipper -filter. Feed-

ing the plate current to the final amplifier

through a choke rather than through the

secondary of the modulation transformer

will help materially.

Even with the normal amount of improvement

which can be attained through the steps men-

tioned above there will still be an amount of

wave cant which must be compensated in some

manner. This compensation can be done in

either of two ways. The first and simpler way

is as follows:

(1) Adjust the speech gain ahead of the clip-

per- filter until with normal talking into

the microphone the distortion being intro-

duced by the clipper -filter circuit is quite

apparent but not objectionable. This amount

of distortion will be apparent to the normal

listener when 10 to 15 db of clipping is

taking place.

( 2) Tune a selective communications receiver

about 15 kc. to one side or the other of the

frequency being transmitted. Use a short

antenna or no antenna at all on the re-

ceiver so that the transmitter is not block-

ing the receiver.

(3) Again with the normal talking into the

microphone adjust the gain following the

clipper -filter to the point where the side -

band splatter is being heard, and then

slightly back off the gain after the clip-

per- filter until the splatter disappears.

If the phase shift in the transmitter or mod-

ulator is not excessive the adjustment proced-

ure given above will allow a clean signal. to be

radiated regardless of any reasonable voice

level being fed into the microphone.

If a cathode -ray oscilloscope is available

the modulated envelope of the transmitter

should be checked with 30 to 70 cycle saw -

tooth waves on the horizontal axis. If the upper

half of the envelope appears in general the

same as the drawing of figure 15C, all is well

and phase -shift is not excessive. However, if

much more slope appears on the tops of the

waves than is illustrated in this figure, it will

be well to apply the second step in compen-

sation in order to insure that sideband splatter

cannot take place and to afford a still higher

average percentage of modulation. This second

step consists of the addition of a high -level

splatter suppressor such as is illustrated in

figure 17.

www.americanradiohistory.com

HANDBOOK Splatter Suppression 301

o

T n

1/ J If i

3000% WAVE

1000% WAVE

3001. WAVE

Figure 16

ILLUSTRATING THE EFFECT OF PHASE

SHIFT AND FILTERED WAVES OF DIF-

FERENT FREQUENCY

Sketch (A) shows the effect of a clipper and

a filter having a cutoff of about 3500 cycles

on o wave of 3000 cycles. Note that no har-

monics ore present in the wave so that phase

shift following the clipper -filter will have

no significant effect on the shape of the

wave. (B) and (C) show the effect of phase

shift on waves well below the cutoff frequen-

cy of the filter. Note that the "cant" placed

upon the top of the wave causes the peak

value to rise higher and higher above the

clipping level as the frequency is lowered.

It is for this reason that bass suppression

before the clipper stage is desirable. Im-

proved low -frequency response following the

clipper -filter will reduce the phase shift and

therefore the canting of the wave at the lower

voice frequencies.

The use of a high -level splatter suppressor

after a clipper -filter system will afford the re-

sult shown in figure 18 since such a device

will not permit the negative -peak clipping

which the wave cant caused by audio -system

phase shift can produce. The high -level splat-

ter suppressor operates by virtue of the fact

that it will not permit the plate voltage on the

modulated amplifier to go completely to zero

regardless of the incoming signal amplitude.

MODULATOR 5R4GY, 1616

836 Cz

z

tit 111

+B MOD.

FIL. TRANS

INSULATED

FOR N.V.

C4 1

TO

PLATE-MODULATED

CLASS -C AMPLIFIER

7500 -10 000 OHMS

LOAD

115 V.A.C. +e R.F. FINAL

Figure 17

HIGH -LEVEL SPLATTER SUPPRESSOR

This circuit is effective in reducing splatter

caused by negative -peak clipping in the mod-

ulated amplifier stage. The use of a two -

section filter as shown is recommended, al-

though either a single m- derived or a con -

stant-k section may be used for greater econ-

omy. Suitable chokes, along with recom-

mended capacitor values, are available from

several manufacturers.

Hence negative -peak clipping with its attend-

ant splatter cannot take place. Such a device

can, of course, also be used in a transmitter

which does not incorporate a clipper- filter sys-

tem. However, the full increase in average

modulation level without serious distortion,

afforded by the clipper- filter system, will not

be obtained.

A word of caution should be noted at this

time in the case of tetrode final modulated

amplifier stages which afford screen voltage

modulation by virtue of a tap or a separate

winding on the modulation transformer such

as is shown in figure 9C of this chapter. If

such a system of modulation is in use, the

high -level splatter suppressor shown in figure

17 will not operate satisfactorily since nega-

tive -peak clipping in the stage can take place

when the screen voltage goes too low.

Clipper Circuits Two effective low -level clip-

per- filter circuits are shown

in figures 19 and 20. The circuit of figure 19

employs a 6J6 double triode as a clipper, each

half of the 6J6 clipping one side of the im-

pressed waveform. The optimum level at which

the clipping operation begins is set by the

value of the cathode resistor. A maximum of

12 to 14 db of clipping may be used with this

circuit, which means that an extra 12 to 14 db

of speech gain must precede the clipper. For

a peak output of 8 volts from the clipper -filter,

a peak audio signal of about 40 volts must be

impressed upon the clipper input circuit. The

6C4 speech amplifier stage must therefore be

considered as a part of the clipper circuit as

www.americanradiohistory.com

302 Amplitude Modulation THE RADIO

100 % POS MODULATION

ZERO AXIS

100 lb NEG. MODULATION

SPLATTER- CAUSING

NEGATIVE OVERMODULATION PEAK

CUT OFF BY "NIGH -LEVEL

SPLATTER SUPPRESSOR"

Figure 18

ACTION OF HIGH -LEVEL

SPLATTER SUPPRESSOR

A high -level splatter suppressor

may be used in a transmitter with-

out a clipper -filter to reduce nega-

tive -peak clipping, or such a unit

may be used following a clipper -

filter to allow a higher average

modulation level by eliminating the

negative -peak clipping which the

wave -cant caused by phase shift

might produce.

it compensates for the 12 to 14 db loss of gain

incurred in the clipping process. A simple low -

pass filter made up of a 20 henry a.c. - d.c

replacement type filter choke and two mica

condensers follows the 616 clipper. This fil-

ter is designed for a cutoff frequency of about

3500 cycles when operating into a load im-

pedance of % megohm. The output level of 8

volts peak is ample to drive a triode speech

amplifier stage, such as a 6C4 or 6J5.

A 6AL5 double diode series clipper is em-

ployed in the circuit of figure 20, and a com-

mercially made low -pass filter is used to give

somewhat better high frequency cutoff char-

acteristics. A double triode is employed as a

speech amplifier ahead of the clipper circuit.

The actual performance of either circuit is

about the same.

To eliminate higher order products that may

be generated in the stages following the clip-

per- filter, it is wise to follow the modulator

with a high -level filter, as shown in figure 21.

Clipper Adjustment These clipper circuits

have two adjustments:

Adjust Gain and Adjust Clipping. The Adj.

Gain control determines the modulation level

of the transmitter. This control should be set

so that over -modulation of the transmitter is

impossible, regardless of the amount of clip-

ping used. Once the Adj. Gain control has

been roughly set, the Adj. Clip. control may

be used to set the modulation level to any per-

centage below 100 %. As the modulation level

is decreased, more and more clipping is intro-

duced into the circuit, until a full 12 db of

clipping is used. This means that the Adj.

Gain control may be advanced some 12 db past

the point where the clipping action started.

Clipping action should start at 85% to 90%

modulation when a sine wave is used for cir-

cuit adjustment purposes.

High -Level Even though we may have cut off

Filters all frequencies above 3000 or 3500

cycles through the use of a filter

system such as is shown in the circuits of fig-

ures 19 and 20, higher frequencies may again

be introduced into the modulated wave by

distortion in stages following the speech am-

plifier. Harmonics of the incoming audio fre-

quencies may be generated in the driver stage

for the modulator; they may be generated in the

plate circuit of the modulator; or they may be

generated by non -linearity in the modulated

amplifier itself.

ATAL

MIC. 4.711

6ÁU6 6C4 6J6 20 N

ADJUST CL/P. jSTANCOR Cilllj ADJUST GAIN

A7oULF SSK AI

5K, K w

Y250 v.

ALL RES /STOPS O.5 WATT UNLESS

OTHERWISE MARKED

ALL CAPACITORS /N {/E UNLESS

OTHERWISE NOTED.

Figure 19

CLIPPER FILTER USING 6J6 DOUBLE TRIODE STAGE

500 11

TO NEXT

GRID

PEAK OUTPUT APPROX.

!V MAX W /TN /Z OE

OF CLIPPING.

www.americanradiohistory.com

HANDBGOK Splatter Suppression 303

4 7n

12AX7 ADJUST GAIN 6AL5 CHICAGO TRANS.

LPF -2 FILTER

SN,IW 00N

ADJUST CLIP.

SeK

TO NEXT

GRID

PEAK OUTPUTAPPROX

5V MAX. WITH 12 DB

OF CLIPPING

ALL RESISTORS 0.5 WATT UNLESS

OTHERWISE MARKED

ALL CAPACITORS IN OF UNLESS

OTHERWISE MARKED.

Figure 20

CLIPPER FILTER USING 6AL5 STAGE

Regardless of the point in the system fol-

lowing the speech amplifier where the high

audio frequencies may be generated, these fre-

quencies can still cause a broad signal to be

transmitted even though all frequencies above

3000 or 3500 cycles have been cut off in the

speech amplifier. The effects of distortion in

the audio system following the speech ampli-

fier can be eliminated quite effectively through

the use of a post- modulator filter. Such a filter

must be used between the modulator plate cir-

cuit and the r -f amplifier which is being mod-

ulated.

This filter may take three general forms in

a normal case of a Class C amplifier plate mod-

ulated by a Class B modulator. The best meth-

od is to use a high level low -pass filter as

CLASS C AMPLIFIER

Figure 21

ADDITIONAL HIGH -LEVEL LOW -PASS FIL-

TER TO FOLLOW MODULATOR WHEN A

LOW -LEVEL CLIPPER FILTER IS USED

Suitable choke, along with recommended ca-

pacitor values, is available from several

manufacturers.

shown in figure 21 and discussed previously.

Another method which will give excellent re-

sults in some cases and poor results in others,

dependent upon the characteristics of the mod-

ulation transformer, is to "build out" the mod-

ulation transformer into a filter section. This

is accomplished as shown in figure 22 by plac-

ing mica capacitors of the correct value across

the primary and secondary of the modulation

transformer. The proper values for the capaci-

CLASS C STAGE

MODULATOR

MODULATION

TRANSFORMER

B+ MOD. BI CLASS C

Figure 22

"BUILDING -OUT" THE MODULATION

TRANSFORMER

This expedient utilizes the leakage react-

ance of the modulation transformer in con-

junction with the capacitors shown to make

up a single- section low -pass filter. In order

to determine exact values for CI and C2 plus

C3, it is necessary to use a measurement

setup such as Is shown in figure 23. How-

ever, experiment has shown in the case of a

number of commercially available modulation

transformers that a value for Cf of 0.002 -µfd.

and C2 plus C3 of 0.004 -µfd. will give satis-

factory results.

www.americanradiohistory.com

304 Amplitude Modulation THE RADIO

AUDIO

OSCILLATOR

Figure 23

TEST SETUP FOR BUILDING -OUT

MODULATION TRANSFORMER

Through the use of a test setup such as is

shown and the method described in the text

it is possible to determine the correct values

for a specified filter characteristic in the

built -out modulation transformer.

tors C1 and C2 must, in the ideal case, be de-

termined by trial and error. Experiment with

a number of modulators has shown, however,

that if a 0.002 pfd. capacitor is used for Cl

and if the sum of C2 and C3 is made 0.004 tad.'

(0.002 pfd. for CZ and 0.002 for C3) the ideal

condition of cutoff above 3000 cycles will be

approached in most cases with the "multiple -

match" type of modulation transformer.

If it is desired to determine the optimum

values of the capacitors across the transformer

this can be determined in several ways, all of

which require the use of a calibrated audio

oscillator. One way is diagrammed in figure 23.

The series resistors R1 and R2 should each be

equal to V2 the value of the recommended plate -

to -plate load resistance for the Class B modu-

lator tubes. Resistor R3 should be equal to the

value of load resistance which the Class C

modulated stage will present to the modulator.

The meter V can be any type of a -c voltmeter.

The indicating instrument on the secondary of

the transformer can be either a cathode -ray

oscilloscope or a high -impedance a -c volt-

meter of the vacuum -tube or rectifier type.

With a set -up as shown in figure 23 a plot of

output voltage against frequency is made, at

all times keeping the voltage across V con-

stant, using various values of capacitance for

C1 and C2 plus C3, When the proper values of

capacitance have been determined which give

substantially constant output up to about 3000

or 3500 cycles and decreasing output at all fre-

quencies above, high -voltage mica capacitors

can be substituted if receiving types were used

in the tests and the transformer connected to

the modulator and Class C amplifier.

With the transformer reconnected in the

transmitter a check of the modulated -wave

output of the transmitter should be made using

an audio oscillator as signal generator and an

oscilloscope coupled to the transmitter output.

With an input signal amplitude fed to the speech

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100 200 300 500 700 1000 2000 3000 5000

FREQUENCY (CPS)

Figure 24

BASE ATTENUATION CHART

Frequency attenuation caused by various

values of coupling capacitor with a grid re-

sistor of 0.5 megohm in the following stage

(Re > RL)

amplifier of such amplitude that limiting does

not take place, a substantially clean sine wave

should be obtained on the carrier of the trans-

mitter at all input frequencies up to the cutoff

frequency of the filter system in the speech

amplifier and of the filter which includes the

modulation transformer. Above these cutoff

frequencies very little modulation of the carrier

wave should be obtained. To obtain a check

on the effectiveness of the "built out" modu-

lation transformer, the capacitors across the

primary and secondary should be removed for

the test. In most cases a marked deterioration

in the waveform output of the modulator will

be noticed with frequencies in the voice range

from 500 to 1500 cycles being fed into the

speech amplifier.

A filter system similar to that shown in fig-

ure 17 may be used between the modulator

and the modulated circuit in a grid -modulated

or screen -modulated transmitter. Lower -voltage

capacitors and low -current chokes may of

course be employed.

Boss Suppression Most of the power repre-

sented by ordinary speech

( particularly the male voice) lies below 1000

cycles. If all frequencies below 400 or 500

cycles are eliminated or substantially atten-

uated, there is a considerable reduction in

power but insignificant reduction in intelligi-

www.americanradiohistory.com

HANDBOOK Bass Suppression 305

bility. This means that the speech level may

be increased considerably without overmodu-

lation or overload of the audio system. In

addition, if speech clipping is used, attenua-

tion of the lower audio frequencies before the

clipper will reduce phase shift and canting of

the clipper output.

A simple method of bass suppression is to

reduce the size of the interstage coupling ca-

pacitors in a resistance coupled amplifier.

Figure 24 shows the frequency characteristics

caused by such a suppression circuit. A sec-

ond simple bass suppression circuit is to

place a small a.c. - d.c. type filter choke from

grid to ground in a speech amplifier stage, as

shown in figure 25.

Modulated Amplifier The systems described

Distortion in the preceding para-

graphs will have no effect

in reducing a broad signal caused by non -

linearity in the modulated amplifier. Even

though the modulating waveform impressed up-

on the modulated stage may be distortion free,

if the modulated amplifier is non -linear dis-

tortion will be generated in the amplifier. The

only way in which this type of distortion may

be corrected is by making the modulated am-

plifier more linear. Degenerative feedback

which includes the modulated amplifier in the

loop will help in this regard.

Plenty of grid excitation and high grid bias

will go a long way toward making a plate -

modulated Class C amplifier linear, although

such operating conditions will make more diffi-

cult the problem of TVI reduction. If this still

does not give adequate linearity, the preced-

ing buffer stage may be modulated 50 per cent

or so at the same time and in the same phase

as the final amplifier. The use of a grid leak

to obtain the majority of the bias for a Class

C stage will improve its linearity.

The linearity of a grid -bias modulated r -f

amplifier can be improved, after proper ad-

justments of excitation, grid bias, and antenna

coupling have been made by modulating the

stage which excites the grid -modulated ampli-

fier. The preceding driver stage may be grid -

bias modulated or it may be plate modulated.

Modulation of the driver stage should be in

the same phase as that of the final modulated

amplifier.

15 -8 The Bias -Shift

Heising Modulator

The simple Class A modulator is limited to

an efficiency of about 30 %, and the tube must

dissipate the full power input during periods

of quiescence. Class AB and class B audio

systems have largely taken the place of the

old Heising modulator because of this great

.01

Figure 25

USE OF PARALLEL INDUCTANCE

FOR BASS SUPPRESSION

waste of power. It is possible, however, to

vary the operating bias of the class A modu-

lator in such a way as to allow class A oper-

ation only when an audio signal is applied to

the grid of the tube. During resting periods,

the bias can be shifted to a higher value,

dropping the resting plate current and plate

dissipation of the tube. When voice waveforms

having 1 o w average power are employed, the

efficiency of the system is comparable to the

popular class B modulator.

The characteristic curve for a class A modu-

lator is shown in figure 26. Normal bias is

used, and the operating point is placed in the

middle of the linear portion of the Eg -Ip curve.

Maximum plate input is limited by the plate

dissipation of the tube under quiescent con-

dition. The bias -shift modulator is biased

close to plate current cut -off under no signal

condition (figure 27). Resting plate current

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RESTING BIAS VOLTAGE

Fìgur° 26

CHARACTERISTIC GRID

VOLTAGE -PLATE CURRENT

CURVE FOR CLASS A

HEISING MODULATOR

+Ec

www.americanradiohistory.com

306 /Inplitude Modulation THE RADIO

EG

+te

PLATE CURRENT

EXCURSION

I P (MAX 5/GNAL)

I P (NO SIGNAL)

+Ec

CUT-OFF

BIAS

BIAS -SNIFT

EXCURSION

_ P

CLASS A OPERATING

BIAS LINE

QUIESCENT

BIAS LINE

Figure 27

BIAS -SHIFT MODULATOR

OPERATING CHARACTERISTICS

Modulator is biased close to plate

current cut -off under no signal,

condition, B. Upon application

of audio signal, the bias of the

stage is shifted toward the class

A operating point, A. Bias -shift

voltage is obtained from audio

signal.

and plate dissipation are therefore quite low.

Upon application of an audio signal, the bias

of the stage is shifted toward the class A

operating point, preventing the negative peaks

of the applied audio voltage from cutting off

the plate current of the tube. As the audio

voltage increases, the operating bias point is

shifted to the right on figure 27 until the class

A operating point is reached at maximum ex-

citation.

The bias -shift voltage may be obtained

directly from the exciting signal by recti-

fication, as shown in figure 28. A simple low

pass filter system is used that will pass only

the syllabic components of speech. Enough

negative bias is applied to the bias -shift modu-

lator to cut the resting plate current to the

desired value, and the output of the bias con-

trol rectifier is polarized so as to "buck"

the fixed bias voltage. No spurious modu-

lation frequencies are generated, since the

modulator operates class A throughout the

audio cycle.

This form of grid pulsing permits the modu-

lator stage to work with an pverall efficiency

of greater than 50 %, comparing favorably with

the class B modulator. The expensive class B

driver and output transformers are not required,

since resistance coupling may be used in the

input circuit of the bias -shift modulator, and

DD- AUDIO

AMPLIFIER

CLA S C

AMPLIFIER

BIAS -SNIFT

MODULATOR ^

Bf

BIAS-SHIFT

REGT FIER

FILTER BIAS -SNIFT

--0 CONTROL

TUBE

NEGATIVE

BIAS

SUPPLY

Figure 28

BLOCK DIAGRAM OF

BIAS -SHIFT MODULATOR

a heavy -duty filter choke will serve as an im-

pedance coupler for the modulated stage.

Series and Parallel The bias -shift system

Control Circuits make take one of several

forms. A "series" control

circuit is shown in figure 29. Resting bias is

applied to the bias -shift modulator tube through

the voltage divider R2 /R4. The bias control

tube is placed across resistor R2. Quiescent

bias for the modulator is set by adjusting R2.

As the internal resistance of the bias control

tube is varied at a syllabic rate the voltage

drop across R2 will vary in unison. The modu-

lator bias, therefore varies at the same rate.

Excitation for the bias control tube is obtained

from the audio signal through potentiometer

RI which regulates the amplitude of the con-

trol signal. The audio signal is rectified by

the bias control rectifier, and filtered by net-

work R3 -Cl in the grid circuit of the bias con-

trol tube.

The "parallel" control system is illustrated

in figure 30. Resting bias for the modulator is

obtained from the voltage divider R2 /R4.

Potentiometer R2 adjusts the resting bias

level, determining the static plate current of

the modulator. Resistor R3 serves as a bias

resistor for the control tube, reducing its plate

current to a low level. When an audio signal

is applied via R1 to the grid of the control

tube the internal resistance is lowered, de-

creasing the shunt resistance across R2. The

negative modulator bias is therefore reduced.

The bias axis of the modulator is shifted from

the cut -off region to a point on the linear

portion of the operating curve. The amount of

bias -shift is controlled by the setting of

potentiometer R1. Capacitor Cl in conjunction

with bias resistor R3 form a syllabic filter for

www.americanradiohistory.com

HANDBOOK Heising Modulator 307

SPEECH

AMPLIFIER

B TO MODULATED

BIAS-SHIFT R-F AMPLIFIER

MODULATOR

ADJUST

OPERAT/N

BIAS

ADJUST

RES T/NG

BIAS

BIAS CONTROL

TUBE

NEGATIVE

MODULATOR BIAS

BIAS CONTROL

RECTIFIER

Figure 29

"SERIES" CONTROL CIRCUIT

FOR BIAS -SHIFT MODULATOR

The internal resistance of the

bias control tube is varied of a

syllabic rate to change the

operating bias of the modulator

tube.

the control bias that is applied to the modu-

lator stage.

A large value of plate dissipation is re-

quired for the bias -shift modulator tube. For

plate voltages below 1500, the 211 (VT -4C)

SPEECH BIAS -SHIFT

AMPLIFIER MODULATOR

B4 TO MODULATED

R -F AMPLIFIER

DJUST

OPERA

TING

BIAS

NEGATIVE FtA

MODULATOR

BIAS

ADJUST REST /NG

81 S 2

BIAS CONTROL

TUBE

Figure 30

"PARALLEL" CONTROL

CIRCUIT FOR BIAS -SHIFT

MODULATOR

The resistance to ground of point

A in the bias network is varied

at a syllabic rate by the bias

control tube.

may be used, while the 304 -TL is suitable for

voltages up to 3000. As with normal class A

amplifiers, low mu tubes function best in this

circuit.

www.americanradiohistory.com

CHAPTER SIXTEEN

Frequency Modulation and

Radioteletype Transmission

Exciter systems for FM and single sideband

transmission are basically similar in that modi-

fication of the signal in accordance with the

intelligence to be transmitted is normally ac-

complished at a relatively low level. Then the

intelligence- bearing signal is amplified to the

desired power level for ultimate transmission.

True, amplifiers for the two types of signals

are basically different; linear amplifiers of the

Class, A or Class B type being used for ssb

signals, while Class C or non -linear Class B

amplifiers may be used for FM amplification.

But the principle of low -level generation and

subsequent amplification is standard for both

types of transmission.

16 -1 Frequency Modulation

The use of frequency modulation and the

allied system of phase modulation has become

of increasing importance in recent years. For

amateur communication frequency and phase

modulation offer important advantages in the

reduction of broadcast and TV interference

and in the elimination of the costly high -level

modulation equipment most commonly employed

with amplitude modulation. For broadcast work

FM offers an improvement in signal -to -noise

ratio for the high field intensities available

in the local -coverage area of FM and TV broad-

cast stations.

In this chapter various points of difference

between FM and amplitude modulation trans-

mission and reception will be discussed and

308

the advantages of FM for certain types of com-

munication pointed out. Since the distinguish-

ing features of the two types of transmission

lie entirely in the modulating circuits at the

transmitter and in the detector and limiter cir-

cuits in the receiver, these parts of the com-

munication system will receive the major por-

tion of attention.

Modulation Modulation is the process of al-

tering a radio wave in accordance

with the intelligence to be transmitted. The

nature of the intelligence is of little impor-

tance as far as the process of modulation is

concerned; it is the method by which this in-

telligence is made to give a distinguishing

characteristic to the radio wave which will

enable the receiver to convert it back into in-

telligence that determines the type of modu-

lation being used.

Figure 1 is a drawing of an r -f carrier am-

plitude modulated by a sine -wave audio volt-

age. After modulation the resultant modulated

r -f wave is seen still to vary about the zero

axis at a constant rate, but the strength of the

individual r -f cycles is proportional to the am-

plitude of the modulation voltage.

In figure 2, the carrier of figure 1 is shown

frequency modulated by the same modulating

voltage. Here it may be seen that modulation

voltage of one polarity causes the carrier fre-

quency to decrease, as shown by the fact that

the individual r -f cycles of the carrier are

spaced farther apart. A modulating voltage of

the opposite polarity causes the frequency to

www.americanradiohistory.com

Frequency Modulation 309

f\,

FIGURE FIGURE 2

AM AND FM WAVES

Figure 1 shows a sketch of the scope pattern of

an amplitude modulated wave at the bottom. The

center sketch shows the modulating wave and the

upper sketch shows the carrier wave.

Figure 2 shows at the bottom a sketch of a fre-

quency modulated wave. In this case the center

sketch also shows the modulating wave and the

Laper sketch shows the carrier wave. Note that

the carrier wave and the modulating wave are the

same in either case, but that the waveform of the

modulated wave is quite different in the two

cases.

J

increase, and this is shown by the r -f cycles

being squeezed together to allow more of them

to be completed in a given time interval.

Figures 1 and 2 reveal two very important

characteristics about amplitude- and frequen-

cy- modulated waves. First, it is seen that

while the amplitude (power) of the signal is

varied in AM transmission, no such variation

takes place in FM. In many cases this advan-

tage of FM is probably of equal or greater im-

portance than the widely publicized noise re-

duction capabilities of the system. When 100

per cent amplitude modulation is obtained, the

average power output of the transmitter must

be increased by 50 per cent. This additional

output must be supplied either by the modu-

lator itself, in the high -level system, or by

operating one or more of the transmitter stages

at such a low output level that they are capa-

ble of producing the additional output without

distortion, in the low -level system. On the

other hand, a frequency -modulated transmitter

requires an insignificant amount of power from

the modulator and needs no provision for in-

creased power output on modulation peaks.

All of the stages between the oscillator and

the antenna may be operated as high- efficiency

Class B or Class C amplifiers or frequency

multipliers.

UNMODULATED CARRIER AMPLITUDE

CARRIER

SIDO FREQUENCY t SIDE FREQUENCY

FREQUENCY

Figure 3

AM SIDE FREQUENCIES

For each AM modulating frequency, a pair of side

frequencies is produced. The side frequencies are

spaced away from the carrier by an amount equal

to the modulation frequency, and their amplitude

is directly proportional to the amplitude of the

modulation. The amplitude of the carrier does not

change under modulation.

Carrier -Wove The second characteristic of FM

Distortion and AM waves revealed by fig-

ures 1 and 2 is that both types

of modulation result in distortion of the r -f

carrier. That is, after modulation, the r -f cy-

cles are no longer sine waves, as they would

be if no frequencies other than the fundamen-

tal carrier frequency were present. It may be

shown in the amplitude modulation case illus-

trated, that there are only two additional fre-

quencies present, and these are the familiar

side /requencies, one located on each side of

the carrier, and each spaced from the carrier

by a frequency interval equal to the modula-

tion frequency. in regard to frequency and am-

plitude, the situation is as shown in figure 3.

The strength of the carrier itself does not vary

during modulation, but the strength of the side

frequencies depends upon the percentage of

modulation. At 100 per cent modulation the

power in the side frequencies is equal to half

that of the carrier.

Under frequency modulation, the carrier wave

again becomes distorted, as shown in figure

2. But, in this case, many more than two addi-

tional frequencies are formed. The first two of

these frequencies are spaced from the carrier

by the modulation frequency, and the additional

side frequencies are located out on each side

of the carrier and are also spaced from each

other by an amount equal to the modulation fre-

quency. Theoretically, there are an infinite

number of side frequencies formed, but, for-

tunately, the strength of those beyond the fre-

quency swing of the transmitter under modula-

tion is relatively low.

One set of side frequencies that might be

formed by frequency modulation is shown in

figure 4. Unlike amplitude modulation, the

www.americanradiohistory.com

310 FM Transmission THE RADIO

UNMODULATED CARRIER AMPLITUDE

CARRIER

SIDE

FREQUENCIES FREQUENCIES

IIli Ill ill

FREQUENCY

Figure 4

FM SIDE FREQUENCIES

With FM each modulation frequency component

causes a large number of side frequencies to be

produced. The side frequencies are separated

from each other and the carrier by an amount equal

to the modulation frequency, but their anplitude

varies greatly as the amount of modulation is

changed. The carrier strength also varies greatly

with frequency modulation. The side frequencies

shown represent a case where the deviation each

side of the "carrier" frequency is equal to five

times the modulating frequency. Other amounts of

deviation with the same modulation frequency

would cause the relative strengths of the various

sidebands to change widely.

strength of the component at the carrier fre-

quency varies widely in FM and it may even

disappear entirely under certain conditions.

The variation of strength of the carrier com-

ponent is useful in measuring the amount of

frequency modulation, and will be discussed

in detail later in this chapter.

One of the great advantages of FM over AM

is the reduction in noise at the receiver which

the system allows. If the receiver is made re-

sponsive only to changes in frequency, a con-

siderable increase in signal -to -noise ratio is

made possible through the use of FM, when

the signal is of greater strength than the noise.

The noise reducing capabilities of FM arise

from the inability of noise to cause appreciable

frequency modulation of the noise -plus -signal

voltage which is applied to the detector in the

receiver.

FM Terms Unlike amplitude modulation, the

term percentage modulation means

little in FM practice, unless the receiver char-

acteristics are specified. There are, however,

three terms, deviation, modulation index, and

deviation ratio, which convey considerable

information concerning the character of the

FM wave.

Deviation is the amount of frequency shift

each side of the unmodulated carrier frequency

which occurs when the transmitter is modu-

lated. Deviation is ordinarily measured in kilo-

cycles, and in a properly operating FM trans-

mitter it will be directly proportional to the

amplitude of the modulating signal. When a

symmetrical modulating signal is applied to

the transmitter, equal deviation each side of

the resting frequency is obtained during each

cycle of the modulating signal, and the total

frequency range covered by the FM transmitter

is sometimes known as the suing. If, for in-

stance, a transmitter operating on 1000 kc.

has its frequency shifted from 1000 kc. to 1010

kc., back to 1000 kc., then to 990 kc., and

again back to 1000 kc. during one cycle of the

modulating wave, the deviation would be 10

kc. and the swing 20 kc.

The modulation index of an FM signal is

the ratio of the deviation to the audio modu-

lating frequency, when both are expressed in

the same units. Thus, in the example above

if the signal is varied from 1000 kc. to 1010

kc. to 990 kc., and back to 1000 kc. at a rate

(frequency) of 2000 times. a second, the mod-

ulation index would be 5, since the deviation

(10 kc.) is 5 times the modulating frequency

(2000 cycles, or 2 kc.).

The relative strengths of the FM carrier and

the various side frequencies depend directly

upon the modulation index, these relative

strengths varying widely as the modulation

index is varied. In the preceding example, for

instance, side frequencies occur on the high

side of 1000 kc. at 1002, 1004, 1006, 1008,

1010, 1012, etc., and on the low frequency

side at 998, 996, 994, 992, 990, 988, etc. In

proportion to the unmodulated carrier strength

(100 per cent), these side frequencies have

the following strengths, as indicated by a

modulation index of 5: 1002 and 998 -33 per

cent, 1004 and 996 -5 per cent, 1006 and 994-

36 per cent, 1008 and 992 -39 per cent, 1010

and 990 -26 per cent, 1012 and 988 -13 per

cent. The carrier strength (1000 kc.) will be

18 per cent of its unmodulated value. Chang-

ing the amplitude of the modulating signal will

change the deviation, and thus the modulation

index will be changed, with the result that the

side frequencies, while still located in the

same places, will have different strength values

from those given above.

The deviation ratio is similar to the modu-

lation index in that it involves the ratio be-

tween a modulating frequency and deviation.

In this case, however, the deviation in ques-

tion is the peak frequency shift obtained under

full modulation, and the audio frequency to be

considered is the maximum audio frequency to

be transmitted. When the maximum audio fre-

quency to be transmitted is 5000 cycles, for

example, a deviation ratio of 3 would call for

a peak deviation of 3 x 5000, or 15 kc. at full

modulation. The noise -suppression capabili-

ties of FM are directly related to the devia-

tion ratio. As the deviation ratio is increased,

www.americanradiohistory.com

HANDBOOK Narrow Band FM 311

the noise suppression becomes better if the

signal is somewhat stronger than the noise.

Where the noise approaches the signal in

strength, however, low deviation ratios allow

communication to be maintained in many cases

where high- deviation -ratio FM and conven-

tional AM are incapable of giving service.

This assumes that a narrow -band FM receiver

is in use. For each value of r -f signal -to -noise

ratio at the receiver, there is a maximum de-

viation ratio which may be used, beyond which

the output audio signal -to -noise ratio de-

creases. Up to this critical deviation ratio,

however, the noise suppression becomes pro-

gressively better as the deviation ratio is in-

creased.

For high- fidelity FM broadcasting purposes,

a deviation ratio of 5 is ordinarily used, the

maximum audio frequency being 15,000 cycles,

and the peak deviation at full modulation be-

ing 75 kc. Since a swing of 150 kc. is covered

by the transmitter, it is obvious that wide -

band FM transmission must necessarily be

confined to the v -h -f range or higher, where

room for the signals is available.

In the case of television sound, the devia-

tion ratio is 1.67; the maximum modulation

frequency is 15,000 cycles, and the trans-

mitter deviation for full modulation is 25 kc.

The sound carrier frequency in a standard TV

signal is located exactly 4.5 Mc. higher than

the picture carrier frequency. In the inter -

carrier TV sound system, which recently has

become quite widely used, this constant differ-

ence between the picture carrier and the sound

carrier is employed within the receiver to ob-

tain an FM sub -carrier at 4.5 Mc. This 4.5

Mc. sub -carrier then is demodulated by the FM

detector to obtain the sound signal which

accompanies the picture.

Narrow -Band Narrow -band FM t r a n s-

FM Transmission mission has become stand-

ardized for use by the mo-

bile services such as police, fire, and taxi-

cab communication, and also on the basis of

a temporary authorization for amateur work in

portions of each of the amateur radiotelephone

bands. A maximum deviation of 15 kc. has

been standardized for the mobile and commer-

cial communication services, while a maxi-

mum deviation of 3 kc. is authorized for ama-

teur NBFM communication.

Bandwidth Re- As the above discussion has

quired by FM indicated, many side frequen-

cies are set up when a radio -

frequency carrier is frequency modulated; theo-

retically, in fact, an infinite number of side

frequencies is formed. Fortunately, however,

the amplitudes of those side frequencies fall-

ing outside the frequency range over which

the transmitter is swung are so small that

most of them may be ignored. In FM trans-

mission, when a complex modulating wave

(speech or music) is used, still additional

side frequencies resulting from a beating to-

gether of the various frequency components

in the modulating wave are formed. This is a

situation that does not occur in amplitude

modulation and it might be thought that the

large number of side frequencies thus formed

might make the frequency spectrum produced

by an FM transmitter prohibitively wide. Analy-

sis shows, however, that the additional side

frequencies are of very small amplitude, and,

instead of increasing the bandwidth, modula-

tion by a complex wave actually reduces the

effective bandwidth of the FM wave. This is

especially true when speech modulation is

used, since most of the power in voiced sounds

is concentrated at low frequencies in the vicin-

ity of 400 cycles.

The bandwidth required in an FM receiver

is a function of a number of factors, both theo-

retical and practical. Basically, the bandwidth

required is a function of the deviation ratio

and the maximum frequency of modulation,

although the practical consideration of drift

and ease of receiver tuning also must be con-

sidered. Shown in figure 5 are the frequency

spectra (carrier and sideband frequencies)

associated with the standard FM broadcast

signal, the TV sound signal, and an amateur -

band narrow -band FM signal with full modula-

tion using the highest permissible modulating

frequency in each case. It will be seen that

for low deviation ratios the receiver band-

width should be at least four times the maxi-

mum frequency deviation, but for a deviation

ratio of 5 the receiver bandwidth need be only

about 2.5 times the maximum frequency de-

viation.

16 -2 Direct FM Circuits

Frequency modulation may be obtained either

by the direct method, in which the frequency

of an oscillator is changed directly by the

modulating signal, or by the indirect method

which makes use of phase modulation. Phase -

modulation circuits will be discussed in sec-

tion 16 -3.

A successful frequency modulated trans-

mitter must meet two requirements: (1) The

frequency deviation must be symmetrical about

a fixed frequency, for symmetrical modulation

voltage. (2) The deviation must be directly

proportional to the amplitude of the modula-

tion, and independent of the modulation fre-

quency. There are several methods of direct

frequency modulation which will fulfill these

www.americanradiohistory.com

312 FM Transmission THE RADIO

OA FM BROADCAST DEVIATION - 75 KC.

MOD. FREQ.-15 KC.

MOD. INDEX - 5

Iy l IS V I!

R

f

} y

n Ñ .11 1! ^

-105 -90 -75 -60 -45 -30 -15 +15 +30 +45 +60 +75 +90 +105

1

O TV SOUND DEVIATION- 25 KC.

MOO. FREQ.- 15 KC.

MOD. INDEX -1.67

ati R II*

,r Ei; fl Vlt fl IN

-4514C -30KC. -15 KC +15 KC 4-30 KC +45 KC.

-6KC. -3KC.

<

DEVIATION - 3 KC.

MOD. EREQ.- 3 KC.

MOD. INDEX - I

CENTER

FREQUENCY

I + 3 KC.

Figure 5

EFFECT OF FM MODULATION INDEX

Showing the side - frequency amplitude and distri-

bution for the three most conrnon modulation indi-

ces used in FM work. The maximum modulating

frequency and maximum deviation are shown in

each case.

requirements. Some of these methods will be

described in the following paragraphs.

Reactance -Tube One of the most practical

Modulators ways of obtaining direct fre-

quency modulation is through

the use of a reactance -tube modulator. In this

arrangement the modulator plate- cathode cir-

cuit is connected across the oscillator tank

circuit, and made to appear as either a capaci-

tive or inductive reactance by exciting the

modulator grid with a voltage which either

leads or lags the oscillator tank voltage by

90 degrees. The leading or lagging grid volt-

age causes a corresponding leading or lagging

plate current, and the plate- cathode circuit

appears as a capacitive or inductive reactance

across the oscillator tank circuit. When the

transconductance of the modulator tube is

varied, by varying one of the element voltages,

the magnitude of the reactance across the os-

AUDIO

IN

1 e C2 47

6BA6

OSCILLATOR IN

1.75 MC. RANGE

R,10K

100K

-T-

00 K- C31.,

POT.

470 K .0068 +150 -200V.

REGULATED

Figure 6

REACTANCE -TUBE MODULATOR

This circuit is convenient for direct frequen-

cy modulation of on oscillator in the 1.75 -Mc.

range. Capacitor C, may be only the input

capacitance of the tube, or a small trimmer

capacitor may be included to permit a varia-

tion in the sensitivity of the reactance tube.

cillator tank is varied. By applying audio mod-

ulating voltage to one of the elements, the

transconductance, and hence the frequency,

may be varied at an audio rate. When properly

designed and operated, the reactance -tube

modulator gives linear frequency modulation,

and is capable of producing large amounts of

deviation.

There are numerous possible configurations

of the reactance -tube modulator circuit. The

difference in the various arrangements lies

principally in the type of phase -shifting cir-

cuit used to give a grid voltage which is in

phase quadrature with the r -f voltage at the

modulator plate.

Figure 6 is a diagram of one of the most

popular forms of reactance -tube modulators.

The modulator tube, which is usually a pen-

tode such as a 6BA6, 6ÁU6, or 6CL6, has its

plate coupled through a blocking capacitor,

C to the "hot" side of the oscillator grid

circuit. Another blocking capacitor, C2, feeds

r.f. to the phase shifting network R -C, in the

modulator grid circuit. If the resistance of R

is made large in comparison with the react-

ance of C, at the oscillator frequency, the cur-

rent through the R -C, combination will be

nearly in phase with the voltage across the

tank circuit, and the voltage across C, will

lag the oscillator tank voltage by almost 90

degrees. The result of the 90- degree lagging

voltage on the modulator grid is that its plate

current lags the tank voltage by 90 degrees,

and the reactance tube appears as an induct-

ance in shunt with the oscillator inductance,

thus raising the oscillator frequency.

The phase- shifting capacitor C, can consist

of the input capacitance of the modulator tube

and stray capacitance between grid and ground.

www.americanradiohistory.com

HANDBOOK Reactance Tube 313

AUDIO

IN +150 -2D0 V.

REGULATED

Figure 7

ALTERNATIVE REACTANCE -TUBE

MODULATOR

This circuit is often preferable for use in the

lower frequency range, although it may be

used at 1.75 Mc. and above if desired. In the

schematic above the reactance tube is shown

connected across the voltage- divider capaci-

tors of a Clapp oscillator, although the modu-

lator circuit may be used with any common

type of oscillator.

However, better control of the operating con-

ditions of the modulator may be had through

the use of a variable capacitor as C,. Resist-

ance R will usually have a value of between

4700 and 100,000 ohms. Either resistance or

transformer coupling may be used to feed audio

voltage to the modulator grid. When a resist-

ance coupling is used, it is necessary to shield

the grid circuit adequately, since the high im-

pedance grid circuit is prone to pick up stray

r -f and low frequency a -c voltage, and cause

undesired frequency modulation.

An alternative reactance modulator circuit

is shown in figure 7. The operating conditions

are generally the same, except that the r -f

excitation voltage to the grid of the reactance

tube is obtained effectively through reversing

the R and C, of figure 6. In this circuit a small

capacitance is used to couple r.f. into the grid

of the reactance tube, with a relatively small

value of resistance from grid to ground. This

circuit has the advantage that the grid of the

tube is at relatively low impedance with re-

spect to r.f. However, the circuit normally is

not suitable for operation above a few mega-

cycles due to the shunting capacitance within

the tube from grid to ground.

Either of the reactance -tube circuits may

be used with any of the common types of os-

cillators. The reactance modulator of figure

6 is shown connected to the high- impedance

point of a conventional hot -cathode Hartley

oscillator, while that of figure 7 is shown con-

nected across the low- impedance capacitors

of a series -tuned Clapp oscillator.

There are several possible variations of the

basic reactance -tube modulator circuits shown

in figures 6 and 7. The audio input may be

applied to the suppressor grid, rather than the

control grid, if desired. Another modification

is to apply the audio to a grid other than the

control grid in a mixer or pentagrid converter

tube which is used as the modulator. Gener-

ally, it will be found that the transconductance

variation per volt of control -element voltage

variation will be greatest when the control

(audio) voltage is applied to the control grid.

In cases where it is desirable to separate com-

pletely the audio and r -f circuits, however,

applying audio voltage to one of the other ele-

ments will often be found advantageous de-

spite the somewhat lower sensitivity.

Adjusting the One of the simplest methods

Phase Shift of adjusting the phase shift to

the correct amount is to place

a pair of earphones in series with the oscilla-

tor cathode -to- ground circuit and adjust the

phase -shift network until minimum sound is

heard in the phones when frequency modula-

tion is taking place. If an electron -coupled or

Hartley oscillator is used, this method re-

quires that the cathode circuit of the oscilla-

tor be inductively or capacitively coupled to

the grid circuit, rather than tapped on the grid

coil. The phones should be adequately by-

passed for r.f. of course.

Stabilization Due to the presence of the react-

ance -tube frequency modulator,

the stabilization of an FM oscillator in regard

to voltage changes is considerably more in-

volved than in the case of a simple self -con-

trolled oscillator for transmitter frequency

control. If desired, the oscillator itself may be

made perfectly stable under voltage changes,

but the presence of the frequency modulator

destroys the beneficial effect of any such

stabilization. It thus becomes desirable to

apply the stabilizing arrangement to the modu-

lator as well as the oscillator. If the oscilla-

tor itself is stable under voltage changes, it

is only necessary to apply voltage- frequency

compensation to the modulator.

Reactance -Tube Two simple reactance -t u b e

Modulators modulators that may be applied

to an existing v.f.o. are illus-

trated in figures 8 and 9. The circuit of figure

8 is extremely simple, yet effective. Only two

tubes are used exclusive of the voltage regu-

lator tubes which perhaps may be already in-

corporated in the v.f.o. A 6AU6 serves as a

high -gain voltage amplifier stage, and a 6CL6

is used as the reactance modulator since its

high value of transconductance will permit a

large value of lagging current to be drawn

under modulation swing. The unit should be

www.americanradiohistory.com

314 FM Transmission THE RADIO

4.7

6AU6 6CL6

68 ULF ( 50 r L U

GRFi D OR CATHODE

RFC

2.814H NOTE: ALL RESISTORS 0.5 WATT UNLESS

OTHERWISE NOTED

ALL CAPACITORS IN 1./F UNLESS

OTHERWISE NOTED

ADJUST FOR CORRECT

VR CURRENT

Figure 8

SIMPLE FM REACTANCE -TUBE MODULATOR

mounted in close proximity to the v.f.o. so that

the lead from the 6CL6 to the grid circuit of

the oscillator can be as short as possible. A

practical solution is to mount the reactance

modulator in a small box on the side of the

v -f -o cabinet.

By incorporating speech clipping in the re-

actance modulator unit, a much more effective

use is made of a given amount of deviation.

When the FM signal is received on an AM re-

ceiver by means of slope detection, the use

of speech clipping will be noticed by the great-

ly increased modulation level of the FM sig-

nal, and the attenuation of the center frequency

null of no modulation. In many cases, it is

difficult to tell a speech -clipped FM signal

from the usual AM signal.

A more complex FM reactance modulator in-

corporating a speech clipper is shown in fig-

ure 9. A 12AX7 double triode speech amplifier

provides enough gain for proper clipper action

when a high level crystal microphone is used.

A double diode 6AL5 speech clipper is used,

the clipping level being set by the potentiome-

ter controlling the plate voltage applied to the

diode. A 6CL6 serves as the reactance modu-

lator.

The reactance modulator may best be ad-

justed by listening to the signal of the v -f -o

exciter at the operating frequency and adjust-

ing the gain and clipping controls for the best

modulation level consistent with minimum side -

band splatter. Minimum clipping occurs when

the Adj. Clip. potentiometer is set for mayimum

voltage on the plates of the 6AL5 clipper tube.

As with the case of all reactance modulators,

a voltage regulated plate supply is required.

Linearity Test It is almost a necessity to run

a static test on the reactance -

tube frequency modulator to determine its line-

arity and effectiveness, since small changes

in the values of components, and in stray ca-

pacitances will almost certainly alter the modu-

lator characteristics. A frequency- versus -con-

trol -voltage curve should be plotted to ascer-

tain that equal increments in control voltage,

both in a positive and a negative direction,

cause equal changes in frequency. If the curve

shows that the modulator has an appreciable

amount of non -linearity, changes in bias, elec-

trode voltages, r -f excitation, and resistance

12AX7 DJUST GAIN 6AL5 6CL6

66LÚ

4.7A CHICAGO TRANS.

LPF -2 FILTER

500 ULF

ri vFO.

TO GRID OR

CAT/IODE OF

5 .01

ADJUST

100A CLIPPING

NOTE'. ALL CAPACITORS IN OF UNLESS

OTHERWISE NOTED

ALL RESISTORS 0.5 WATT UNLESS

OTHERWISE NOTED

.01

Figure 9

FM REACTANCE MODULATOR WITH SPEECH CLIPPER

R FC

2.SMH

ADJUST FOR

CORRECT VR CURRENT

f e+

www.americanradiohistory.com

HANDBOOK Phase Modulation 315

TO MODULATOR

CONTROL ELEMENT

Figure 10

REACTANCE -TUBE LINEARITY CHECKER

values may be made to obtain a straight -line

characteristic.

Figure 10 shows a method of connecting

two 4/2 -volt C batteries and a potentiometer

to plot the characteristic of the modulator. It

will be necessary to use a zero -center volt-

meter to measure the grid voltage, or else re-

verse the voltmeter leads when changing from

positive to negative grid voltage. When a

straight -line characteristic for the modulator

is obtained by the static test method, the ca-

pacitances of the various by -pass capacitors

in the circuit must be kept small to retain this

characteristic when an audio voltage is used

to vary the frequency in place of the d -c volt-

age with which the characteristic was plotted.

16 -3 Phase Modulation

By means of phase modulation (PM) it is

possible to dispense with self -controlled os-

cillators and to obtain directly crystal -con-

trolled FM. In the final analysis, PM is sim-

ply frequency modulation in which the devia-

tion is directly proportional to the modulation

frequency. If an audio signal of 1000 cycles

causes a deviation of % kc., for example, a

2000 -cycle modulating signal of the same am-

plitude will give a deviation of 1 kc., and so

on. To produce an FM signal, it is necessary

to make the deviation independent of the modu-

lation frequency, and proportional only to the

modulating signal. With PM this is done by

including a frequency correcting network in

the transmitter. The audio correction network

must have an attenuation that varies directly

with frequency, and this requirement is easily

met by a very simple resistance- capacity net-

work. The only disadvantage of PM, as compared

to direct FM such as is obtained through the

use of a reactance -tube modulator, is the fact

that very little frequency deviation is pro-

duced directly by the phase modulator. The

deviation produced by a phase modulator is

independent of the actual carrier frequency on

which the modulator operates, but is depend-

ent only upon the phase deviation which is

being produced and upon the modulation fre-

quency. Expressed as an equation:

Fd = MP modulating frequency

Where Fd is the frequency deviation one way

from the mean value of the carrier, and M, is

the phase deviation accompanying modulation

expressed in radians(a radian is approximately

57.3 °). Thus, to take an example, if the phase

deviation is % radian and the modulating fre-

quency is 1000 cycles, the frequency deviation

applied to the carrier being passed through

the phase modulator will be 500 cycles.

It is easy to see that an enormous amount

of multiplication of the carrier frequency is

required in order to obtain from a phase modu-

lator the frequency deviation of 75 kc. required

for commercial FM broadcasting. However, for

amateur and commercial narrow -band FM work

(NBFM) only a quite reasonable number of

multiplier stages are required to obtain a de-

viation ratio of approximately one. Actually,

phase modulation of approximately one -half

radian on the output of a crystal oscillator in

the 80 -meter band will give adequate deviation

for 29 -Mc. NBFM radiotelephony. For example;

if the crystal frequency is 3700 kc., the de-

viation in phase produced is t/ radian, and the

modulating frequency is 500 cycles, the devia-

tion in the 80 -meter band will be 250 cycles.

But when the crystal frequency is multiplied

on up to 29,600 kc. the frequency deviation

will also be multiplied by 8 so that the result-

ing deviation on the 10 -meter band will be 2 kc.

either side of the carrier for a total swing in

carrier frequency of 4 kc. This amount of de-

viation is quite adequate for NBFM work.

Odd -harmonic distortion is produced when

FM is obtained by the phase- modulation meth-

od, and the amount of this distortion that can

be tolerated is the limiting factor in determin-

ing the amount of PM that can be used. Since

the aforementioned frequency- correcting net-

work causes the lowest modulating frequency

to have the greatest amplitude, maximum phase

modulation takes place at the lowest modu-

lating frequency, and the amount of distortion

that can be tolerated at this frequency deter-

mines the maximum deviation that can be ob-

tained by the PM method. For high -fidelity

broadcasting, the deviation produced by PM is

limited to an amount equal to about one -third

of the lowest modulating frequency. But for

NBFM work the deviation may be as high as

0.6 of the modulating frequency before distor-

tion becomes objectionable on voice modula-

tion. In other terms this means that phase de-

viations as high as 0.6 radian may be used for

amateur and commercial NBFM transmission.

www.americanradiohistory.com

316 FM Transmission THE RADIO

REACTANCE

TUBE

CRYSTAL

OSCILLATOR

TUBE r -- y NE %T

STAGE

LOW-C

AUDIO

Figure 11

REACTANCE -TUBE MODULATION OF

CRYSTAL OSCILLATOR STAGE

Phase -Modulation A simple reactance modula -

Circuits cor normally used for FM

may also be used for PM by

connecting it to the plate circuit of a crystal

oscillator stage as shown in figure 11.

Another PM circuit, suitable for operation

on 20, 15 and 10 meters with the use of 80

meter crystals is shown in figure 12. A double

triode 12AX7 is used as a combination Pierce

crystal oscillator and phase modulator. C,

should not be thought of as a neutralizing con-

denser, but rather as an adjustment for the

phase of the r -f voltage acting between the

grid and plate of the 12AX7 phase modulator.

C2 acts as a phase angle and magnitude con-

trol, and both these condensers should be ad-

justed for maximum phase modulation capabili-

ties of the circuit. Resonance of the circuit is

established by the iron slug of coil L, -L,. A

6CL6 is used as a doubler to 7 Mc. and de-

livers approximately 2 watts on this band. Ad-

ditional doubler stages may be added after the

6CL6 stage to reach the desired band of opera-

tion.

Still another PM circuit, which is quite wide-

ly used commercially, is shown in figure 13.

In this circuit L and C are made resonant at a

frequency which is 0.707 times the operating

frequency. Hence at the operating frequency

the inductive reactance is twice the capacitive

reactance. A cathode follower tube acts as a

variable resistance in series with the L and

C which go to make up the tank circuit. The

operating point of the cathode follower should

be chosen so that the effective resistance in

series with the tank circuit (made up of the

resistance of the cathode- follower tube in par-

allel with the cathode bias resistor of the cath-

ode follower) is equal to the capacitive react-

ance of the tank capacitor at the operating fre-

quency. The circuit is capable of about plus

or minus % radian deviation with tolerable dis-

tortion.

Measurement When a single- frequency mod -

of Deviation ulating voltage is used with an

FM transmitter, the relative

amplitudes of the various sidebands and the

carrier vary widely as the deviation is varied

by increasing or decreasing the amount of mod-

ulation. Since the relationship between the

amplitudes of the various sidebands and car-

rier to the audio modulating frequency and the

deviation is known, a simple method of meas-

uring the deviation of a frequency modulated

transmitter is possible. In making the meas-

urement, the result is given in the form of the

modulation index for a certain amount of audio

input. As previously described, the modulation

index is the ratio of the peak frequency devia-

tion to the frequency of the audio modulation.

The measurement is made by applying a

sine -wave audio voltage of known frequency

to the transmitter, and increasing the modula-

tion until the amplitude of the carrier compo-

nent of the frequency modulated wave reaches

zero. The modulation index for zero carrier

may then be determined from the table below.

As may be seen from the table, the first point

of zero carrier is obtained when the modulation

index has a value of 2.405, -in other words,

when the deviation is 2.405 times the modula-

tion frequency. For example, if a modulation

frequency of 1000 cycles is used, and the

modulation is increased until the first carrier

null is obtained, the deviation will then be

2.405 times the modulation frequency, or 2.405

kc. If the modulating frequency happened to be

2000 cycles, the deviation at the first null

would be 4.810 kc. Other carrier nulls will be

obtained when the index is 5.52, 8.654, and at

increasing values separated approximately by

rr. The following is a listing of the modulation

index at successive carrier nulls up to the

tenth: Zero carrier

point no. Modulation

index

1 2.405

2 5.520

3 8.654

4 11.792

5 14.931

6 18.071

7 21.212

8 24353

9 27.494

10 30.635

The only equipment required for making the

measurements is a calibrated audio oscillator

of good wave form, and a communication re-

ceiver equipped with a beat oscillator and

crystal filter. The receiver should be used

with its crystal filter set for minimum band-

width to exclude sidebands spaced from the

carrier by the modulation frequency. The un-

www.americanradiohistory.com

HANDBOOK FM Reception 317

12AX7

250 100F

6C L6

TO DOUBLER STAGES

B+

300 V.

L I -SET RSl E. }SPACED 71 APART ON FORM

L 2-Is T. 36E. r POWDERED IRON COR

L3 -377'. H20E. CLOSE-SPACED I O/A.

NOTE. ALL RESISTORS 0.5 WA rr UNLESS

OTHERWISE NOTED

ALL CAPACITORS IN 4/F UNLESS

OTHERWISE NOTED

Figure 12

REACTANCE MODULATOR FOR 10, 15 AND 20 METER OPERATION

modulated carrier is accurately tuned in on the

receiver with the beat oscillator operating.

Then modulation from the audio oscillator is

applied to the transmitter, and the modulation

is increased until the first carrier null is ob-

tained. This carrier null will correspond to a

modulation index of 2.405, as previously men-

tioned. Successive null points will correspond

to the indices listed in the table.

A volume indicator in the transmitter audio

system may be used to measure the audio level

required for different amounts of deviation,

and the indicator thus calibrated in terms of

frequency deviation. If the measurements are

made at the fundamental frequency of the os-

cillator, it will be necessary to multiply the

frequency deviation by the harmonic upon which

the transmitter is operating, of course. It will

probably be most convenient to make the deter-

mination at some frequency intermediate be-

tween that of the oscillator and that at which

the transmitter is operating, and then to mul-

tiply the result by the frequency multiplication

between that frequency and the transmitter

output frequency.

16 -4 Reception of FM

Signals

A conventional communications receiver may

be used to receive narrow -band FM transmis-

sions, although performance will be much poor-

er than can be obtained with an NBFM receiver

or adapter. However, a receiver specifically

designed for FM reception must be used when

it is desired to receive high deviation FM such

as used by FM broadcast stations, TV sound,

and mobile communications FM.

The FM receiver must have, first of all, a

bandwidth sufficient to pass the range of fre-

quencies generated by the FM transmitter. And

since the receiver must be a superheterodyne

if it is to have good sensitivity at the frequen-

cies to which FM is restricted, i -f bandwidth

is an important factor in its design.

The second requirement of the FM receiver

is that it incorporate some sort of device for

converting frequency changes into amplitude

changes, in other words, a detector operating

on frequency variations rather than amplitude

variations. The third requirement, and one which

is necessary if the full noise reducing capa-

R F. INPUT

fo

65.17

01

XL ABOUT 1500n AT fo

XC ABOUT T50 R. AT f0

PHASE- MODULATED

OUTPUT

+B 200 V.

AUDIO IN

Figure 13

CATHODE -FOLLOWER PHASE

MODULATOR

The phase modulator illustrated above is

quite satisfactory when the stage is to be

operated on a single frequency or over a nar-

row range of frequencies.

www.americanradiohistory.com

318 FM Transmission THE RADIO

MIXE

T I. r.

AMPLIFIER

OSCILLATOR

LIMITER FREQUENCY AUDIO

DETECTOR AMP.

(DISCRIMINATOR)

Figure 14

FM RECEIVER BLOCK DIAGRAM

Up to the amplitude limiter stage, the FM

receiver is similar to an AM receiver, except

for a somewhat wider i -f bandwidth. The lim-

iter removes any amplitude modulation, and

the frequency detector following the limiter

converts frequency variations into amplitude

variations.

F R E Q U E N C Y

Figure 15

SLOPE DETECTION OF FM SIGNAL

One side of the response characteristic of a

tuned circuit or of on i -f amplifier may be

used as shown to convert frequency varia-

tions of an incoming signal into amplitude

variations.

bilities of the FM system of transmission are

desired, is a limiting device to eliminate am-

plitude variations before they reach the de-

tector. A block diagram of the essential parts

of an FM receiver is shown in figure 14.

The Frequency The simplest device for con -

Detector venting frequency variations

to amplitude variations is an

"off- tune" resonant circuit, as illustrated in

figure 15. With the carrier tuned in at point

"A," a certain amount of r -f voltage will be

developed across the tuned circuit, and, as

the frequency is varied either side of this fre-

quency by the modulation, the r -f voltage will

increase and decrease to points "C" and "B"

in accordance with the modulation. If the volt-

age across the tuned circuit is applied to an

ordinary detector, the detector output will vary

in accordance with the modulation, the ampli-

tude of the variation being proportional to the

deviation of the signal, and the rate being

equal to the modulation frequency. It is obvious

from figure 15 that only a small portion of the

resonance curve is usable for linear conversion

Figure 16

TRAVIS DISCRIMINATOR

This type of discriminator makes use of two

off -tuned resonant circuits coupled to a sin-

gle primary winding. The circuit is capable

of excellent linearity, but is difficult to a-

lign.

of frequency variations into amplitude varia-

tions, since the linear portion of the curve is

rather short. Any frequency variation which

exceeds the linear portion will cause distor-

tion of the recovered audio. It is also obvious

by inspection of figure 15 that an AM receiver

used in this manner is wide open to signals

on the peak of the resonance curve and also

to signals on the other side of the resonance

curve. Further, no noise limiting action is af-

forded by this type of reception. This system,

therefore, is not recommended for FM recep-

tion, although widely used by amateurs for

occasional NBFA1 reception.

Travis Discriminator Another form of frequen-

cy detector or discrimi-

nator, is shown in figure 16. In this arrange-

ment two tuned circuits are used, one tuned

on each side of the i -f amplifier frequency,

and with their resonant frequencies spaced

slightly more than the expected transmitter

swing. Their outputs are combined in a differ-

ential rectifier so that the voltage across the

series load resistors, R, and R2, is equal to

the algebraic sum of the individual output

voltages of each rectifier. When a signal at the

At its "center" fre-

quency t he discrimi-

n

discrimi-

nator produces zero

output voltage. On

either side of this

frequency It gives

a voltage of a polar-

ity and magnitude

which depend on the

direction and amount

of frequency shift. Figure

DISCRIMINATOR VOLTAGE -FREQUENCY

CURVE

17

FREQUENCY

www.americanradiohistory.com

HANDBOOK FM Reception 319

Figure 18

FOSTER -SEELEY DISCRIMINATOR

This discriminator is the most widely used

circuit since it is capable of excellent lin-

earity and is relatively simple to align when

proper test equipment is available.

i -f mid -frequency is received, the voltages

across the load resistors are equal and oppo-

site, and the sum voltage is zero. As the r -f

signal varies from the mid -frequency, however,

these individual voltages become unequal, and

a voltage having the polarity of the larger volt-

age and equal to the difference between the

two voltages appears across the series resis-

tors, and is applied to the audio amplifier.

The relationship between frequency and dis-

criminator output voltage is shown in figure

17. The separation of the discriminator peaks

and the linearity of the output voltage vs. fre-

quency curve depend upon the discriminator

frequency, the Q of the tuned circuits, and the

value of the diode load resistors. As the inter-

mediate (and discriminator) frequency is in-

creased, the peaks must be separated further to

secure good linearity and output. Within limits,

as the diode load resistance or the Q is re-

duced, the linearity improves, and the separa-

tion between the peaks must be greater.

Foster -Seeley The most widely used form of

Discriminator discriminator is that shown in

figure 18. This type of discrimi-

nator yields an output- voltage- versus -frequen-

cy characteristic similar to that shown in fig-

ure 19. Here, again, the output voltage is equal

to the algebraic sum of the voltages developed

across the load resistors of the two diodes,

the resistors being connected in series to

ground. However, this Foster-Seeley discrim-

inator requires only two tuned circuits instead

of the three used in the previous discriminator.

The operation of the circuit results from the

phase relationships existing in a transformer

having a tuned secondary. In effect, as a close

examination of the circuit will reveal, the pri-

mary circuit is in series, for r.f., with each

half of the secondary to ground. When the re-

ceived signal is at the resonant frequency of

the secondary, the r -f voltage across the sec-

ondary is 90 degrees out of phase with that

across the primary. Since each diode is con-

nected across one half of the secondary wind-

SECONDARY VOLTAGE

Figure 19

DISCRIMINATOR VECTOR DIAGRAM

A signal at the resonant frequency of the

secondary will cause the secondary voltage

to be 90 degrees out of phase with the pri-

mary voltage, as shown at A, and the result-

ant voltages R and R' are equal. If the sig-

nal frequency changes, the phase relation-

ship also changes, and the resultant voltages

are no longer equal, as shown at B. A differ -

ential rectifier is used to give an output volt-

age proportional to the difference between

R and R'.

ing and the primary winding in series, the re-

sultant r -f voltages applied to each are equal,

and the voltages developed across each diode

load resistor are equal and of opposite polar-

ity. Hence, the net voltage between the top of

the load resistors and ground is zero. This is

shown vectorially in figure 19A where the re-

sultant voltages R and R which are applied

to the two diodes are shown to be equal when

the phase angle between primary and second-

ary voltages is 90 degrees. If, however, the

signal varies from the resonant frequency, the

90- degree phase relationship no longer exists

between primary and secondary. The result of

this effect is shown in figure 1913 where the

secondary r -f voltage is no longer 90 degrees

out of phase with respect to the primary volt-

age. The resultant voltages applied to the two

diodes are now no longer equal, and a d -c

voltage proportional to the difference between

the r -f voltages applied to the two diodes will

exist across the series load resistors. As the

signal frequency varies back and forth across

the resonant frequency of the discriminator,

an a -c voltage of the same frequency as the

original modulation, and proportional to the

deviation, is developed and passed on to the

audio amplifier.

Ratio One of the more recent types of FM

Detector detector circuits, called the ratio

detector is diagrammed in figure 20.

The input transformer can be designed so that

the parallel input voltage to the diodes can be

taken from a tap on the primary of the trans-

www.americanradiohistory.com

320 FM Transmission THE RADIO

.0001 RFC

A.F. OUTPUT

Figure 20

RATIO DETECTOR CIRCUIT

The parallel voltage to the diodes in a ratio

detector may be obtained from a tap on the

primary winding of the transformer or from

o third winding. Note that one of the diodes

is reversed from the system used with the

Foster -Seeley discriminator, and that the

output circuit is completely different. The

ratio detector does not have to be preceded

by o limiter, but is more difficult to align for

distortion -free output than the conventional

discriminator.

former, or this voltage may be obtained from a

tertiary winding coupled to the primary. The

r -f choke used must have high impedance at

the intermediate frequency used in the receiver,

although this choke is not needed if the trans-

former has a tertiary winding.

The circuit of the ratio detector appears

very similar to that of the more conventional

discriminator arrangement. However, it will be

noted that the two diodes in the ratio detector

are poled so that their d -c output voltages add,

as contrasted to the Foster -Seeley circuit

wherein the diodes are poled so that the d -c

output voltages buck each other. At the center

frequency to which the discriminator trans-

former is tuned the voltage appearing at the

top of the 1- megohm potentiometer will be one -

half the d -c voltage appearing at the a -v -c out-

put terminal -since the contribution of each

diode will be the same. However, as the input

frequency varies to one side or the other of

the tuned value (while remaining within the

pass band of the i -f amplifier feeding the de-

tector) the relative contributions of the two

diodes will be different. The voltage appearing

at the top of the 1- megohm volume control will

increase for frequency deviations in one direc-

tion and will decrease for frequency deviations

in the other direction from the mean or tuned

value of the transformer. The audio output volt-

age is equal to the ratio of the relative contri-

butions of the two diodes, hence the name

ratio detector.

The ratio detector offers several advantages

over the simple discriminator circuit. The cir-

cuit does not require the use of a limiter pre-

ceding the detector since the circuit is inher-

ently insensitive to amplitude modulation on

6SJ 7

+250

TO

015C R I M-

INATOR

Figure 21

LIMITER CIRCUIT

One, or sometimes two, limiter stages nor -

mally precede the discriminator so that a con-

stant signal level will be fed to the FM de-

tector. This procedure eliminates amplitude

variations in the signal fed to the discrimi-

nator, so that it will respond only to frequen-

cy changes.

an incoming signal. This factor alone means

that the r -f and i -f gain ahead of the detector

can be much less than the conventional dis-

criminator for the same overall sensitivity.

Further, the circuit provides a -v -c voltage for

controlling the gain of the preceding r -f and

i -f stages. The ratio detector is, however, sus-

ceptible to variations in the amplitude of the

incoming signal as is any other detector cir-

cuit except the discriminator with a limiter

preceding it, so that a -v -c should be used on

the stages preceding the detector.

Limiters The limiter of an FM receiver using

a conventional discriminator serves

to remove amplitude modulation and pass on

to the discriminator a frequency modulated

signal of constant amplitude; a typical circuit

is shown in figure 21. The limiter tube is oper-

ated as an i -f stage with very low plate volt-

age and with grid leak bias, so that it over-

loads quite easily. Up to a certain point the

output of the limiter will increase with an in-

crease in signal. Above this point, however,

the limiter becomes overloaded, and further

large increases in signal will not give any in-

crease in output. To operate successfully, the

limiter must be supplied with a large amount

of signal, so that the amplitude of its output

will not change for rather wide variations in

amplitude of the signal. Noise, which causes

little frequency modulation but much ampli-

tude modulation of the received signal, is vir-

tually wiped out in the limiter.

The voltage across the grid resistor varies

with the amplitude of the received signal. For

this reason, conventional amplitude modulated

signals may be received on the FM receiver

by connecting the input of the audio amplifier

to the top of this resistor, rather than to the

discriminator output. When properly filtered.

www.americanradiohistory.com

HANDBOOK NBFM Adapter 321

by a simple R -C circuit, the voltage across

the grid resistor may also be used as a -v -c

voltage for the receiver. When the limiter is

operating properly, a.v.c. is neither necessary

nor desirable, however, for FM reception alone.

Receiver Design One of the most important

Considerations factors in the design of an

FM receiver is the frequency

swing which it is intended to handle. It will

be apparent from figure 17 that if the straight

portion of the discriminator circuit covers a

wider range of frequencies than those gener-

ated by the transmitter, the audio output will

be reduced from the maximum value of which

the receiver is capable.

In this respect, the term "modulation per-

centage" is more applicable to the FM receiver

than it is to the transmitter, since the modula-

tion capability of the communication system

is limited by the receiver bandwidth and the

discriminator characteristic; full utilization of

the linear portion of the characteristic amounts,

in effect, to 100 per cent modulation. This

means that some sort of standard must be agreed

upon, for any particular type of communication,

to make it unnecessary to vary the transmitter

swing to accommodate different receivers.

Two considerations influence the receiver

bandwidth necessary for any particular type of

communication. These are the maximum audio

frequency which the system will handle, and

the deviation ratio which will be employed.

For voice communication, the maximum audio

frequency is more or less fixed at 3000 to 4000

cycles. In the matter of deviation ratio, how-

ever, the amount of noise suppression which

the FM system will provide is influenced by

the ratio chosen, since the improvement in

signal -to -noise ratio which the FM system

shows over amplitude modulation is equiva-

lent to a constant multiplied by the deviation

ratio. This assumes that the signal is some-

what stronger than the noise at the receiver,

however, as the advantages of wideband FM

in regard to noise suppression disappear when

the signal -to -noise ratio approaches unity.

On the other hand, a low deviation ratio is

more satisfactory for strictly communication

work, where readability at low signal -to -noise

ratios is more important than additional noise

suppression when the signal is already appre-

ciably stronger than the noise.

As mentioned previously, broadcast FM prac-

tice is to use a deviation ratio of 5. When this

ratio is applied to a voice -communication sys-

tem, the total swing becomes 30 to 40 kc. With

lower deviation ratios, such as are most fre-

quently used for voice work, the swing becomes

proportionally less, until at a deviation ratio

of 1 the swing is equal to twice the highest

audio frequency. Actually, however, the re-

FROM

DISCRIMINATOR f C T

TO AUDIO GRID

ME'.

L

R' 220 , C. 340 uuF

R2100 K, C' 730 uuF

R'47It, C' IE00uur.

R' 22 R, C' 3400 SWF.

Figure 22

75- MICROSECOND DE- EMPHASIS

CIRCUITS

The audio signal transmitted by FM and TV

stations has received high -frequency pre-

emphasis, so that a de- emphasis c i r c u i t

should be included between the output of the

FM detector and the input of the audio

system.

ceiver bandwidth must be slightly greater than

the expected transmitter swing, since for dis-

tortionless reception the receiver must pass

the complete band of energy generated by the

transmitter, and this band will always cover a

range somewhat wider than the transmitter

swing.

Pre -Emphasis Standards in FM broadcast

and De- Emphasis and TV sound work call for

the pre- emphasis of all au-

dio modulating frequencies above about 2000

cycles, with a rising slope such as would be

produced by a 75- microsecond RL network.

Thus the FM receiver should include a com-

pensating de- emphasis RC network with a time

constant of 75 microseconds so that the over-

all frequency response from microphone to

loudspeaker will approach linearity. The use

of pre- emphasis and de- emphasis in this man-

ner results in a considerable improvement in

the overall signal -to -noise ratio of an FM sys-

tem. Appropriate values for the de- emphasis

network, for different values of circuit imped-

ance are given in figure 22.

A NBFM 455 -kc. The unit diagrammed in figure

Adapter Unit 23 is designed to provide

NBFM reception when attached

to any communication receiver having a 455 -kc.

i -f amplifier. Although NBFM can be received

on an AM receiver by tuning the receiver to

one side or the other of the incoming signal,

a tremendous improvement in signal -to -noise

ratio and in signal to amplitude ratio will be

obtained by the use of a true FM detector sys-

tem. The adapter uses two tubes. A 6AU6 is

used as a limiter, and a 6AL5 as a discrimi-

nator. The audio level is approximately 10

www.americanradiohistory.com

322 FM Transmission Radio Teletype

ro /JUT

6AÚ6 6AL5

SODUF IOOK loo AUDIO

UL OUT

100K

405 KC.

I.F. IN

220K

0.1 2 W T I -J.W. MILLEN 0i2-C3

100

LUF

1D0(,

VOLT-

METER

e+ 250 V

AT 3 MA.

NOTE: ALL CAPACITORS IN /JP UNLESS OTHERWISE NOTED

ALL RESISTORS 0.5 WATT UNLESS OTHERWISE NOTED

Figure 23

NBFM ADAPTER FOR 455 -KC. I -F SYSTEM

volts peak for the maximum deviation which

can be handled by a conventional 455 -kc. i -f

system. The unit may be tuned by placing a

high resistance d -c voltmeter across R, and

tuning the trimmers of the i -f transformer for

maximum voltage when an unmodulated signal

is injected into the i -f strip of the receiver.

The voltmeter should next be connected across

the audio output terminal of the discriminator.

The receiver is now tuned back and forth a-

cross the frequency of the incoming signal,

and the movement of the voltmeter noted. When

the receiver is exactly tuned on the signal the

voltmeter reading should be zero. When the re-

ceiver is tuned to one side of center, the volt-

meter reading should increase to a maximum

value and then decrease gradually to zero as

the signal is tuned out of the passband of the

receiver. When the receiver is tuned to the

other side of the signal the voltmeter should

increase to the same maximum value but in

the opposite direction or polarity, and then

fall to zero as the signal is tuned out of the

passband. It may be necessary to make small

adjustments to C, and C, to make the volt-

meter read zero when the signal is tuned in

the center of the passband.

16 -5 Radio Teletype

The teletype machine is an electric type-

writer that is stimulated by d.c. pulses origi-

nated by the action of a second machine. The

pulses may be transmitted from one machine

to another by wire, or by a radio signal. When

radio transmission is used, the system is

termed radio teletype (RTTY).

The d.c. pulses that comprise the teletype

signal may be converted into three basic types

of emission suitable for radio transmission.

These are: 1- Frequency shift keying (FSK),

designated as F1 emission; 2- Make -break

keying (MBK),designated as Al emission. and;

3- Audio frequency shift keying (AFSK),

designated as F2 emission.

Frequency shift keying is obtained by vary-

ing the transmitted frequency of the radio

signal a fixed amount (usually 850 cycles)

during the keying process. The shift is ac-

complished in discrete intervals designated

mark and space. Both types of intervals convey

information to the teletype printer. Make -break

keying is analogous to simple c -w transmission

in that the radio carrier conveys information

by changing from an off to an on condition.

Early RTTY circuits employed MBK equipment,

which is rapidly becoming obsolete since it

is inferior to the frequency shift system.

Audio frequency shift keying employs a

steady radio carrier modulated by an audio

tone that is shifted in frequency according to

the RTTY pulses. Other forms of information

transmission may be employed by a RTTY

system which also encompass the translation

of RTTY pulses into r -f signals.

Teletype The RTTY code consists of

Coding the 26 letters of the alphabet,

the space, the line feed, the

carriage return, the bell, the upper case shift,

and the lower case shift; making a total of 32

coded groups. Numerals, punctuation, and

symbols may be taken care of in the case shift,

since all transmitted letters are capitals.

The FSK system normally employs the higher

radio frequency as the mark, and the lower

frequency as the space. This relationship

holds true in the AFSK system also. The lower

audio frequency (mark) is normally 2125 cycles

and the higher audio tone (space) is 2975

cycles, giving a frequency difference of 850

cycles.

The Teletype A simple FSK teletype system

System may be added to any c -w trans-

mitter. The teletype keyboard

prints the keyed letters on a tape, and at the

same time generates the electrical code group

that describes the letter. The d.c. pulses are

impressed upon a distributor unit which ar-

ranges the typing and spacing pulses in proper

sequence. The resulting series of impulses

are applied to the transmitter frequency con-

trol device, which may be a reactance modu-

lator, actuated by a polar relay.

The received signal is hetrodyned against

a beat oscillator to provide the two audio tones

which are limited in amplitude and passed

through audio filters to separate them. Recti-

fication of the tones permits operation of a

polar relay which can provide d.c. pulses

suitable for operation of the tele- typewriter.

www.americanradiohistory.com

CHAPTER SEVENTEEN

Sideband Transmission

While single- sideband transmission (SSB)

has attracted significant interest on amateur

frequencies only in the past few years, the prin-

ciples have been recognized and put to use in

various commercial applications for many years.

Expansion of single -sideband for both com-

mercial and amateur communication has await-

ed the development of economical components

possessing the required characteristics (such as

sharp cutoff filters and high stability crystals)

demanded by SSB techniques. The availability

of such components and precision test equip-

ment now makes possible the economical test-

ing, adjustment and use of SSB equipment on

a wider scale than before. Many of the seem-

ingly insurmountable obstacles of past years

no longer prevent the amateur from achieving

the advantages of SSB for his class of oper-

ation.

17 -1 Commercial

Applications of SSB

Before discussion of amateur SSB equipment,

it is helpful to review some of the commercial

applications of SSB in an effort to avoid prob-

lems that are already solved.

The first and only large scale use of SSB

has been for multiplexing additional voice cir-

cuits on long distance telephone toll wires.

Carrier systems came into wide use during the

30's, accompanied by the development of high

Q toroids and copper oxide ring modulators

of controlled characteristics.

The problem solved by the carrier system

was that of translating the 300 -3000 cycle

voice band of frequencies to a higher frequen-

cy (for example, 40.3 to 43.0 kc.) for trans-

mission on the toll wires, and then to reverse

the translation process at the receiving termi-

nal. It was possible in some short -haul equip-

ment to amplitude modulate a 40 kilocycle

carrier with the voice frequencies, in which

case the resulting signal would occupy a band

of frequencies between 37 and 43 kilocycles.

Since the transmission properties of wires and

cable deteriorate rapidly with increasing fre-

quency, most systems required the bandwidth

conservation characteristics of single -sideband

transmission. In addition, the carrier wave was

generally suppressed to reduce the power

handling capability of the repeater amplifiers

and diode modulators. A substantial body of

literature on the components and circuit tech-

niques of SSB has been generated by the large

and continuing development effort to produce

economical carrier telephone systems.

The use of SSB for overseas radiotelephony

has been practiced for several years though

the number of such circuits has been numeri-

cally small. However, the economic value of

such circuits has been great enough to war-

rant elaborate station equipment. It is from

these stations that the impression has been ob-

tained that SSB is too complicated for all but

a corps of engineers and technicians to handle.

Components such as lattice filters with 40

or more crystals have suggested astronomical

expense.

323

www.americanradiohistory.com

324 Sideband Transmission T H E R A D I O

,10lllllllllllu IIIIIIIIIIIII1III

LOWER UPPER

SIDEBAND I SIDEDAND

CARRIER

FRED.

e

FREQUENCY SPECTRUM WITH CARRIER ENVELOPE WITH

COMPLEX MODULATING WA, COMBLE+ MODULATING WAVE

Figure 1

REPRESENTATION OF A

CONVENTIONAL AM SIGNAL

More recently, SSB techniques have been

used to multiplex large numbers of voice chan-

nels on a microwave radio band using equip-

ment principally developed for telephone car-

rier applications. It should be noted that all

production equipment employed in these ser-

vices uses the filter method of generating the

single -sideband signal, though there is a wide

variation in the types of filters actually used.

The SSB signal is generated at a low fre-

quency and at a low level, and then trans-

lated and linearly amplified to a high level

at the operating frequency.

Considerable development effort has been

expended on high level phasing type trans-

mitters wherein the problems of linear ampli-

fication are exchanged for the problems of

accurately controlled phase shifts. Such equip-

ment has featured automatic tuning circuits,

servo- driven to facilitate frequency changing,

but no transmitter of this type has been suffi-

ciently attractive to warrant appreciable pro-

duction.

17 -2 Derivation of

Single -Sideband Signals

The single -sideband method of communica-

tion is, essentially, a procedure for obtaining

more efficient use of available frequency spec-

trum and of available transmitter capability.

As a starting point for the discussion of sin-

gle- sideband signals, let us take a conven-

tional AM signal, such as shown in figure 1,

as representing the most common method for

transmitting complex intelligence such as

voice or music.

It will be noted in figure 1 that there are

three distinct portions to the signal: the car-

rier, and the upper and the lower sideband

group. These three portions always are present

in a conventional AM signal. Of all these por-

tions the carrier is the least necessary and

the most expensive to transmit. It is an actual

fact, and it can be proved mathematically (and

physically with a highly selective receiver)

that the carrier of an AM signal remains un-

changed in amplitude, whether it is being mod-

ulated or not. Of course the carrier appears

to be modulated when we observe the modu-

lated signal on a receiving system or indicator

which passes a sufficiently wide band that

the carrier and the modulation sidebands are

viewed at the same time. This apparent change

in the amplitude of the carrier with modula-

tion is simply the result of the sidebands beat-

ing with the carrier. However, if we receive

the signal on a highly selective receiver, and if

we modulate the carrier with a sine wave of

3000 to 5000 cycles, we will readily see that

the carrier, or either of the sidebands can be

tuned in separately; the carrier amplitude, as

observed on a signal strength meter, will re-

main constant, while the amplitude of the side -

bands will vary in direct proportion to the

modulation percentage.

Elimination of It is obvious from the pre -

the Carrier and vious discussion that the

One Sideband carrier is superfluous so far

as the transmission of in-

telligence is concerned. It is obviously a con-

venience, however, since it provides a signal

at the receiving end for the sidebands to beat

with and thus to reproduce the original mod-

ulating signal. It is equally true that the trans-

mission of both sidebands under ordinary con-

ditions is superfluous since identically the

same intelligence is contained in both side -

bands. Several systems for carrier and side-

band elimination will be discussed in this

chapter.

Power Advantage Single sideband is a very

of SSB over AM efficient form of voice

communication by radio.

The amount of radio frequency spectrum oc-

cupied can be no greater than the frequency

range of the audio or speech signal transmitted,

whereas other forms of radio transmission re-

quire from two to several times as much spec-

trum space. The r -f power in the transmitted

SSB signal is directly proportional to the power

in the original audio signal and no strong

carrier is transmitted. Except for a weak pilot

carrier present in some commercial usage,

there is no r -f output when there is no audio

input. The power output rating of a SSB trans-

mitter is given in terms of peak envelope

pcwer (PEP) . This may be defined as the

r -m -s power at the crest of the modulation

www.americanradiohistory.com

HANDBOOK Derivation 325

envelope. The peak envelope power of a con-

ventional amplitude modulated signal at 100%

modulation is four times the carrier power.

The average power input to a SSB transmitter

is therefore a very small fraction of the power

input to a conventional amplitude modulated

transmitter of the same power rating.

Single sideband is well suited for long -

range communications because of its spectrum

and power economy and because it is less sus-

ceptible to the effects of selective fading and

interference than amplitude modulation. The

principal advantages of SSB arise from the

elimination of the high- energy carrier and

from further reduction in sideband power per-

mitted by the improved performance of SSB

under unfavorable propagation conditions.

In the presence of narrow band man -made

interference, the narrower bandwidth of SSB

reduces the probability of destructive inter-

ference .A statistical study of the distribution

of signals on the air versus the signal strength

shows that the probability of successful com-

munication will be the same if the SSB power

is equal to one -half the power of one of the

two a -m sidebands. Thus SSB can give from 0

to 9 db improvement under various conditions

when the total sideband power is equal in

SSB and a -m. In general, it may be assumed

that 3 db of the possible 9 db advantage will

be realized on the average contact. In this case,

the SSB -power required for equivalent per-

formance is equal to the power in one of the

a -m sidebands. For example, this would rate

a 100 -watt SSB and a 400 watt (carrier) a -m

transmitter as having equal performance. It

should be noted that in this comparison it is

assumed that the receiver bandwidth is just

sufficient to accept the transmitted intelligence

in each case.

To help evaluate other methods of compari-

son the following points should be considered.

In conventional amplitude modulation two

sidebands are transmitted, each having a peak

envelope power equal to 1/4-carrier power. For

example, a 100 -watt a -m signal will have 25-

watt peak envelope power in each sideband,

or a total of 50 watts. When the receiver de-

tects this signal, the voltages of the two side -

bands are added in the detector. Thus the de-

tector output voltage is equivalent to that of

a 100 -watt SSB signal. This method of com-

parison says that a 100 watt SSB transmitter

is just equivalent to a 100 -watt a -m trans-

mitter. This assumption is valid only when the

receiver bandwidth used for SSB is the same

as that required for amplitude modulation

KC

AUDIO SPECTRUM

KC. 4004RC' `3996 KC 4000 KC

SSB SPECTRUM SSB SPECTRUM

(UPPER SIOEIAMO) (LOWER S/OEBAAO )

Figure 2

RELATIONSHIP OF AUDIO AND

SSB SPECTRUMS

The single sideband components are the same

as the original audio components except that

the frequency of each is raised by the fre-

quency of the carrier. The relative amplitude

of the various components remains the same.

(e.g., 6 kilocycles) , when there is no noise or

interference other than broadband noise, and

if the a -m signal is not degraded by propaga-

tion. By using half the bandwidth for SSB

reception ( e.g., 3 kilocycles) the noise is re-

duced 3 db so the 100 watt SSB signal be-

comes equivalent to a 200 watt carrier a -m

signal. It is also possible for the a -m signal

to be degraded another 3 db on the average

due to narrow band interference and poor

propagation conditions, giving a possible 4

to 1 power advantage to the SSB signal.

It should be noted that 3 db signal -to -noise

ratio is lost when receiving only one sideband

of an a -m signal. The narrower receiving band-

width reduces the noise by 3 db but the 6 db

advantage of coherent detection is lost, leaving

a net loss of 3 db. Poor propagation will de-

grade this "one sideband" reception of an a -m

signal less than double sideband reception,

however. Also under severe narrow band in-

terference conditions (e.g., an adjacent strong

signal) the ability to reject all interference on

one side of the carrier is a great advantage.

The Nature of o The nature of a single

SSB Signal sideband signal is easily

visualized by noting that

the SSB signal components are exactly the same

as the original audio components except that

the frequency of each is raised by the frequen-

cy of the carrier. The relative amplitude of the

various components remains the same, how-

ever. (The first statement is only true for the

upper sideband since the lower sideband fre-

quency components are the difference between

the carrier and the original audio signal).

Figure 2A, B, and C shows how the audio

spectrum is simply moved up into the radio

spectrum to give the upper sideband. The

lower sideband is the same except inverted, as

shown in figure 2C. Either sideband may be

used. It is apparent that the carrier frequency

www.americanradiohistory.com

326 Sideband Transmission THE RADIO

SINGLE TONE

Figure 3

A SINGLE SINE WAVE TONE INPUT

TO A SSB TRANSMITTER RESULTS

IN A STEADY SINGLE SINE WAVE

R -F OUTFIT (A). TWO AUDIO TONES

OF EQUAL AMPLITUDE BEAT

TOGETHER TO PRODUCE HALF -SINE

WAVES AS SHOWN IN (B).

of a SSB signal can only be changed by add-

ing or subtracting to the original carrier fre-

quency. This is done by heterodyning, using

converter or mixer circuits similar to those

employed in a superheterodyne receiver.

It is noted that a single sine wave tone in-

put to a SSB transmitter results in a single

steady sine wave r -f ouput, as shown in figure

3A. Since it is difficult to measure the per-

formance of a linear amplifier with a single

tone, it has become standard practice to use

two tones of equal amplitude for test pur-

poses. The two radio frequencies thus pro -

duced beat together to give the SSB envelope

shown in figure 3B. This figure has the shape

of half sine waves, and from one null to the

next represents one full cycle of the difference

frequency. How this envelope is generated is

shown more fully in figures 4A and 4B. f,

and f2 represent the two tone signals. When

a vector representing the lower frequency tone

signal is used as a reference, the other vector

rotates around it as shown, and this action

r,

FREQUENCY fi 12

OF

CRRiER

Figure 4

VECTOR REPRESENTATION OF

TWO -TONE SSB ENVELOPE

Figure 5

TWO -TONE SSB

ENVELOPE WHEN

ONE TONE HAS

TWICE THE

AMPLITUDE OF

THE OTHER.

Figure 6

THREE -TONE SSB

ENVELOPE WHEN

EQUAL TONES OF

EQUAL FREQUENCY

SPACINGS

ARE USED.

generates the SSB envelope When the two

vectors are exactly opposite in phase, the out-

put is zero and this causes the null in the en-

velope. If one tone has twice the amplitude of

the other, the envelope shape is shown in

figure 5. Figure 6 shows the SSB envelope of

three equal tones of equal frequency spacings

and at one particular phase relationship. Figure

7A shows the SSB envelope of four equal

tones with equal frequency spacings and at

one particular phase relationship. The phase

relationships chosen are such that at some in-

stant the vectors representing the several tones

are all in phase. Figure 7B shows a SSB envel-

ope of a square wave. A pure square wave re-

quires infinite bandwidth, so its SSB envelope

requires infinite amplitude. This emphasizes

the point that the SSB envelope shape is not

the same as the original audio wave shape, and

usually bears no similarity to it. This is be-

cause the percentage difference between the

radio frequencies is very small, even though

one audio tone may be several times the other

in terms of frequency. Speech clipping as used

©

Figure 7A Figure 7B

FOUR TONE SSB ENVELOPE

SSB ENVELOPE OF A SQUARE

when equal tones WAVE.

with equal frequency Peak of wave reaches

spacings are used infinite amplitude.

www.americanradiohistory.com

HANDBOOK Derivation 327

in amplitude modulation is of no practical

value in SSB because the SSB r -f envelopes

are so different than the audio envelopes. A

heavily clipped wave approaches a square wave

and a square wave gives a SSB envelope with

peaks of infinite amplitude as shown in figure

7B.

Carrier Frequency Reception of a SSB

Stability Requirements signal is accom-

plished by simply

heterodyning the carrier down to zero fre-

quency. (The conversion frequency used in the

last heterodyne step is often called the rein-

serted carrier). If the SSB signal is not hetero-

dyned down to exactly zero frequency, each

frequency component of the detected audio

signal will be high or low by the amount of

this error. An error of 10 to 20 c.p s. for speech

signals is acceptable from an intelligibility

standpoint, but an error of the order of 50

c.p.s. seriously degrades the intelligibility. An

error of 20 c.p.s. is not acceptable for the

transmission of music, however, because the

harmonic relationship of the notes would be

destroyed. For example, the harmonics of 220

c.p.s. are 440, 660, 880, etc., but a 10 c.p s.

error gives 230, 450, 670, 890, etc., or 210,

430, 650, 870, etc., if the original error is on

the other side. This error would destroy the

original sound of the tones, and the harmony

between the tones.

Suppression of the carrier is common in ama-

teur SSB work, so the combined frequency sta-

bilities of all oscillators in both the transmit-

ting and receiving equipment add together to

give the frequency error found in detection.

In order to overcome much of the frequency

stability problem, it is common commercial

practice to transmit a pilot carrier at a re-

duced amplitude. This is usually 20 db below

one tone of a two -tone signal, or 26 db below

the peak envelope power rating of the trans-

mitter. This pilot carrier is filtered out from

the other signals at the receiver and either am-

plified and used for the reinserted carrier or

used to control the frequency of a local oscil-

lator. By this means, the frequency drift of

the carrier is eliminated as an error in detec-

tion.

Advantage of SSB On long distance corn -

with Selective Fading munication circuits

using a -m, selective

fading often causes severe distortion and at

times makes the signal unintelligible. When

one sideband is weaker than the other, distor-

PUSH -PULL

AUDIO IN

R G

o T

R r

0

OUT

Figure 8

SHOWING TWO COMMON TYPES

OF BALANCED MODULATORS

Notice that o balanced modulator changes

the circuit condition from single ended to

push -pull, or vice versa. Choice of circuit de-

pends upon external circuit conditions since

both the (A) and B: arrangements can give

satisfactory generation of a double -sideband

suppressed- carrier signal.

tien results; but when the carrier becomes

weak and the sidebands are strong, the distor-

tion is extremely severe and the signal may

sound like "monkey chatter." This is because

a carrier of at least twice the amplitude of

either sideband is necessary to demodulate the

signal properly. This can be overcome by us-

ing exalted carrier reception in which the car-

rier is amplified separately and then reinserted

before the signal is demodulated or detected.

This is a great help, but the reinserted carrier

must be very close to the same phase as the

original carrier. For example, if the reinserted

carrier were 90 degrees from the original

source, the a -m signal would be converted to

phase modulation and the usual a -m detector

would deliver no output.

The phase of the reinserted carrier is of no

importance in SSB reception and by using a

strong reinserted carrier, exalted carrier recep-

tion is in effect realized. Selective fading with

one sideband simply changes the amplitude

and the frequency response of the system and

very seldom causes the signal to become unin-

telligible. Thus the receiving techniques used

with SSB are those which inherently greatly

minimize distortion due to selective fading.

www.americanradiohistory.com

328 Sideband Transmission THE RADIO

MO

VOLTAGE. -,)

BRIDGE

MODULATOR

CARRIER

vOLTAGE

slDE-

22 DAN:

OUTPUT

z,

SHUNT-QUAD

MODULATOR

RING

MODULATOR - DOUBLE-BALANCED

MODULATOR

SiDE-

BAND

7 OUTPUT

z,

GAI ggiER I

VOLTAGE

Figure 9

TWO TYPES OF DIODE BALANCED

MODULATOR

Such balanced modulator circuits are com-

monly used in carrier telephone work and in

single -sideband systems where the carrier

frequency and modulating frequency are rela-

tively close together. Vacuum diodes, copper -

oxide rectifiers, or crystal diodes may be

used in the circuits.

17 -3 Carrier Elimination

Circuits

Various circuits may be employed to elimi-

nate the carrier to provide a double sideband

signal. A selective filter may follow the carrier

elimination circuit to produce a single side -

band signal.

Two modulated amplifiers may be connected

with the carrier inputs 180° out of phase, and

with the carrier outputs in parallel. The car-

rier will be balanced out of the output circuit,

leaving only the two sidebands. Such a cir-

cuit is called a balanced modulator.

Any non -linear element will produce modu-

lation. That is, if two signals are put in, sum

and difference frequencies as well as the orig-

inal frequencies appear in the output. This

phenomenon is objectionable in amplifiers and

desirable in modulators or mixers.

In addition to the sum and difference fre-

quencies, other outputs (such as twice one

frequency plus the other) may appear. All

combinations of all harmonics of each input

frequency may appear, but in general these are

of decreasing amplitude with increasing order

of harmonic. These outputs are usually re-

jected by selective circuits following the mod-

ulator. All modulators are not alike in the

magnitude of these higher order outputs. Bal-

anced diode rings operating in the square law

region are fairly good and pentagrid converters

much poorer. Excessive carrier level in tube

mixers will increase the relative magnitude

of the higher order outputs. Two types of

triode balanced modulators are shown in figure

8, and two types of diode modulators in figure

9. Balanced modulators employing vacuum

tubes may be made to work very easily to a

point. Circuits may be devised wherein both

input signals may be applied to a high im-

pedance grid, simplifying isolation and load-

ing problems. The most important difficulties

with these vacuum tube modulator circuits

are: (1) Balance is not independent of signal

level. (2) Balance drifts with time and envir-

onment. (3) The carrier level for low "high -

order output" is critical, and (4) Such circuits

have limited dynamic range.

A number of typical circuits are shown in

figure 10. Of the group the most satisfactory

performance is to be had from plate modulated

triodes.

PLATE MODULATED BALANCED

TRIODE MODULATOR

IDO

o-.1

R.r

0.I CAR-

O T IRR

IN

BALANCED TRIODE MODULATOR

WITH SINGLE ENDED INPUT CIRCUITS

Figure 10

BALANCED MODULATORS

PU SII PULL

AUDIO IN

BALANCED PENTAGRID CON-

VERTER MODULATOR

www.americanradiohistory.com

HANDBOOK Carrier Elimination 329

MODULA T.

VOLTAGE SIDEBAND

OurPUr

CARRIER VOLTAGE

HIGH Z

MODULATING

VOLTAGE

HIGH Z

SIDEBAND

OUTPUT

CARRIER VOLTAGE

DOUBLE- BALANCED RING MODULATOR SHUN.' QUAD MODAL ATOR

Figure 11

DIODE RING MODULATORS

LOW Z

MODULATING

VOLTAGE

LOW Z

SIDEBAND

OUTPUT

CARRIER VOLTAGE

SERIES -QUAD MODULATOR

Diode Ring

Modulators Modulation in telephone car-

rier equipment has been very

successfully accomplished with

copper -oxide double balanced ring modulators.

More recently, germanium diodes have been

applied to similar circuits. The basic diode

ring circuits are shown in figure 11. The most

widely applied is the double balanced ring

(A) . Both carrier and input are balanced with

respect to the output, which is advantageous

when the output frequency is not sufficiently

different from the inputs to allow ready sep-

aration by filters. It should be noted that the

carrier must pass through the balanced input

and output transformers. Care must be taken

in adapting this circuit to minimize the carrier

power that will be lost in these elements. The

shunt and series quad circuits are usable when

the output frequencies are entirely different

(i.e.: audio and r.f.) . The shunt quad (B)

is used with high source and load impedances

and the series quad (C) with low source and

load impedances. These two circuits may be

adapted to use only two diodes, substituting a

balanced transformer for one side of the

bridge, as shown in figure 12. It should be

noted that these circuits present a half -wave

load to the carrier source. In applying any of

these circuits, r -f chokes and capacitors must

be employed to control the path of signal and

carrier currents. In the shunt pair, for example,

a blocking capacitor is used to prevent the r -f

load from shorting the audio input.

To a first approximation, the source and

load impedances should be an arithmetical

mean of the forward and back resistances of

the diodes employed. A workable rule of

thumb is that the source and load impedances

be ten to twenty times the forward resistance

for semi -conductor rings. The high frequency

limit of operation in the case of junction and

copper -oxide diodes may be appreciably ex-

tended by the use of very low source and load

impedances.

Copper -oxide diodes suitable for carrier

work are normally manufactured to order. They

offer no particular advantage to the amateur,

though their excellent long -term stability is

important in commercial applications. Recti-

fier types intended to be used as meter recti-

fiers are not likely to have the balance or high

frequency response desirable in amateur SSB

transmitters.

Vacuum diodes such as the 6AL5 may be

used as modulators. Balancing the heater -

cathode capacity is a major difficulty except

when the 6AL5 is used at low source and load

impedance levels. In addition, contact poten-

tials of the order of a few tenths of a volt may

also disturb low level applications (figure 13).

The double diode circuits appear attractive,

but in general it is more difficult to balance a

transformer at carrier frequency than an addi-

tional pair of diodes. Balancing potentiometers

may be employed, but the actual cause of the

unbalance is far more subtile, and cannot be

adequately corrected with a single adjustment.

A signal produced by any of the above cir-

cuits may be classified as a double sideband,

suppressed- carrier signal.

MODULATING SIDEBAND

VOLTAGE I_ OUTPUT

SHUNT -PAIR

MODULATOR

CARRIER VOLTAGE

SERIES -PAIR

MODULATOR

O

MODULATI NC

VOLTAGE

ARRIER VOLTAGE

Figure 12

DOUBLE -DIODE PAIRED MODULATORS

www.americanradiohistory.com

330 Sideband Transmission THE RADIO

B

i

PUSH-PULL R F

CARRIER IN

R.F OUT

O

OA SERIES- BALANCED DIODE MODULATOR

USING 6AL5 TUBE

6AL5

R.F.CAR

RIES IN

o-

AUDIO

IN

R.F. OUT

001

RFC

OB RING -DIODE MODULATOR USING 6AL5 TUBE

Figure 13

VACUUM DIODE MODULATOR CIRCUITS

17 -4 Generation of

Single -Sideband Signals

In general, there are two commonly used

methods by which a single -sideband signal may

be generated. These systems are: (I) The Fil-

ter Method, and (2) The Phasing Method.

The systems may be used singly or in com-

bination, and either method, in theory, may

be used at the operating frequency of the

transmitter or at some other frequency with

the signal at the operating frequency being ob-

tained through the use of frequency changers

(mixers) .

The Filter The filter method for obtaining

Method a SSB signal is the classic meth-

od which has been in use by the

telephone companies for many years both for

-6 -5 -4 -3 -2 -1 0

KILOCYCLES DEVIATION

Figure 15

BANDPASS CHARACTERISTIC OF

BURNELL S -15000 SINGLE

SIDEBAND FILTER

land -line and radio communications. The mode

of operation of the filter method is diagram-

med in figure 14, in terms of components and

filters which normally would be available to

the amateur or experimenter. The output of

the speech amplifier passes through a con-

ventional speech filter to limit the frequency

range of the speech to about 200 to 3000

cycles. This signal then is fed to a balanced

modulator along with a 50,000 -cycle first car-

rier from a self- excited oscillator. A low -fre-

quency balanced modulator of this type most

conveniently may be made up of four diodes

of the vacuum or crystal type cross connected

in a balanced bridge or ring modulator circuit.

Such a modulator passes only the sideband

components resulting from the sum and dif-

ference between the two signals being fed to

the balanced modulator. The audio signal and

the 50 -kc. carrier signal from the oscillator

both cancel out in the balanced modulator so

that a band of frequencies between 47 and 50

kc. and another band of frequencies between

50 and 53 kc. appear in the output.

The signals from the first balanced modu-

lator are then fed through the most critical

100-10000 '1. ZOO-50001. I7-SO NC

SPE ECM

AMPLIFIER SPEECH

FILTER BALANCED

MODULATOR

50-53 KC

SO KC.

OSCILLATOR

47 -50 KC

SIDEBAND

FILTER

47-SO KC. 47-5014C

PHASE

INVERTER BALANCED

MODULATOR

1750-1950 KC.

OSCILLATOR

Figure 14

BLOCK DIAGRAM OF FILTER EXCITER EMPLOYING A 50 -K.C.

SIDEBAND FILTER

HIGH -O TUNEDCIRCUIT

FOR OUTPUT IN

1100 - 2000 AC. SAND

www.americanradiohistory.com

HANDBOOK Generation of S.S.B. 331

MIC

6AU6 2 12AU7 6AL5 i 12AU7

SNUNTDIODE MODULATOR RF AMPLIFIER

01 15MH 250LU1F

( SO RC.)

33111f LO

P G

SO KCR .

FILTE

B r

6C4

AMASE-INVERTER

.01

002-- 0.35 005

{ I.

100

12AU7

SO KC OSCILLATOR

00A

+350 V

SOMA

PUSH -PULL R.F. TO BALANCED

MODULATOR FOR CONVERSION

TO 160 METERS

NOTE : UNLESS OTHERWISE SPECIFIED,

RESISTORS ARE 0.5 WATT.

CAPACITORS /AI 4/F.

Figure 16

OPERATIONAL CIRCUIT FOR SSB EXCITER USING THE BURNELL

50 -KC. SIDEBAND FILTER

component in the whole system -the first side-

band filter. It is the function of this first side-

band filter to separate the desired 47 to 50 kc.

sideband from the unneeded and undesired 50

to 53 kc. sideband. Hence this filter must have

low attenuation in the region between 47 and

50 kc., a very rapid slope in the vicinity of

50 kc., and a very high attenuation to the

sideband components falling between 50 and

53 kilocycles.

Burnell & Co., Inc., of Yonkers, New York

produce such a filter, designated as Burnell

S -1 5,000. The passband of this filter is shown

in figure 15.

Appearing, then, at the output of the filter

is a single sideband of 47 kc. to 50 kc. This

sideband may be passed through a phase in-

verter to obtain a balanced output, and then

fed to a balanced mixer. A local oscillator

operating in the range of 1750 kc. to 1950 kc.

is used as the conversion oscillator. Additional

conversion stages may now be added to trans-

late the SSB signal to the desired frequency.

Since only linear amplification may be used,

it is not possible to use frequency multiplying

stages. Any frequency changing must be done

by the beating- oscillator technique. An oper-

ational circuit of this type of SSB exciter is

shown in figure 16.

A second type of filter -exciter for SSB may

be built around the Collins Mechanical Filter.

Such an exciter is diagrammed in figure 17.

Voice frequencies in the range of 200 -3000

cycles are amplified and fed to a low imped-

ance phase -inverter to furnish balanced audio.

This audio, together with a suitably chosen

r -f signal, is mixed in a ring modulator, made

up of small germanium diodes. Depending

upon the choice of frequency of the r -f oscilla-

tor, either the upper or lower sideband may be

applied to the input of the mechanical filter.

The carrier, to some extent, has been rejected

by the ring modulator. Additional carrier re-

jection is afforded by the excellent passband

200-30001 200-3000

SPEECH

AMPLIFIER 111111 LOW 2

PHASE

INVERTER

453 -158 RC.

SHUNT -QUAD

RING

MODULATOR - 453-458 RC. 3953 RC.

455 K C.

MECHANICAL

F ILTER

450-453 KC.

453 K.C.

OSCILLATOR

CONVERTER

3500 K.C.

OSCILLATOR

Figure 17

BLOCK DIAGRAM OF FILTER EXCITER EMPLOYING A 455 -KC.

MECHANICAL FILTER FOR SIDEBAND SELECTION

R.F. AMPLIFIER

WITH HIGH -Q

TUNED CIRCUIT

FOR OUTPUT ON

3953 R C.

www.americanradiohistory.com

332 Sideband Transmission THE RADIO

CMSO CMSO

CM SO CMSO

fT -EII CHANNEL If CRYSTAL 4II. J NC.

AT-241 CHANNEL SO CRYSTAL 465.PMC.

Figure 18

SIMPLE CRYSTAL LATTICE FILTER

0

10

20

< e0

70 459 480 461 462 463 464

FREQUENCY (KC.)

.- CARRIER

FREQUENCY

characteristics of the mechanical filter. For

simplicity, the mixing and filtering operation

usually takes place at a frequency of 455 kilo-

cycles. The single -sideband signal appearing

at the output of the mechanical filter may be

translated directly to a higher operating fre-

quency. Suitable tuned circuits must follow

the conversion stage to eliminate the signal

from the conversion oscillator.

Wave Filters The heart of a filter -type SSB

exciter is the sideband filter.

Conventional coils and capacitors may be

used to construct a filter based upon standard

wave filter techniques. The Q of the filter in-

ductances must be high when compared with

the reciprocal of the fractional bandwidth. If

a bandwidth of 3 kc. is needed at a carrier fre-

quency of 50 kc., the bandwidth expressed in

terms of the carrier frequency is 3/50 or 6 %.

This is expressed in terms of fractional band-

width as 1/16. For satisfactory operation, the

o

10

30

60

I 7WE UPPER SI MOAN

CARRIER

FREQUENCY

246 247 248 249 250 251 252 253 254 235

FREQUENCY (K.C.)

Figure 19

PASSBAND OF LOWER AND UPPER

SIDEBAND MECHANICAL FILTER

Q of the filter inductances should be 10 times

the reciprocal of this, or 160. Appropriate Q is

generally obtained from toroidal inductances,

though there is some possibility of using iron

core solenoids between 10 kc. and 20 kc. A

characteristic impedance below 1000 ohms

should be selected to prevent distributed ca-

pacity of the inductances from spoiling overall

performance. Paper capacitors intended for

bypass work may not be trusted for stability

or low loss and should not be used in filter

circuits. Care should be taken that the levels

of both accepted and rejected signals are low

enough so that saturation of the filter induct-

ances does not occur.

Crystal Filters The best known filter re-

sponses have been obtained

with crystal filters. Types designed for pro-

gram carrier service cut -off 80 db in less than

50 cycles. More than 80 crystals are used in

this type of filter. The crystals are cut to con-

trol reactance and resistance as well as the

resonant frequency. The circuits used are based

on full lattices.

The war -surplus low frequency crystals may

be adapted to this type of filter with some

success. Experimental designs usually syn-

thesize a selectivity curve by grouping sharp

notches at the side of the passband. Where the

width of the passband is greater than twice

the spacing of the series and parallel reso-

nance of the crystals, special circuit techni-

ques must be used. A typical crystal filter using

these surplus crystals, and its approximate pass-

band is shown in figure 18.

Mechanical Filters Filters using mechanical

resonators ha v e been

studied by a number of companies and are

offered commercially by the Collins Radio Co.

They are available in a variety of bandwidths

www.americanradiohistory.com

HANDBOOK Generation of S.S.B. 333

20Q JOOD

BALANCED

MODULATO

TO POWER AMPLIFIER STAGES

AUDIO

PHASE

N I

SPC SPEECH

AMPLIFIEEEH R FILTER SPLITTING

NETWORKS

OR DIRECTLY TO ANTENNA SYSTEM

Q2 BALANCED

MODULATOF

N. 2

PHASE DIFFERENCE BETWEEN O / AND 15=110*

PHASE DIFFERENCE BETWEEN RI AND RI = PO. ei 92

RADIO FRED.

PHASE

SPLITTING

NETWORK

RADIO FRED.

SIGNAL AT

CARRIER FRED.

Figure 20

BLOCK DIAGRAM OF THE "PHASING" METHOD

The phasing method of obtaining a single -sideband signal is simpler thon the filter system in regard to

the number of tubes and circuits required. The system is also less expensive in regard to the components

required, but is more critical in regard to adjustments for the transmission of a pure single -sideband signal.

at center frequencies of 250 kc. and 455 kc.

The 250 kc. series is specifically intended for

sideband selection. The selectivity attained by

these filters is intermediate between good LC

filters at low center frequencies and engineered

quartz crystal filters. A passband of two 250

kc. filters is shown in figure 19. In application

of the mechanical filters some special precau-

tions are necessary. The driving and pick -up

coils should be carefully resonated to the op-

erating frequency. If circuit capacities are un-

known, trimmer capacitors should be used

across the coils. Maladjustment of these tuned

coils will increase insertion loss and the peak -

to- valley ratio. On high impedance filters ( ten

to twenty thousand ohms) signals greater than

2 volts at the input should be avoided. D -c

should be blocked out of the end coils. While

the filters are rated for 5 ma. of coil current,

they are not rated for d -c plate voltage.

The Phasing There are a number of points

System of view from which the op-

eration of the phasing system

of SSB generation may be described. We may

state that we generate two double -sideband

suppressed carrier signals, each in its own bal-

anced modulator, that both the r -f phase and

the audio phase of the two signals differ by

90 degrees, and that the outputs of the two

balanced modulators are added with the result

that one sideband is increased in amplitude

and the other one is cancelled. This, of course,

is a true description of the action that takes

place. But it is much easier to consider the

phasing system as a method simply of adding

(or of subtracting) the desired modulation

frequency and the nominal carrier frequency.

The carrier frequency of course is not trans-

mitted, as is the case with all SSB transmis-

sions, but only the sum or the difference of the

modulation band from the nominal carrier is

transmitted ( figure 20 ) .

The phasing system has the obvious advan-

tage that all the electrical circuits which give

rise to the single sideband can operate in a

practical transmitter at the nominal output fre-

quency of the transmiter. That is to say that

if we desire to produce a single sideband whose

nominal carrier frequency is 3.9 Mc., the

balanced modulators are fed with a 3.9 -Mc.

signal and with the audio signal from the

phase splitters. It is not necessary to go through

several frequency conversions in order to ob-

tain a sideband at the desired output fre-

quency, as in the case with the filter method

of sideband generation.

Assuming that we feed a speech signal to

the balanced modulators along with the 3900 -

kc. carrier (3.9 Mc.) we will obtain in the out-

put of the balanced modulators a signal which

is either the sum of the carrier signal and the

speech band, or the difference between the car-

rier and the speech band. Thus if our speech

signal covers the band from 200 to 3000

cycles, we will obtain in the output a band of

frequencies from 3900.2 to 3903 kc. (the sum

of the two, or the "upper" sideband), or a band

from 3897 to 3899.8 kc. (the difference be-

tween the two or the "lower" sideband) . A

further advantage of the phasing system of

sideband generation is the fact that it is a very

simple matter to select either the upper side-

band or the lower sideband for transmission.

A simple double -pole double -throw reversing

switch in two of the four audio leads to the

balanced modulators is all that is required.

www.americanradiohistory.com

334 Sideband Transmission T H E R A D I O

\ 180690270/

FOUR-PHASE A F.

INDUCTIVE

COUPLING

R.F

OUT o

0 IBC' ar 270

FOUR -PHASE A. F.

Figure 21

TWO CIRCUITS FOR SINGLE

SIDEBAND GENERATION BY THE

PHASING METHOD.

The circuit of (A) offers the advantages of

simplicity in the single -ended input circuits

plus a push -pull output circuit. Circuit (8) re-

quires double -ended input circuits but allows

all the plates to be connected in parallel for

the output circuit.

High -Level The plate -circuit effi-

Phasing vs. ciency of the four tubes

Low -Level Phasing usually used to make up

the two balanced modu-

lators of the phasing system may run as high

as 50 to 70 per cent, depending upon the oper-

ating angle of plate current flow. Hence it is

possible to operate the double balanced modu-

lator directly into the antenna system as the

output stage of the transmitter.

The alternative arrangement is to generate

the SSB signal at a lower level and then to

amplify this signal to the level desired by

means of class A or class B r -f power ampli-

fiers. If the SSB signal is generated at a level

of a few milliwatts it is most common to make

the first stage in the amplifier chain a class

A amplifier, then to use one or more class B

linear amplifiers to bring the output up to the

desired level.

Balanced Illustrated in figure 8 are

the two basic balanced

modulator circuits which

give good results with a radio frequency car-

rier and an audio modulating signal. Note that

one push -pull and one single ended tank cir-

cuit is required, but that the push -pull circuit

may be placed either in the plate or the grid

circuit. Also, the audio modulating voltage al-

ways is fed into the stage in push -pull, and

the tubes normally are operated Class A.

When combining two balanced modulators

to make up a double balanced modulator as

used in the generation of an SSB signal by the

phasing system, only one plate circuit is re-

quired for the two balanced modulators. How-

ever, separate grid circuits are required since

the grid circuits of the two balanced modula-

tors operate at an r -f phase difference of 90

degrees. Shown in figure 21 are the two types

of double balanced modulator circuits used for

generation of an SSB signal. Note that the cir-

cuit of figure 21A is derived from the bal-

anced modulator of figure 8A, and similarly

figure 21B is derived from figure 8B.

Another circuit that gives excellent perform-

ance and is very easy to adjust is shown in

figure 22. The adjustments for carrier balance

are made by adjusting the potentiometer for

voltage balance and then the small variable ca-

pacitor for exact phase balance of the balanced

carrier voltage feeding the diode modulator.

Modulator Circuits

MECHANICAL

FILTER

0 1 VOLT

SSB

OUTPUT

R-F CARRIER

2.3 VOLTS

Figure 22

BALANCED MODULATOR FOR USE

WITH MECHANICAL FILTER

www.americanradiohistory.com

HANDBOOK Generation of S.S.B. 335

B

Figure 23

LOW -Q R -F PHASE -SHIFT NETWORK

The r -f phase -shift system illustrated above

is convenient in a case where it is desired to

make small changes in the operating fre-

quency of the system without the necessity

of being precise in the adjustment of two

coupled circuits as used for r -f phase shift

in the circuit of figure 21.

Radio -Frequency A single -sideband genera -

Phasing tor of the phasing type

requires that the two bal-

anced modulators be fed with r -f signals hav-

ing a 90- degree phase difference. This r -f

phase difference may be obtained through the

use of two loosely coupled resonant circuits,

such as illustrated in figures 21A and 21B.

The r -f signal is coupled directly or inductive-

ly to one of the tuned circuits, and the coupling

between the two circuits is varied until, at

resonance of both circuits, the r -f voltages

developed across each circuit have the same

amplitude and a 90- degree phase difference.

The 90- degree r -f phase difference also may

be obtained through the use of a low -Q phase

shifting network, such as illustrated in figure

23; or it may be obtained through the use of

a lumped -constant quarter -wave line. The low -

Q phase- shifting system has proved quite prac-

ticable for use in single -sideband systems,

particularly on the lower frequencies. In such

an arrangement the two resistances R have

the same value, usually in the range between

100 and a few thousand ohms. Capacitor C, in

shunt with the input capacitances of the tubes

and circuit capacitances, has a reactance at

the operating frequency equal to the value of

the resistor R. Also, inductor L has a net in-

ductive reactance equal in value at the oper-

ating frequency to resistance R.

The inductance chosen for use at L must

take into account the cancelling effect of the

input capacitance of the tubes and the circuit

capacitance; hence the inductance should be

AUDIO

SIGNAL

.024220I1

fISO V.

3900

TO BAL.

MOO. I

3900

TO SAL

9100 .2

7

00l3! 20 K

.150 v.

Figure 24

DOME AUDIO -PHASE -SHIFT NETWORK

This circuit arrangement is convenient for ob-

taining the audio phase shift when it is desired

to use a minimum of circuit components and

tube elements.

variable and should have a lower value of in-

ductance than that value of inductance which

would have the same reactance as resistor R.

Inductor L may be considered as being made

up of two values of inductance in parallel; (a)

a value of inductance which will resonate at

the operating frequency with the circuit and

tube capacitances, and ( b) the value of induct-

ance which is equal in reactance to the resist-

ance R. In a network such as shown in figure

23, equal and opposite 45- degree phase shifts

are provided by the RL and RC circuits, thus

providing a 90- degree phase difference be-

tween the excitation voltages applied to the

two balanced modulators.

Audio -Frequency The audio -frequency phase -

Phasing shifting networks used in

generating a single -side-

band signal by the phasing method usually are

based on those described by Dome in an ar-

ticle in the December, 1946, Electronics. A

relatively simple network for accomplishing

the 90- degree phase shift over the range from

160 to 3500 cycles is illustrated in figure 24.

The values of resistance and capacitance must

be carefully checked to insure minimum devia-

tion from a 90- degree phase shift over the 200

to 3000 cycle range.

Another version of the Dome network is

shown in figure 25. This network employs

three 12AU7 tubes and provides balanced out-

put for the two balanced modulators. As with

the previous network, values of the resistances

within the network must be held to very close

tolerances. It is necessary to restrict the speech

range to 300 to 3000 cycles with this network.

Audio frequencies outside this range will not

have the necessary phase -shift at the output

www.americanradiohistory.com

336 Sideband Transmission THE RADIO

.01

AUD

INPUT

12AÚ7

R

12AU7 12AU7

+105 V. REGULATED

TO SAL.

N MODSI

343 K

2

B 11M

30

TO SAL.

M0D2

1%

+105 V. REGULATED

Figure 25

A VERSION OF THE DOME

AUDIO -PHASE -SHIFT

NETWORK

of the network and will show up as spurious

emissions on the sideband signal, and also

in the region of the rejected sideband. A low -

pass 3500 cycle speech filter, such as the

Chicago Transformer Co. LPF -2 should be

used ahead of this phase -shift network.

A passive audio phase -shift network that

employs no tubes is shown in figure 26. This

network has the same type of operating re-

strictions as those described above. Additional

information concerning phase -shift networks

will be found in Single Sideband Techniques

published by the Cowan Publishing Corp.,

New York, and The Single Sideband Digest

published by the American Radio Relay

League. A comprehensive sideband review is

contained in the December, 1956 issue of

Proceedings of the I.R.E.

Comparison of Filter Either the filter or the

and Phasing Methods phasing method of

of SSB Generation single -sideband gener-

ation is theoretically

capable of a high degree of performance.

In general, it may be said that a high degree

of unwanted signal rejection may be attained

with less expense and circuit complexity with

the filter method. The selective circuits for

rejection of unwanted frequencies operate at a

relativly low frequency, are designed for this

one frequency and have a relatively high order

of Q. Carrier rejection of the order of 50 db or

so may be obtained with a relatively simple

filter and a balanced modulator, and unwanted

sideband rejection in the region of 60 db is

economically possible.

The phasing method of SSB generation ex-

changes the problems of high -Q circuits and

linear amplification for the problems of accu-

rately controlled phase -shift networks. If the

0.5 TO BAL.

MOD Y I

PUSH -PULL

AUDIO

INPUT

0.5 2450 607

VLF

Figure 26

PASSIVE AUDIO -PHASE -SHIFT

NETWORK, USEFUL OVER RANGE

OF 300 TO 3000 CYCLES.

TO BAL

1)]]B MOD II 2

1!s Ill

phasing method is employed on the actual

transmitting frequency, change of frequency

must be accompanied by a corresponding re-

balance of the phasing networks. In addition,

it is difficult to obtain a phase balance with

ordinary equipment within 2% over a band of

audio frequencies. This means that carrier

suppression is limited to a maximum of 40 db

or so. However, when a relatively simple SSB

transmitter is needed for spot frequency opera-

tion, a phasing unit will perform in a satis-

factory manner.

Where a high degree of performance in the

SSB exciter is desired, the filter method

and the phasing method may be combined.

Through the use of the phasing method in the

first balanced modulator those undesired side -

band components lying within 1000 cycles of

the carrier may be given a much higher degree

of rejection than is attainable with the filter

method alone, with any reasonable amount of

complexity in the sideband filter. Then the

sideband filter may be used in its normal way

to attain very high attenuation of all undesired

sideband components lying perhaps further

than 500 cycles away from the carrier, and to

restrict the sideband width on the desired side

of the carrier to the specified frequency limit.

17 -5 Single Sideband

Frequency Conversion Systems

In many instances the band of sideband

frequencies generated by a low level SSB trans-

mitter must be heterodyned up to the desired

carrier frequency. In receivers the circuits

which perform this function are called con-

verters or mixers. In sideband work they are

usually termed mixers or modulators.

Mixer Stages One circuit which can be used

for this purpose employs a

receiving -type mixer tube, such as the 6BE6.

The output signal from the SSB generator is

fed into the #1 grid and the conversion fre-

www.americanradiohistory.com

HANDBOOK Frequency Conversion 337

2000 KC.

CONVERSION

FREQUENCY

(S.S )

68E6

250 KC. SSB

SIGNAL

(0.25V. )

TUNE TO SELECT

2000 + 250 =2250 KC.

ON

2000 -220 1750 KC.

Figure 27

PENTAGRID MIXER CIRCUIT FOR

SSB FREQUENCY CONVERSION

quency into the #3 grid. This is the reverse of

the usual grid connections, but it offers about

10 db improvement in distortion. The plate

circuit is tuned to select the desired output

frequency product. Actually, the output of the

mixer tube contains all harmonics of the two

input signals and all possible combinations of

the sum and difference frequencies of all the

harmonics. In order to avoid distortion of the

SSB signal, it is fed to the mixer at a low

level, such as 0.1 to 0.2 volts. The conversion

frequency is fed in at a level about 20 db

higher, or about 2 volts. By this means, har-

monics of the incoming SSB signal generated

in the mixer tube will be very low. Usually

the desired output frequency is either the sum

or the difference of the SSB generator carrier

frequency and the conversion frequency. For

example, using a SSB generator carrier fre-

quency of 250 kc. and a conversion injection

frequency of 2000 kc. as shown in figure 27,

the output may be tuned to select either 2250

kc. or 1750 kc.

Not only is it necessary to select the desired

mixing product in the mixer output but also

the undesired products must be highly attenu-

ated to avoid having spurious output signals

from the transmitter. In general, all spurious

signals that appear within the assigned fre-

quency channel should be at least 60 db below

the desired signal, and those appearing out-

side of the assigned frequency channel at least

80 db below the signal level.

When mixing 250 kc. with 2000 kc. as in

the above example, the desired product is the

2250 kc. signal, but the 2000 kc. injection

frequency will appear in the output about 20

db stronger than the desired signal. To reduce

it to a level 80 db below the desired signal

means that it must be attenuated 100 db.

The principal advantage of using balanced

modulator mixer stages is that the injection

frequency theoretically does not appear in the

output. In practice, when a considerable fre-

quency range must be tuned by the balanced

modulator and it is not practical to trim the

0.2 VOLT

SIGNAL INPUT

100

SSB OUTPUT

12AU7

2.0 VOLT

CONVERSION

SIGNAL

Figure 28

TWIN TRIODE MIXER CIRCUIT FOR

SSB FREQUENCY CONVERSION

push -pull circuits and the tubes into exact

amplitude and phase balance, about 20 db

of injection frequency cancellation is all that

can be depended upon. With suitable trim-

ming adjustments the cancellation can be made

as high as 40 db, however, in fixed frequency

circuit s.

The Twin Triode Mixer The mixer circuit

shown in figure 28

has about 10 db lower distortion than the con-

ventional 6BE6 converter tube. It has a lower

voltage gain of about unity and a lower out-

put impedance which loads the first tuned

circuit and reduces its selectivity. In some ap-

plications the lower gain is of no consequence

but the lower distortion level is important

enough to warrant its use in high performance

equipment. The signal -to- distortion ratio of

this mixer is of the order of 70 db compared

to approximately 60 db for a 6BE6 mixer

when the level of each of two tone signals is

0.5 volt. With stronger signals, the 6BE6

distortion increases very rapidly, whereas the

12AU7 distortion is much better compara-

tively.

6AS6's

001

SSB

SIGNAL

INPUT

-BIAS CARRIER + 120 V.

IN ATSMA.

Figure 29

BALANCED MODULATOR CIRCUIT

FOR SSB FREQUENCY CONVERSION

www.americanradiohistory.com

338 Sideband Transmission THE RADIO

1

7

6

3

9

8

7

6

.

3

2

7

9

8

6

i

2

6

Ell

. .

! . . . ... . .

.IéíNS

IIIII1111EMY

MURMUR

®UME=E__N

2:EN

BEM

0o

io 64

Figure 30

RESPONSE OF "N" NUMBER OF TUNED CIRCUITS,

ASSUMING EACH CIRCUIT Q IS 50

www.americanradiohistory.com

HANDBOOK Frequency Conversion 339

In practical equipment where the injection

frequency is variable and trimming adjust-

ments and tube selection cannot be used, it

may be easier and more economical to obtain

this extra 20 db of attenuation by using an

extra tuned circuit in the output than by using

a balanced modulator circuit. A balanced mod-

ulator circuit of interest is shown in figure

29, providing a minimum of 20 db of carrier

attenuation with no balancing adjustment.

Selective Tuned Circuits The selectivity re-

quirements of the

tuned circuits following a mixer stage often

become quite severe. For example, using an

input signal at 250 kc. and a conversion in-

jection frequency of 4000 kc. the desired out-

put may be 4250 kc. Passing the 4250 kc.

signal and the associated sidebands without

attenuation and realizing 100 db of attenuation

at 4000 kc. (which is only 250 kc. away) is

a practical example. Adding the requirement

that this selective circuit must tune from 2250

kc. to 4250 kc. further complicates the basic

requirement. The best solution is to cascade a

number of tuned circuits. Since a large num-

ber of such circuits may be required, the most

practical solution is to use permeability tun-

ing, with the circuits tracked together. An ex-

ample of such circuitry is found in the Collins

KWS -1 sideband transmitter.

If an amplifier tube is placed between each

tuned circuit, the overall response will be the

sum of one stage multiplied by the number

of stages (assuming identical tuned circuits).

Figure 30 is a chart which may be used to

determine the number of tuned circuits re-

quired for a certain degree of attenuation at

some nearby frequency. The Q of the circuits

is assumed to be 50, which is normally realized

in small permeability tuned coils. The number

of tuned circuits with a Q of 50 required for

providing 100 db of attenuation at 4000 kc.

while passing 4250 kc. may be found as fol-

lows:

of is 4250 -4000 =250 kc.

fr is the resonant frequency, 4250 kc.

and ff = 4250 - 0.059

The point on the chart where .059 inter-

sects 100 db is between the curves for 6 and 7

tuned circuits, so 7 tuned circuits are required.

Another point which must be considered in

practice is the tuning and tracking error of

the circuits. For example, if the circuits were

actually tuned to 4220 kc. instead of 4250 kc.,

the f'f would be 4220 or 0.0522. Checking

the curves shows that 7 circuits would just

barely provide 100 db of attenuation. This

illustrates the need for very accurate tuning

and tracking in circuits having high attenua-

tion properties.

Coupled Tuned When as many as 7 tuned

Circuits circuits are required for pro-

per attenuation, it is not

necessary to have the gain that 6 isolating am-

plifier tubes would provide. Several vacuum

tubes can be eliminated by using two or three

coupled circuits between the amplifiers. With

a coefficient of coupling between circuits 0.5

of critical coupling, the overall response is

very nearly the same as isolated circuits. The

gain through a pair of circuits having 0.5

coupling is only eight -tenths that of two criti-

cally coupled circuits, however. If critical

coupling is used between two tuned circuits,

the nose of the response curve is broadened

and about 6 db is lost on the skirts of each

pair of critically coupled circuits. In some

cases it may be necessary to broaden the nose

of the response curve to avoid adversely af-

fecting the frequency response of the desired

passband. Another tuned circuit may be re-

quired to make up for the loss of attenuation

on the skirts of critically coupled circuits.

Frequency Conversion The example in the

Problems previous section shows

the difficult selectivi-

ty problem encountered when strong undesired

signals appear near the desired frequency. A

high frequency SSB transmitter may be re-

quired to operate at any carrier frequency in

the range of 1.75 Mc. to 30 Mc. The problem

is to find a practical and economical means of

heterodyning the generated SSB frequency to

any carrier frequency in this range. There are

many modulation products in the output of the

mixer and a frequency scheme must be found

that will not have undesired output of appre-

ciable amplitude at or near the desired signal.

When tuning across a frequency range some

products may "cross over" the desired fre-

quency. These undesired crossover frequencies

should be at least 60 db below the desired

signal to meet modern standards. The ampli-

tude of the undesired products depends upon

the particular characteristics of the mixer and

the particular order of the product. In general,

most products of the 7th order and higher

will be at least 60 db down. Thus any cross-

www.americanradiohistory.com

340 Sideband Transmission THE RADIO

II SP-4Q 4P-3R 11

3P 2V 2A-Q

"SIGNAL TO

DISTORTION

(S /D) RATIO

P Q 2Q7

I

3Q4j 431. 5t41.

Figure 31

SSB DISTORTION PRODUCTS,

SHOWN UP TO NINTH ORDER

over frequency lower than the 7th must be

avoided since there is no way of attenuating

them if they appear within the desired pass -

band. The General Electric Ham News, volume

11 #6 of Nov. -Dec., 1956 covers the subject

of spurious products and incorporates a "mix -

selector" chart that is useful in determining

spurious products for various different mixing

schemes.

In general, for most applications when the

intelligence bearing frequency is lower than

the conversion frequency, it is desirable that

the ratio of the two frequencies be between 5

to 1 and 10 to 1. This is a compromise be-

tween avoiding low order harmonics of this

signal input appearing in the output, and

minimizing the selectivity requirements of the

circuits following the mixer stage.

17 -6 Distortion Products

Due to Nonlinearity of

R -F Amplifiers

When the SSB envelope of a voice signal

is distorted, a great many new frequencies are

generated. These represent all of the possible

combinations of the sum and difference fre-

quencies of all harmonics of the original fre-

quencies. For purposes of test and analysis,

two equal amplitude tones are used as the

SSB audio source. Since the SSB radio fre-

quency amplifiers use tank circuits, all distor-

tion products are filtered out except those

which lie close to the desired frequencies.

These are all odd order products; third order,

fifth order, etc.. The third order products are

2p -q and 2q -p where p and q represent the

two SSB r -f tone frequencies. The fifth order

products are 3p -2q and 3q -2p. These and

some higher order products are shown in

figure 31. It should be noted that the fre-

quency spacings are always equal to the dif-

ference frequency of the two original tones.

Thus when a SSB amplifier is badly over-

FROM SSB

GENERATOR GAIN CONTROL

PREAMPLIFIER y

CONTROL BIAS

POWER

AMPLIFIER

STAGE +TO ANT.

R-F FROM P-A

PLATE CIRCUIT

R F

RECTIFIER

t

DELAY BIAS VOLTAGE

FROM POWER SUPPLY

Figure 32

BLOCK DIAGRAM OF AUTOMATIC

LOAD CONTROL (A.L.C.) SYSTEM

loaded, these spurious frequencies can extend

far outside the original channel width and

cause an unintelligible "splatter" type of in-

terference in adjacent channels. This is usually

of far more importance than the distortion of

the original tones with regard to intelligibility

or fidelity. To avoid interference in another

channel, these distortion products should be

down at least 40 db below adjacent channel

signal. Using a two -tone test, the distortion is

given as the ratio of the amplitude of one

test tone to the amplitude of a third order

product. This is called the signal -to- distortion

ratio (S /D) and is usually given in decibels.

The use of feedback r -f amplifiers make S/D

ratios of greater than 40 db possible and prac-

tical.

Automatic Two means may be used to

Load Control keep the amplitude of these

distortion products down to

acceptable levels. One is to design the ampli-

fier for excellent linearity over its amplitude

or power range. The other is to employ a

means of limiting the amplitude of the SSB

envelope to the capabilities of the amplifier.

An automatic load control ssytem (ALC) may

be used to accomplish this result. It should be

noted that the r.f wave shapes of the SSB sig-

nal are always sine waves because the tank cir-

cuits make them so. It is the change in gain

with signal level in an amplifier that distorts

the SSB envelope and generates unwanted dis-

tortion products. An ALC system may be used

to limit the input signal to an amplifier to

prevent a change in gain level caused by ex-

cessive input level.

The ALC system is adjusted so the power

amplifier is operating near its maximum power

capability and at the same time is protected

from being over -driven. In amplitude modu-

lated systems it is common to use speech com-

pressors and speech clipping systems to per-

form this function. These methods are not

www.americanradiohistory.com

HANDBOOK Distortion Products 341

DB SIGNAL LEVEL INPUT

Figure 33

PERFORMANCE CURVE

A.L.C. CIRCUIT OF

equally useful in SSB. The reason for this is

that the SSB envelope is different from the

audio envelope and the SSB peaks do not

necessarily correspond with the audio peaks

as explained earlier in this chapter. For this

reason a "compressor" of some sort located

between the SSB generator and the power am-

plifier is most effective because it is controlled

by SSB envelope peaks rather than audio peaks.

Such a "SSB signal compressor" and the means

of obtaining its control voltage comprises a

satisfactory ALC system.

The ALC Circuit A block diagram of an ALC

circuit is shown in figure

32. The compressor or gain control part of

this circuit uses one or two stages of remote

cutoff tubes such as 6BA6, operating very sim-

ilarly to the intermediate frequency stages of

a receiver having automatic volume control.

The grid bias voltage which controls the

gain of the tubes is obtained from a voltage

detector circuit connected to the power am-

plifier tube plate circuit. A large delay bias is

used so that no gain reduction takes place until

the signal is nearly up to the full power capa-

bility of the amplifier. At this signal level,

the rectified output overcomes the delay bias

and the gain of the preamplifier is reduced

rapidly with increasing signal so that there is

very little rise in output power above the

threshold of gain control.

When a signal peak arrives that would nor-

mally overload the power amplifier, it is de-

sireable that the gain of the ALC amplifier be

reduced in a few milliseconds to a value where

overloading of the power amplifier is over-

come. After the signal peak passes, the gain

should return to the normal value in about

one -tenth second. These attack and release

times are commonly used for voice communi-

cations. For this type of work, a dynamic range

of at least 10 db is desirable. Input peaks as

high as 20 db above the threshold of com-

pression should not cause loss of control al-

though some increase in distortion in the up-

per range of compression can be tolerated be-

cause peaks in this range are infrequent. An-

other limitation is that the preceding SSB

generator must be capable of passing signals

above full power output by the amount of

compression desired. Since the signal level

through the SSB generator should be main-

tained within a limited range, it is unlikely

that more than 12 db ALC action will be

useful. If the input signal varies more than

this, a speech compressor should be used to

limit the range of the signal fed into the SSB

generator.

Figure 33 shows the effectiveness of the

ALC in limiting the output signal to the cap-

abilities of the power amplifier. An adjustment

of the delay bias will place the threshold of

compression at the desired power output. Fig-

ure 34 shows a simplified schematic of an ALC

system. This ALC uses two variable gain am-

Figure 34 ee

INPUT

SIMPLIFIED SCHEMA-

TIC OF AUTOMATIC

LOAD CONTROL AM-

PLIFIER. OPERATING

POINT OF ALC

CIRCUIT MAY BE

SET BY VARYING

BLOCKING BIAS ON

CATHODE OF 6X4

SIGNAL RECTIFIER

6BA6 6BA6

SENS.

ALC ZERO

=COMPRESSION ADJ. _

INDICATOR

www.americanradiohistory.com

342 Sideband Transmission T H E R A D I O

Figure 35

SSB JR. MODULATOR CIRCUIT

R -F and A -F sources are applied in series

to balanced modulator.

plifier stages and the maximum overall gain is

about 20 db. A meter is incorporated which is

calibrated in db of compression. This is use-

ful in adjusting the gain for the desired

amount of load control. A capacity voltage

divider is used to step down the r -f voltage

at the plate of the amplifier tube to about 50

volts for the ALC rectifier. The output of the

ALC rectifier passes through R -C networks

to obtain the desired attack and release times

and through r -f filter capacitors. The 3.3K

resistor and 0.1 µfd. capacitor across the recti-

fier output stabilizes the gain around the ALC

loop to prevent "motor- boating."

17 -7 Sideband Exciters

Some of the most popular sideband exciters

in use today are variations of the simple phas-

ing circuit introduced in the November, 1950

issue of General Electric Ham Netes. Called the

SSB, Jr., this simple exciter is the basis for

many of the phasing transmitters now in use.

Employing only three tubes, the SSB, Jr. is a

classic example of sideband generation re-

duced to its simplest form.

The SSB, Jr. This phasing exciter employs

audio and r -f phasing circuits

to produce a SSB signal at one spot frequency.

The circuit of one of the balanced modulator

stages is shown in figure 35. The audio signal

and r -f source are applied in series to two ger-

manium diodes serving as balanced modulators

AUDIO

INPUT

PHASE SHIFT NETWORK

r

XTAL

T2 voF O. i 12AU7

BLV FEO-WN/TE

TWIST

-10.5V. C+,6-

C2A,B,C,D =EACH SECTION 20LF, 450 V. ELECTROLYTIC G1,

C7= 2430 UUFD (.002 UFD MICA f 5% WITH 170 -700 UUFD TRIMMER) LI,

CA =4600 UUFD. (.0043 UFD MICA 8.5% WITH 170-760 UUFD TRIMMER)

C9= 1215 UUFD (.001 UFO MICA ±5% WITH 50 -360 UUFD TRIMMER)

C10 =607.5 UUFD (500 UUFD MICA t 5% WITH 5 -100 UUFD TRIMMER)L3

C16= 35018UFD 600V. MICA *10% (2S0UUFD AND 100UUFD PARALLEL)

R7,RID= 133,300 OHMS, 1/2 WATT t 1% L4

Re R9. 100,000 OHMS, 1/2 WATT t 1% LS

T1= STANCON A -53C TRANSFORMER.

T2,T3= UTC R -364 TRANSFORMER.

S1= DPDT TOGGLE SWITCH

000

RFC 250 f 101b

0.5 MH ,,

6.3 V.

12ÁU7

I 2A17

6AG7

2,3,4= INS GERMANIUM DIODE OR EQUIVALENT

L2= 33 T. N21 E. WIRE CLOSEWOUND ON MILLEN N69046

IRON CORE ADJUSTABLE SLUG COIL FORM. LINK OF 6

TURNS OF HOOKUP WIRE WOUND ON OPEN END.

=16 T. N'19 E. WIRE SPACED TO FILL MILLEN M. 69046

COIL FORM. TAP AT 6 TURNS. LINK OF 1 TURN AT CENTER.

=SAME AS L, EXCEPT NO LINK USED.

= 26 T. OF N19 E. WIRE. LINK ON END TO MATCH LOAD.

(4 TURN LINK MATCHES 72 OHM LOAD)

R- = MOUNTING ENO OF COILS

Figure 36

SCHEMATIC, SSB, JR.

www.americanradiohistory.com

HANDBOOK S.B. Exc iters 343

having a push -pull output circuit tuned to the

r -f "carrier" frequency. The modulator drives

a linear amplifier directly at the output fre-

quency. The complete circuit of the exciter

is shown in figure 36.

The first tube, a 12AU7, is a twin -triode

serving as a speech amplifier and a crystal

oscillator. The second tube is a I2AT7, act-

ing as a twin channel audio amplifier follow-

ing the phase -shift audio network. The linear

amplifier stage is a 6AG7, capable of a peak

power output of 5 watts.

Sideband switching is accomplished by the

reversal of audio polarity in one of the audio

channels (switch SI), and provision is made

for equalization of gain in the audio channels

(R,z) . This adjustment is necessary in order

to achieve normal sideband cancellation, which

may be of the order of 35 db or better. Phase -

shift network adjustment may be achieved by

adjusting potentiometer R5. Stable modulator

balance is achieved by the balance potentio-

meters R,° and R in conjunction with the

germanium diodes.

The SSB, Jr. is designed for spot frequency

operation. Note that when changing frequency

L,, L_, L.,, L, and L should be readjusted,

since these circuits constitute the tuning ad-

justments of the rig. The principal effect of

mistuning Ls, L., and L. will be lower output.

The principal effect of mistuning L , however,

will be degraded sideband suppression.

Power requirements of the SSB, Jr. are 300

volts at 60 ma., and -10.5 volts at 1 ma.

Under load the total plate current will rise to

about 80 ma. at full level with a single tone

input. With speech input, the total current

will rise from the resting value of 60 ma. to

about 70 ma., depending upon the voice wave-

form.

The "Ten -A" The Model 10 -A phasing ex-

Exciter citer produced by Central

Electronics, Inc. is an ad-

vanced version of the SSB, Jr. incorporating

extra features such as VFO control, voice op-

eration, and multi -band operation. A simpli-

fied schematic of the Model 10A is shown in

figure 37. The 12AX7 two stage speech am-

plifier excites a transformer coupled 1/2 -12BH7

low impedance driver stage and a voice oper-

ated (VOX) relay system employing a 12AX7

and a 6AL5. A transformer coupled 12AT7

follows the audio phasing network, providing

two audio channels having a 90- degree phase

difference. A simple 90- degree r -f phase shift

network in the plate circuit of the 9 Mc. crys-

tal oscillator stage works into the matched,

balanced modulator consisting of four 1N48

diodes.

The resulting 9 Mc. SSB signal may be con-

verted to the desired operating frequency in a

6BA7 mixer stage. Eight volts of r -f from an

external v -f -o injected on grid #1 of the

6BA7 is sufficient for good conversion effic-

iency and low distortion. The plate circuit of

the 6BA7 is tuned to the sum or difference

mixing frequency and the resulting signal is

amplified in a 6AG7 linear amplifier stage.

Two "tweet" traps are incorporated in the

6BA7 stage to reduce unwanted responses of

the mixer which are apparent when the unit is

operating in the 14 Mc. band. Band -changing

is accomplished by changing coils L. and L°

and the frequency of the external mixing sig-

nal. Maximum power output is of the order

of 5 watts at any operating frequency.

A Simple 80 Meter A SSB exciter employ -

Phasing Exciter ing r -f and audio phas-

ing circuits is shown

in figure 38. Since the r -f phasing circuits are

balanced only at one frequency of operation,

the phasing exciter is necessarily a single fre-

quency transmitter unless provisions are made

to re- balance the phasing circuits every time a

frequency shift is made. However for mobile

operation, or spot frequency operation a rela-

tively simple phasing exciter may be made to

perform in a satisfactory manner.

A 12AU7 is employed as a Pierce crystal

oscillator, operating directly on the chosen

SSB frequency in the 80 meter band. The sec-

ond section of this tube is used as an isolation

stage, with a tuned plate circuit, L. The out-

put of the oscillator stage is link coupled to a

90° r -f phase -shift network wherein the audio

signal from the audio phasing network is com-

bined with the r -f signals. Carrier balance

is accomplished by adjustment of the two 1000

ohm potentiometers in the r -f phase network.

The output of the r -f phasing network is cou-

pled through L_ to a single 6CL6 linear am-

plifier which delivers a 3 watt peak SSB sig-

nal on 80 meters.

A cascade 12AT7 and a single 6C4 comprise

the speech amplifier used to drive the audio

phase shift network. A small inter -stage trans-

former is used to provide the necessary 180°

audio phase shift required by the network. The

output of the audio phasing network is cou-

pled to a 12AU7 dual cathode follower which

provides the necessary low impedance circuit

to match the r -f phasing network. A double-

www.americanradiohistory.com

344 Sideband Transmission THE RADIO

Figure 37

SIMPLIFIED SCHEMATIC OF "TEN -A" EXCITER

www.americanradiohistory.com

HANDBOOK S.B. Exciters 345

12 AU) Boo K

111,8 so* 6CL6

3

f100 150

K Tuur

L2 L3

3W PEAK

SSB

L1, L2,L38 24T0/22E.0w

XR-30 PORN

(0.3. O/A.)

CO/LS ARE T EACH.

EN-D4-1N71

12AT7

MIC. 2

.001 .001 NOTE. UNLESS OTHERWISE SPECIFIED/

ALL RESISTORS 0.3 WATT

ALL CAPACITORS /N//P.

ASTERISK AFTER CAPACITOR OR

RESISTOR VALUE IND/CATES

1.0 PRECIS /ON UNIT EAACT VALUE

CRITICAL ONLY /N T/IAT /T SHOULD

MATCH THE MAT /NG UNIT CLOSELY.

CARRIER

INJECT.

70K 2511

200 K 500K

6C 4

SIDE BAN D

SELECTOR

SWITCH

STANCOR

A53-C 12AÚ7

1 -5 e

500 7

14 10.2, r.

Figure 38

SIMPLE 3 -WATT PHASING TYPE SSB EXCITER

M

+300 V.

C

pole double -throw switch in the output circuit

of the cathode follower permits sideband se-

lection.

A Filter -Type Exciter A simple SSB filter -

for 80 and 40 Meters type exciter employ-

ing the Collins me-

chanical filter illustrates many of the basic

principles of sideband generation. Such an ex-

citer is shown in figure 39. The exciter is de-

signed for operation in the 80 or 40 meter

phone bands and delivers sufficient output to

drive a class ABI tetrode such as the 2E26, 807,

or 6146. A conversion crystal may be em-

ployed, or a separate conversion v -f -o can be

used as indicated on the schematic illustration.

The exciter employs five tubes, exclusive

of power supply. They are: 6U8 low frequency

oscillator and r -f phase inverter, 6BA6 i -f am-

plifier, 6BA7 high frequency mixer, 6AG7

linear amplifier, and 12AU7 speech amplifier

and cathode follower. The heart of the exciter

is the balanced modulator employing two

1N81 germanium diodes and the 455 kc.,

3500 -cycle bandwidth mechanical filter. The

input and output circuits of the filter are re-

sonated to 455 kc. by means of small padding

capacitors.

A series -tuned Clapp oscillator covers the

range of 452 kc. - 457 kc. permitting the

carrier frequency to be adjusted to the "20

decibel" points on the response curve of the

filter, as shown in figure 40. Proper r -f sig-

nal balance to the diode modulator may be

obtained by adjustment of the padding capaci-

tor in the cathode circuit of the triode section

of the 6U8 r -f tube. Carrier balance is set by

means of a 50K potentiometer placed across

the balanced modulator.

One half of a 12AU7 serves as a speech am-

plifier delivering sufficient output from a

high level crystal microphone to drive the sec-

ond half of the tube as a low impedance cath-

ode follower, which is coupled to the balanced

modulator. The two 1N81 diodes act as an

electronic switch, impressing a double side-

band, suppressed- carrier signal upon the me-

chanical filter. By the proper choice of fre-

quency of the beating oscillator, the unwanted

sideband may be made to fall outside the pass -

band of the mechanical filter. Thus a single

sideband suppressed- carrier signal appears at

the output of the filter. The 455 kc. SSB sig-

nal is amplified by a 6BA6 pentode stage, and

is then converted to a frequency in the 80

meter or 40 meter band by a 6BA7 mixer

stage. Either a crystal or an external v -f -o may

be used for the mixing signal.

To reduce spurious signals, a double tuned

www.americanradiohistory.com

I p

N (

III Ç 1,11

> I O OVN.

F LL

3

, In "

Figure 39

SCHEMATIC, FILTER -TYPE SSB EXCITER

FOR 80 OR 40 METER OPERATION

p &MIN B.r =WM

e

w- WM EV I ,

'o ME

OWE MOW

I

,D

452 453 454 455 456

FREQUENCY (KC.)

Figure 40

THE "TWENTY DB" CARRIER

POINTS ON THE FILTER CURVE

The beating oscillator should be adjusted so

that its frequency corresponds to the 20 db

attenuation points of the mechanical filter

passbond. The carrier of the SSB signal is

thus attenuated 20 db in addition to the

inherent carrier attenuation of the balanced

mixer. A total carrier attenuation of 50 db

is achieved. Unwanted sideband rejection is

of the some order.

457 456

500 RC.

CARRIER

INJECTION

6BE 6 R FC

556

SIGNAL

FROM

F AMP.

AUDIO

SIGNAL

Figure 41

THE PRODUCT DETECTOR

The above configuration resembles pentagrid

converter circuit.

® ICYCLEOF -

MWAV FORM

ODULATIN

ß.

LYWER UPPER

SIDEBAND SIDEBAND

CARRIER

FREQ.

FREQUENCY SPECTRUM WITN

COMPLEX MODULATING WAVE DOUBLE SIDE -BAND OUTPUT

FROM BALANCED MODULATOR

WITH SINE-WAVE MODULATION

Figure 42

DOUBLE -SIDEBAND

SUPPRESSED- CARRIER SIGNAL

The envelope shown at B also is obtained on

the oscilloscope when two audio frequencies

of the some amplitude are fed to the input

of a single -sideband transmitter.

www.americanradiohistory.com

S.B. Exciters 347

transformer is placed between the mixer stage

and the 6AG7 output stage. A maximum sig-

nal of 3 watts may be obtained from the 6AG7

linear amplifier.

Selection of the upper or lower sideband is

accomplished by tuning the 6U8 beating os-

cillator across the passband of the mechanical

filter, as shown in figure 40. If the 80 meter

conversion oscillator is placed on the low fre-

quency side of the SSB signal, placing the

6U8 beating oscillator on the low frequency

side of the passband of the mechanical filter

will produce the upper sideband on 80 meters.

When the beating oscillator is placed on the

high frequency side of the passband of the

mechanical filter the lower sideband will be

generated on 80 meters. If the 80 meter con-

version oscillator is placed on the high fre-

quency side of the SSB signal, the sidebands

will be reversed from the above. The variable

oscillator should be set at approximately the

20 db suppression point of the passband of

the mechanical filter for best operation, as

shown in figure 40. If the oscillator is closer

in frequency to the filter passband than this,

carrier rejection will suffer. If the oscillator is

moved farther away in frequency from the

passband, the lower voice frequencies will be

attenuated, and the SSB signal will sound high -

pitched and tinny. A little practice in setting

the frequency of the beating oscillator while

monitoring the 80 meter SSB signal in the

station receiver will quickly acquaint the oper-

ator with the proper frequency setting of the

beating oscillator control for transmission of

either sideband.

If desired, an amplitude modulated signal

with full carrier and one sideband may be

transmitted by placing the 6U8 low frequency

oscillator just inside either edge of the pass-

band of the filter (designated "AM point',

figure 40) .

After the 6U8 oscillator is operating over

the proper frequency range it should be possi-

ble to tune the beating oscillator tuning capac-

itor across the passband of the mechanical fil-

ter and obtain a reading on the S -meter of a

receiver tuned to the filter frequency and cou-

pled to the input grid of the 6BA6 i -f ampli-

fier tube. The two carrier balance controls of

the 6U8 phase inverter section should be ad-

justed for a null reading of the S -meter when

the oscillator is placed in the center of the

filter passband. The 6BA6 stage is now

checked for operation, and transformed T1

aligned to the carrier frequency. It may be

necessary to unbalance temporarily potentio-

meter #2 of the 6U8 phase inverter in order

to obtain a sufficiently strong signal for prop-

er alignment of Ti.

A conversion crystal is next plugged in the

6BA7 conversion oscillator circuit, and the op-

eration of the oscillator is checked by moni-

toring the crystal frequency with a nearby re-

ceiver. The SSB "carrier" produced by the un-

balance of potentiometer #2 should be heard

at the proper sideband frequency in either the

80 meter or 40 meter band. The coupled cir-

cuit between the 6BA7 and the 6AG7 is re-

sonated for maximum carrier voltage at the

grid of the amplifier stage. Care should be

taken that this circuit is tuned to the sideband

frequency and not to the frequency of the con-

version oscillator. Finally, the 6AG7 stage is

tuned for maximum output. When these ad-

justments have been completed, the 455 kc.

beating oscillator should be moved just out

of the passband of the mechanical filter. The

80 meter "carrier" will disappear. If it does

not, there is either energy leaking around the

filter, or the amplifier stages are oscillating.

Careful attention to shielding (and neutrali-

zation) should cure this difficulty.

Audio excitation is now applied to the ex-

citer, and the S -meter of the receiver should

kick up with speech, but the audio output of

the receiver should be unintelligible. As the

frequency of the beating oscillator is adjusted

so as to bring the oscillator frequency within

the passband of the mechanical filter the mo-

dulation should become intelligible. A single

sideband a.m. signal is now being generated.

The BFO of the receiver should now be turned

on, and the beating oscillator of the exciter

moved out of the filter passband. When the

receiver is correctly tuned, clean, crisp speech

should be heard. The oscillator should be set

at one of the "20 decibel" points of the filter

curve, as shown in figure 40 and all adjust-

ments trimmed for maximum carrier suppres-

sion.

17 -8 Reception of

Single Sideband Signals

Single -sideband signals may be received,

after a certain degree of practice in the tech-

nique, in a quite adequate and satisfactory

manner with a good communications receiver.

However, the receiver must have quite good

frequency stability both in the high- frequency

oscillator and in the beat oscillator. For this

reason, receivers which use a crystal -con-

trolled first oscillator are likely to offer a

www.americanradiohistory.com

348 Sideband Transmission T H E R A D I O

greater degree of satisfaction than the more

common type which uses a self -controlled os-

cillator.

Beat oscillator stability in most receivers

is usually quite adequate, but many receivers

do not have a sufficient amplitude of beat os-

cillator injection to allow reception of strong

SSB signals without distortion. In such re-

ceivers it is necessary either to increase the

amount of beat- oscillator injection into the

diode detector, or the manual gain control of

the receiver must be turned down quite low.

The tuning procedure for SSB signals is as

follows: The SSB signals may first be located

by tuning over the band with receiver set

for the reception of c -w.; that is, with the man-

ual gain at a moderate level and with the beat

oscillator operating. By tuning in this manner

SSB signals may be located when they are far

below the amplitude of conventional AM sig-

nals on the frequency band. Then after a sig-

nal has been located, the beat oscillator should

be turned off and the receiver put on a.v.c.

Following this the receiver should be tuned

for maximum swing of the S meter with modu-

lation of the SSB signal. It will not be possible

to understand the SSB signal at this time, but

the receiver may be tuned for maximum deflec-

tion. Then the receiver is put back on manual

gain control, the beat oscillator is turned on

again, the manual gain is turned down until

the background noise level is quite low, and

the beat oscillator control is varied until the

signal sounds natural.

The procedure in the preceding paragraph

may sound involved, but actually all the steps

except the last one can be done in a moment.

However, the last step is the one which will

require some practice. In the first place, it is

not known in advance whether the upper or

lower sideband is being transmitted. So it will

be best to start tuning the beat oscillator from

one side of the pass band of the receiver to

the other, rather than starting with the beat

oscillator near the center of the pass band as

is normal for c -w reception.

With the beat oscillator on the wrong side

of the sideband, the speech will sound inverted;

that is to say that low- frequency modulation

tones will have a high pitch and high- frequen-

cy modulation tones will have a low pitch -

and the speech will be quite unintelligible.

With the beat oscillator on the correct side of

the sideband but too far from the correct posi-

tion, the speech will have some intelligibility

but the voice will sound quite high pitched.

Then as the correct setting for the beat oscilla-

tor is approached the voice will begin to sound

natural but will have a background growl on

each syllable. At the correct frequency for the

beat oscillator the speech will clear complete-

ly and the voice will have a clean, crisp qual-

ity. It should also be mentioned that there is

a narrow region of tuning of the beat oscillator

a small distance on the wrong side of the side -

band where the voice will sound quite bassy

and difficult to understand.

With a little experience it will be possible

to identify the sound associated with improper

settings of the beat -oscillator control so that

corrections in the setting of the control can be

made. Note that the main tuning control of the

receiver is not changed after the sideband

once is tuned into the pass band of the re-

ceiver. All the fine tuning should be done with

the beat oscillator control. Also, it is very im-

portant that the r -f gain control be turned to

quite a low level during the tuning process.

Then after the signal has been tuned properly

the r -f gain may be increased for good signal

level, or until the point is reached where best

oscillator injection becomes insufficient and

the signal begins to distort.

Single -Sideband Greatly simplified tuning,

Receivers and coupled with strong atten-

Adopters uation of undesired sig-

nals, can be obtained

through the use of a single -sideband receiver

or receiver adapter. The exalted carrier prin-

ciple usually is employed in such receivers, with

a phase -sensitive system sometimes included

for locking the local oscillator to the frequency

of the carrier of the incoming signal. In order

for the locking system to operate, some carrier

must be transmitted along with the SSB signal.

Such receivers and adapters include a means

for selecting the upper or lower sideband by

the simple operation of a switch. For the re-

ception of a single -sideband signal the switch

obviously must be placed in the correct posi-

tion. But for the reception of a conventional

AM or phase- modulated signal, either sideband

may be selected, allowing the sideband with

the least interference to be used.

The Product Detector An unusually satis-

factory form of de-

modulator for SSB service is the product de-

tector, shown in one form in figure 41. This

circuit is preferred since it reduces intermodu-

lation products and does not require a large

local carrier voltage, as contrasted to the more

common diode envelope detector. This product

detector operates much in the same manner as

www.americanradiohistory.com

HANDBOOK S.S.B. Reception 349

605

MODULATORS

4 -250A

200 00=

.001 S4WS850

TURRET

10 N V. ANT.

250 1500

IS KV.

STANCOR

A-782

(!/SE PRIMARY

AS SECONDARY

O01

10Kv

200 - 4-250A h_

ES 4000 V.

Figure 43

HIGH -LEVEL DSB BALANCED MODULATOR

a multi -grid mixer tube. The SSB signal is

applied to the control grid of the tube and

the locally generated carrier is impressed upon

the other control grid. The desired audio out-

put signal is recovered across the plate resist-

ance of the demodulator tube. Since the cath-

ode current of the tube is controlled by the

simultaneous action of the two grids, the cur-

rent will contain frequencies equal to the sum

and difference between the sideband signal and

the carrier. Other frequencies are suppressed by

the low -pass r -f filter in the plate circuit of the

stage, while the audio frequency is recovered

from the i -f sideband signal.

17 -9 Double Sideband

Transmission

Many systems of intelligence transmission

lie in the region between amplitude modula-

tion on the one hand and single sideband sup-

pressed- carrier transmission on the other hand.

One system of interest to the amateur is the

Synchronous Communications System, popular-

ly known as "double sideband" (DSB -) trans-

mission, wherein a suppressed -carrier double

sideband signal is transmitted (figure 42) .

Reception of such a signal is possible by util-

izing a local oscillator phase -control system

which derives carrier phase information from

the sidebands alone and does not require the

use of any pilot carrier.

The DSB Transmitter A balanced modulator

of the type shown in

figure 8 may be employed to create a DSB

signal. For higher operating levels, a pair of

class -C type tetrode amplifier tubes may be

screen modulated by a push -pull audio system

and excited from a push -pull r -f source. The

plates of such a modulator are connected in

parallel to the tank circuit, as shown in figure

43. This DSB modulator is capable of 1 -kilo-

watt peak power output at a plate potential of

4000 volts. The circuit is self- neutralizing and

the tune -up process is much the same as with

any other class -C amplifier stage. As in the

case of SSB, the DSB signal may also be gen-

erated at a low level and amplified in linear

stages following the modulator.

Synchronous A DSB signal may be re-

Detection ceived with difficulty on a

conventional receiver, a n d

one of the two sidebands may easily be received

on a single sideband receiver. For best recep-

tion, however, a phase- locked local oscillator

and a synchronous detector should be em-

ployed. This operation may be performed eith-

er at the frequency of reception or at a con-

venient intermediate frequency. A block dia-

I -SYN-

CHRONOUS

DETECTOR

I Low

PASS

FILTER

E- DS 8

SIGNAL LOCAL

"OSCILLATOR

V0

PHASE

SHIFTER

0. -SYN-

CHRONOUS

DETECTOR

I-AUDIO

AMPLIFIER

FREQUENCY

CONTROL

Q -Low

PASS

FILTER

AUDIO

0- PHASE

DISCR IMIN.

Q -AUDIO

AMPLIFIER

AUDIO

AMPLIFIER

Figure 44

BLOCK DIAGRAM OF DSB

RECEIVING ADAPTER

www.americanradiohistory.com

350 Sideband Transmission

gram of a DSB synchronous receiver is shown

in figure 44. The DSB signal is applied to

two detectors having their local oscillator con-

version voltages in phase quadrature to each

other so that the audio contributions of the

upper and lower sidebands reinforce one an-

other. The in -phase oscillator voltage is ad-

justed to have the same phase as the suppressed

carrier of the transmitted signal. The I- ampli-

fier audio output, therefore, will contain the

demodulated audio signal, while the Q- ampli-

fier (supplied with quadrature oscillator volt-

age) will produce no output due to the quad-

rature null. Any frequency change of the local

oscillator will produce some audio output in

the Q- amplifier, while the I- amplifier is rela-

tively unaffected. The Q- amplifier audio will

have the same polarity as the I- channel audio

for one direction of oscillator drift, and oppo-

site polarity for oscillator drift in the opposite

direction. The Q- amplifier signal level is pro-

portional to the magnitude of the local oscilla-

tor phase angle error (the oscillator drift) for

small errors. By combining the I- signal and

the Q- signal in the audio phase discriminator

a d -c control voltage is developed which auto-

matically corrects for local oscillator phase er-

rors. The reactance tube therefore locks the

local oscillator to the correct phase. Phase con-

trol information is derived entirely from the

sideband component of the signal and the

carrier (if present) is not employed. Phase

control ceases with no modulation of the sig-

nal and is reestablished with the reappearance

of modulation.

Interference Rejection Interference falling

within the passband

of the receiver can be reduced by proper com-

bination of the I- and Q- audio signals. Under

phase lock conditions, the I- signal is com-

posed of the audio signal plus the undesired

interference, whereas the Q- signal contains

only the interference component. Phase can-

cellation obtained by combining the two sig-

nals will reduce the interference while still

adding the desired information contained in

both side -bands. The degree of interference re-

jection is dependent upon the ratio of inter-

ference falling upon the two sidebands of the

received signal and upon the basic design of

the audio networks. A schematic and descrip-

tion of a complete DSB receiving adapter is

shown in the June, 1957 issue of CQ maga-

zine.

17 -10 The Beam

Deflection Modulator

A recent development in the single side -

band field is the beam deflection tube (type

7360). This miniature tube employs a simple

electron "gun" which generates, controls, and

accelerates a beam of electrons directed toward

identical plates. The total plate current is de-

termined by the voltages applied to the control

grid and screen grid of the "gun ". The divi-

sion of plate current between the two plates

is determined by the difference in voltage be-

tween two deflecting electrodes placed between

the "gun" and the plates. R.f. voltage is used

to modulate the control grid of the electron

"gun" and the electron stream within the tube

may be switched between the plates by means

of an audio signal applied to the deflecting

electrodes. The 7360 makes an excellent

balanced modulator (figure 45) or product

detector having high impedance input

circuits, low distortion, and excellent carrier

suppression.

.00, I.

SO

R.F

IN 470 K

7360

- PUSH-PULL AUDIO 1.265 V.

Figure 45.

BALANCED MODULATOR CIRCUIT

USING 7360 BEAM DEFLECTION TUBE.

www.americanradiohistory.com

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www.americanradiohistory.com

CHAPTER EIGHTEEN

Transmitter Design

The excellence of a transmitter is a func-

tion of the design, and is dependent upon the

execution of the design and the proper choice

of components. This chapter deals with the

study of transmitter circuitry and of the basic

components that go to make up this circuitry.

Modern components are far from faultless. Re-

sistors have inductance and distributed ca-

pacity. Capacitors have inductance and re-

sistance, and inductors have resistance and

distributed capacity. None of these residual

attributes show up on circuit diagrams, yet

they are as much responsible for the success

or failure of the transmitter as are the neces-

sary and vital bits of resistance, capacitance

and inductance. Because of these unwanted

attributes, the job of translating a circuit on

paper into a working piece of equipment often

becomes an impossible task to those individ-

uals who disregard such important trivia.

Rarely do circuit diagrams show such pitfalls

as ground loops and residual inductive cou-

pling between stages. Parasitic resonant cir-

cuits are rarely visible from a study of the

schematic. Too many times radio equipment is

rushed into service before it has been entirely

checked. The immediate and only too apparent

results of this enthusiasm are transmitter in-

stability, difficulty of neutralization, r.f. wan-

dering all over the equipment, and a general

"touchiness" of adjustment. Hand in glove

with these problems go the more serious ones

of TVI, key -clicks, and parasitics. By paying

352

attention to detail, with a good working know-

ledge of the limitations of the components,

and with a basic conception of the actions of

ground currents, the average amateur will be

able to build equipment that will work "just

like the book says."

The twin problems of TVI and parasitics

are an outgrowth of the major problem of over-

all circuit design. If close attention is paid

to the cardinal points of circuitry design, the

secondary problems of TVI and parasitics

will in themselves be solved.

18-1 Resistors

The resistance of a conductor is a function

of the material, the form the material takes,

the temperature of operation, and the frequen-

cy of the current passing through the resist-

ance. In general, the variation in resistance

due to temperature is directly proportional to

the temperature change. With most wire -wound

resistors, the resistance increases with tem-

perature and returns to its original value when

the temperature drops to normal. So- called

composition or carbon resistors have less

reliable temperature /resistance characteris-

tics. They usually have a positive tempera-

ture coefficient, but the retrace curve as the

resistor is cooled is often erratic, and in

www.americanradiohistory.com

Resistors 353

+3

+4

+3

+2

Z

Ñ +1

N_

UI 0

w z -I

W 2

2 i -3

U

-S -30

R A

-20 - 0 O 10 20 30 40 50 SO

DEGREES CENTIGRADE 70 !O

HEAT CYCLE OF UNCONDITIONED

COMPOSITION RESISTORS

SO 100

+!

+4

0

2

Figure 1

b -20 - O 0 10 20 30 40 50 !O

DEGREES CENTIGRADE

70 SO

HEAT CYCLE OF CONDITIONED

COMPOSITION RESISTORS

SO 100

many cases the resistance does not return to

its original value after a heat cycle. It is for

this reason that care must be taken when sol-

dering composition resistors in circuits that

require close control of the resistance value.

Matched resistors used in phase- inverter serv-

ice can be heated out of tolerance by the act

of soldering them into the circuit. Long leads

should be left on the resistors and a long -

nose pliers should grip the lead between the

iron and the body of the resistor to act as a

heat block. General temperature character-

istics of typical carbon resistors are shown

in figure 1. The behavior of an individual re-

sistor will vary from these curves depending

upon the manufacturer, the size and wattage

of the resistor, etc.

Inductance of Every resistor because of its

Resistors physical size has in addition

to its desired resistance, less

desirable amounts of inductance and distrib-

uted capacitance. These quantities are illus-

trated in figure 2A, the general equivalent cir-

cuit of a resistor. This circuit represents the

actual impedance network of a resistor at any

frequency. At a certain specified frequency

Figure 2

Roc L

-Ms--1001

C- 6-O

EQUIVALENT CIRCUIT OF A RESISTOR

AT A PARTICULAR FREQUENCY

a ``IZEIII--

_"` ,__-_-` _-_"mg_

` __-=_- `- _

,--`

M.

6

A

3

2

0

S IO

FREQUENCY (MC.)

Figure 3

FREQUENCY EFFECTS ON SAMPLE

COMPOSITION RESISTORS

IS

www.americanradiohistory.com

354 Transmitter Design THE RADIO

R=55000n .

so

s WEI! -""

i-1 I

xo =MINIM

5 i0

FREQUENCY (MC.)

Figure 4

CURVES OF THE IMPEDANCE OF WIRE -

WOUND RESISTORS AT RADIO

FREQUENCIES

20

the impedance of the resistor may be thought

of as a series reactance (X,) as shown in fig-

ure 2B. This reactance may be either induc-

tive or capacitive depending upon whether the

residual inductance or the distributed capaci-

tance of the resistor is the dominating factor.

As a rule, skin effect tends to increase the

reactance with frequency, while the capacity

between turns of a wire -wound resistor, or ca-

pacity between the granules of a composition

resistor tends to cause the reactance and re-

sistance to drop with frequency. The behavior

of various types of composition resistors over

a large frequency range is shown in figure 3.

By proper component design, non -inductive re-

sistors having a minimum of residual react-

ance characteristics may be constructed. Even

these have reactive effects that cannot be ig-

nored at high frequencies.

Wirewound resistors act as low -Q inductors

at radio frequencies. Figure 4 shows typical

curves of the high frequency characteristics

of cylindrical wirewound resistors. In addition

to resistance variations wirewound resistors

exhibit both capacitive and inductive react-

ance, depending upon the type of resistor and

the operating frequency. In fact, such resis-

tors perform in a fashion as low -Q r -f chokes

below their parallel self -resonant frequency.

18-2 Capacitors

The inherent residual characteristics of ca-

pacitors include series resistance, series in-

ductance and shunt resistance, as shown in

figure 5. The series resistance and inductance

RsHUN'

M;

o---

C L RSERIES

---

Figure 5

EQUIVALENT CIRCUIT OF A CAPACITOR

depend to a large extent upon the physical

configuration of the capacitor and upon the

material of which it is made. Of great interest

to the amateur constructor is the series in-

ductance of the capacitor. At a certain fre-

quency the series inductive reactance of the

capacitor and the capacitive reactance are

equal and opposite, and the capacitor is in

itself series resonant at this frequency. As

the operating frequency of the circuit in which

the capacitor is used is increased above the

series resonant frequency, the effectiveness

of the capacitor as a by- passing element de-

teriorates until the unit is about as effective

as a block of wood.

By -Pass The usual forms of by -pass ca-

Capacitors pacitors have dielectrics of paper,

mica, or ceramic. For audio work,

and low frequency r -f work up to perhaps 2 Mc.

or so, the paper capacitors are satisfactory

as their relatively high internal inductance has

little effect upon the proper operation of the

circuit. The actual amount of internal induct-

ance will vary widely with the manufacturing

process, and some types of paper capacitors

have satisfactory characteristics up to a fre-

quency of 5 Mc. or so.

When considering the design of transmitting

equipment, it must be remembered that while

the transmitter is operating at some relatively

low frequency of, say, 7 Mc., there will be

harmonic currents flowing through the various

by -pass capacitors of the order of 10 to 20

times the operating frequency. A capacitor

that behaves properly at 7 Mc. however, may

offer considerable impedance to the flow of

these harmonic currents. For minimum har-

monic generation and radiation, it is obviously

of greatest importance to employ by-pass ca-

pacitors having the lowest possible internal

inductance.

Mica dielectric capacitors have much less

internal inductance than do most paper con-

densers. Figure 6 lists self- resonant frequen-

cies of various mica capacitors having various

lead lengths. It can be seen from inspection

of this table that most mica capacitors be-

come self- resonant in the 12 -Mc. to 50 -Mc.

region. The inductive reactance they would

offer to harmonic currents of 100 Mc. or so

www.americanradiohistory.com

HANDBOOK Capacitors 355

CONDENSER LEAD LENGTHS RESONANT FREQ.

.02 LF MICA NONE 44.5 MC.

.002 OF MICA NONE 23.5 MC.

.01 LF MICA f 10 MC.

.000911F MICA f 55 MC.

.002 LF CERAMIC f 24 MC

.001 LF CERAMIC T 55 MC.

500 1111F BUTTON NONE 220 MC.

.001 LF CERAMIC f NO MC.

.01 1JF CERAMIC f 14.5 MC.

Figure 6

SELF -RESONANT FREQUENCIES OF

VARIOUS CAPACITORS WITH

RANDOM LEAD LENGTH

would be of considerable magnitude. In certain

instances it is possible to deliberately series -

resonate a mica capacitor to a certain frequen-

cy somewhat below its normal self -resonant

frequency by trimming the leads to a critical

length. This is sometimes done for maximum

by- passing effect in the region of 40 Mc. to

60 Mc.

The recently developed button -mica capaci-

tors shown in figure 7 are especially designed

to have extremely low internal inductance.

Certain types of button -mica capacitors of

small physical size have a self -resonant fre-

quency in the region of 600 Mc.

Ceramic dielectric capacitors in general

have the lowest amount of series inductance

per unit of capacitance of these three univer-

sally used types of by -pass capacitors. Typi-

cal resonant frequencies of various ceramic

units are listed in figure 6. Ceramic capaci-

tors are available in various voltage and capa-

city ratings and different physical configura-

tions. Stand -off types such as shown in figure

7 are useful for by- passing socket and trans-

former terminals. Two of these capacitors may

be mounted in close proximity on a chassis

and connected together by an r -f choke to form

a highly effective r -f filter. The inexpensive

"clamshell" type of ceramic capacitor is

recommended for general by- passing in r -f cir-

cuitry, as it is effective as a by -pass unit to

well over 100 Mc.

The large TV "doorknob" capacitors are

useful as by -pass units for high voltage lines.

These capacitors have a value of 500 micro -

microfarads, and are available in voltage rat-

ings up to 40,000 volts. The dielectric of

these capacitors is usually titanium- dioxide.

This material exhibits piezo -electric effects,

and capacitors employing it for a dielectric

will tend to "talk- back" when a -c voltages

are applied across them. When these capaci-

tors are used as plate bypass units in a modu-

lated transmitter they will cause acoustical

noise. Otherwise they are excellent for gen-

eral r -f work.

A recent addition to the varied line of ca-

pacitors is the coaxial or " Hypass" type of

capacitor. These capacitors exhibit superior

by- passing qualities at frequencies up to 200

Mc. and the bulkhead type are especially ef-

fective when used to filter leads passing

through partition walls between two stages.

Variable Air Even though air is the perfect

Capacitors dielectric, air capacitors exhibit

losses because of the inherent

resistance of the metallic parts that make up

the capacitor. In addition, the leakage loss

across the insulating supports may become of

some consequence at high frequencies. Of

greater concern is the inductance of the ca-

pacitor at high frequencies. Since the capaci-

tor must be of finite size, it will have tie -rods

and metallic braces and end plates, all of which

contribute to the inductance of the unit. The

actual amount of the inductance will depend

upon the physical size of the capacitor and

the methbd used to make contact to the stator

and rotor plates. This inductance may be cut

to a minimum value by using as small a ca-

pacitor as is practical, by using insulated tie -

rods to prevent the formation of closed induc-

tive loops in the frame of the unit, and by

making connections to the centers of the plate

assemblies rather than to the ends as is com-

.i

y

d

Figure 7

TYPES OF CERAMIC AND MICA CAPACI-

TORS SUITABLE FOR HIGH -FREQUENCY

BYPASSING

The Centralab 858S (1000 mad) is recoin.

mended for screen and plate circuits of tet-

rode tubes.

www.americanradiohistory.com

356 Transmitter Design THE RADIO

monly done. A large transmitting capacitor

may have an inherent inductance as large as

0.1 microhenry, making the capacitor suscep-

tible to parasitic resonances in the 50 Mc. to

150 Mc. range of frequencies.

The question of optimum C/L ratiq and ca-

pacitor plate spacing is covered in Chapter

Thirteen. For all -band operation of a high power

stage, it is recommended that a capacitor just

large enough for 40 -meter phone operation be

chosen. (This will have sufficient capacitance

for phone operation on all higher frequency

bands.) Then use fixed padding capacitors for

operation on 80 meters. Such padding capaci-

tors are available in air, ceramic, and vacuum

types.

Specially designed variable capacitors are

recommended for u -h -f work; ordinary capaci-

tors often have "loops" in the metal frame

which may resonate near the operating fre-

quency.

Variable Vacuum Variable vacuum capacitors

Capacitors because of their small phy-

sical size have less inher-

ent inductance per unit of capacity than do

variable air capacitors. Their losses are ex-

tremely low, and their dielectric strength is

high. Because of increased production the

cost of such units is now within the reach of

the designer of amateur equipment, and their

use is highly recommended in high power tank

circuits.

18 -3 Wire and Inductors

Any length of wire, no matter how short,

has a certain value of inductance. This prop-

erty is of great help in making coils and in-

ductors, but may be of great hindrance when

it is not taken into account in circuit design

and construction. Connecting circuit elements

(themselves having residual inductance) to-

gether with a conductor possessing additional

inductance can often lead to puzzling difficul-

ties. A piece of no. 10 copper wire ten inches

long (a not uncommon length for a plate lead

in a transmitter) can have a self- inductance of

0.15 microhenries. This inductance and that

of the plate tuning capacitor together with the

plate -to- ground capacity of the vacuum tube

can form a resonant circuit which may lead to

parasitic oscillations in the v -h -f regions. To

keep the self- inductance at a minimum, all r -f

carrying leads should be as short as possible

and should be made out of as heavy material

as possible.

At the higher frequencies, solid enamelled

copper wire is most efficient for r -f leads.

Tinned or stranded wire will show greater

losses at these frequencies. Tank coil and

tank capacitor leads should be of heavier wire

than other r -f leads.

The best type of flexible lead from the en-

velope of a tube to a terminal is thin copper

strip, cut from thin sheet copper. Heavy, rigid

leads to these terminals may crack the enve-

lope glass when a tube heats or cools.

Wires carrying only a.f. or d.c. should be

chosen with the voltage and current in mind.

Some of the low- filament- voltage transmitting

tubes draw heavy current, and heavy wire must

be used to avoid voltage drop. The voltage is

low, and hence not much insulation is required.

Filament and heater leads are usually twisted

together. An initial check should be made on

the filament voltage of all tubes of 25 watts

or more plate dissipation rating. This voltage

should be measured right at the tube sockets.

If it is low, the filament transformer voltage

should be raised. If this is impossible, heavier

or parallel wires should be used for filament

leads, cutting down their length if possible.

Coaxial cable may be used for high voltage

leads when it is desirable to shield them from

r -f fields. RG -8 /U cable may be used at d -c

potentials up to 8000 volts, and the lighter

RG -17 /U may be used to potentials of 3000

volts. Spark -plug type high- tension wire may

be used for unshielded leads, and will with-

stand 10,000 volts.

If this cable is used, the high- voltage leads

may be cabled with filament and other low -

voltage leads. For high -voltage leads in low -

poc/ r exciters, where the plate voltage is not

over 450 volts, ordinary radio hookup wire of

good quality will serve the purpose.

No r -f leads should be cabled; in fact it is

better to use enamelled or bare copper wire

for r -f leads and rely upon spacing for insula-

tion. All r -f joints should be soldered, and the

joint should be a good mechanical junction

before solder is applied.

The efficiency and Q of air coils commonly

used in amateur equipment is a factor of the

shape of the coil, the proximity of the coil to

other objects (including the coil form) and the

material of which the coil is made. Dielectric

losses in so- called "air wound" coils are

low and the Q of such coils runs in the neigh-

borhood of 300 to 500 at medium frequencies.

Unfortunately, most of the transmitting type

plug -in coils on the market designed for link

coupling have far too small a pick up link for

proper operation at 7 Mc. and 3.5 Mc. The co-

efficient of coupling of these coils is about

0.5, and additional means must be employed

to provide satisfactory coupling at these low

frequencies. Additional inductance in series

with the pick up link, the whole being reso-

www.americanradiohistory.com

HANDBOOK Inductors 357

Rc L Rc L

C DISTRIBUTED

Rc C ¡_ L

"°-z'

Figure 8

ELECTRICAL EQUIVALENT OF R -F CHOKE AT VARIOUS FREQUENCIES

nated to the operating frequency will often

permit satisfactory coupling.

Coil Placement For best Q a coil should be

in the form of a solenoid with

length from one to two times the diameter. For

minimum interstage coupling, coils should be

made as small physically as is practicable.

The coils should then be placed so that ad-

joining coils are oriented for minimum mutual

coupling. To determine if this condition exists,

apply the following test: the axis of one of the

two coils must lie in the plane formed by the

center turn of the other coil. If this condition

is not met, there will be appreciable coupling

unless the unshielded coils are very small in

diameter or are spaced a considerable dis-

tance from each other.

Insulation On frequencies above 7 Mc., cera-

mic, polystyrene, or blycalex in-

sulation is to be recommended. Cold flow must

be considered when using polystyrene (Am-

phenol 912, etc.). Bakelite has low losses on

the lower frequencies but should never be

used in the field of high- frequency tank cir-

cuits.

Lucite (or Plexiglas), which is available

in rods, sheets, or tubing, is satisfactory for

use at all radio frequencies where the r -J volt-

ages are not especially high. It is very easy

to work with ordinary tools and is not expen-

sive. The loss factor depends to a consider-

able extent upon the amount and kind of plas-

ticizer used.

The most important thing to keep in mind

regarding insulation is that the best insulation

is air. If it is necessary to reinforce air -wound

coils to keep turns from vibrating or touching,

use strips of Lucite or polystyrene cemented

in place with Amphenol 912 coil dope. This

will result in lower losses than the commonly

used celluloid ribs and Duco cement.

Radio Frequency R -f chokes may be consid-

Chokes ered to be special induct-

ances designed to have a

high value of impedance over a large range of

frequencies. A practical r -f choke has induct-

ance, distributed capacitance, and resistance.

At low frequencies, the distributed capacity

has little effect and the electrical equivalent

circuit of the r -f choke is as shown in figure

8A. As the operating frequency of the choke

is raised the effect of the distributed capacity

becomes more evident until at some particular

frequency the distributed capacity resonates

with the inductance of the choke and a parallel

resonant circuit is formed. This point is shown

in figure 8B. As the frequency of operation

is further increased the overall reactance of

the choke becomes capacitive, and finally a

point of series resonance is reached (figure

8C.). This cycle repeats itself as the operating

frequency is raised above the series resonant

point, the impedance of the choke rapidly be-

coming lower on each successive cycle. A

chart of this action is shown in figure 9. It

can be seen that as the r -f choke approaches

and leaves a condition of series resonance,

the performance of the choke is seriously im-

paired. The condition of series resonance

may easily be found by shorting the terminals

of the r -f choke in question with a piece of

wire and exploring the windings of the choke

with a grid -dip oscillator. Most commercial

transmitting type chokes have series reso-

nances in the vicinity of 11 Mc. or 24 Mc.

p.o 111111111V".

1111111/11116/11

, :ill=

/ /Iv

E

5 IS 20

FREQUENCY (MC.)

25

Figure 9

FREQUENCY- IMPEDANCE CHARACTERIS-

TICS FOR TYPICAL PIE -WOUND

R -F CHOKES

www.americanradiohistory.com

358 Transmitter Design THE RADIO

LEAD

NDUCTANCE

Figure 10

GROUND LOOPS IN AMPLIFIER STAGES

A. Using chassis return

8. Common ground point

18 -4 Grounds

At frequencies of 30 Mc. and below, a chas-

sis may be considered as a fixed ground refer-

ence, since its dimensions are only a fraction

of a wavelength. As the frequency is increased

above 30 Mc., the chassis must be considered

as a conducting sheet on which there are

points of maximum current and potential. How-

ever, for the lower amateur frequencies, an

object may be assumed to be at ground poten-

tial when it is affixed to the chassis.

In transmitter stages, two important current

loops exist. One loop consists of the grid cir-

cuit and chassis return, and the other loop

consists of the plate circuit and chassis re-

turn. These two loops are shown in figure 10A.

It can be seen that the chassis forms a return

for both the grid and plate circuits, and that

ground currents flow in the chassis towards

the cathode circuit of the stage. For some

years the theory has been to separate these

ground currents from the chassis by returning

all ground leads to one point, usually the

cathode of the tube for the stage in question.

This is well and good if the ground leads are

of minute length and do not introduce cross

couplings between the leads. Such a technique

is illustrated in figure 1013. wherein all stage

components are grounded to the cathode pin

of the stage socket. However, in transmitter

construction the physical size of the compon-

ents prevent such close grouping. It is nec-

essary to spread the components of such a

stage over a fairly large area. In this case it

is best to ground items directly to the chassis

at the nearest possible point, with short, direct

grounding leads. The ground currents will

flow from these points through the low induct-

ance chassis to the cathode return of the

stage. Components grounded on the top of the

chassis have their ground currents flow through

holes to the cathode circuit which is usually

located on the bottom of the chassis, since

such currents travel on the surface of the chas-

sis. The usual "top to bottom" ground path

is through the hole cut in the chassis for the

tube socket. When the gain per stage is rela-

tively low, or there are only a small number

of stages on a chassis this universal ground-

ing system is ideal. It is only in high gain

stages (i -f strips) where the "gain per inch"

is very high that circulating ground currents

will cause operational instability.

Intercoupling of It is important to prevent in-

Ground Currents tercoupling of various differ-

ent ground currents when the

chassis is used as a common ground return.

To keep this intercoupling at a minimum, the

stage should be completely shielded. This

will prevent external fields from generating

spurious ground currents, and prevent the

ground currents of the stage from upsetting

the action of nearby stages. Since the ground

currents travel on the surface of the metal,

the stage should be enclosed in an electrically

tight box. When this is done, all ground cur-

rents generated inside the box will remain in

the box. The only possible means of escape

for fundamental and harmonic currents are im-

perfections in this electrically tight box. When-

ever we bring a wire lead into the box, make

a ventilation hole, or bring a control shaft

through the box we create an imperfection. It

is important that the effect of these imperfec-

tions be reduced to a minimum.

18 -5 Holes, Leads and Shafts

Large size holes for ventilation may be put

in an electrically tight box provided they are

properly screened. Perforated metal stock hav-

ing many small, closely spaced holes is the

best screening material. Copper wire screen

may be used provided the screen wires are

bonded together every few inches. As the wire

corrodes, an insulating film prevents contact

between the individual wires, and the attenua-

tion of the screening suffers. The screening

material should be carefully soldered to the

www.americanradiohistory.com

HANDBOOK Shielding 359

TIN CAN BOTTOM WITH

FLUTED EDGE PRESSED

AGAINST PANEL

RUBBER GROMMET

METER NUT

.001 CERAMIC

RFC

,OOI CERAMIC

PANEL

METER

LEAD

HOLES FOR

METER STUDS

PLUTEO EDGES TO MAKE

000D ELECTRICAL CON-

TACT WITH PANEL

Figure 11A

SIMPLE METER SHIELD

box, or bolted with a spacing of not less than

two inches between bolts. Mating surfaces of

the box and the screening should be clean.

A screened ventilation opening should be

roughly three times the size of an equivalent

unscreened opening, since the screening rep-

resents about a 70 per cent coverage of the

area. Careful attention must be paid to equip-

ment heating when an electrically tight box

is used.

Commercially available panels having half -

inch ventilating holes may be used as part of

the box. These holes have much less attenua-

tion than does screening, but will perform in a

satisfactory manner in all but the areas of

weakest TV reception. If it is desired to re-

duce leakage from these panels to a minimum,

the back of the grille must be covered with

screening tightly bonded to the panel.

Doors may be placed in electrically tight

boxes provided there is no r -f leakage around

the seams of the door. Electronic weather-

stripping or metal "finger stock" may be used

to seal these doors. A long, narrow slot in a

closed box has the tendency to act as a slot

antenna and harmonic energy may pass more

readily through such an opening than it would

through a much larger circular hole.

Variable capacitor shafts or switch shafts

may act as antennas, picking up currents in-

side the box and re- radiating them outside of

the box. It is necessary either to ground the

shaft securely as it leaves the box, or else to

make the shaft of some insulating material.

A two or three inch panel meter requires a

large leakage hole if it is mounted in the wall

of an electrically tight box. To minimize leak -

age, the meter leads should be by- passed and

shielded. The meter should be encased with a

metal shield that makes contact to the box

entirely around the meter. The connecting

studs of the meter may project through the

back of the metal shield. Such a shield may

be made out of the end of a tin can of correct

COAXIAL SOCKET

COAXIAL PLUG

EXTERNAL FIELD

CENTER

CONDUCTOR .

HOLE

RIGHT

-OREN- BOX

EXTERNAL

FIELD

INTERNAL GROUND

CURRENTS

[[ LL (( J

OÑ EXTE0.1EOR ÓF BO1AlL CURRENTS

WRONG

Figure 11B

Use of coaxial connectors on electrically

tight box prevents escape of ground currents

from interior of box. At the same time exter-

nal fields are not conducted into the interior

of the box.

diameter, cut to fit the depth of the meter.

This complete shield assembly is shown in

figure 11A.

Careful attention should be paid to leads

entering and leaving the electrically tight

box. Harmonic currents generated within the

box can easily flow out of the box on power

or control leads, or even on the outer shields

of coaxially shielded wires. Figure 11B illus-

trates the correct method of bringing shielded

cables into a box where it is desired to pre-

serve the continuity of the shielding.

Unshielded leads entering the box must be

carefully filtered to prevent fundamental and

harmonic energy from escaping down the lead.

Combinations of r -f chokes and low inductance

by -pass capacitors should be used in power

leads. If the current in the lead is high, the

chokes must be wound of large gauge wire.

Composition resistors may be substituted for

the r -f chokes in high impedance circuits.

Bulkhead or feed - through type capacitors are

preferable when passing a lead through a

shield partition. A summary of lead leakage

with various filter arrangements is shown in

figure 12.

Internal Leads Leads that connect two points

within an electrically tight box

www.americanradiohistory.com

360 Transmitter Design THE RADIO

FIELD STRENGTH

TEST IN UV SHIELDED OSCILLATOR

NO.

I 12000 ,o-SMALL HOLE IN SHIELD TO

B I OSC.

CI

2 10000

3 630

600

S 150

6 70

7 140

600

110

10 SO

25

12 TRACE

WELDED HOOK-UP WIRE

C2 RI _2

c. RFC

C RiC CI RFC

x

C4 RFC 4 RFC _

J.c3 IEID D WZRE

L--_1---- r-i

R I z

R I - MOOR CARBON

RFC-OHMITE 2-50

CI- 75MLFCERAMIC

FEED-THROUGH

C2 -.005 DISC CERAMIC

C3 - .01 SPRAGUE HI -PASS

C4 - 005 CERAMIC

FEED -THROUGH

Figure 12

LEAD LEAKAGE WITH VARIOUS

LEAD FILTERING SYSTEMS

(COURTESY WIDBM)

may pick up fundamental and harmonic cur-

rents if they are located in a strong field of

flux. Any lead forming a closed loop with it-

self will pick up such currents, as shown in

figure 13. This effect is enhanced if the lead

happens to be self -resonant at the frequency

at which the exciting energy is supplied. The

solution for all of this is to by -pass all inter-

nal power leads and control leads at each

end, and to shield these leads their entire

length. All filament, bias, and meter leads

should be so treated. This will make the job

of filtering the leads as they leave the box

much easier, since normally "cool" leads

within the box will not have picked up spur-

ious currents from nearby "hot" leads.

18 -6 Parasitic Resonances

Filament leads within vacuum tubes may

resonate with the filament by -pass capacitors

at some particular frequency and cause insta-

bility in an amplifier stage. Large tubes of

the 810 and 250TH type are prone to this spur-

ious effect. In particular, a push -pull 810 am-

plifier using .001 -µfd. filament by -pass capac-

itors had a filament resonant loop that fell in

the 7 -Mc. amateur band. When the amplifier

was operated near this frequency marked in-

stability was noted, and the filaments of the

810 tubes increased in brilliance when plate

voltage was applied to the amplifier, indicat-

ing the presence of r.f. in the filament circuit.

Changing the filament by -pass capacitors to

.01-pfd. lowered the filament resonance fre-

quency to 2.2 Mc. and cured this effect. A

ceramic capacitor of .01 -pfd. used as a fila-

ment by -pass capacitor on each filament leg

seems to be satisfactory from both a resonant

and a TVI point of view. Filament by -pass

capacitors smaller in value than .01-pfd.

should be used with caution.

Various parasitic resonances are also found

in plate and grid tank circuits. Push -pull tank

circuits are prone to double resonances, as

shown in figure 14. The parasitic resonance

circuit is usually several megacycles higher

than the actual resonant frequency of the full

tank circuit. The cure for such a double reso-

nance is the inclusion of an r -f choke in the

center tap lead to the split coil.

Chassis Material From a point of view of elec-

trical properties, aluminum

is a poor chassis material. It is difficult to

make a soldered joint to it, and all grounds

must rely upon a pressure joint. These pres-

Figure 13

SHIELDED COMPARTMENT

RADIATION

FIELD \ \.

SHIELDED COMPARTMENT

RE-RADIATED

HOLE 1 FIELD

ICKU' RADIATION

LOOP LOOP

BY-PASS

CAPACITOR BY -PASS

CAPACITOR

WRONG

ILLUSTRATION OF HOW A SUPPOSEDLY

GROUNDED POWER LEAD CAN COUPLE

ENERGY FROM ONE COMPARTMENT

TO ANOTHER

R IGHT

LECTRICALLY -TIGHT ELECTRICALLY -TIGHT

OMPARTMENT COMPARTMENT

RADIATION

FIELD \ BULKHEAD TYPE

\ fBY -PASS CAPACITOR

ILLUSTRATION OF LEAD ISOLATION BY

PROPER USE OF BULKHEAD BYPASS

CAPACITOR

www.americanradiohistory.com

HANDBOOK Parasitic Oscillations 361

WRONG RIGHT

Figure 14

DOUBLE RESONANCE EFFECTS IN PUSH -

PULL TANK CIRCUIT MAY BE ELIMI-

NATED BY THE INSERTION OF ANY

R -F CHOKE IN THE COIL CENTER

TAP LEAD

sure joints are prone to give trouble at a later

date because of high resistivity caused by

the formation of oxides from eletrolytic action

in the joint. However, the ease of working

and forming the aluminum material far out-

weighs the electrical shortcomings, and alu-

minum chassis and shielding may be used

with good results provided care is taken in

making all grounding connections. Cadmium

and zinc plated chassis are preferable from a

corrosion standpoint, but are much more diffi-

cult to handle in the home workshop.

18 -7 Parasitic Oscillation

in R -F Amplifiers

Parasitics (as distinguished from sell- oscil-

lation on the normal tuned frequency of the

amplifier) are undesirable oscillations either

of very high or very low frequencies which

may occur in radio -frequency amplifiers.

They may cause spurious signals (which

are often rough in tone) other than normal har-

monics, hash on each side of a modulated car-

rier, key clicks, voltage breakdown or flash-

over, instability or inefficiency, and shortened

life or failure of the tubes. They may be damped

and stop by themselves after keying or modu-

lation peaks, or they may be undamped and

build up during ordinary unmodulated trans-

mission, continuing if the excitation is re-

moved. They may result from series or par-

allel resonant circuits of all types. Due to neu-

tralizing lead length and the nature of most

parasitic circuits, the amplifier usually is not

neutralized for the parasitic frequency.

Sometimes the fact that the plate supply is

keyed will obscure parasitic oscillations in a

final amplifier stage that might be very severe

if the plate voltage were left on and the exci-

tation were keyed.

In some cases, an all -wave receiver will

prove helpful in locating v -h -f spurious oscil-

lations, but it may be necessary to check from

several hundred megacycles downward in fre-

quency to the operating range. A normal har-

monic is weaker than the fundamental but of

good tone; a strong harmonic or a rough note

at any frequency generally indicates a para-

sitic. In general, the cure for parasitic oscillation

is two-fold: The oscillatory circuit is damped

until sustained oscillation is impossible, or

it is detuned until oscillation ceases. An ex-

amination of the various types of parasitic os-

cillations and of the parasitic oscillatory cir-

cuits will prove handy in applying the correct

cure.

Low Frequency One type of unwanted

Parasitic Oscillations oscillation often occurs

in shunt -fed circuits in

which the grid and plate chokes resonate, cou-

pled through the tube's interelectrode capaci-

tance. This also can happen with series feed.

This oscillation is generally at a much lower

frequency than the operating frequency and

will cause additional carriers to appear, spaced

from perhaps twenty to a few hundred kilo-

cycles on either side of the main wave. Such a

circuit is illustrated in figure 15. In this case,

RFC, and RFC2 form the grid and plate induct-

ances of the parasitic oscillator. The neutral-

izing capacitor, no longer providing out -of-

phase feedback to the grid circuit actually en-

hances the low frequency oscillation. Because

of the low Q of the r -f chokes, they will usu-

ally run warm when this type of parasitic os-

cillation is present and may actually char and

burn up. A neon bulb held near the oscillatory

circuit will glow a bright yellow, the color

appearing near the glass of the neon bulb and

not between the electrodes.

One cure for this type of oscillation is to

change the type of choke in either the plate

or the grid circuit. This is a marginal cure,

because the amplifier may again break into the

same type of oscillation when the plate volt-

age is raised slightly. The best cure is to re-

move the grid r -f choke entirely and replace

it with a wirewound resistor of sufficient watt-

age to carry the amplifier grid current. If the

inclusion of such a resistor upsets the operat-

ing bias of the stage, an r -f choke may be

used, with a 100 -ohm 2 -watt carbon resistor

in series with the choke to lower the operating

Q of the choke. If this expedient does not

eliminate the condition, and the stage under

investigation uses a beam -tetrode tube, nega-

tive resistance can exist in the screen circuit

www.americanradiohistory.com

362 Transmitter Design THE RADIO

2

RFC RFC2

R F. CIRCUIT

RFC!

GRID

TANIÇ

RFCz

vLATE

PARASITIC CIRCUIT FOR

LOW FREQ. OSCILLATION

Figure 15

THE CAUSE AND CURE OF LOW FREQUENCY PARASITICS

CURE

of such tubes. Try larger and smaller screen

by -pass capacitors to determine whether or not

they have any effect. If the condition is corn-

ing from the screen circuit an audio choke with

a resistor across it in series with the screen

feed lead will often eliminate the trouble.

Low -frequency parasitic oscillations can

often take place in the audio system of an AM

transmitter, and their presence will not be

known until the transmitter is checked on a

receiver. It is easy to determine whether or

not the oscillations are coming from the modu-

lator simply by switching off the modulator

tubes. If the oscillations are coming from the

modulator, the stage in which they are being

generated can be determined by removing tubes

successively, starting with the first speech

amplification stage, until the oscillation stops.

When the stage has been found, remedial steps

can be taken on that stage.

If the stage causing the oscillation is a low-

level speech stage it is possible that the

trouble is coming from r -f or power- supply

feedback, or it may be coming about as a re-

sult of inductive coupling between two trans-

formers. If the oscillation is taking place in

a high -level audio stage, it is possible that

inductive or capacitive coupling is taking

place back to one of the low -level speech

stages. It is also possible, in certain cases,

that parasitic push -pull oscillation can take

place in a Class B or Class AB modulator as

a result of the grid -to -plate capacitance with-

in the tubes and in the stage wiring. This con-

dition is more likely to occur if capacitors

have been placed across the secondary of the

driver transformer and across the primary of

the modulation transformer to act in the reduc-

tion of the amplitude of the higher audio fre-

quencies. Relocation of wiring or actual neu-

tralization of the audio stage in the manner

used for r -f stages may be required.

It may be said in general that the presence

of low- frequency parasitics indicates that

somewhere in the oscillating circuit there is

an impedance which is high at a frequency in

the upper audio or low r -f range. This imped-

ance may include one or more r -f chokes of

the conventional variety, power supply chokes,

modulation components, or the high impedance

may be presented simply by an RC circuit

such as might be found in the screen -feed cir-

cuit of a beam -tetrode amplifier stage.

18 -8 Elimination of V -H -F

Parasitic Oscillations

V -h -f parasitic oscillations are often diffi-

cult to locate and difficult to eliminate since

their frequency often is only moderately above

the desired frequency of operation. But it may

be said that v -h -f parasitics always may be

eliminated if the operating frequency is appre-

ciably below the upper frequency limit for the

tubes used in the stage. However, the elimi-

nation of a persistent parasitic oscillation on

a frequency only moderately higher than the

desired operating frequency will involve a

sacrifice in either the power output or the

power sensitivity of the stage, or in both.

Beam- tetrode stages, particularly those

using 807 type tubes, will almost invariably

have one or more v -h -f parasitic oscillations

unless adequate precautions have been taken

in advance. Many of the units described in

the constructional section of this edition had

parasitic oscillations when first constructed.

But these oscillations were eliminated in each

case; hence, the expedients used in these

equipments should be studied. V -h -f parasitics

may be readily identified, as they cause a

www.americanradiohistory.com

HANDBOOK Parasitic Oscillations 363

neon lamp to have a purple glow close to the

electrodes when it is excited by the parasitic

energy.

Parasitic Oscillations Triode stages are less

with Triodes subject to parasitic os-

cillations primarily be-

cause of the much lower power sensitivity of

such tubes as compared to beam tetrodes. But

such oscillations can and do take place. Usual-

ly, however, it is not necessary to incorporate

losser resistors as normally is the case with

beam tetrodes, unless the triodes are operated

quite near to their upper frequency limit, or

the tubes are characterized by a relatively

high transconductance. Triode v -h -f parasitic

oscillations normally may be eliminated by ad-

justment of the lengths and effective induct-

ance of the leads to the elements of the tubes.

In the case of triodes, v -h -f parasitic oscil-

lations often come about as a result of induct-

ance in the neutralizing leads. This is partic-

ularly true in the case of push -pull amplifiers.

The cure for this effect will usually be found

in reducing the length of the neutralizing leads

and increasing their diameter. Both the reduc-

tion in length and increase in diameter will

reduce the inductance of the leads and tend

to raise the parasitic oscillation frequency

until it is out of the range at which the tubes

will oscillate. The use of straightforward cir-

cuit design with short leads will assist in

forestalling this trouble at the outset. Butter-

fly -type tank capacitors with the neutralizing

capacitors built into the unit (such as the

B &W type) are effective in this regard.

V -h -f parasitic oscillations may take place

as a result of inadequate by- passing or long

by -pass leads in the filament, grid- return and

plate -return circuits. Such oscillations also

can take place when long leads exist between

the grids and the grid tuning capacitor or be-

tween the plates and the plate tuning capaci-

tor. The grid and plate leads should be kept

short, but the leads from the tuning capacitors

to the tank coils can be of any reasonable

length insofar as parasitic oscillations are

concerned. In an amplifier where oscillations

have been traced to the grid or plate leads,

their elimination can often be effected by mak-

ing the grid leads much longer than the plate

leads or vice versa. Sometimes parasitic os-

cillations can be eliminated by using iron or

nichrome wire for the grid or plate leads, or

for the neutralizing leads. But in any event

it will always be found best to make the neu-

tralizing leads as short and of as heavy con-

ductor as is practicable.

In cases where it has been found that in-

creased length in the grid leads for an ampli-

fier is required, this increased length can often

be wound into the form of a small coil and still

Figure 16

GRID PARASITIC SUPPRESSORS IN PUSH -

PULL TRIODE STAGE

obtain the desired effect. Winding these small

coils of iron or nichrome wire may sometimes

be of assistance.

To increase losses at the parasitic frequency,

the parasitic coils may be wound on 100 -ohm

2 -watt resistors. These "lossy" suppressors

should be placed in the grid leads of the tubes

close to the grid connection, as shown in fig-

ure 16.

Parasitics with Where beam -tetrode tubes are

Beam Tetrodes used in the stage which has

been found to be generating

the parasitic oscillation, all the foregoing

suggestions apply in general. However, there

are certain additional considerations involved

in elimination of parasitics from beam -tetrode

amplifier stages. These considerations involve

the facts that a beam -tetrode amplifier stage

has greater power sensitivity than an equiva-

lent triode amplifier, such a stage has a cer-

tain amount of screen -lead inductance which

may give rise to trouble, and such stages have

a small amount of feedback capacitance.

Beam - tetrode stages often will require the

inclusion of a neutralizing circuit to eliminate

oscillation on the operating frequency. How-

ever, oscillation on the operating frequency

normally is not called a parasitic oscillation,

and different measures are required to elimi-

nate the condition.

Basically, parasitic oscillations in beam -

tetrode amplifier stages fall into two classes:

cathode -grid- screen oscillations, and cathode -

screen -plate oscillations. Both these types of

oscillation can be eliminated through the use

of a parasitic suppressor in the lead between

the screen terminal of the tube and the screen

by -pass suppressor, as shown in figure 17.

Such a suppressor has negligible effect on the

by- passing effect of the screen at the operat-

ing frequency. The method of connecting this

www.americanradiohistory.com

364 Transmitter Design THE RADIO

PC=ST *18E. ON

52/1, 2W CAR-

BON RESISTOR

RFC=ONM/TE Z50 OR

EQUIVALENT

Figure 17

SCREEN PARASITIC SUPPRESSION CIR-

CUIT FOR TETRODE TUBES

suppressor to tubes having dual screen leads

is shown in figure 18. At the higher frequen-

cies at which parasitics occur, the screen is

no longer at ground potential. It is therefore

necessary to include an r -f choke by -pass con-

denser filter in the screen lead after the para-

sitic suppressor. The screen lead, in addition,

should be shielded for best results.

During parasitic oscillations, considerable

r -f voltage appears on the screen of a tetrode

tube, and the screen by -pass condenser can

easily be damaged. It is best, therefore, to

employ screen by -pass condensers whose d -c

working voltage is equal to twice the maximum

applied screen voltage.

The grid- screen oscillations may occasion-

ally be eliminated through the use of a para-

sitic suppressor in series with the grid lead

of the tube. The screen plate oscillations may

also be eliminated by inclusion of a parasitic

suppressor in series with the plate lead of the

tube. A suitable grid suppressor may be made

of a 22 -ohm 2 -watt Ohmite or Allen- Bradley

resistor wound with 8 turns of no. 18 enameled

wire. A plate circuit suppressor is more of a

problem, since it must dissipate a quantity of

power that is dependent upon just how close

the parasitic frequency is to the operating fre-

quency of the tube. If the two frequencies are

close, the suppressor will absorb some of the

fundamental plate circuit power. For kilowatt

stages operating no higher than 30 Mc. a satis-

factory plate circuit suppressor may be made

of five 570 -ohm 2 -watt carbon resistors in par-

allel, shunted by 5 turns of no. 16 enameled

wire, % inch diameter and % inch long (figure

19A and B).

The parasitic suppressor for the plate cir-

cuit of a small tube such as the 5763, 2E26,

807, 6146 or similar type normally may con-

sist of a 47 -ohm carbon resistor of 2 -watt size

with 6 turns of no. 18 enameled wire wound

around the resistor. However, for operation

above 30 Mc., special tailoring of the value

""``""ftmeYrnrw. lj

Figure 18

PHOTO OF APPLICATION OF SCREEN

PARASITIC SUPPRESSION CIRCUIT

OF FIGURE 17

of the resistor and the size of the coil wound

around it will be required in order to attain

satisfactory parasitic suppression without ex-

cessive power loss in the parasitic suppressor.

Tetrode Screening Isolation between the grid

and plate circuits of a tet-

rode tube is not perfect. For maximum stabili-

ty, it is recommended that the tetrode stage

be neutralized. Neutralization is absolutely

necessary unless the grid and plate circuits

of the tetrode stage are each completely iso-

lated from each other in electrically tight

boxes. Even when this is done, the stage will

show signs of regeneration when the plate

and grid tank circuits are tuned to the same

frequency. Neutralization will eliminate this

regeneration. Any of the neutralization cir-

cuits described in the chapter Generation of

R -F Energy may be used.

18 -9 Checking for -Parasitic

Oscillations

It is an unusual transmitter which harbors

no parasitic oscillations when first constructed

www.americanradiohistory.com

HANDBOOK Parasitic Oscillations 365

PC

PC =! T R/eE. ON

22d.2W. COM-

POSITION RESISTOR

PC

O

FOR 507, ETC.

PC =ArI /IE.ON47/1,2W

COMPOSITION RESISTOR

FOR 4 -250A, ETC.

PC = S -570!2, 2W COMPOSITION

RESISTORS IN PARALLEL WITH

ST. 0 /eE. //4.

Figure 19

PLATE AND GRID PARASITIC SUPPRESSION IN TETRODE TUBES

and tested. Therefore it is always wise to fol-

low a definite procedure in checking a new

transmitter for parasitic oscillations.

Parasitic oscillations of all types are most

easily found when the stage in question is

running by itself, with full plate (and screen)

voltage, sufficient protective bias to limit the

plate current to a safe value, and no excita-

tion. One stage should be tested at a time,

and the complete transmitter should never be

put on the air until all stages have been thor-

oughly checked for parasitics.

To protect tetrode tubes during tests for

parasitics, the screen voltage should be ap-

plied through a series resistor which will limit

the screen current to a safe value in case the

plate voltage of the tetrode is suddenly re-

moved when the screen supply is on. The cor-

rect procedure for parasitic testing is as fol-

lows (figure 20):

1. The stage in question should be coupled

to a dumpy load, and tuned up in correct oper-

ating shape. Sufficient protective bias should

be applied to the tube at all times. For pro-

tection of the stage under test, a lamp bulb

should be added in series with one leg of the

primary circuit of the high voltage power sup-

ply. As the plate supply load increases during

a period of parasitic oscillation, the voltage

drop across the lamp increases, and the effec-

tive plate voltage drops. Bulbs of various

size may be tried to adjust the voltage under

testing conditions to the correct amount. If a

Variac or Powerstat is at hand, it may be used

in place of the bulbs for smoother voltage con-

trol. Don't test for parasitics unless some type

of voltage control is used on the high voltage

supply! When a stage breaks into parasitic

oscillations, the plate current increases vio-

lently, and some protection to the tube under

test must be used.

2. The r -f excitation to the tube should now

be removed. When this is done, the grid, screen

and plate currents of the tube should drop to

zero. Grid and plate tuning condensers should

be tuned to minimum capacity. No change in

resting grid, screen or plate current should

be observed. If a parasitic is present, grid cur-

rent will flow, and there will be an abrupt in-

crease in plate current. The size of the lamp

bulb in series with the high voltage supply may

be varied until the stage can oscillate contin-

uously, without exceeding the rated plate dis-

sipation of the tube.

3. The frequency of the parasitic may now

be determined by means of an absorption wave

meter, or a neon bulb. Low frequency oscilla-

tions will cause a neon bulb to glow yellow.

High frequency oscillations will cause the

bulb to have a soft, violet glow. Once the fre-

quency of oscillation is determined, the cures

suggested in this chapter may be applied to

the stage.

4. When the stage can pass the above test

with no signs of parasitics, the bias supply of

the tube in question should be decreased until

the tube is dissipating its full plate rating

when full plate voltage is applied, with no r -f

EXCITER

EXCITER CONTROL

SWITCH

BIAS SUPPLY

AMPLIFIER STAGE

TO BE TESTED

FOR PARASITICS

HIGH VOLTAGE

POWER SUPPLY

n l

°

DUMMY

LOAD

VARIAC OR

LIGHT BULBS 4

A.C.

SUPPLY

Figure 20

SUGGESTED TEST SETUP FOR PARASITIC

TESTS

www.americanradiohistory.com

366 Transmitter Design

excitation. Excitation may now be applied and

the stage loaded to full input into a dummy

load. The signal should now be monitored in

a nearby receiver which has the antenna ter-

minals grounded or otherwise shorted out. A

series of rapid dots should be sent, and the

frequency spectrum for several megacycles

each side of the carrier frequency carefully

searched. If any vestige of parasitic is left,

it will show up as an occasional "pop" on a

keyed dot. This "pop" may be enhanced by a

slight detuning of either the grid or plate cir-

cuit.

5. If such a parasitic shows up, it means

that the stage is still not stable, and further

measures must be applied to the circuit. Para-

sitic suppressors may be needed in both screen

and grid leads of a tetrode, or perhaps in both

grid and neutralizing leads of a triode stage.

As a last resort, a 10,000 -ohm 25 -watt wire -

wound resistor may be shunted across the grid

coil, or grid tuning condenser of a high pow-

ered stage. This strategy removed a keying

pop that showed up in a commercial transmit-

ter, operating at a plate voltage of 5000.

Test for Parasitic It is common experience

Tendency in Tetrodo to develop an engineer -

Amplifiers ing model of a new

equipment that is ap-

parently free of parasitics and then find trouble-

some oscillations showing up in production

units. The reason for this is that the equipment

has a parasitic tendency that remains below

the verge of oscillation until some change in

a component, tube gain, or operating condition

raises the gain of the parasitic circuit enough

to start oscillation.

In most high frequency transmitters there

are a great many resonances in the tank cir-

cuits at frequencies other than the desired

SIGNAL GENERATOR

100CC -20O MC

Figure 21

PARASITIC GAIN MEASUREMENT

Grid -dip oscillator and vacuum tube

voltmeter may be used to measure para-

sitic stage gain over IOOkc -200mc

region.

operating frequency. Most of these parasitic

resonant circuits are not coupled to the tube

and have no significant tendency to oscillate.

A few, however, are coupled to the tube in

some form of oscillatory circuit. If the regener-

ation is great enough, oscillation at the para-

sitic frequency results. Those spurious circuits

existing just below oscillation must be found

and suppressed to a safe level.

One test method is to feed a signal from a

grid -dip oscillator into the grid of a stage and

measure the resulting signal level in the plate

circuit of the stage, as shown in figure 21. The

test is made with all operating voltages applied

to the tubes. Class C stages should have bias

reduced so a reasonable amount of static plate

current flows. The grid -dip oscillator is tuned

over the range of 100 kc to 200 mc. and the

relative level of the r -f voltmeter is watched

and the frequencies at which voltage peaks

occur are noted. Each significant peak in volt-

age gain in the stage must be investigated. Cir-

cuit changes or suppression must then be added

to reduce all peaks by 10 db or more in ampli-

tude.

www.americanradiohistory.com

CHAPTER NINETEEN

Television and Broadcast

Interference

The problem of interference to television

reception is best approached by the philoso-

phy discussed in Chapter Eighteen. By correct

design procedure, spurious harmonic genera-

tion in low frequency transmitters may be held

to a minimum. The remaining problem is two-

fold: to make sure that the residual harmonics

generated by the transmitter are not radiated,

and to make sure that the fundamental signal

of the transmitter does not overload the tele-

vision receiver by reason of the proximity of

one to the other.

In an area of high TV- signal field intensity

the TVI problem is capable of complete solu-

tion with routine measures both at the amateur

transmitter and at the affected receivers. But

in fringe areas of low TV- signal field strength

the complete elimination of TVI is a difficult

and challenging problem. The fundamentals

illustrated in Chapter Fifteen must be closely

followed, and additional antenna filtering of

the transmitter is required.

19 -1 Types of Television

Interference

There are three main types of TVI which

may be caused singly or in combination by the

emissions from an amateur transmitter. These

types of interference are:

(1) Overloading of the TV set by the trans-

mitter fundamental

(2) Impairment of the picture by spurious

emissions

(3) Impairment of the picture by the radia-

tion of harmonics .

367

TV Set Even if the amateur transmitter

Overloading were perfect and had no har-

monic radiation or spurious

emissions whatever, it still would be likely to

cause overloading of TV sets whose antennas

were within a few hundred feet of the trans-

mitting antenna. This type of overloading is

essentially the same as the common type of

BCI encountered when operating a medium -

power or high -power amateur transmitter with-

in a few hundred feet of the normal type of

BCL receiver. The field intensity in the im-

mediate vicinity of the transmitting antenna

is sufficiently high that the amateur signal

will get into the BC or TV set either through

overloading of the front end, or through the

i -f, video, or audio system. A characteristic

of this type of interference is that it always

will be eliminated when the transmitter tem-

porarily is operated into a dummy antenna.

Another characteristic of this type of over-

loading is that its effects will be substan-

tially continuous over the entire frequency

coverage of the BC or TV receiver. Channels

2 through 13 will be affected in approximately

the same manner.

With the overloading type of interference

the problem is simply to keep the fundamental

of the transmitter out of the affected receiver.

Other types of interference may or may not

show up when the fundamental is taken out of

the TV set (they probably will appear), but at

least the fundamental must be eliminated first.

The elimination of the transmitter fundamen-

tal from the TV set is normally the only opera-

tion performed on or in the vicinity of the TV

receiver. After the fundamental has been elimi-

www.americanradiohistory.com

368 TV and Broadcast Interference THE RADIO

300-OHM

LINE FROM

ANTENNA

CI

TO INPUT

OF TV SE S

300 OHM

LINE FROM

ANTENNA

pA FOR

0

I

TO TV ANTENNA

TO ANTENNA

TERMINALS

OF TV SET

Figure 1

TUNED TRAPS FOR THE TRANSMITTER

FUNDAMENTAL

The arrangement at (A) has proven to be ef-

fective in eliminating the condition of gen-

eral blocking as caused by a 28 -Mc. trans-

mitter in the vicinity of a TV receiver. The

tuned circuits LI -CI are resonated separate-

ly to the frequency of transmission. The ad-

justment may be done at the station, or it

may be accomplished at the TV receiver by

tuning for minimum interference on the TV

screen.

Shown at (B) is an alternative arrangement

with a series -tuned circuit across the anten-

na terminals of the TV set. The tuned cir-

cuit should be resonated to the operating

frequency of the transmitter. This arrange-

ment gives less attenuation of the interfer-

ing signal than that at (A); the circuit has

proven effective with interference from trans-

mitters on the 50 -Mc. band, and with low -

power 28 -Mc. transmitters.

SHIELD BOX

L2 ` Li

(SNORT

LEADS

TO

_ ANTENNA

TERM ON

TV SET

300 -OHM LINE, SHIELDED OR UNSHIELDED

C2 Jl

COAx

FIiTING II-E--d --F

L3 I C2

3

COAX

FITTING

Figure 2

HIGH -PASS TRANSMISSION LINE FILTERS

The arrangement at (A) will stop the passing

of all signals below about 45 Mc. from the

antenna transmission line into the TV set.

Coils LI ore each 1.2 microhenrys (17 turns

no. 24 enam. closewound on 1/2-inch dia. poly-

styrene rod) with the center tap grounded.

It will be found best to scrape, twist, and

solder the center tap before winding the coil.

The number of turns each side of the tap may

then be varied until the tap is in the exact

center of the winding. Coil L2 is 0.6 micro-

henry (12 turns no. 24 enam. closewound on

1/2-inch dia. polystyrene rod). The capacitors

should be about 16.5 Wald., but either 15 or

20 pµtd. ceramic capacitors will give satis-

factory results. A similar filter for coaxial

antenna transmission line is shown at (B).

Both coils should be 0.12 microhenry (7 turns

no. 18 enam. spaced to % inch on 1/2-inch dia.

polystyrene rod). Capacitors C2 should be

75 µµtd. midget ceramics, while C3 should

be a 40 -µµtd. ceramic.

naced as a source of interference to reception,

work may then be begun on or in the vicinity

of the transmitter toward eliminating the other

two types of interference.

Taking Out More or less standard BCI-

the Fundamental type practice is most com-

monly used in t a k i n g out

fundamental interference. Wavetraps and fil-

ters are installed, and the antenna system may

or may not be modified so as to offer less re-

sponse to the signal from the amateur trans-

mitter. In regard to a comparison between

wavetraps and filters, the same considerations

apply as have been effective in regard to BCI

for many years; wavetraps are quite effective

when properly installed and adjusted, but they

must be readjusted whenever the band of oper-

ation is changed, or even when moving from

one extreme end of a band to the other. Hence,

wavetraps are not recommended except when

operation will be confined to a relatively nar-

row portion of one amateur band. However, fig-

ure 1 shows two of the most common signal

trapping arrangements.

High -Pass Filters High -pass filters in the

antenna lead of the TV set

have proven to be quite sat i s factory as a

means of eliminating TVI of the overloading

type. In many cases when the interfering trans-

mitter is operated only on the bands below

7.3 Mc., the use of a high -pass filter in the

antenna lead has completely eliminated all

www.americanradiohistory.com

HANDBOOK Harmonic Radiation 369

TVI. In some cases the installation of a high -

pass filter in the antenna transmission line

and an a -c line filter of a standard variety has

proven to be completely effective in eliminat-

ing the interference from a transmitter operat-

ing in one of the lower frequency amateur

bands.

In general, it is suggested that commercial-

ly manufactured high -pass filters be purchased.

Such units are available from a number of manu-

facturers at a relatively moderate cost. How-

ever, such units may be home constructed;

suggested designs are given in figures 2 and

3. Types for use both with coaxial and with

balanced transmission lines have been shown.

In most cases the filters may be constructed

in one of the small shield boxes which are

now on the market. Input and output terminals

may be standard connectors, or the inexpen-

sive type of terminal strips usually used on

BC and TV sets may be employed. Coaxial

terminals should of course be employed when

a coaxial feed line is used to the antenna. In

any event the leads from the filter box to the

TV set should be very short, including both

the antenna lead and the ground lead to the

box itself. If the leads from the box to the set

have much length, they may pick up enough

signal to nullify the effects of the high -pass

filter.

Blocking from Operation on the 50 -Mc. ama-

50-Me. Signals teur band in an area where

channel 2 is in use for TV

imposes a special problem in the matter of

blocking. The input circuits of most TV sets

are sufficiently broad so that an amateur sig-

nal on the 50 -Mc. band will ride through with

little attenuation. Also, the normal TV antenna

will have a quite large response to a signal

in the 50 -Mc. band since the lower limit of

channel 2 is 54 Mc.

High -pass filters of the normal type simply

are not capable of giving sufficient attenua-

tion to a signal whose frequency is so close

to the necessary pass band of the filter. Hence,

a resonant circuit element, as illustrated in

figure :, must be used to trap out the amateur

field at the input of the TV set. The trap must

be tuned or the section of transmission line

cut, if a section of line is to be used for a

particular frequency in the 50 -Mc. band.

This frequency will have to be near the lower

frequency limit of the 50 -Mc. band to obtain

adequate rejection of the amateur signal while

still not materially affecting the response of

the receiver to channel 2.

Elimination of All spurious e m i s s i o n s

Spurious Emissions from amateur transmitters

(ignoring harmonic signals

for the time being) must be eliminated to corn-

Figure 3

SERIES -DERIVED HIGH -PASS FILTER

This filter is designed for use in the

300 -ohm transmission line from the TV

antenna to the TV receiver. Nominal cut-

off frequency is 36 Mc. and maximum re-

jection is at about 29 Mc.

Ct,C6- 15 -µµfd. zero- coefficient ceramic

C2,C3,C4,C5-20 -i fd. zero -coefficint cra-

mic

L1,Lt -2.0 µh. About 24 turns no. 28 d.c.c.

wound to ?éi on '4" diameter polystyrene

rod. Turns should be adjusted until the

coil resonates to 29 Mc. with the asoci-

ated 15- ptµfd. capacitor.

L2-0.66 i h., 14 turns no. 28 d.c.c. wound

to Se" on t/ " die. polystyrene rod. Adjust

turns to resonate externally to 20 Mc.

with an auxiliary 100- µ{tfd. capacitor

whose value is accurately known.

ply with FCC r e g u l a t i o n s. But in the past

many amateur transmitters have emitted spur-

ious signals as a result of key clicks, para-

sities, and overmodulation transients. In most

cases the operators of the transmitters were

not aware of these emissions since they were

radiated only for a short distance and hence

were not brought to his attention. But with

one or more TV sets in the neighborhood it

is probable that such spurious signals will

be brought quickly to his attention.

19 -2 Harmonic Radiation

After any condition of blocking at the TV

receiver has been eliminated, and when the

transmitter is completely free of transients

and parasitic oscillations, it is probable that

TVI will be eliminated in certain cases. Cer-

tainly general interference should be elimi-

nated, particularly if the transmitter is a well

designed affair operated on one of the lower

frequency bands, and the station is in a high -

signal TV area. But when the transmitter is

to be operated on one of the higher frequency

bands, and particularly in a marginal TV area,

the job of TVI -proofing will just have begun.

The elimination of harmonic radiation from

the transmitter is a difficult and tedious job

which must be done in an orderly manner if

completely satisfactory results are to be ob-

tained.

www.americanradiohistory.com

370 TV and Broadcast Interference THE RADIO

TRANSMITTER

FUNDAMENTAL 2ND 3RD 4TH 5TH 6TH 7TH 8TH 9TH 107H

7.0 21 -21 9 42-44 56-56 4 63-65 7 70-73

TV I.F NEW CHA NEL ELC EL

7 3 701F 2

14.0 2 -43 56 -57.6 70 -72 84 -664 96-100.6

NEW CHANNEL CHANNEL CHANNEL FM

14!4 TV 1. F. 02 pd © BROAD CAST

21.0 63 -64 35 64 -65 6 105' 110725 169.193 210-214 5

cHt5NEL CHAfNNEL FM S C EL

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( V1 F5)

26.96 53.92- 60.66- 107.64- 169 216

271 23 54 46 61 69

CH.jtJNEL

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10692 CHANNEL

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28.0 56 -59 4 B4 -89.1 166-1762199-2079

291 7 CHANNEL CHANNEL

O

CHANNEL 1,1E}á

I`11, 1

50.0 100-105 200.216 50 -486 500-540

5410 FM HAy

BROAD- ST 1

3 POSSIBLE INTERFERENCE

TO U-H-F CHANNELS

Figure 4

HARMONICS OF THE AMATEUR BANDS

Shown are the harmonic frequency ranges of the amateur bands between 7 and 54 Mc., with the

TV channels (and TV i -f systems) which are most likely to receive interference from these har-

monics. Under certain conditions amateur signals in the 1.8 and 3.5 Mc. bands con cause inter-

ference as a result of direct pickup in the video systems of TV receivers which are not ade-

quately shielded.

First it is well to become familiar with the

TV channels presently assigned, with the TV

intermediate frequencies commonly used, and

with the channels which will receive inter-

ference from harmonics of the various ama-

teur bands. Figures 4 and 5 give this infor-

mation.

Even a short inspection of figures 4 and 5

will make obvious the seriousness of the in-

terference which can be caused by harmonics

of amateur signals in the higher frequency

bands. With any sort of reasonable precautions

in the design and shielding of the transmitter

it is not likely that harmonics higher than the

6th will be encountered. hence the main of-

fenders in the way of harmonic interference

will be those bands above 14 -Mc.

Nature of Investigations into the

Harmonic Interference nature of the interfer-

ence caused by ama-

teur signals on the TV screen, assuming that

blocking has been eliminated as described

earlier in this chapter, have revealed the fol-

lowing facts:

1. An unmodulated carrier, such as a c -w

signal with the key down or an AM sig-

nal without m o d u l at ion, will give a

cross -hatch or herringbone pattern on

the TV screen. This same general type

of picture also will occur in the case of

a narrow -band FM signal either with or

without modulation.

2. A relatively strong AM signal will give

in addition to the herringbone a very

serious succession of light and dark

bands across the TV picture.

3. A moderate strength c -w signal without

transients, in the absence of overload-

ing of the TV set, will result merely in

the turning on and off of the herringbone

on the picture.

To discuss condition (1) above, the herring-

bone is a result of the beat note between the

TV video carrier and the amateur harmonic.

Hence the higher the beat note the less ob-

vious will be the resulting cross -hatch. Fur-

ther, it has been shown that a much stronger

signal is required to produce a discernible

herringbone when the interfering harmonic is

as far away as possible from the video car-

rier, without running into the sound carrier.

Thus, as a last resort, or to eliminate the last

vestige of interference after all corrective

measures have been taken, operate the trans-

mitter on a frequency such that the interfer-

www.americanradiohistory.com

HANDBOOK Harmonic Interference 371

1

VIDEO

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SOUND

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192 198

HIGH BAND

204 210 216

Figure 5

FREQUENCIES OF THE V -H -F TV CHANNELS

Showing the frequency ranges of TV channels 2 through 13, with the picture carrier and sound

carrier frequencies also shown.

ing harmonic will fall as far as possible from

the picture carrier. The worst possible inter-

ference to the picture from a continuous car-

rier will be obtained when the interfering sig-

nal is very close in frequency to the video

carrier.

Isolating Throughout the testing proce-

the Source of dure it will be necessary to

the Interference have some sort of indicating

device as a means of deter-

mining harmonic field intensities. The best

indicator for field intensities some distance

from the transmitting antenna will probably be

the TV receiver of some neighbor with whom

friendly relations are still maintained. This

person will then be able to give a check, oc-

casionally, on the relative nature of the inter-

ference. But it will probably be necessary to

go and check yourself periodically the results

obtained, since the neighbor probably will not

be able to give any sort of a quantitative anal-

ysis of the progress which has been made.

An additional device for checking relative-

ly high field intensities in the vicinity of the

transmitter will be almost a necessity. A sim-

ple crystal diode wavemeter, shown in figure

6 will accomplish this function. Also, it will

be very helpful to have a receiver, with an S

meter, capable of covering at least the 50 to

100 Mc. range and preferably the range to 216

Mc. This device may consist merely of the

station receiver and a simple converter using

the two halves of a 6J6 as oscillator and

mixer.

The first check can best be made with the

neighbor who is receiving the most serious

or the most general interference. Turn on the

transmitter and check all channels to deter-

mine the extent of the interference and the

number of channels affected. Then disconnect

the antenna and substitute a group of 100 -watt

lamps as a dummy load for the transmitter. Ex-

perience has shown that 8 100 -watt lamps con-

nected in two seriesed groups of four in par-

allel will take the output of a kilowatt trans-

mitter on 28 Mc. if connections are made sym-

metrically to the group of lamps. Then note

the interference. Now remove plate voltage

from the final amplifier and determine the ex-

tent of interference caused by the exciter

stages.

In the average case, when the final ampli-

fier is a beam tetrode stage and the exciter is

10' PICKUP WIRE

5T . X0.3 DIA.18E, 0.5 LONG

IN 34

COVERAGE -35-140 MC.

Figure 6

Crystal -diode wavemeter suitable for check-

ing high -intensity harmonics in TV region.

www.americanradiohistory.com

372 TV and Broadcast Interference THE RADIO

relatively low powered and adequately shield-

ed, it will be found that the interference drops

materially when the antenna is removed and a

dummy load substituted. It will also be found

in such an average case that the interference

will stop when the exciter only is operating.

Transmitter It should be made clear at this

Power Level point that the l e v e l of power

used at the transmitter is not of

great significance in the basic harmonic re-

duction problem. The difference in power level

between a 20 -watt transmitter and one rated

at a kilowatt is only a matter of about 17 db.

Yet the degree of harmonic attenuation re-

quired to eliminate interference caused by

harmonic radiation is from 80 to 120 db, de-

pending upon the TV signal strength in the

vicinity. This is not to say that it is not a

simpler job to eliminate harmonic interference

from a low -power transmitter than from a kilo-

watt equipment. It is simpler to suppress har-

monic radiation from a low -power transmitter

simply because it is a much easier problem to

shield a low -power unit, and the filters for the

leads which enter the transmitter enclosure

may be constructed less expensively and

smaller for a low -power unit.

19 -3 Low -Pass Filters

After the transmitter has been shielded, and

all power leads have been filtered in such a

manner that the transmitter shielding has not

been rendered ineffective, the only remaining

available exit for harmonic energy lies in the

antenna transmission line. Hence the main

burden of harmonic attenuation will fall on the

low -pass filter installed between the output

of the transmitter and the antenna system.

Experience has shown that the low -pass

filter can best be installed externally to the

main transmitter enclosure, and that the trans-

mission line from the transmitter to the low -

pass filter should be of the coaxial type.

Hence the majority of low -pass filters are de-

signed for a characteristic impedance of 52

ohms, so that RG -8 /U cable (or RG -58 /U for

a small transmitter) may be used between the

output of the transmitter and the antenna trans-

mission line or the antenna tuner.

Transmitting -type low -pass filters for ama-

teur use usually are designed in such a man-

ner as to pass frequencies up to about 30 Mc.

without attenuation. The nominal cutoff fre-

quency of the filters is usually between 38

and 45 Mc., and m- derived sections with maxi-

mum attenuation in channel 2 usually are in-

cluded. Well- designed filters capable of carry-

ing any power level up to one kilowatt are

Lx L3 L4 L5

Lip l Cz C3 Ca

T

Figure 7

LOW -PASS FILTER SCHEMATIC DIAGRAMS

The filter illustrated at (A) uses m-

derived terminating half sections at each

end, with three constant -k mid -sections.

The filter at (B) is essentially the same

except that the center section has been

changed to act as on m- derived section

which can be designed to offer maximum

attenuation to channels 2, 4, 5, or 6 in

accordance with the constants given be-

low. Cutoff frequency is 45 Mc. in all

cases. All coils, except L4 in (B) above,

are wound 1/2 "i.d. with 8 turns per inch.

The (A) Filter

C,,C6 -41.5µµ4d. (40 µµfd. will be found suit-

able.)

C2, C3, C4-136 µµfd. (130 to 140 µµid. may be

used.)

L,, L6 -0.2 ph; 3 S t. no. 14

L2, L4-0.3 ph; 5 t. no. 12

L3, L4, -0.37 ph; 615 t. no. 12

The (B) Filter with Mid- Section tuned to Channel

2 (58 Mc..)

C C6 -41.5 pfd.

C2, Ca -136

C3 -87 µpfd. (50 µpfd. fixed and 75 µpfd. vari-

able in parallel.)

L3, L7-0.2 ph; 3 Si t. no. 14

L2, L3, L3 , L6 -0.3 ph; 5 t. no. 12

L4-0.09 ph; 2 t. no. 14 55" die. by '4 " long

The (B) Filter with Mid -Sction tuned to Channel

4 (71 Mc.). All components same except that:

C3 -106

L3, L3 -0.33 ph; 6 t. no. 12

L4 -0.05 ph; 1)5 t. no. 14, 3 '8 " dia. by 3/8 "

long.

The (B) Filter with Mid -Section tuned to Channel

5 (81 Mc.). Change the following:

C3 -113 µpfd.

L,, L4-0.34 ph; 6 t. no. 12

L4-0.033 ph; 1 t. no. 14 3/8" dia.

The (B) Filter with Mid -Section tuned to Channel

6 (86 Mc.). All comp is are essentially

the same except that the theoretical value of

La Is changed to 0.03 ph., and the capacitance

of C3 is changed to 117 µpfd.

www.americanradiohistory.com

HANDBOOK Low Pass Filters 373

available commercially from several manufac-

turers. Alternatively, filters in kit form are

available from several manufacturers at a

somewhat lower price. Effective filters may

be home constructed, if the test equipment is

available and if sufficient care is taken in the

construction of the assembly.

Construction of Figures 7, 8 and 9 illustrate

Low -Pass Filters high- performance low -pass

filters which are suitable

for home construction. All are constructed in

slip -cover aluminum boxes (ICA no. 29110)

with dimensions of 17 by 3 by 2% inches. Five

aluminum baffle plates have been installed in

the chassis to make six shielded sections

within the enclosure. Feed -through bushings

between the shielded sections are Johnson no.

135 -55.

Both the (A) and (B) f i 1 t e r types are de-

signed for a nominal cut -off frequency of 45

Mc., with a frequency of maximum rejection

at about 57 Mc. as established by the termi-

nating half- sections at each end. Characteris-

tic impedance is 52 ohms in all cases. The

alternative filter designs diagrammed in figure

7B have provision for an additional rejection

trap in the center of the filter unit which may

be designed to offer maximum r ejection in

channel 2, 4, 5, or 6, depending upon which

channel is likely to be received in the area in

question. The only components which must be

changed when changing the frequency of the

maximum rejection notch in the center of the

filter unit are inductors L L4, and L,, and

capacitor C,. A trimmer capacitor has been in-

cluded as a portion of C, so that the frequency

of maximum rejection can be tuned accurately

to the desired value. Reference to figures 5

and 6 will show the amateur bands which are

.I

Figure 8

PHOTOGRAPH OF THE (B) FILTER WITH

THE COVER IN PLACE

most likely to cause interference to specific

TV channels.

Either high -power or low -power components

may be used in the filters diagrammed in figure

7. With the small Centralab TCZ zero- coeffi-

cient ceramic capacitors used in the filter

units of figure 7A or figure 7B, power levels

up to 200 watts output may be used without

danger of damage to the capacitors, provided

the filter is feeding a 52 -ohm resistive load.

It may be practicable to use higher levels of

power with this type of ceramic capacitor in

the filter, but at a power level of 200 watts on

the 28 -Mc. band the capacitors run just per-

ceptibly warm to the touch. As a point of in-

terest, it is the current rating which is of sig-

nificance to the capacitors used in filter s

such as illustrated. Since current ratings for

small capacitors such as these are not readily

available, it is not possible to establish an

accurate power rating for such a unit. The

high -power unit illustrated in figure 9, which

uses Centralab type 850S and 854S capacitors,

Figure 9

PHOTOGRAPH OF THE (B) FILTER WITH COVER REMOVED

The mid -section in this filter is adjusted for maximum rejection of channel 4. Note that the main

coils of the filter are mounted at on angle of about 45 degrees so that there will be minimum

inductive coupling from one section to the next through the holes in the aluminum partitions.

Mounting the coils in this manner was found to give a measurable improvement in the attenua-

tion characteristics of the filter.

www.americanradiohistory.com

374 TV and Broadcast Interference THE RADIO

has proven quite suitable for power levels up

to one kilowatt.

Capacitors C C2, C4, and C, can be stand-

ard manufactured units with normal 5 per cent

tolerance. The coils for the end sections can

be wound to the dimensions given (Lt, L6, and

L,). Then the resonant frequency of the series

resonant end sections should be checked with

a grid -dip meter, after the adjacent input or

output terminal has been shorted with a very

short lead. The coils should be squeezed or

spread until resonance occurs at 57 Mc.

The intermediate m- derived section in the

filter of figure 7B may also be checked with a

grid -dip meter for resonance at the correct re-

jection frequency, after the hot end of L4 has

been temporarily grounded with a low- induct-

ance lead. The variable capacitor portion of

C, can be tuned until resonance at the correct

frequency has been obtained. Note that there

is so little difference between the constants

of this intermediate section for channels 5 and

6 that variation in the setting of C, will tune

to either channel without materially changing

the operation of the filter.

The coils in the intermediate sections of

the filter (Lr, L L4, and L3 in figure 7A, and

L2, L3, L4, and L6 in figure 7B) may be checked

most conveniently outside the filter unit with

the aid of a small ceramic capacitor of known

value and a grid -dip meter. The ceramic ca-

pacitor is paralleled a c r o s s the small coil

with the shortest possible leads. Then the as-

sembly is placed atop a cardboard box and the

resonant frequency checked with a grid -dip

meter. A Shure reactance slide rule may be

used to ascertain the correct resonant frequen-

cy for the desired L -C combination and the

coil altered until the desired resonant frequen-

cy is attained. The coil may then be installed

in the filter unit, making sure that it is not

squeezed or compressed as it is being in-

stalled. However, if the coils are wound ex-

actly as given under figure 10, the filter may

be assembled with reasonable assurance that

it will operate as designed.

Using Low -Pass The low -pass filter con -

Filters nected in the output trans-

mission line of the trans-

mitter is capable of affording an enormous de-

gree of harmonic attenuation. However, the

filter must be operated in the correct manner

or the results obtained will not be up to ex-

pectations.

In the first place, all direct radiation from

the transmitter and its control and power leads

must be suppressed. This subject has been

discussed in the previous section. Secondly,

the filter must be operated into a load imped-

ance approximately equal to its design char-

acteristic impedance. The filter itself will

Figure 10

SCHEMATIC OF THE SINGLE -SECTION

HALF -WAVE FILTER

The constants given below are for a char-

acteristic impedance of 52 ohms, for use with

RG -8 /U and RG -58/1/ cable. Coil Lt should

be checked for resonance at the operating

frequency with Cr, and the same with L2 and

C4. This check can be mode by soldering a

low- inductance grounding strap to the lead

between L1 and L2 where it passes through

the shield. When the coils have been trimmed

to resonance with a grid -dip meter, the

grounding strap should of course be removed.

This filter type will give an attenuation of

about 30 db to the second harmonic, about

48 db to the third, about 60 db to the fourth,

67 to the fifth, and so on increasing at a

rate of about 30 db per octave.

Ct,C2,C3,C4- Silver mica or small ceramic for low

power, transmitting type ceramic for high power.

Capacitance for different bands is given below:

160 meters -1700 µµfd.

80 meters -850 µµfd.

40 meters -440 µµfd.

20 meters -220 µµid.

10 meters -110 µµfd.

6 meters -60 µµfd.

L L, -May be made up of sections of B&W Mini -

ductor for power levels below 250 watts, or of

no. 12 enom. for power up to one kilowatt. Ap-

proximate dimensions for the coils are given

below, but the coils should be trimmed to reso-

nate at the proper frequency with a grid -dip me-

ter as discussed above. All coils except the

ones for 160 meters are wound 8 turns per Inch.

160 meters -4.2 µh; 22 turns no. 16 enam., 1

dia. 2" long

80 meters -2.1 µh; 13 t. 1" dia. (No. 3014 Mini -

ductor or no. 12)

40 meters -1.1 µh; 8 t. 1" dla. (No. 3014 or no.

12 at 8 t.p.i.)

20 meters -0.55 µh; 7 t',l" dia. (No. 3010 or

no. 12 at 8 t.p.i.)

10 meters -0.3 µh; 6 t. S4" dia. (No. 3002 or

no. 12 of 8 tpi.)

6 meters -0.17 µh; 4 t. 14" dia. (No. 3002 or

no. 12 at 8 t.p.i.)

have very low losses (usually less than 0.5

db) when operated into its nominal value of re-

sistive load. But if the filter is mis- terminated

its losses will become excessive, and it will

not present the correct value of load imped-

ance to the transmitter.

If a f i l t e r, being fed from a high -power

transmitter, is operated into an incorrect ter-

mination it may be damaged; the coils may be

overheated and the capacitors destroyed as a

result of excessive r -f currents. Hence it is

wise, when first installing a low -pass filter,

www.americanradiohistory.com

HANDBOOK Broadcast Interference 375

Figure 11

HALF -WAVE FILTER

FOR THE 28 -MC. BAND

Showing one possible type

of construction of o 52 -ohm

half -wave filter for relative-

ly low power operation on

the 28 -Mc. bond.

to check the standing -wave ratio of the load

being presented to the output of the filter with

a standing -wave meter of any of the conven-

tional types. Then the antenna termination or

the antenna coupled should be adjusted, with

low power on the transmitter, until the s.w.r.

of the load being presented to the filter is

less than 2.0, and preferably below 1.5.

Half -Wove Filters Half -wave filters ( "Har-

monikers") have been dis-

cussed in various publications including the

Nov. -Dec. 1949 GE Ham News. Such filters

are relatively simple and offer the advantage

that they present the same value of impedance

at their input terminals as appears as load

across their output terminals. Such filters nor-

mally are used as one -band affairs, and they

offer high attenuation only to the third and

higher harmonics. Design data on the half -

wave filter is given in figure 10. Construction

of half -wave filters is illustrated in figure 11.

19 -4 Broadcast

Interference

Interference to the reception of signals in

the broadcast band (540 to 1600 kc.) or in the

FM broadcast band (88 to 108 Mc.) by amateur

transmissions is a serious matter to those

amateurs living in densely populated areas.

Although broadcast interference has recently

been overshadowed by the seriousness of tele-

vision interference, the condition of BCI is

still present.

In general, signals from a transmitter oper-

ating properly are not picked up by receivers

tuned to other frequencies unless the receiver

is of inferior design, or is in poor condition.

Therefore, if the receiver is of good design

and is in good repair, the burden of rectifying

the trouble rests with the owner of the inter-

fering station. Phone and c -w stations both

are capable of causing broadcast interference,

key -click annoyance from the code transmitters

being particularly objectionable.

A knowledge of each of the several types

of broadcast interference, their cause, and

methods of eliminating them is necessary for

the successful disposition of this trouble. An

effective method of combating one variety of

interference is often of no value whatever in

the correction of another type. Broadcast in-

terference seldom can be cured by "rule of

thumb" procedure.

Broadcast interference, as covered in this

section refers primarily to standard (amplitude

modulated, 550 -1600 kc.) broadcast. Interfer-

ence with FM broadcast reception is much

less common, due to the wide separation in

frequency between the FM broadcast band and

the more popular amateur bands, and due also

to the limiting action which exists in all types

of FM receivers. Occasional interference with

FM broadcast by a harmonic of an amateur

transmitter has been reported; if this condi-

tion is encountered, it may be eliminated by

the procedures discussed in the first portion

of this chapter under Television Interference.

The use of frequency -modulation transmis-

sion by an amateur station is likely to result

in much less interference to broadcast recep-

tion than either amplitude -modulated telephony

or straight keyed c.w. This is true because,

insofar as the broadcast receiver is concerned,

the amateur FM transmission will consist of a

plain unmodulated carrier. There will be no

key clicks or voice reception picked up by

the b-c-1 set (unless it happens to be an FM

receiver which might pick up a harmonic of

the signal), although there might be a slight

click when the transmitter is put on or taken

www.americanradiohistory.com

376 TV and Broadcast Interference THE RADIO

Figure 12

WAVE -TRAP CIRCUITS

The circuit at (A) is the most common ar-

rangement, but the circuit at (B) may give

improved results under certain conditions.

Manufactured wave traps for the desired band

of operation may be purchased or the traps

may be assembled from the data given in

figure 14.

off the air. This is one reason why narrow -

band FM has become so popular with phone

enthusiasts who reside in densely populated

areas.

Interference Depending upon whether it is

Classifications traceable directly to causes

within the station or within

the receiver, broadcast interference may be

divided into two main classes. For example,

that type of interference due to transmitter

over -modulation is at once listed as b e i n g

caused by improper operation, while an inter-

fering signal that tunes in and out with a

broadcast station is probably an indication of

cross modulation or image response in the re-

ceiver, and the poorly- designed input stage

of the receiver is held liable. The various

types of interference and recommended cures

will be discussed in the following paragraphs.

Blanketing This is not a tunable effect, but

a total blocking of the receiver.

A more or less complete "washout" covers

the entire receiver range when the carrier is

switched on. This produces either a complete

blotting out of all broadcast stations, or else

knocks down their volume several decibels -

depending upon the severity of the interfer-

ence. Voice modulation of the carrier causing

the blanketing will be highly distorted or even

Figure 13

HIGH -ATTENUATION WAVE -TRAP

CIRCUIT

The two circuits may be tuned to the same

frequency for highest attenuation of a strong

signal, or the two traps may be tuned sep-

arately for different bands of operation.

unintelligible. Keying of the carrier which

produces the blanketing will cause an annoy-

ing fluctuation in the volume of the broadcast

signals.

Blanketing generally occurs in the imme-

diate neighborhood (inductive field) of a pow-

erful transmitter, the affected area being di-

rectly proportional to the power of the trans-

mitter. Also it is more prevalent with trans-

mitters which operate in the 160 -meter and

80 -meter bands, as compared to those on the

higher frequencies.

The remedies are to (1) shorten the receiv-

ing antenna and thereby shift its resonant fre-

quency, or (2) remove it to the interior of the

building, (3) change the direction of either

the receiving or transmitting antenna to mini-

mize their mutual coupling, or (4) keep the

interfering signal from entering the receiver

input circuit by installing a wavetrap tuned

to the signal frequency (see figure 12) or a

low -pass filter as shown in figure 21.

A suitable wave -trap is quite simple in con-

struction, consisting only of a coil and midget

variable capacitor. When the trap circuit is

tuned to the frequency of the interfering sig-

nal, little of the interfering voltage reaches

the grid of the first tube. Commercially manu-

factured wave -traps are available from several

concerns, including the J. W. Miller Co. in

Los Angeles. However, the majority of ama-

teurs prefer to construct the traps from spare

components selected from the "junk box."

The circuit shown in figure 13 is particu-

larly effective because it consists of two

traps. The shunt trap blocks or rejects the

frequency to which it is tuned, while the

series trap across the antenna and ground ter-

minals of the receiver provides a very low im-

pedance path to ground at the frequency to

www.americanradiohistory.com

HANDBOOK Wavetraps 377

BAND COIL, L CAPACITOR, C

1.8 Mc. 1 Inch no. 30 enom. 75 -,..ofd. var.

closewound on 1" form

3.5 Mc. 42 turns no. 30 enam. 50 -cAfd. var.

closewound on 1" form

7.0 Mc. 23 turns no. 24 enam. 50 -µsfd. var.

closewound on 1" form

14 Mc. 10 turns no. 24 enam. 50 -Aµfd. var.

closewound on 1" form

21 Mc. 7 turns no. 24 enam. 50 -ccfd. var.

closewound on 1" form

28 Mc. 4 turns no. 24 enam. 25 -µµfd. vor.

closewound on 1" form

50 Mc. 3 t no. 24 cram. 25 -Acfd. var.

spaced I /2" on 1" form

Figure 14

COIL AND CAPACITOR TABLE FOR

AMATEUR -BAND WAVETRAPS

Figure 15

MODIFICATION OF THE FIGURE 13

CIRCUIT

In this circuit arrangement the paralll -tuned

tank is inductively coupled to the antenna

lead with a 3 to 6 turn link instead of being

placed directly in series with the antenna

lead.

which it is tuned and by- passes the signal to

ground. In moderate interference cases, either

the shunt or series trap may be used alone,

while similarly, one trap may be tuned to one

of the frequencies of the interfering trans-

mitter and the other trap to a different inter-

fering frequency. In either case, each trap is

effective over but a small frequency range

and must be readjusted for other frequencies.

The wave -trap must be installed as close

to the receiver antenna terminal as practic-

able, hence it should be as small in size as

possible. The variable capacitor may be a

midget air -tuned trimmer type, and the coil

may be wound on a 1 -inch dia. form. The table

of figure 14 gives winding data for wave -traps

built around standard variable capacitors. For

best results, both a shunt and a series trap

should be employed as shown.

Figure 15 shows a two- circuit coup 1 e d

wave -trap that is somewhat sharper in tuning

and more efficacious. The specifications for

the secondary coil L, may be obtained from

the table of figure 14. The primary coil of the

shunt trap consists of 3 to 5 closewound turns

of the same size wire wound in the same di-

rection on the same form as L, and separated

from the latter by '4 of an inch.

Overmodulation A carrier modulated in excess

of 100 per cent acquires

sharp cutoff periods which give rise to tran-

sients. These transients create a broad signal

and generate spurious responses. Transients

caused by overmodulation of a radio -telephone

signal may at the same time bring about im-

pact or shock excitation of nearby receiving

antennas and power lines, generating inter-

fering signals in that manner.

Broadcast interference due to overmodula-

tion is frequently encountered. The remedy is

to reduce the modulation percentage or to use

a clipper -filter system or a high -level splatter

suppressor in the speech circuit of the trans-

mitter.

Cross Cross modulation or cross talk is

Modulation characterized by the amateur sig-

nal riding in on top of a strong

broadcast signal. There is usually no hetero-

dyne note, the amateur signal being tuned in

and out with the program carriers.

This effect is due frequently to a faulty in-

put stage in the affected receiver. Modulation

of the interfering carrier will swing the oper-

ating point of the input tube. This type of

trouble is seldom experienced when a varia-

ble-µ tube is used in the input stage.

Where the receiver is too ancient to incor-

porate such a tube, and is probably poorly

shielded at the same time, it will be better to

attach a wave -trap of the type shown in figure

12 rather than to attempt rebuilding of the re-

ceiver. The addition of a good ground and a

shield can over the input tube often adds to

the effectiveness of the wave -trap.

Transmission via A small amount of ca-

Capacitive Coupling pacitive coupling is now

widely used in receiver

r.f. and antenna transformers as a gain booster

at the high- frequency end of the tuning range.

The coupling capacitance is obtained by

means of a small loop of wire cemented close

to the grid end of the secondary winding, with

one end directly connected to the plate or an-

tenna end of the primary winding. (See figure

16.)

www.americanradiohistory.com

378 TV and Broadcast Interference THE RADIO

CAPACITIVE

COUPLING LOOP

Figure 16

CAPACITIVE BOOST COUPLING

CIRCUIT

Such circuits, included within the broadcast

receiver to bring up the stage gain at the

high -frequency end of the tuning ronge, have

a tendency to increase the susceptibility of

the receiver to interference from amateur -

band transmissions.

It is easily seen that a small capacitor at

this position will favor the coupling of the

higher frequencies. This type of capacitive

coupling in the receiver coils will tend to

pass amateur high- frequency signals into a re-

ceiver tuned to broadcast frequencies.

The amount of capacitive coupling may be

reduced to eliminate interference by moving

the coupling turn further away from the sec-

ondary coil. However, a simple wave -trap of

the type shown in figure 12, inserted at the

antenna input terminal, will generally accom-

plish the same result and is more to be recom-

mended than reducing the amount of capaci-

tive coupling (which lowers the receiver gain

at the high- frequency end of the broadcast

band). Should the wave -trap alone not suffice,

it will be necessary to resort to a reduction

in the coupling capacitance.

In some simple broadcast receivers, capaci-

tive coupling is obtained by closely coupled

primary and secondary coils, or as a result of

running a long primary or antenna lead close

to the secondary coil of an unshielded anten-

na coupler.

Phantoms With two strong local carriers ap-

plied to a non -linear impedance,

the beat note resulting from cross -modulation

between them may fall on some frequency

within the broadcast band and will be audible

at that point. If such a "phantom" signal falls

on a local broadcast frequency, there will be

heterodyne interference as well. This is a

common occurence with broadcast receivers in

the neighborhood of two amateur stations, or

an amateur and a police station. It also some-

times occurs when only one of the stations is

located in the immediate vicinity.

As an example: an amateur signal on 3514

kc. might beat with a local 2414 -kc. police

carrier to produce a 1100 -kc. phantom. If the

two carriers are strong enough in the vicinity

of a circuit which can cause rectification, the

1100 -kc. phantom will be heard in the broad-

cast band. A poor contact between two oxi-

dized wires can produce rectification.

Two stations must be transmitting simulta-

neously to produce a phantom signal; when

either station goes off the air the phantom

disappears. Hence, this type of interference

is apt to be reported as highly intermittent and

might be difficult to duplicate unless a test

oscillator is used "on location" to simulate

the missing station. Such interference cannot

be remedied at the transmitter, and often the

rectification takes place some distance from

the receivers. In such occurrences it is most

difficult to locate the source of the trouble.

It will also be apparent that a phantom

might fall on the intermediate frequency of a

simple superhet receiver and cause interfer-

ence of the untunable variety if the manufac-

turer has not provided an i -f wave -trap in the

antenna circuit.

This particular type of phantom may, in

addition to causing i -f interference, generate

harmonics which may be tuned in and out with

heterodyne whistles from one end of the re-

ceiver dial to the other. It is in this manner

that birdies often result from the operation of

nearby amateur stations.

G hen one component of a phantom is a

steady, unmodulated carrier, only the intelli-

gence present on the other carrier is conveyed

to the broadcast receiver.

Phantom signals almost always may be

identified by the suddenness with which they

are interrupted, signalizing withdrawal of one

party to the union. This is especially baffling

to the inexperienced interference -locater, who

observes that the interference suddenly disap-

pears, even though his own transmitter re-

mains in operation.

If the mixing or rectification is taking place

in the receiver itself, a phantom signal may

be eliminated by removing either one of the

contributing signals from the receiver input

circuit. A wave -trap of the type shown in fig-

ure 12, tuned to either signal, will do t h e

trick. If the rectification is taking place out-

side the receiver, the wave -trap should be

tuned to the frequency of the phantom, instead

of to one of its components. I -f wave -traps

may be built around a 2.5- millihenry r -f choke

as the inductor, and a compression -type mica

padding capacitor. The capacitor should have

a capacitance range of 250 -525 µµfd. for the

175- and 206 -kc. intermediate frequencies;

65 -175 µµfd. for 260 -kc. and other intermedi-

www.americanradiohistory.com

HANDBOOK Audio Rectification 379

ates lying between 250- and 400 -kc; and

17 -80 µµEd. for 456 -, 465-, 495 -, and 500 -kc.

Slightly more capacitance will be required for

resonance with a 2.1 millihenry choke.

Spurious This sort of interference arises

Emissions from the transmitter itself. The

radiation of any signal (other than

the intended carrier frequency) by an amateur

station is prohibited by FCC regulations. Spu-

rious radiation may be traced to imperfect neu-

tralization, parasitic oscillations in the r -f or

modulator stage s, or to "broadcast- band"

variable- frequency oscillators or e.c.o.'s.

Low- frequency parasitics may actually oc-

cur on broadcast frequencies or their near sub -

harmonics, causing direct interference to pro-

grams. An all -wave monitor operated in the

vicinity of the transmitter will detect these

spurious signals.

The remedy will be obvious in individual

cases. Elsewhere in this book are discussed

methods of complete neutralization and the

suppression of parasitic oscillations in r -f

and audio stages.

A -c /d -e Receivers Inexpensive tab 1 e- model

a -c /d -c receivers are par-

ticularly susceptible to interference from ama-

teur transmissions. In fact, it may be said

with a fair degree of assurance that the major-

ity of BCI encountered by amateurs operating

in the 1.8 -Mc. to 29 -Mc. range is a result of

these inexpensive receivers. In most cases

the receivers are at fault; but this does not

absolve the amateur of his responsibility in

attempting to eliminate the interference.

Stray Receiver In most cases of interference

Rectification to inexpensive receivers, par-

ticularly those of the a -c /d -c

type, it will be found that stray receiver recti-

fication is causing the trouble. The offending

stage usually will be found to be a high -mu

triode as the first audio stage following the

second detector. Tubes of this type are quite

non -linear in their grid characteristic, and

hence will readily rectify any r -f signal ap-

pearing between grid and cathode. The r -f sig-

nal may get to the tube as a result of direct

signal pickup due to the lack of shielding, but

more commonly will be fed to the tube from

the power line as a result of the series heater

string.

The remedy for this condition is simply to

insure that the cathode and grid of the high -mu

audio tube (usually a 12SQ7 or equivalent) are

at the same r -f potential. This is accomplished

by placing an r -f by -pass capacitor with the

shortest possible leads directly from grid to

cathode, and then adding an impedance in the

lead from the volume control to the grid of the

HIGH -MU TUBE

SUCH AS 12507

Figure 17

CIRCUITS FOR ELIMINATING AUDIO -

STAGE RECTIFICATION

audio tube. The impedance may be an amateur

band r -f choke (such as a National R -I00U)

for best results, but for a majority of cases

it will be found that a 47,000 -ohm V2-watt re-

sistor in series with this lead will giv.e satis-

factory operation. Suitable circuits for such

an operation on the receiver are given in fig-

ure 17.

In many a.c. -d.c. receivers there is no r -f

by -pass Included across the plate supply recti-

fier for the set. If th e r e is an appreciable

level of r -f signal on the power line feeding

the re ce i v e r, r -f rectification in the power

rectifier of the receiver can cause a particu-

larly bad type of interference which may be

received on other broadcast receivers in the

vicinity in addition to the one causing the

rectification. The soldering of a 0.01 -pfd. disc

ceramic capacitor directly from anode to cath-

ode of the power rectifier (whether it is of the

vacuum -tube or selenium- rectifier type) usual-

ly will by -pass the r -f signal across the recti-

fier and thus eliminate the difficulty.

"Floating" Volume Several sets have been

Control Shafts encountered where there

was only a slightly inter-

fering signal; but, upon placing one's hand up

to the volume control, the signal would great-

ly increase. Investigation revealed that the

volume control was installed with its shaft

insulated from ground. The control itself was

connected to a critical part of a circuit, in

many instances to the grid of a high -gain au-

dio stage. The cure is to install a volume con-

trol with all the terminals insulated from the

shaft, and then to ground the shaft.

www.americanradiohistory.com

380 TV and Broadcast Interference THE RADIO

BAND COIL, L CAPACITOR, C

3.5 Mc. 17 turns

no. 14 enameled 100-;.,.íd.

3 -inch diameter

23/4-inch length

variable

7.0 Mc.

11 turns

no. 14 enameled

212 -inch 100- n,.fd.

diameter

l'2 -inch length

variable

14 and

21 Mc.

4 turns

no. 10 enameled 100 -,. ;.td.

3 -inch diameter

118 -inch length

variable

27 and

28 Mc.

3 turns

I ¡ -inch o.d.

copper tubing 100- ,,id.

2 -inch diameter

1 -inch length

variable

Figure 18

COIL AND CAPACITOR TABLE

FOR A -C LINE TRAPS

Power -Line a hen radio - frequency energy

Pickup from a radio transmitter enters a

broadcast receiver through the

a -c power lines, it has either been fed back

into the lighting system by the offending trans-

mitter, or picked up from the air by over -head

power lines. Underground lines are seldom re-

sponsible for spreading this interference.

To check the path whereby the interfering

signals reach the line, it is only necessary to

replace the transmitting antenna with a dum-

my antenna and adjust the transmitter for max-

imum output. If the interference then ceases,

overhead lines have been picking up the en-

ergy. The trouble can be cleared up by install-

ing a wave -trap or a commercial line filter in

the power lines at the receiver. If the receiver

is reasonably close to the transmitter, it is

very doubtful that changing the direction of

the transmitting antenna to right angles with

the overhead lines will eliminate the trouble.

If, on the contrary, the interference con-

tinues when the transmitter is connected to

the dummy antenna, radio- frequency energy is

being fed directly into the power line by the

transmitter, and the station must be inspected

to determine the cause.

One of the following reasons for the trouble

will usually be found: (1) the r -f stages are

not sufficiently bypassed and /or choked, (2)

the antenna coupling system is not performing

efficiently, (3) the power transformers have

no electrostatic shields; or, if shields are pre-

sent, they are ungrounded, (4) power lines are

running too close to an antenna or r -f circuits

carrying high currents. If none of these causes

r-i -

METAL BO

LOV

TO A.C. C I TO TRANSMITTER

LINE C I OR RECEIVER

Qv

I r SHIELD BRAID

- L J

SHIELD BRAID

Figure 19

RESONANT POWER -LINE

WAVE -TRAP CIRCUIT

The resonant type of power -line filter is

more effective than the more conventional

"brute force type of line filter, but requires

tuning to the operating frequency of the

transmitter.

apply, wave -traps must be installed in the

power lines at the transmitter to remove r -f

energy passing back into the lighting system.

The wave -traps used in the power lines at

transmitter or receiver must be capable of

passing relatively high current. The coils are

accordingly wound with heavy wire. Figure 18

lists the specifications for power line wave -

trap coils, while figure 19 illustrates the meth-

od of connecting these wave- traps. Observe

that these traps are enclosed in a shield box

of heavy iron or steel, well grounded.

All -Wave Each complete- coverage home re-

Receivers ceiver is a potential source of an-

noyance to the transmitting ama-

teur. The novice short -wave broadcast- listener

who tunes in an amateur station often con-

siders it an interfering signal, and complains

accordingly.

Neither selectivity nor image rejection in

most of these sets is comparable to t ho s e

properties in a communication receiver. The

result is that an amateur signal will occupy

too much dial space and appear at more than

one point, giving rise to interference on ad-

jacent channels and distant channels as well.

If carrier- frequency harmonics are present

in the amateur transmission, serious interfer-

ence will result at the all -wave receiver. The

harmonics may, if the carrier frequency has

been so unfortunately chosen, fall directly

upon a favorite short -wave broadcast station

and arouse warranted objection.

The amateur is apt to be blamed, too, for

transmissions for which he is not responsible,

so great is the public ignorance of short -wave

allocations and signals. Owners of all -wave

receivers have been quick to ascribe to ama-

teur stations all signals they hear from tape

machines and V- wheels, as well as stray tones

and heterodyne flutters.

www.americanradiohistory.com

HANDBOOK Image Interference 381

The amateur cannot be held responsible

when his carrier is deliberately tuned in on

an all -wave receiver. Neither is he account-

able for the width of his signal on the receiver

dial, or for the strength of image repeat points,

if it can be proven that the receiver design

does not afford good selectivity and image re-

jection.

If he so desires, the amateur (or the owner

of the receiver) might sharpen up the received

signal somewhat by shortening the receiving

antenna. Set retailers often supply quite a

sizeable antenna with all -wave receivers, but

most of the time these sets perform almost as

well with a few feet of inside antenna.

The amateur is accountable for harmonics

of his carrier frequency. Such emissions are

unlawful in the first place, and he must take

all steps necessary to their suppression. Prac-

tical suggestions for the elimination of har-

monics have been given earlier in this chap-

ter under Television Interference.

Image Interference In addition to those types

of interference al read y

discussed, there are two more which are com-

mon to superhet receivers. The prevalence of

these types is of great concern to the ama-

teur, although the responsibility for their ex-

istence more properly rests with the broadcast

receiver.

The mechanism whereby image production

takes place may be explained in the following

manner: when the first detector is set to the

frequency of an incoming signal, the high -fre-

quency oscillator is operating on another fre-

quency which differs from the signal by the

number of kilocycles of the intermediate fre-

quency. Now, with the setting of these two

stages undisturbed, there is another signal

which will beat with the high- frequency oscil-

lator to produce an i -f signal. This other sig-

nal is the so- called image, which is separated

from the desired signal by twice the inter-

mediate frequency.

Thus, in a receiver with 175 -kc. i.f., tuned

to 1000 kc.: the h -f oscillator is operating on

1175 kc., and a signal on 1350 kc. (1000 kc.

plus 2 x 175 kc.) will beat with this 1175 kc.

oscillator frequency to produce the 175 -kc. i -f

signal. Similarly, when the same receiver is

tuned to 1400 kc., an amateur signal on 1750

kc. can come through.

If the image appears only a few cycles or

kilocycles from a broadcast carrier, heterodyne

interference will be present as well. Other-

wise, it will be tuned in and out in the manner

of a station operating in the broadcast band.

Sharpness of tuning will be comparable to that

of broadcast stations producing the same a -v -c

voltage at the receiver.

The second variety of superhet interference

is the result of harmonics of the receiver h -f

oscillator beating with amateur carriers to pro-

duce the intermediate frequency of the receiv-

er. The amateur transmitter will always be

found to be on a frequency equal to some har-

monic of the receiver h -f oscillator, plus or

minus the intermediate frequency.

As an example: when a broadcast superhet

with 465 -kc. i.f. is tuned to 1000 kc., its high -

frequency oscillator operates on 1465 kc. The

third harmonic of this oscillator frequency is

4395 kc., which will beat with an amateur sig-

nal on 3930 kc. to send a signal through the

i -f amplifier. The 3930 kc. signal would be

tuned in at the 1000 -kc. point on the dial.

Some oscillator harmonics are so related to

amateur frequencies that more than one point

of interference will occur on the receiver dial.

Thus, a 3500 -kc. signal may be tuned in at

six points on the dial of a nearby broadcast

superhet having 175 kc. i.f. and no r -f stage.

Insofar as remedies for image and harmonic

superhet interference are concerned, it is well

to remember that if the amateur signal did not

in the first place reach the input stage of the

receiver, the annoyance would not have been

created. It is therefore good policy to try to

eliminate it by means of a wave -trap or low -

pass filter. Broadcast superhets are not al-

ways the acme of good shielding, however,

and the amateur signal is apt to enter the cir-

cuit through channels other than the input cir-

cuit. If a wave -trap or filter will not cure the

trouble, the only alternative will be to attempt

INPUT

O T

OUTPUT

O

CONSTANT K TYPE

INPUT

O T T

OUTPUT

M- DERIVED TYPE

FREQUENCY

Figure 20

TYPES OF LOW -PASS FILTERS

Filters such as these may be used in the

circuit between the antenna and the input

of the receiver.

www.americanradiohistory.com

382 TV and Broadcast Interference

ANT L, L2

TO RECEIVER ANT. POST

T`

GND.

TC2 Cs TCA

1 O TO RECEivER CND. POST

Figure 21

COMPOSITE LOW -PASS FILTER

CIRCUIT

This filter is highly effective in reducing

broadcast interference from all high frequen-

cy stations, and requires no tuning. Con-

stants for 400 ohm terminal impedance and

1600 kc. cutoff are as follows: L,, 65 turns

no. 22 d.c.c. closewound on 1%2 in. dia. form.

L2, 41 turns ditto, not coupled to Lr. C,,

250 µµfd. fixed mica capacitor. C2, 400 Auld.

fixed mica capacitor. C3 and C4, ISO Auld.

fixed mica capacitors, former of S% toler-

ance. With some receivers, better results

will be obtained with o 200 ohm carbon re-

sistor inserted between the filter and an-

tenna post on the receiver. With other re-

ceivers the effectiveness will be improved

with a 600 ohm carbon resistor placed from

the antenna post to the ground post on the

receiver. The filter should be placed as

close to the receiver terminals os possible.

to select a transmitter frequency such that

neither image nor harmonic interference will be

set up on favorite stations in the susceptible

receivers. The equation given earlier may be

used to determine the proper frequencies.

Low Pass Filters The greatest drawback of

the wave -trap is the fact

that it is a single- frequency device; i.e. -it

may be set to reject at one time only one fre-

quency (or, at best, an extremely narrow band

of frequencies). Each time the frequency of

the interfering transmitter is changed, every

wave -trap tuned to it must be retuned. A much

more satisfactory device is the wave filler

which requires no tuning. One type, the low -

pass filter, passes all frequencies below one

critical frequency, and eliminates all higher

frequencies. It is this property that makes the

device ideal for the task of removing amateur

frequencies from broadcast receivers.

A good low -pass filter designed for maxi-

mum attenuation around 1700 kc. will pass

all broadcast carriers, but will reject signals

originating in any amateur band. Naturally

such a device should be installed only in

standard broadcast receivers, never in all -

wave sets.

Two types of low -pass filter sections are

shown in figure 20. A composite arrangement

comprising a section of each type is more

effective than either type operating alone. A

composite filter composed of one K- section

and one shunt -derived M- section is shown in

figure 21, and is highly recommended. The

M- section is designed to have maximum atten-

uation at 1700 kc., and for that reason C,

should be of the "close tolerance" variety.

Likewise, C, should not be stuffed down in-

side L, in the interest of compactness, as

this will alter the inductance of the coil appre-

ciably, and likewise the resonant frequency.

If a fixed 150 µµfd. mica capacitor of 5 per

cent tolerance is not available for C1r a com-

pression trimmer covering the range of 125-

175 µµEd. may be substituted and adjusted to

give maximum attenuation at about 1700 kc.

19 -5 HI -FI Interference

The rapid growth of high -fidelity sound systems

in the home has brought about many cases of

interference from a nearby amateur transmitter.

In most cases, the interference is caused by

stray pickup of the r -f signal by the interconnect-

ing leads of the hi -fi system and audio rectifi-

cation in the.low level stages of the amplifier.

The solution to this difficulty, in general, is to

bypass and filter all speaker and power leads

to the hi -fi amplifier and preamplifier. A com-

bination of a VHF choke and 500 µpfd ceramic

disc capacitors in each power and speaker lead

will eliminate r -f pickup in the high level section

of the amplifier. A filter such as shown in figure

17A placed in the input circuit of the first audio

stage of the preamplifier will reduce the level of

the r -f signal reaching the input circuit of the

amplifier. To prevent loss of the higher audio

frequencies it may be necessary to decrease the

value of the grid bypass capacitor to 50 ppfd

or so.

Shielded leads should be employed between the

amplifier and the turntable or f -m tuner. The

shield should be. grounded at both ends of the

line to the chassis of the equipment, and care

should be taken to see that the line does not

approach an electrical half -wavelength of the

radio signal causing the interference. In some

instances, shielding the power cable to the hi -fi

equipment will aid in reducing interference. The

framework of the phonograph turntable should be

grounded to the chassis of the amplifier to reduce

stray r -f pickup in the turntable equipment.

www.americanradiohistory.com

562 Receivers and Transceivers THE RADIO

(figure 37). The front panel has the same

dimensions as the outside of this box, and

takes the form of a shallow pan, about 3/4 -inch

deep (figure 36). The panel is affixed to two

angle brackets mounted on the edges of the

sub -panel. The two meters are mounted to the

sub -panel, as is the dial mechanism and the

pilot lamp. The various potentiometers are

mounted to small L- shaped plates spaced away

from the sub -panel. The pan -shaped panel is

merely a decorative cover that finishes the

appearance of the unit.

The dial is home -made and is driven by a

35 -1 gear train made from re- mounted parts

of a surplus BC -453 ( "Command ") receiver

dial (figure 37). The dial drive and pointer

may be made from a broadcast -type slide rule

dial and the escutcheon is cut and formed

from a piece of bakelite and is suitably

engraved.

Figure 38

UNDER -CHASSIS VIEW OF TRANSCEIVER

Neat wiring and use of cabling techniques makes "clean" looking assembly. Power leads and

long "runs" are laced into main cable passing in a square about r.f. section. Small components are

soldered directly to socket pins, or are mounted on phenolic terminal boards, as is the case of the

squelch components. R.I. coils and padding capacitors are at center of layout, beneath main

tuning capacitor gang. Individual shield sections separate the r.f. stages. Change -over relay RY

is mounted to rear wall of chassis next to antenna receptacle.

www.americanradiohistory.com

HANDBOOK Deluxe Mobile Transceiver 561

Figure 37

VIEW OF TRANSCEIVER WITH FRONT PANEL REMOVED

The various panel controls are mounted on L- shaped brackets attached to the sub -panel by means

of metal bushings. Meters are mounted to the sub -panel by means of encircling straps. The dial

mechanism is made from geared portions of "command" receiver dial drive fixed to a thin

phenolic plate.

switches the 250 -volt supply from the receiver

section to the transmitter section and section B

transfers the antenna from the receiver to the

transmitter. It is necessary to remove the B -plus

from the modulator and power amplifier of

the transmitter during reception, and this may

be accomplished by switching off the high

voltage supply by means of an auxiliary relay

whose actuator coil is paralleled with the coil

of relay RY1. The auxiliary relay should be

located at the power supply.

Transceiver

Construction This transceiver is an excellent

example of the fine workman-

ship possible by an amateur

adept in sheet metal work and who has the

necessary shop facilities. The chassis -cabinet is

made of 14 -gauge sheet durai, cut and bent to

size by a sheet metal shop. The assembly is

made up of six pieces: A wrap- around back

and side piece, removable top and bottom

plates, the chassis, the sub -panel, and the front

panel. Ventilation holes are drilled in the top

plate and the wrap -around section to venti-

late the unit, as a considerable amount of heat

is generated by the tubes.

The chassis is constructed with a 1/2 -inch

lip around the edges which is bolted to the

wrap- around piece and the sub -panel. In order

to conserve height, the chassis has a "step" in

it to allow room for the taller tubes (6146

and 6BQ6 -GT's) and the modulation trans-

former. Less room above the chassis is re-

quired for the receiver section, and a corres-

pondingly greater area beneath the chassis

allows room for the receiver coils and stage

shields. The "step" can be seen in figure 36,

running from the front to the back of the

chassis, immediately to the right of the ganged

tuning capacitors.

The chassis, the wrap- around piece, and the

sub -panel make up a complete TVI -proof box

www.americanradiohistory.com

560 Receivers and Transceivers THE RADIO

transformer coupled to two 6BQ6 -GT pen-

todes connected as zero bias class B modu-

lators. No grid bias or screen voltage is

required for the modulator, and the audio

driving voltage is applied to the screens of

the tubes. The control grid is connected to

Figure 36

TOP VIEW OF TRANSCEIVER SHOWS

PLACEMENT OF MAJOR COMPONENTS

The receiver section of the unit occupies the

left -hand section of the chassis, with the

transmitter section at right. The "step" in the

chassis is at the right of the main tuning

gang, running parallel to it, from the front to

the rear of the chassis.

Receiver i.f. section runs along left edge of

chassis, with squelch, regulator tube, and

second conversion oscillator in the adjacent

row. R.I. and audio stages are next to tuning

gang. The three sections of capacitors nearest

the front panel are for the receiver portion,

while the rear capacitors are for the trans-

mitter section. The 6146 plate tank coil is at

the rear of the chassis.

Modulator section occupies right -hand portion

of chassis, with transmitter r.f. stages imme-

diately to the left. 6CL6 buffer tube is

shielded and directly in front of the 6146.

the cathode. This simple circuit is capable of

over 40 watts of audio output. Negative peak

control is exercised by a silicon rectifier placed

in series with the secondary winding of the

modulation transformer, and a simple low

pass audio filter composed of the leakage

reactance of the modulation transformer plus

the plate bypass capacitor of the r.f. amplifier

stage reduces the higher order audio harmonics

generated by this system. A high level of "talk

power" is thus insured.

Filament and The transmitter is designed

Control Circuits for either 6- or 12 -volt

operation. The circuit of

figures 34 -35 shows 6 -volt configuration. For

12 -volt operation, it is only necessary to re-

wire the power plugs as shown and filament

switching is automatic.

Change -over from receive to transmit is

accomplished by means of relay RY1, which is

actuated by the microphone button, and which

has a d.c. coil suited to the voltage of the auto-

mobile battery. Contact section A of this relay

www.americanradiohistory.com

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www.americanradiohistory.com

HANDBOOK Deluxe Mobile Transceiver 557

stability. The oscillator runs continuously and

is voltage regulated.

A single i.f. transformer (T1) provides

sufficient image selectivity at 4.26 Mc. and

no additional amplification or tuned circuits

are required. The second intermediate fre-

quency is 260 kc., and a 6BE6 multi -grid

converter tube is used as a mixer to this fre-

quency. The local oscillator is crystal con-

trolled at 4.52 Mc., and makes use of the

6BE6 as a "hot cathode' crystal oscillator.

Precise adjustment of the oscillator frequency

may be made by means of the variable induc-

tance (L;) in the grid- cathode circuit of the

mixer tube. The choice of frequency of the

mixing oscillator is important in that no har-

monic frequencies of the oscillator should fall

into the 10 meter band, or into its "image"

frequency band. This insures that undesired

"birdies" or spurious responses of the receiver

are reduced to an absolute minimum.

Two stages of i.f. amplification employing low

filament drain 6BJ6 tubes provide sufficient re-

ceiver gain, and are followed by a 6AL5 de-

tector/a.v.c. rectifier stage. A two -stage noise

limiter patterned after the popular "twin -

noise squelch (TNS) circuit" provides maxi-

mum noise rejection with minimum audio

distortion. A 12AX7 and 6AL5 are used in

this portion of the receiver. A 12AT7 tube

serves a dual purpose as a first audio stage

and v.t.v.m. -type S -meter amplifier, followed

by a 6ÁQ5 audio output stage. The S -meter

circuit makes use of a "backwards reading"

meter that rests at full scale. The a.v.c. voltage

applied to the amplifier tube reduces the meter

current in accordance with the strength of the

incoming signal.

The Transmitter Section. The transmitter sec-

tion of the transceiver is shown in figure 35,

and in outline form in figure 32. A 12AT7

dual triode serves as a mixer -oscillator stage,

beating the receiver v.f.o. with a 4.26 Mc.

crystal (equal to the receiver intermediate fre-

quency) . The sum of these two frequencies is

the transmitting frequency, which is equal to

the frequency of reception. Following the

mixer -oscillator are two gang -tuned r.f. am-

plifier stages employing high gain 6CL6 pen-

tode tubes. The second stage is neutralized for

maximum stability. The power amplifier stage

uses a single 6146 in a pi- network output

circuit, which is also gang -tuned in conjunc-

tion with the exciter and v.f.o.

Tuning and loading controls of the power

amplifier stage are located on the rear of

the chassis and need not be readjusted unless

a change is made in the antenna system (figure

39) . Antenna change -over is controlled by a

section of relay RYI. Grid and plate currents

of the 6146 are monitored by meter M2.

The Modulator Section. The modulator is de-

signed to work with either a ceramic -type

crystal microphone, or a high impedance dy-

namic unit. A 12AT7 serves as a two stage

resistance coupled amplifier, exciting a parallel

connected 12ÁU7 driver. This, in turn, is

Figure 33

MINIATURE

POWERHOUSE

PACKS PLENTY

OF PUNCH!

The transceiver is built

in a custom -made case

which permits maximum

utilization of available

space. Mounting flanges

may be seen attached to

upper portion of trans-

ceiver case. At left of

main tuning dial are vol-

ume control (with on -off

switch), r.f. gain control,

and squelch. At right are

microphone level control,

meter switch, and micro-

phone receptacle.

riwgr-

www.americanradiohistory.com

556 Receivers and Transceivers THE RADIO

ANT

R.F. AMP.

(26 Mc.) 1ST MIX

(4.26 AK.) 2NDI%. I.F I.F.

(280 RC.

DET. GAIN AUDIO AUDIO

AVC 5-METER

® ® ® ® ® ® 12AT7 ®'I

(4.26MC)

SPEECH AMP. DRIVER

MIC J

Figure 32

BLOCK DIAGRAM OF THE TRANSCEIVER

The tuning oscillator of the unit covers the range of 23.74 -25.44 megacycles. The transmitter

conversion crystal (4.26 Mc.) is the same frequency as the first i.f. of the receiver, thus placing

receiver and transmitter operating frequencies at the same soot on the tuning dial. Receiver

selectivity is obtained by use of two i.f. stages at 260 kc. R.I. circuits of both transceiver sections

are ganged for single dial control.

SPMR

Circuit A block diagram of the trans -

Description ceiver is shown in figure 32.

The circuit utilizes a double

conversion receiver employing eleven tubes

and a voltage regulator, and a v.f.o.- controlled

amplitude modulated transmitter having eight

tubes. A feature of the unit is that transmitting

and receiving frequencies are locked together

and controlled by one master oscillator. All

variable r.f. circuits are tracked for single con-

trol tuning. The operator merely tunes the

transceiver to the station he desires to contact,

pushes the microphone control button and the

transmitter is tuned to the same frequency,

ready to "talk."

The Receiver Section. The receiver portion of

the transceiver is shown in figure 34, and in

outline form in figure 32. Double conversion

is used, with the second conversion oscillator

crystal controlled. The first conversion oscil-

lator is also the v.f.o. for the transmitter sec-

tion, as explained later. The three r.f. circuits

of the receiver section (r.f. stage, mixer, and

oscillator) are gang -tuned for proper tracking

across the 10 meter band.

The r.f. stage utilizes a 6BZ6 high gain,

semi- remote cutoff pentode to achieve maxi-

mum signal gain without troublesome cross -

modulation effects from strong nearby signals.

The circuit of this stage is conventional, except

that the cathode return may be removed from

the gain buss by switch S) for optimum weak

signal response, if desired. Partial a.v.c. is ap-

plied to the 6BZ6 by means of a high im-

pedance voltage divider in the a.v.c. system.

A 6BA7 multi -grid converter tube is used

as a mixer from the operating frequency to

the first intermediate frequency of 4.26 Mc.

Mixer injection voltage is applied to the #1

grid of the 6BA7. The local oscillator em-

ploys a 6ÁH6 and tunes the range of 23,740-

25,400 kc., with a slight overlap at both ends

of the range. A high -C "hot cathode" oscillator

circuit is employed for maximum frequency

www.americanradiohistory.com

HANDBOOK Deluxe Mobile Transceiver 555

shown in figure 30. This unit is sufficient

to run one converter at a time.

27 -6 A Deluxe

Mobile Transceiver

The modern automobile leaves little room

for radio equipment mounted in proximity

to the driver. Mobile equipment, as a result,

must be built more compactly in order to fit

in the dashboard firewall area available for

auxiliary equipment. The amateur having sheet

Figure 31

COMPACT TRANSCEIVER OFFERS

ULTIMATE IN MOBILE

COMMUNICATION

This compact a.m. transceiver is a complete

10 meter station, packaged so that it will fit

into all but the most cramped automobiles.

The transmitter section runs up to 70 watts

input and is designed for "on frequency"

operation with the receiver section. The easy -

to -read dial controls the master oscillator for

both transmission and reception. The operator

merely tunes the transceiver to the station he

desires to contact and the transmitter is

automatically tuned to the correct frequency.

The transceiver is mounted in the car by

means of dashboard clamps fastened to the

top of the unit by means of a sliding fixture.

Top and bottom plates are removable by

means of snap fasteners, and are perforated

for good ventilation. Simplicity of operation

permits transceiver to be operated without the

driver taking his eyes from the road.

560 10N

Ti = 125 V., 50 MA.

6.3v.,2A

STANCOR PI-6121

SR= SELENIUM RECTI FIER. S0 MA.

Figure 30

SCHEMATIC, CONVERTER POWER

SUPPLY

8+ 70 V.

B+RCG

6.3 V.

GND

metal working facilities at hand is indeed

fortunate, as he may custom -form his equip-

ment chassis and cabinet to fit the space pro-

vided in his particular automobile.

Described in this section is a deluxe trans-

ceiver, designed and built by W7JNC which

will fit easily into all but the most cramped

automobiles. The unit is a complete 10 meter

station capable of running up to 70 watts

input, having a sensitive double conversion

receiver, and packaged in a cabinet measuring

only 11 inches wide, 4 inches high, and 8

inches deep. The transceiver is suited for

either mobile or fixed -station operation.

www.americanradiohistory.com

554 Receivers and Transceivers THE RADIO

Figure 28

PLACEMENT OF MAJOR COMPONENTS ABOVE THE CHASSIS

I, 1

Figure 29

CLOSE -UP OF

2 -METER R.F.

AMPLIFIER STAGE

A shield partition passes

across the center of Nu-

vistor socket. The grid

compartment is at the

right, and plate cornpart-

ment at the left. Coil L,

is wound on high value

composition resistor. Six -

meter r.f. section is iden-

tical except for coil

changes.

www.americanradiohistory.com

552 Receivers and Transceivers

verter. Adjust the converter output coil

(L6 /L16) for maximum receiver noise, making

sure that you are not tuning to the image fre-

quency of the receiver. Connection to the

receiver should be made by means of a short

length of coaxial line to prevent spurious

signal pick -up in the 28 -30 Mc. range.

With these preliminary adjustments made,

the r.f. stage is ready for test and alignment.

Start with the 144 Mc. section. Remove the

B+ to coil L3 and insert a 6CW4 in the r.f.

socket. Connect a temporary antenna to the

converter and tune in a strong local test

signal. Make sure signal pickup is via the an-

tenna and not by indirect pickup via coils L3

or L5. Roughly peak coils L3, L ;, and L6 for

Figure 26

UNDER -CHASSIS

VIEW OF "SIAMESE"

CONVERTER

The converter chassis has been removed from

the end plates for this photograph. The two

crystal oscillators are at the center of the

chassis, with the mixer stages adjacent to

them. At the ends of the chassis are the r.f.

amplifiers. Note that a T- shaped shield iso-

lates the input and output circuits of the r.f.

amplifier from the remainder of the circuitry.

The shields are made up of thin flashing

copper and are about 112 inches high. The

small leg of the shield passes across the center

of the Nuvistor socket, and the grid -plate

blocking capacitor passes through a hole

drilled in this partition.

maximum signal. Now, carefully spread and

adjust the turns of coil L2 for minimum re-

ceived signal. The neutralization point will be

a sharp and almost complete signal null. If

neutralization is obscure, add or remove a

turn or two of wire from coil L2.

Now, reconnect the B -plus lead to the plate

coil of the r.f. stage and tune in a weak signal

near the center of the desired tuning range.

Peak coils L3, L5, and L6. Coil L1 will tune

very broadly. Recheck the neutralization once

again (after removing the r.f. B -plus lead)

and secure the turns of coil L2 with a spot of

cellulose cement or colorless nail polish. As a

final check, measure the plate current of the

r.f. stage. It should run approximately 8 ma.

and should not vary when the antenna is dis-

connected from the stage. A variation in plate

current indicates oscillation of the r.f.

amplifier.

If a noise generator is available, coil L1 and

the antenna tap can be adjusted for a one-

decibel or so improvement in noise figure

after the above adjustments are completed.

Adjustment of the six -meter converter is

identical to the above outline.

The Converter

Power Supply

and 105 volts

the mixer and

Plate power requirements of

each converter are 70 volts

at 8 ma. for the r.f. stage,

at approximately 10 ma. for

oscillator. A suitable supply is

www.americanradiohistory.com

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www.americanradiohistory.com

550 Receivers and Transceivers

SCHEMATIC, "SIAMESE"

Figure 24

CONVERTER FOR 2- AND 6- METERS

Coil data: LI, L3 -5 turns :26 e., on 14 -inch

diameter polystyrene rod or form. Wind

3/8 -inch long, top 2 turns from ground end

on L,. Adjust by spreading turns.

L2- Neutralizing coil. 20 turns :30 e. on

5 :32" diameter 10 megohm resistor, close

wound. Adjust by spreading turns.

L4 -I turn hookup wire over 8 -plus end of

coil L3.

L5-6 turns, some as L,. Tap 2 turns from

ground end.

L6, 116 -26 turns :32 e. on 1/4-inch slug -tuned

form, close wound. (Cambridge :PLS -6 v.h.f.

form with green colored slug.)

L7, LIT -2 turns hookup wire over 8 -plus end of

Coils L6 and L,6.

/44-146 MC

L2

vl

6Cwa

A. F

C2,500

LI L3 LS

L8 -512 turns hookup wire close wound about

the body of 6 Id. trimmer capacitor.

L9 -19/s turns :32 e. close wound about the

body of 6 ,o.fd. trimmer capacitor.

L,0 -16 turns :28 e., tap 51/2 turns from

ground. Some as 1.1.

L,1- Neutralizing coil. 50 turns :36 e., wound

some as L2.

L,2 -19 turns :28 e., some construction os L6.

L,3 -1 turn hookup wire over B -plus end of

coil L12.

L14 -Some as L,3, wound over ground end of

coil L15.

L,5 -17 turns :28, same construction as L6.

L18 -25 turns :32 e., same construction as L6.

L19 -2 turns hookup wire over 8 -plus end of

coil L16. See text.

vz

6AK5

MIA

4

50 -52 MC.

Lio

Ln

°411. L

Ca, 1- C3

500 = 1000

V4

6Cw4

A.F.

C12,500

L6

C 20

500

O

9

C6 000

2B-30 MC.

V3

6AK5

Yac

X1 c

38.66 MC

Li2

t O Q p L13

° -3

= C23 Cis

500 -T- 1000

C7

1000

Lu

P,

Lis

Vs

6AK5

MIA.

5

IMF MC.I

` 51

% --o A- I

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Lis

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Sz

Cu

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Re

2206

NOTES

I. ALL RESISTORS i /2-WATT.

2.500 ULF CAPACITORS ERIE GP -500 SILVCR MICA BUTTON.

3. 1000 LUF CAPACITORS, ERIE GP -500 SILVER MICA BUTTON.

4. 61.UF TRIMMER CAPACITORS CENrRALAB B2P -6 OR EOUIV.

5. NUVISTOR SOCKETS CINCH-JONES l33-65 -10 -001.

6. FILAMENT BYPASS CAPACITORS: CENTRALAB DISC 10501

Cis

I1

000

Ve

6AK5

OSC.

Br 70V.

8+105V

RCS.

GND.

6.3V.

2 8-30 MC.

X2

22.0 MC.

0

L19 HIR;

C25 100K=

Cie

1000

500

Re, 470

www.americanradiohistory.com

HANDBOOK 2 -6 M. "Siamese" Converter 549

of the triode -connected 6AK5 operating with

grid injection. The link circuit from the r.f.

amplifier stage is tapped directly to the 144

Mc. mixer coil to obtain optimum coupling.

The Local Oscillator Stage. A 6AK5 tube is

used as the crystal controlled oscillator in the

2 -meter converter. A 38.66 Mc. overtone

crystal oscillates in a grid- screen circuit, with

the plate circuit tuned to the third harmonic

(116 Mc.). The oscillator is capacitively

coupled to the grid circuit of the mixer stage.

The six -meter converter makes use of a

6AK5 overtone oscillator using a 22 Mc.

crystal. The tube is connected as a triode, and

the oscillator is inductively coupled to the

cathode circuit of the mixer stage. This con-

figuration is required to obtain sufficient in-

jection voltage without permitting the 22 Mc.

frequency to appear in the broadly tuned plate

circuit of the mixer. While the pentode mixer

is undoubtedly noisier than the triode, the

overall noise figure of the converter is much

less than the atmospheric noise at 50 Mc. so

this configuration does not tend to degrade

the usable sensitivity of the converter.

Converter

Construction As each converter is extremely

small in size, it is simple to

construct both of them upon a

single chassis. The two units are therefore

mounted on a small copper plate measuring

3" x 7" in area, having a 1/2-inch turned down

lip running along the edges. A drilling tem-

plate for the chassis is shown in figure 25.

If the sheet copper is not available, a phenolic

"printed circuit board" covered with a thin

layer of copper may be used as a substitute.

Figure 23

THE "SIAMESE" CONVERTER

PROVIDES SUPERIOR V.H.F.

PERFORMANCE ON TWO BANDS

This dual converter has a noise figure better

than 3 decibels on 2- and 6- meters. Utilizing

crystal control for maximum frequency sta-

bility and the new Nuvistor triode, superior

performance is achieved at minimugp cost. The

converter is built upon a small copper chassis

mounted to an aluminum panel by means of

two end plates. Panel size is 31/4" x 81/2 ".

Plate voltage is applied to both converters,

and filament voltage is controlled by the

panel mounted toggle switches. Nuvistor tube

(right) is compared to conventional 6AK5 in

foreground.

All chassis holes are drilled, the major com-

ponents mounted in place, and then the auxi-

liary shields are soldered to the chassis. Place-

ment of parts may be seen in figures 26 -29. The

six tube sockets lie along the center line of

the chassis and all wiring is done in a point -

to -point fashion. The 500 -µpfd. ceramic grid -

plate blocking capacitors pass through small

holes drilled in the interstage partitions and

are supported between the top terminal of the

r.f. stage plate coil and one lead of neutralizing

coil L_ Lii. The neutralizing coil, in turn, is

attached to the top (grid) terminal of the r.f.

stage grid coil. Every effort should be made

to make all leads in the r.f. stages as short and

direct as possible.

Mixer stage wiring is straightforward. The

cathode injection coil of the 50 Mc. mixer

may be made of a length of small hook -up

wire run from pin #2 of the 6AK5 socket,

looping twice around oscillator coil L1R, then

back to the 6AK5 socket, to be soldered to the

grounded filament terminal of the socket.

Testing the Wiring should be checked and

Converters the mixer and oscilator tubes

placed in their sockets. Power

is applied to the converter and the oscillator

stage adjusted for operation. A grid -dip oscil-

lator or a nearby receiver will serve as a handy

indicator of oscillation. You can temporarily

unground the 220K grid resistor of each mixer

stage and insert a low range micro- ammeter

in the circuit, tuning the oscillator controls

for maximum mixer grid current. If a v.t.m.

is handy, it may be attached to the grid pin of

the mixer stage and the oscillator controls

adjusted for maximum negative grid voltage.

Voltage should measure between -1 and -2

volts. Next, connect a receiver capable of tuning

the 28 -30 Mc. range to the output of the con-

www.americanradiohistory.com

548 Receivers and Transceivers THE RADIO

Figure 21

EASE OF MOUNTING

IN YOUR

AUTOMOBILE IS

FEATURED IN

THESE UNITS

The transceiver and

power supply have low

profile so that they may

be placed in line beneath

the dashboard of your

automobile. Tuning con-

trols are easily accessible

to driver of car.

of these converters is better than 3.5 decibels,

which compares favorably with units employ-

ing the expensive 417A low noise triode, and

may only be surpassed by use of the costly

416B tube.

For simplicity and ease of operation, the

two miniature converters are built on one

panel- chassis combination approximately 8" x

31/2" in size. The units may be powered from

the communications receiver, or may be run

from a separate supply as desired.

Circuit The circuits of the two con -

Description verters are similar except for

minor details ( figure 24). A

6CW4 is used as a grid driven, neutralized

r.f. stage, link coupled to a 6AK5 mixer stage.

A second 6AK5 serves as a crystal -controlled

local oscillator. The intermediate frequency

range is 28 to 30 Mcs. The choice of a high

i.f. eliminates image problems and permits use

of a simple slug -tuned coupling circuit be-

tween the converter and the companion

receiver.

The R.F. Stage. The r.f. stage of each converter

consists of a single 6CW4 Nuvistor triode.

Inductive neutralization is used (L2 and L11)

incorporating a series blocking capacitor to

remove plate voltage from the circuit. This

simple configuration provides above 20 deci-

bels of usable gain, which is more than suffi-

cient to override mixer noise. A single stage

such as this is noticeably less susceptible to

cross -modulation from strong local signals than

is a double stage (6BQ7A, for example) , or

two cascaded high gain stages.

The Mixer Stage. A pentode- connected 6AK5

serves as a grid biased mixer for the 50 Mc.

converter. Cathode injection from the crystal

controlled local oscillator is used to achieve

proper mixing voltage. Use of a triode mixer

stage is not recommended as the reduction in

conversion noise of the triode over the pentode

is minimal at 50 Mc. and there is tendency of

the triode to regenerate as the frequency of

the injection oscillator is quite close to the in-

termediate frequency. The 144 Mc. mixer stage

takes advantage of the lower mixer noise level

Figure 22

THE RCA "NUVISTOR" VHF TUBE

The miniature RCA Nuvistor triode provides

high gain, low noise performance in the v.h.f.

spectrum at low cost. Intended for TV use,

this small tube shows excellent results in the

2- and 6 -meter converter described in this

section.

www.americanradiohistory.com

HANDBOOK 2 -6 M. "Siamese" Converter 547

a 0 -100 d.c. milliameter across the plate "test"

points. A 20 watt lamp bulb may be attached

to the antenna receptacle as a dummy load.

Power is now applied and the pi- network cir-

cuit is adjusted for maximum glow of the

lamp. A 0 -10 d.c. milliameter placed across

the grid "test" points may be used to adjust

the excitation level to the 2E26. Grid current

should run between 2 and 3 ma., and plate

current is approximately 50 ma.

For 21 Mc. operation, the "grid tuning"

capacitor is resonated to 21 Mc. and the pi-

network retuned to this band. Slight adjust-

ment of L3 and L4 will permit the two bands

to be properly tuned by swinging the reson-

ating capacitors from minimum to maximum

capacitance.

The last step is to switch to v.f.o. operation,

and adjust the slug of coil L1 for proper dial

calibration. The slug should be permanently

fixed in position with a drop of nail polish

to prevent mechanical instability during mobile

operation of the unit.

The "magic eye" tube can be used to indi-

cate amplifier resonance, but an external plate

meter is recommended for v.f.o. operation,

since loading must be readjusted as the trans-

mitter frequency is varied. The "eye" tube

can be used for loading adjustment, but it

takes practice to interpret variations in the

pattern.

The Power Supply An inexpensive power

supply suitable for a.c.

operation is shown in figure 20. Voltage regu-

lation is employed for maximum stability.

Mobile supplies, such as the transistor types

shown in the Power Supply chapter are suit-

able for mobile operation. Low voltage re-

quired for operation of the v.f.o. and receiver

may be obtained from a dropping resistor and

regulator tube.

27 -5 "Siamese"

Converter for

Six and Two Meters

The new R.C.A. Nuvistor series of minia-

ture tubes brings low noise level v.h.f. recep-

tion within the economic capability of the

average radio amateur. Described in this sec-

tion are twin crystal controlled converters for

50 and 144 Mc. that make use of the 6CW4

Nuvistor v.h.f. triode. The inherent noise level

Figure 20

HOME MADE POWER

SUPPLY FOR

TRANSCEIVER FITS

IN MATCHING

CABINET WITH

SPEAKER

Simple transformer-oper -

ated a.c. supply is used

for home station work.

VR -ISO provides regu-

lated voltage for maxi-

mum stability. Dynamic

speaker is included in

enclosure.

www.americanradiohistory.com

546 Receivers and Transceivers THE RADIO

speaker to the audio jack. Light the filaments

and apply plate voltage. Transformers T1, T2,

and T3 can be aligned by loosely coupling a

2050 kc. signal from an external source to the

plate circuit (pin #6) of the 6CG8 mixer

tube. Next, the bandswitch is placed in the 10

meter position and a 28 Mc. signal is applied

to the input circuit of the receiver. Proper

tracking is achieved in the usual manner, with

the oscillator padding capacitor determining

the calibration at the high frequency end of

the dial, and the variable slug of the oscillator

coil (L6) being used to set the edge of the

band at the low frequency end of the dial. An

Figure 19

REAR VIEW OF UNDER -CHASSIS

AREA

Point to point wiring is used, with many small

components soldered directly to the tube

socket pins. Coaxial antenna receptacle, micro-

phone receptacle, and crystal- v.f.o. switch are

mounted on back apron of chassis. Pilot lamp

receptacles are bolted to frame of tuning

capacitor which is dropped below the chassis

deck by means of cut -out in deck.

antenna can now be connected to the antenna

jack of the transceiver and signals should be

heard. Mixer plate coil L, is peaked for maxi-

mum signal response near the center of the

band and adjustment of the transmitter pi-

network circuit can be made for greatest re-

ceiver sensitivity.

Bandswitch S2 is now placed in the 15

meter position and the oscillator padding ca-

pacitor is adjusted to correctly position the

high frequency end of the 15 meter band when

the tuning capacitor is at minimum setting.

The mixer padding capacitor is adjusted for

maximum receiver sensitivity at the same

frequency.

The transmitter portion should now be

aligned. Place the 6AU6 and 6CL6 tubes in

their respective sockets and insert a 7 Mc.

crystal in socket X2. Throw S1 to the transmit

position and adjust Ln for proper crystal oscil-

lation. Next, plate coil L3 of the 6CL6 stage

is adjusted to 28 Mc. with the "grid tuning"

capacitor nearly open. Plate voltage is removed

and the 2E26 is inserted in its socket. Place

www.americanradiohistory.com

HANDBOOK 10 -15 Meter Transceiver 545

of the stator support bars leaving them at-

tached to one bar. Cut the rear plate so that

it is supported only by the other bar. A small

planetary unit is placed between the capacitor

and the dial for ease of tuning. A second plane-

tary unit is used for the transmitter v.f.o.

All sockets, terminal strips, and trimmer

capacitors are mounted in place using 4 -40

hardware with soldering lugs placed .,eneath

the nuts in various convenient positions.

Transceiver The wiring of the unit is quite

Wiring simple if done in the proper

sequence. The under -chassis

area contains many small components but these

need not be crowded, provided proper care is

taken in the layout and installation of parts.

The smaller components (capacitors and resis-

tors) are installed between the socket pins of

the various tubes. Socket ground connections

are made before the wiring is done, filament

wiring is done next, then the socket -mounted

components are placed in position. Number 22

stranded thermoplastic insulated wire (0.07"

diameter, Consolidated #737) is recommended

for all leads except the filament circuit. Num-

ber 18 wire should be used for these leads.

Small diameter, insulated ' phono- type" shield-

ed wire is used for the lead running from

pin #2 of the 12AX7 to the receiver volume

control capacitor. The filament circuit is wired

in a series -parallel arrangement so that either

6- or 12 -volt operation may be chosen at the

power plug.

Before i.f. transformer T1 is mounted in

position, it should be modified so that it tunes

to 2050 kc. Some makes of transformers will

reach that frequency with no modification.

Others will require that some turns be removed

from the primary and secondary windings, or

that the value of internal fixed capacitance

be reduced accordingly.

The three small 15 meter variable padding

capacitors are mounted below the chassis in

close proximity to the bandswitch and may be

seen in the center of the chassis (figure 17) .

Crystal socket X1 is mounted horizontally on

a small metal bracket under the rear of the

chassis so that the type FT -243 crystal may be

inserted and removed from the rear of the

transceiver. The buffer tuning capacitor

(marked "grid tuning" on the front panel) is

mounted on a small aluminum bracket at the

middle of the chassis. Oscillator coil L1 is posi-

- REAR PANEL

REAR EDGE Or CHASSIS

xl

T1

T -R I

SWITCH

ii DIRONT 5UBPANEL L U LIRONT EDGE Or CABINET

FRONT PANEL

Ta

TUNING

CAPACITOR

RDC6 6E5-M I PLATE

!TUNING/

1

Figure 18

LAYOUT OF MAJOR COMPONENTS

ABOVE THE CHASSIS

tioned between the buffer capacitor and the

v.f.o. capacitor, and is mounted in a small

aluminum shield cut down from an i.f. trans-

former can.

A dust plate is bolted to the rear lip of the

chassis and adds extra strength to the assem-

bly by virtue of the two angle brackets bolted

to the plate and chassis. The 2E26 socket is

mounted on this plate, as are the v.f.o., crystal

switch, power plug, and antenna receptacle.

A slot is cut along the top edge of the dust

plate to insure adequate ventilation.

Transceiver Coils Only six coils are required

for the transceiver, four of

them in the transmitter section. Because of the

compact construction and the influence of

nearby objects, it is wise to grid -dip each coil

to the proper frequency after installation.

Oscillator plate coil (L2) is resonated by the

internal capacitance of the associated tubes and

stray circuit capacitance, and should be grid -

dipped with the oscillator and buffer tubes in

their respective sockets.

Testing the The transceiver should be tested

Transceiver a section at a time. Start with

the receiver and audio portion.

Insert the tubes in the sockets and place crystal

X1 in the holder and connect a temporary

www.americanradiohistory.com

544 Receivers and Transceivers THE RADIO

tank circuit is very short. Directly behind the

tuning capacitor is the modulation transformer.

The 2E26 transmitting amplifier tube is

mounted in a horizontal position at the right

of the chassis, as shown in figure 15. The

transmitter pi- network output circuit is panel

mounted, directly in front of the plate cap of

the 2E26. The 6ME -10 tuning "eye" is panel

Figure 17

UNDERCHASSIS VIEW OF

TRANSCEIVER

V.f.o. tuning capacitor and coil (in shield) are

at top edge of chassis. "Grid" tuning capacitor

is recessed behind panel and driven with shaft

extension. The 21 Mc. padding capacitors are

directly behind bandswitch at center of chassis.

1.1. amplifier is along chassis edge in fore-

ground.

mounted between the pi- network components

and the receiver tuning capacitor. Placement

of receiver components is conventional, with

the 6CG8 mixer stage mounted near the tuning

capacitor.

The receiver tuning capacitor is a two -

section unit, converted from a single section

Johnson 167 -3 variable capacitor. Using a

small hacksaw or jeweler's saw, the two stator

rods are cut so that a front group of plates are

supported by one post, and a rear plate is held

by the other post. This is the way you do this

operation: Leave the front two stator plates

and the rear stator plate in position, removing

the three other plates in between. Next, cut

the remaining two front plates away from one

www.americanradiohistory.com

3

www.americanradiohistory.com

542 Receivers and Transceivers

section. During the transmission the "eye"

indicates proper amplifier adjustment.

The three stage audio amplifier serves as a

modulator for the transmitter as well as an

audio system for the receiver. During trans-

mission, both sections of a 12AX7 serve as a

voltage amplifier, driving two 6CM6 pentode

tubes in a parallel class A modulator circuit.

A simple resistive feedback circuit from the

plates of the modulator to the plate of the

driver stage improves speech quality and re-

duces distortion. The audio output transformer

T4 serves as a modulation choke when switch

section S1.1) opens the return circuit of the

loud speaker jack. Switch section SI_c couples

the modulator to the plate circuit of the r.f.

amplifier stage.

In the receiving mode, the audio signal

from the diode second detector circuit is ap-

plied through the volume control to the grid

circuit of the second section of the 12AX7

speech amplifier. The cathode circuit of the

first section of the 12AX7 is opened by switch

section S1.1) during reception.

Transceiver Layout Figures 13, 15, 17 and

and Assembly 19 illustrate the general

plan of the transceiver.

The panel layout of controls is shown in figure

13, and parts placement above the chassis is

illustrated in figures 15 and 19. The transceiver

is built upon an aluminum chassis 67/8" x

55/8" x 1" in size. This assembly fits within a

steel wrap- around type cabinet .33,4" high,

63/8" deep and 7" wide. This cabinet was

custom -made to allow absolute minimum size

of the transceiver. A manufactured cabinet can

be used at a sacrifice in compactness. The Cali-

fornia Chassis Co. type LTC -464 cabinet and

chassis, with an over -all measurement of 41/2"

x 91/8" x 71/8" is suitable and less expensive

than a custom package.

The transceiver makes use of a dual front

panel. Both panels are made of 1/16 inch

clear plastic sheet. The sub -panel is bolted

directly to the chassis and is painted black to

provide a good background for the tuning

dial. The front panel is a similar piece of

plastic, spaced about 1/4 inch in front of the

sub -panel by means of four bolts and metal

spacers. This panel is painted and lettered as

shown. For decorative purposes, a thin strip

of aluminum is run across the bottom of the

panel to provide a pleasing color contrast to

the eye.

Layout of principal parts above the chassis

can be observed by comparing the photographs

with figure 18. Viewed from the top front,

the receiver occupies the left portion of the

chassis and the transmitter occupies the right

half. The external plugs and receptacles are

mounted on the rear apron of the chassis.

The receiver tuning capacitor is centered on

the chassis, with the 6DC6 r.f. amplifier tube

mounted horizontally above it on a bracket.

The socket is oriented so that the grid con-

nection between the tube and the amplifier

Figure 16

SCHEMATIC OF TRANSCEIVER

Receiver tuning capacitor -Oscillator section

3 -18 ppId. Detector section 3 -8 µµId. (See

text for details.)

Pi- network loading capacitor -400 µµId. Allied

Radio Co., Chicago, 111. #61 -H -009.

L, -1 1H. 1/2" diam. form, 1" long, tuned

with adjustable 1/4-20 iron core slug. (7.0-

7.42 Mc.) Wind with # 18 e.

L2 -25 µh. Tunes to 7 Mc. with circuit capa-

citance. 5/16" diam., 1/2" long with ad-

justable 1/4 -20 iron core slug. Wind with

22 e.

L3-0.9 pH. Tunes to 21 Mc. with tuning

capacitor at maximum, and 29.7 Mc. with

capacitor at minimum. 5/16" diam., 1/="

long with adjustable 1/4 -20 iron core slug.

Wind with »22 e.

L4-0.9 µh. Tunes to 21 Mc. with tuning

capacitor at maximum, and 29.7 Mc. with

capacitor at minimum. B&W coil, 3/4" diam.,

8 turns per inch #18 wire.

L5-1.5 µH. Tunes to 21 Mc. with tuning

capacitor at maximum and auxiliary pad-

ding capacitor in circuit, and 29.7 Mc. with

tuning capacitor at minimum and auxiliary

capacitor out of circuit.

L6 -Two windings. Tuned winding: 0.4 pH,

wound on 5/16" diam. form., 1/2" long with

adjustable 1/4 -20 iron core slug. Wind with

.-.18 e. Secondary winding: 0.4 pH scramble

wound, spaced 1/8 -inch from tuned winding,

#22 d.c.c. Tunes 25.95 -27.65 Mc. for 10

meters, 22.05 -22.5 Mc. for 15 meters.

Note: Coils may be wound on J. W. Miller Co.

#41 -A000 -CBI ceramic forms, with type R

slug. Alternatively, J. W. Miller Co. #20A

and #21A series adjustable r.f. coils may

be substituted. All coils should be adjusted

to frequency with o grid -dip oscillator.

T1 -2050 kc. i.?. transformer. J. W. Miller Co.

#13-WI. Remove turns from 1500 kc. wind-

ings to resonate at 2050 kc.

T2T3 -265 kc. i.f. transformer. J .W. Miller

Co. #12-H1.

74- Primary, 5000 ohms. Secondary 4 ohms.

10 -watt.

Dial -Made up of Jackson Bros. planetary

drive. (Arrow Electronics Co., 6S Cortland

St., New York 7, N.Y.)

Tuning Eye: 6ME -10 (midget) or EM -84. (See

tube manual for pin connections.)

www.americanradiohistory.com

HANDBOOK 10 -15 Meter Transceiver 541

semi - remote cutoff pentode is used as an r.f.

amplifier with the input grid connected direct-

ly to the r.f. circuit of the transmitter power

amplifier. Thus, when the transmitter is prop-

erly adjusted and loaded to the antenna system,

the receiver input circuit is automatically tuned

to the same frequency. This eliminates the

components and space normally required for a

tuned r.f. input circuit. The coupling capa-

citor and grid resistor of the 6DC6 stage are

chosen so that the tube blocks itself off during

transmission periods. The relatively large po-

tential developed on the grid of the r.f. stage

does no harm. A 27 -ohm composition resistor

is placed in the plate lead of the r.f. amplifier

to suppress a parasitic oscillation that often

shows up in such circuitry. The resistor has no

effect upon the operation of the amplifier

stage.

Figure 15

OBLIQUE VIEW OF TRANSCEIVER

CHASSIS

The 2(26 power amplifier tube is mounted in

a horizontal position, supported by bracket at

rear of the chassis. Chassis is perforated be-

low tube to permit passage of air around tube.

Pi- network components are in front of tube

cap. Antenna coaxial receptacle is at rear of

chassis on bracket. Construction of multiple

front panel may be seen in this view.

A triode -pentode (6CG8) is used as the

first mixer stage. The triode section operates

as a "hot plate" oscillator 2050 kc. below the

signal frequency. Grid injection is used to the

pentode mixer section. Parallel padding capa-

citors are switched across the mixer and oscil-

lator circuits in order to tune the 15 meter

band. A 6BE6 pentagrid tube is employed as

a second mixer from 2050 kc. to 265 kc. The

#1 grid acts as the anode of a cathode feed-

back crystal oscillator, with the degree of feed-

back controlled by a capacity bridge placed be-

tween #1 grid, cathode and ground.

A single 6BÁ6 provides sufficient i.f. gain

at 265 k.c., and two transformers produce ex-

cellent "skirt" selectivity and adjacent signal

separation. The r.f. stage, the second mixer,

and the i.f. stage are all controlled by the

a.v.c. circuit, operating from one -half of a

6AL5 tube. The second diode section of the

6AL5 acts as the second detector and auto-

matic noise limiter. The a.n.l. circuit has a

very low distortion level, and is in the circuit

at all times. A 6ME -10 miniature "magic eye"

tube serves as a signal strength indicator,

operating from the a.v.c. line of the receiver

www.americanradiohistory.com

540 Receivers and Transceivers THE RADIO

MIC

V --O

VIAL 05C. MULTIPLIER A-P AMP.

()MC) (21ORjCMC)

VOL.

B.6300 V. ei 1'O V.

Figure 14

BLOCK DIAGRAM OF TRANSCEIVER

External power supply is used, so transceiver may be operated from a.c. supply or from mobile

power pack. Tuning adjustments are accomplished by means of 6E5 miniature "magic eye." The

"eye" tube shown is the imported type 6ME -10 (Concord Electronics Co., 809 No. Cahuenga Blvd.,

Los Angeles 38, Calif.). The FM -type EM -84 may be substituted.

has arisen for a compact transceiver that will

work well either in the car or at the home

station. The unit described in this section has

been designed to meet this need.

This compact transceiver package covers the

10 and 15 meter bands, and employs a stable

superheterodyne receiver and a 20 watt a.m.

transmitter. The transmitter may be either

crystal controlled, or driven by the internal

v.f.o. A 10 watt audio system operates from

a crystal microphone and provides 100% modu-

lation of the transmitter. During reception the

audio stages deliver sufficient power to drive

an external speaker well above the noise level

of the automobile.

Small enough to fit comfortably under the

dash of today's car, the transceiver delivers a

well modulated signal at a maximum plate

power load of 300 volts at 170 milliamperes

and 150 volts at 20 milliamperes.

Transceiver A block diagram of the trans -

Circuit ceiver circuit is shown in

figure 14. Twelve tubes are

employed, three in the transmitter section,

three in the audio portion, and six in the re-

ceiver section. Change -over from receive to

transmit is accomplished by a four -pole, two -

position switch (S1), mounted in the upper

left corner of the front panel (figure 13).

One section of this switch (Six) removes the

plate and screen voltage from the transmitter

amplifier stage, another section (S).p) trans-

fers the low voltage from the transmitter ex-

citer stages to the receiver circuits, a third

switch section (Sl.D) activates either the

speech amplifier or the loud speaker jack, and

the fourth section (S)_B) sensitizes the "magic

eye" indicator tube for reception.

The receiver portion employs a double con-

version circuit to achieve maximum image

rejection with adequate selectivity. A 6DC6

www.americanradiohistory.com

HANDBOOK 10 -15 Meter Transceiver 539

fairly broad as is the r.f. stage tuning although

the antenna trimmer will have to be repeaked

when going from one end of the 80 meter

band to the other. However, if the mixer

trimmer capacity has to be changed at the low

end of a band to obtain an increase in S -meter

reading, it means that the coil will have to be

altered. If the trimmer has to be increased in

capacity, it indicates that more inductance is

needed and the tap will have to be moved

farther up on the coil. Conversely, if the

trimmer has to be decreased in capacity, less

inductance is needed and the tap is moved

down on the coil. The Erie trimmer capacitors

pass from maximum to minimum capacity in

180 degrees of rotation and are at minimum

capacity when the lettering on the cap is

adjacent to the mounting bolts.

The final dial calibration must be made

with a bottom shield on the chassis -a tem-

porary piece of screen wire is satisfactory - in

order that the calibration will be correct when

the receiver is placed in the cabinet. 100 kc.

points are marked off on the dial from har-

monics of the crystal calibrator and in between

points may be marked off with dividers since

the dial is fairly linear. The dial scale is re-

moved for inking the calibration points and

when permanently reinstalled, it is covered

with a 1/16 inch piece of clear polystyrene

sheet the same size as the scale. This will keep

the dial clean and prevent warping of the

paper scale. A permanent pointer is made from

a long scrap of polystyrene sheet with an inked

line scribed down the center of the pointer.

It can be shaped to fit over the planetary drive

by holding the plastic under hot water until it

is soft enough to be bent to shape.

The front panel is fastened to the chassis

by means of the hexagonal nuts holding the

bandswitch and toggle switches. Another 1/16

inch sheet of polystyrene or plexiglass is cut

to fit over the dial opening with the cutout

for the shaft of the planetary drive allowing

clearance for the tuning knob. Lettering decals

are used to mark the controls.

Any well filtered power supply that delivers

about 250 volts at 80 ma. is suitable for the

receiver. For 6 volt operation terminal 4 on

the power plug is jumpered to terminal 2

(ground) and power is applied to terminal 3.

For 12 volt operation terminal 3 is left open

and 12 volts applied to terminal 4.

27 -4 A Compact

Transceiver for

10 and 15 Meters

Regardless of "conditions" and the sunspot

cycle, the 10 and 15 meter bands are exceed-

ingly popular with a large group of amateurs.

Many stations on these bands employ low

power, and the amateur using a low power

transmitter, inexpensive receiver and modest

antenna suffers no great handicap.

In addition, the growth of mobile operation

on these bands has been rapid and the need

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Figure 13

POCKET -SIZE

TRANSCEIVER!

This miniature trans-

ceiver is designed for

top -performance on the

21 and 28 Mc. amateur

bands. Receiver section

utilizes double conversion

for suppression of images

and for maximum selec-

tivity. Modulated ampli-

fier stage of transmitter

employs a 2E26. Power

supply and speaker are

contained in auxiliary

cabinet shown sitting

atop transceiver.

www.americanradiohistory.com

538 Receivers and Transceivers THE RADIO

Coil Assembly The next step in the assem-

bly of the receiver is to

mount the coil partitions with the attached

switch sections. The shaft is connected to the

switch index (previously mounted on the front

of the chassis) with a 1/4 -inch shaft coupling

and the switch index is rotated so that the

switch contacts are in proper alignment. The

nut holding the index can then be tightened

down to hold it in place.

The r.f. coils are wound as shown in the

coil table (figure 11) and the two grid coils

(L1 and L3) can be wired in without further

attention since they tune separately with the

antenna trimmer. The third set of contacts on

the grid coil switch wafer (SW1.c) are used

to switch in the additional 680 ohm cathode

resistor used to limit the excessive gain de-

veloped on the 80 meter band. This prevents

overloading the triode mixer with a strong

signal. The mixer and oscillator coils are next

wound and installed and the r.f. section is

ready to be tuned up.

Receiver Before proceeding with the

Calibration alignment of the oscillator and

mixer coils, a dial scale is made

up from stiff paper or white cardboard and

fastened to the dial back plate. The semi-

circular scales are made with a compass using

black india ink. A temporary pointer is made

of light aluminum or plastic scrap and is

attached to the planetary drive dial plate by

means of the small screws holding the dial

plate. The section of the pointer extending

over the scale is cut in half along the center

line so that only half of the pointer remains

to be used as a guide line for marking off the

calibration points on the dial. A rough align-

ment of the oscillator and mixer coils can be

made with a grid dip meter using the tuning

range data given in the coil table. The fre-

quency of the oscillator circuit is higher than

the frequency of the mixer circuit by 3000 kc.

(the frequency of the i.f.) except in the case

of the 15 meter band where the oscillator

frequency is lower than the mixer frequency

by 3000 kc. As a starting point, the tuning

capacitor is set at almost full mesh (near maxi-

mum capacity) and each oscillator coil is set

at its lowest frequency by adjusting its asso-

ciated trimmer to maximum capacity. The

mixer coils can be set in the same fashion on

the low frequency edge of each amateur band.

The low band edges can be found by having

the receiver turned on for operation and apply-

ing a signal to the antenna input from a signal

generator or your transmitter v.f.o. The an-

tenna trimmer will have to be peaked for each

band.

Once the low band edges have been found,

the rest of the alignment and dial calibration

is easily done using the built in 100 kc. cali-

brator. A short piece of wire is clipped to the

plate of the crystal calibrator tube and brought

near the antenna coils to get a fairly strong

signal to work with. A crystal harmonic should

fall at the low end of each band at the point

on the dial found previously with the external

signal. With this as a starting point, the tuning

capacitor can be rotated over the dial range

and 100 kc. points counted off to check the

coverage of each band on the dial. The coil

and capacitor combination given in the coil

table is designed to spread each band over

almost the entire 180 degrees of the dial using

a 15 µµfd. tuning capacitor. If an entire band

cannot be covered, the turns on the particular

oscillator coil must be squeezed together

slightly to increase the inductance and the

alignment procedure repeated by resetting the

oscillator trimmer so that the dial pointer

will align with the original calibration point

at the low end of the band. If the bandspread

is insufficient, the coil turns must be spread

to decrease the inductance. In the case of the

80 meter band it may mean removing one or

two turns from the oscillator coil.

Tracking When the band coverage has

Adjustments been set by the above adjust-

ment of the oscillator coils and

associated trimmers, the mixer coils are ad-

justed for proper tracking. Using the same

harmonics from the 100 kc. calibrator set the

tuning dial at the high frequency end of the

band, peak the antenna trimmer for a maxi-

mum reading on the S -meter and likewise peak

the trimmer capacitor on the mixer coil. Rotate

the tuning capacitor to the low frequency end

of the band and again adjust the same mixer

trimmer for a maximum reading on the S-

meter. If no improvement can be made in the

meter reading, the mixer is tracking with the

oscillator and no further adjustment need be

made. The 80 and 40 meter mixer tuning is

www.americanradiohistory.com

HANDBOOK Bandpass- Filter Receiver 537

0

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Figure 12

PLACEMENT OF

MAJOR

COMPONENTS

BENEATH THE

CHASSIS

The 6BZ6 r.f. tube socket

is located so that the

rear wall of the coil com-

partment passes over the

center of the socket, iso-

lating the input and out-

put circuits. The 6U8

mixer socket is mounted

in the some fashion with

the mixer pins (I, 8, and

9) falling within the coil

compartment and the

oscillator pins (2, 6, and

7) placed in the oscillator

coil area of the chassis.

Bandchange switch seg-

ments are mounted to

the walls of the coil com-

partment and are driven

by the switch index af-

fixed to the front wall of

the chassis. The antenna

trimming capacitor and

b.f.o. pitch capacitor are

mounted to extension

"ears" on the rear wall

of the coil compartment.

of the ceramic trimmer capacitors and the coil

leads are connected to the same lugs, with a

heavy wire going to the respective terminals

on the band switch. There is a continuously

shorting deck on the oscillator switch section

that picks up the unused coils and shorts them

all together to prevent any absorption loss

from occurring in the higher frequency coils.

The two 3000 kc. i.f. transformers are made

from standard 1500 kc. units by removing 5

feet of wire from each winding of the trans-

former. The transformers shown are J. W.

Miller # 13 -W 1, but other makes of transform-

ers could also be altered to tune to 3000 kc.

The proper frequency is easy to check with a

grid dip meter. The b.f.o. coil is made the easy

way by using a broadcast -type "vari- loopstick."

Twelve inches of wire is removed from the

coil and an additional 18 inches is unwound

to make the cathode tap and is then rewound

back on the coil. The end of the rewound

section of the coil will be the ground end. A

padding capacitor of 68 µµfd. is placed across

this coil and the modified " loopstick" is

mounted in a shield can to match the i.f.

transformers. The slug of the coil tunes the

circuit to the exact frequency and the variable

pitch control capacitor between cathode and

ground provides about 3 kc. variation each

side of zero beat.

Preliminary Checking Most of the wiring

and Adjustment can be done before

the coil partitions and

the r.f. coils are installed. At this point the

tubes can be put in the sockets (figure 12)

and power applied for a preliminary check on

the i.f. and audio circuits. Tuning the i.f.

channel is a simple matter because the center

frequency is determined by the bandpass filter.

A low level 3000 kc. signal from a signal

generator or grid dip meter is applied through

a capacitor to the input of the filter and the

slugs of the i.f. transformers are adjusted for

maximum signal reading on the S- meter. The

S -meter is adjusted with the "zero" control to

read zero with no signal and the meter sensi-

tivity is determined by the value of the meter

ground resistor. The value of 47K used with

the 0 -1 ma. meter will give full scale deflec-

tion with a very strong local signal.

www.americanradiohistory.com

536 Receivers and Transceivers THE RADIO

near that part of the circuit in which they are

used to allow short leads and to facilitate

wiring (figure 10) . Terminal boards can be

made up from one inch wide strips of fiber-

glass sheet or phenolic material using solder-

ing lugs and rivets for the terminals. A plain

piece of the same material is placed between

the terminal board and the side of the chassis

and the two pieces are fastened to the chassis

with 4 -40 hardware. The trimmer capacitors

for the mixer coils are mounted on the coil

partition so that their terminals extend into

the mixer coil section adjacent to the proper

coil. 4 -40 tapped holes in the lip of the par-

tition allow the trimmers to be fastened direct-

ly to the partition. These trimmers can be

wired to the band switch before the partition

is mounted to the chassis, and the leads of

the coils may be soldered to the trimmer capa-

citor terminals when they are installed. Notice

that one terminal of the trimmer capacitors

returns to a ground strap allowing them to be

tuned with a metal screw driver without short-

ing the B -plus voltage appearing on the mixer

coils. The bottom leads of the mixer coils

return to a common bus wire run near the

chassis and terminating on an insulated tie -

point near the 80 meter coil.

Shielded wire (phonograph pickup type) is

used in the audio circuit to bring the leads

from the 6T8 socket to the volume control,

but all r.f. wiring is unshielded, short and

direct. B -plus leads, S -meter wiring and the

a.v.c. and r.f. gain control wires pass around

the inside edge of the chassis where they are

out of the way. The disc -type bypass capacitors

are mounted right at the tube sockets with

short leads. The metal center posts of the r.f.

and i.f. tube sockets are grounded and the

cathode bypass capacitors are positioned to

cross the center of the sockets to further isolate

the input and output circuits of the tubes.

The silver mica padding capacitors in the

oscillator circuit mount directly on the lugs

Figure 11

COIL DATA

All r.f. coils are wound on 138" lengths of !'2" diameter polystyrene rod in a space of 2/4" except

coil Li which is close wound in 9 16" space. The antenna coils are wound at the ground end of

the grid coils and spaced 1 16" below the grid coil. All wire is plain enamel in the sizes shown.

Holes are drilled through the forms to fasten the end of the coils and the forms are tapped for

4 40 bolts at the bottom ends to attach them to the chassis. The tcp point is the number of

turns from the bottom (B -plus) end of the coil.

TRIMMER FIXED PAD

BAND COIL TUNING RANGE TURNS TAP WIRE µµtd. ppfd.

80 L1 Grid 3500 -7300 45 #30 C-1

L., Ant. 17 30

L- Mixer 3500 -4000 80 30 8 -50

1_19 Osc. 6500 -7000 37 26 4 -30 None

40 L, Mixer 7000 -7300 35 6 26 8 -50

L11 Osc. 10 -10.3 Mc. 16 20 4 -30 85

20 L,; Grid 14 -30 Mc. 13 20 C-1

L Ant. 10 30

L Mixer 14 -14.4 Mc. 20 6 20 8-50

L12 Osc. 17 -17.4 Mc. 7 20 4-30 120

15 Lti Mixer 21 -21.5 Mc. 14 6 20 8 -50

L1s Osc. 18 -18.5 Mc. 7 1/2 20 4 -30 110

10 L, Mixer 28 -30 Mc. 9 7 20 8 -50

L1d Osc. 31 -33 Mc. 51/2 20 4 -30 30

L1 B.f.o. coil - BC "vari -loop tick" (see text)

www.americanradiohistory.com

HANDBOOK Bandpass -Filter Receiver 535

SW6 0

o TERMINAL BOARD

POWER

PLUG

REAR APRON

vTCRMI NAL.

BOARD

LEFT SIDE

I I

COIL

COMPARTMENT

I I

L J

TERMINAL

BOARD

RIGHT SIDE

Figure 10

UNDER -CHASSIS

LAYOUT AND

PLACEMENT OF

TERMINAL BOARDS

Terminal boards are

mounted on the left,

right, and rear side walls

of the chassis. A piece of

phenolic material placed

beneath the board insu-

lates the terminals from

the chassis. R.f. bypass

capacitors are mounted

directly to the pins of

the tube sockets.

the dial which is 53/4" long and /8" high. It

has 3/8" lips and takes the form of a shallow

pan. Two holes are drilled in the bottom lip

of the pan to fasten it to the chassis flush with

the front edge. The center of the pan is in

line with the center of the chassis and the

capacitor shaft. Tapped 6 -32 holes in the

chassis make it easy to mount the dial pan.

The center hole for the planetary drive is

found by sliding the back plate of the dial up

against the capacitor shaft and marking the

location of the hole. The drive mechanism is

then positioned on the back plate and bolted

to it. After these parts are permanently

mounted, a couple of braces are affixed from

the back plate of the dial to the chassis so

that the whole dial assembly and tuning

capacitor are held rigid (figure 8).

The Coil Assembly Light sheet aluminum

is used to make the

under -chassis partitions that separate the coils

and act as mounting plates for the bandswitch

sections. The rear partition holding the an-

tenna switch section, the antenna trimmer and

the b.f.o. capacitor measures 71/2" x 17/8"

with 1/4 -inch lips bent at right angles to the

top and bottom (figure 9) . The lips provide

stiffening and permit easy mounting to the

chassis. The front partition holding the oscil-

lator and mixer switch sections measures 51/2"

x 1 %s ", with the same 1/4 -inch lips top and

bottom. The mixer coil trimmers are mounted

on the lip and do not project beyond the depth

of the chassis. Two side pieces attach to the

partitions to hold them rigid. Threaded holes

are cut in the bottom lips of the partitions so

that they can be fastened to the chassis with

4 -40 self- tapping screws. Cutouts are made in

each partition to clear the r.f. and oscillator

tube sockets and two 1/4-inch feed -through

holes are drilled in one side partition for the

B -plus and plate leads to the r.f. stage.

The bandswitch wafers, SW1, SW2, and

SW3 are mounted with bolts and spacers on

the coil partitions on a center line directly

below the tuning capacitor. The antenna

trimmer capacitor and the b.f.o. pitch capa-

citor are mounted on the rear partition three

inches to each side of the band switch center

line. Fiber extension shafts bring the controls

up to the front panel. A flat sided fiber shaft

is passed through the band switch sections and

is connected to the switch index with a metal

shaft coupling. The switch index is mounted

on the front apron of the chassis. The loca-

tion of the holes to be drilled in the panel is

found by placing the drilled chassis against

the panel and marking the holes from the in-

side of the chassis. The large cutout for the

dial and the S -meter can be made with a fine

toothed coping saw.

Wiring the Three small phenolic terminal

Receiver boards are used to mount most

of the miscellaneous resis-

tors and the audio coupling capacitors. The

boards are bolted to the side of the chassis

www.americanradiohistory.com

!

534 Receivers and Transceivers THE RADIO

chassis by 6 -32 bolts in the space between

each capacitor. The ground terminals of the

coils, trimmers, and padding capacitors attach

to ground lugs affixed to the chassis at the

ground bolt of the trimmers. The tuning capa-

citor is mounted on a small L- shaped bracket

and is inverted to place the terminals and

ground strap closer to the chassis. The ground

strap on the capacitor projects through a hole

in the chassis and makes connection to the

ground terminal underneath the chassis. The

tuning capacitor is positioned along the center

line of the chassis at a distance from the front

panel determined by the length of the dial

drive mechanism.

The Dial For smooth tuning the planetary

drive must be lined up carefully

with the shaft of the tuning capacitor. The

drive mechanism mounts on the back plate of

Figure 9

UNDER -CHASSIS VIEW OF RECEIVER

The general layout of components beneath the chassis is shown in this view. The bandswitch passes

through the central coil comportment. Each switch segment is mounted to a wall of the compart-

ment. The two antenna coil forms are mounted behind the coil compartment with the primary

windings connected to the antenna receptacle by a short length of coaxial line. To the right is the

r.f. stage tuning capacitor, and to the left is the b.f.o. pitch capacitor.

The bank of mixer coils is centered in the compartment and the various trimming capacitors are

mounted on the top front flange of the compartment to facilitate adjustment. Note that the

rear wall of the compartment passes across the socket of the r.f. stage, thus isolating the input

and output circuits of the tube.

The bank of oscillator coils is mounted between the compertment and the front wall of the chassis,

with the oscillator switch mounted to the outside wall of the compartment. All wiring is short and

direct, and most small bypass capacitors are mounted directly to the tube socket pins. Resistors

are mounted on terminal boards placed on the walls of the chassis.

www.americanradiohistory.com

Bandpass -Filter Receiver 533

up for in the following i.f. stage using stan-

dard transformers. The bandpass filter re-

quires no special coupling circuits and is

symmetrical as viewed from its terminals -

in other words, either terminal may serve as

input or output at a nominal impedance of

4700 ohms. Mixer plate voltage may be ap-

plied directly to the filter since it is tested

at a voltage far in excess of any value normally

encountered in receivers. The gain control is

in the cathode of the i.f. stage as well as the

r.f. stage for more effective control of the

over -all gain of the receiver. The second 6BJ6

i.f. tube has its screen voltage regulated for

more effective operation of the bridge -type

S -meter in its plate circuit.

The Detector and The detector and audio

Audio Stages stages are conventional

and the noise limiter is

a series diode (part of the 6T8 tube) with a

fixed threshold level that does a good job of

limiting noise without causing apparent dis-

tortion on phone signals. A switch is included

to disable the limiter when it is not needed.

The audio volume control is isolated from d.c.

potentials by coupling capacitors to eliminate

the tendency of these controls to become noisy

when used in current carrying circuits. A

standby switch opens the cathode of the 6AQ5

stage silencing the receiver, but other methods

can be used such as cutting the B -minus of

the separate power supply or relay switching

the B -plus of a mobile power supply.

Receiver

Construction The 81/2" x 11" x 2" chassis

size is just right for building

this receiver and there is

plenty of room for all the components without

crowding, even if the exact parts specified are

not used. Location of the major components

is shown in the photographs and the layout

follows the circuit diagram (figure 7) , pass-

ing around the chassis with the r.f. section

taking up most of the center area. It is a good

idea to make paper templates for the band -

pass filter and the i.f. transformers, marking

the drilling holes on the chassis from the tem-

plates. The various tube sockets should be

oriented so that grid and plate leads do not

have to cross over the sockets. Oscillator

trimmer capacitors are mounted on top of the

chassis in a line above the oscillator coils with

the capacitor leads projecting through 1/4 -inch

holes to the under side of the chassis. The

oscillator coils can then be mounted under the

Figure 8

REAR VIEW OF

BAN DPASS- FILTER

RECEIVER

The audio stages, 100

kc. calibrator crystal, and

b.f.o. are located along

the left edge of the

chassis. Intermediate am-

plifier stages pass across

the back of the chassis,

with the 3 Mc. bandpass

crystal filter and voltage

regulator tube at right.

Oscillator alignment ca-

pacitors are placed di-

rectly behind the home-

made dial, with the main

tuning capacitor centered

on the chassis. The r.f.

and mixer tubes are at

the center of the chassis.

Along the rear apron of

the chassis are (I. to r.):

audio jacks and a.n.l.

switch, antenna recep-

tacle, and power plug.

The 5 -meter "zero"-set

potentiometer is placed

atop the chassis, between

the 100 kc. crystal and

the first i.f. tube. 411

www.americanradiohistory.com

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www.americanradiohistory.com

HANDBOOK Bandpass -Filter Receiver 531

plug and antenna connector. The most -used

controls are on the front panel. The tuning

dial is a very smooth planetary type having a

long plastic pointer which extends over a cali-

brated scale. Indirect lighting of the dial is

provided by two panel bulbs recessed at the

edge of the dial scale. A direct reading S -meter

is connected in the plate circuit of the last

i.f. amplifier tube, and a 100 kc. calibrator

is included.

Receiver Circuit The tube lineup consists of

a high gain 6BZ6 semi -

remote cutoff r.f. amplifier; a 6U8 triode

mixer and oscillator; two 6BJ6 i.f. amplifiers;

a 6T8 detector, noise limiter and audio ampli-

fier; and a 6AQ5 audio output stage. A second

6U8 is used for a combined beat oscillator and

crystal calibrator, and an 0A2 serves as a

voltage regulator.

The circuit is conventional with several

simplifications that make for less work and

easier construction, to say nothing of fewer

parts (figure 7) . The main d.c. supply volt-

age to the plates and screens of the tubes is

divided by resistors to eliminate coupling

through a common voltage source so extra

decoupling networks are not required. A volt-

age regulator tube stabilizes the oscillator and

screen voltages as well as the voltage on the

b.f.o. Bandswitching for the five amateur bands

is accomplished with only three switch wafers

(SWIA,B,c, SW2A_B, and SW3) and the

main tuning dial is used only for the oscillator

circuit (C2B) and plate coil of the r.f. stage

(C.,A) . This calls for only a small inexpensive

two gang capacitor. Only two coils (L1 and

L3) are used in the r.f. grid circuit to cover

five bands. Coil L1 covers the 80 and 40

meter bands and coil L3 covers the 20, 15

and 10 meter bands, trimmed by the 100 µµEd.

antenna capacitor, C1. The r.f. tuning is fairly

broad and the trimmer only requires resetting

when going from one end of a band to the

other.

The plate circuit of the r.f. amplifier (C2A,

L5.0) is resonated using separate coils for

each band which are preadjusted to track with

the oscillator tuning. Except for the 80 meter

band, the plate coils are tapped for proper

oscillator tracking, eliminating the need for

extra series or padding capacitors required

with other tracking methods. The oscillator

circuit (C2B,L10 -L14) uses single winding

coils in a Colpitts arrangement utilizing large

padding capacitors which serve the dual pur-

pose of stabilizing the oscillator as well as

providing proper bandspread for the tuning

ranges. The oscillator tuning capacitor C2B

is as small in capacity as can be used to cover

the desired tuning range. The oscillator cir-

cuit is designed as if it were going to be used

for a v.f.o. in a transmitter - which calls for

mechanical rigidity and use of short leads, a

ceramic switch and tube socket, silver mica

capacitors, and solidly mounted coils. The coils

are wound on polystyrene rods and are bolted

to the chassis. Directly above each coil (on

top of the chassis) are the ceramic trimmer

capacitors (C4 -Cg) used to adjust the tuning

range (figure 8) . The capacitor lugs project

through holes drilled in the chassis and fall

adjacent to their respective coils and switch

contacts. The trimmer capacitors have a nega-

tive temperature coefficient, and the combina-

tion of fixed silver mica padding capacitors

and the negative compensating characteristics

of the adjustable trimmers tends to stabilize

the oscillator frequency with respect to tem-

perature changes. Ceramic capacitors, together

with the small, rigid plates of the tuning

capacitor make the oscillator almost immune

to vibration.

The Mixer Stage A triode mixer is not

commonly used in band -

switching receivers but its low internal noise

and low plate resistance make it ideal for

working into the low impedance of the crystal

bandpass filter. The injection voltage from

the oscillator is fed directly into the cathode

of the triode section of the 6U8. Although the

injection voltage varies from one band to the

next, the value is not critical and is sufficient

on all bands. A v.t.v.m. reading at the grid

of the oscillator (pin 2) will show between

-5 and -8 volts for proper operation. A

5600 ohm resistor is placed across the 80

meter oscillator coil (L10) to reduce the

injection voltage on this band to the proper

level.

The I.F. Amplifier The crystal bandpass

filter (Blackhawk En-

gineering Co.) has a maximum insertion loss

of only 3 db. This loss is more than made

www.americanradiohistory.com

530 Receivers and Transceivers THE RADIO

in the collector circuit. Optimum collector load

for the 2N217 is approximately 500 ohms,

and the 2N217 develops a maximum audio

signal of 75 milliwatts at this load impedance.

Transformer T, matches the transistor circuit

to the 12 ohm miniature loudspeaker. The

receiver draws a maximum signal current of

11 milliamperes from the 9 -volt battery sup-

ply. If no external antenna is used, the re-

ceiver should be moved about to orient the

"loopstick" coil L, for best pickup of each

individual broadcast station. Adjacent channel

interference can often be eliminated by care-

ful rotation of the set to "null out" the offend-

ing signa:. Ample loudspeaker volume will be

obtained from local stations without the use

of an external antenna.

27 -3 An Inexpensive

Bandpass- Filter

Receiver

A very selective high performance amateur

band receiver can be built by using a high

frequency crystal bandpass- filter in the i.f.

system. Selectivity and image rejection are

accomplished without the complex circuitry

and elaborate construction required in a dual

conversion receiver to obtain the same results.

The intermediate frequency used in this re-

ceiver is 300 kc., yet the bandwidth is only

3 kc. at 6 db down and 12 kc. at 60 db down

with sharp skirt selectivity. The receiver covers

all amateur bands between 10 and 80 meters.

To supplement this efficient i.f. system, the

entire receiver has been designed towards the

goal of simplicity both in circuitry and me-

chanical construction without sacrificing any-

thing in performance or leaving out any con-

trols needed in a communication receiver. The

receiver is built on a standard size chassis with

a compact perforated cabinet suitable for use

in the shack, or in the car as a first rate mobile

receiver (figure 6) . The power supplies are

built as a separate unit for this purpose and

the tube filaments are wired in a series -

parallel configuration so that either a 6 or 12

volt d.c. supply may be used. The speaker is

external and the seldom -used noise limiter

switch and headphone jack are on the rear

apron of the chassis, along with the power

Figure 6

HIGH PERFORMANCE AMATEUR BAND RECEIVER

MAKES USE OF 3 MC. CRYSTAL I.F. FILTER

This simple, easy to build receiver achieves a near ultimate in selectivity and sensitivity

without the complex circuitry of a dual conversion receiver.

The bandspread dial is made up from an inexpensive vernier unit, to which a celluloid

pointer has been attached. The dial opening is covered with a thin sheet of lucite, held

in position with 4 -40 sheet metal screws. Directly below the tuning dial is the band -

switch with Vie antenna trimmer, the a.v.c. and b.f.o. switch, and the i.f. gain control

to the left. Above these is the S- meter, mounted to the rear of the panel. To the right

are the b.f.o. pitch control, the standby and calibration switches, and the audio

gain control.

The receiver sits on four rubber feet to prevent the operating table from being marred

by the metal case. The front panel is attached to the receiver chassis, and the ventilated

cabinet is bolted to the rear of the chassis. Receiver size is only 11" x 6 ".

www.americanradiohistory.com

HANDBOOK 529

27 -1 Circuitry and

Components

It is the practice of the editors of this

Handbook to place as much usable information

in the schematic illustration as possible. In

order to simplify the drawing the component

nomenclature of figure 1 is used in all the fol-

lowing construction chapters.

The electrical value of many small circuit

components such as resistors and capacitors is

often indicated by a series of colored bands

or spots placed on the body of the component.

Several color codes have been used in the past,

and are being used in modified form in the

present to indicate component values. The

most important of these color codes are illus-

trated in figure 2. Other radio components

such as power transformers, i -f transformers,

chokes, etc. have their leads color -coded for

easy identification as tabulated in figure 3.

27 -2 A Simple

Transistorized Portable

B -C Receiver

Illustrated in figures 4 and 5 is an easy

to construct two transistor portable broadcast

receiver that is an excellent circuit for the

beginner to build. The receiver covers the

range of 500 kc. to 1500 kc. and needs no

external antenna when used close to a high

power broadcasting station. An external anten-

na may be added for more distant reception.

The receiver is powered from a single 9 -volt

miniature transistor battery and delivers good

speaker volume, yet draws a minimum of cur-

rent permitting good battery life.

Circuit Operation of the receiver may

Description be understood by referring to

the schematic diagram of figure

5. The tuned circuit L1 -C, resonates at the

frequency of the broadcasting station. A por-

tion of the r -f energy is applied to the base of

the 2N112 p -n -p type transistor. A tapped

winding is placed on coil LI to achieve an im-

pedance match to the low base impedance of

the transistor. Emitter bias is used on this

stage, and the amplified signal is capacity-

coupled from the collector circuit to a 1N34

diode rectifier. The rectifier audio signal is

recovered across the 2K diode load resistor,

which takes the form of the audio volume

control of the receiver. The diode operates in

an untuned circuit, the selectivity of the re-

ceiver being determined by the tuned circuit

in the r -f amplifier stage.

The audio signal taken from the arm of the

volume control (R1) is applied to the base of

the 2N112 r -f amplifier which functions sim-

ultaneously as an audio amplifier stage. The

amplified audio signal is recovered across the

2K collector load resistor of the 2N112, and

is capacitively coupled to the base of a 2N217

p -n -p audio transistor. This stage is base and

emitter biased, having the output transformer

Figure 5

INTERIOR VIEW

OF TRANSISTOR

RECEIVER

The speaker and output

transformer are mounted

at the left of the Ma-

sonite chassis. Top, cen-

ter is the "loopstick" r -f

coil, and directly to the

right is the 10 millihenry

r -f choke in the collector

lead of the 2N112 trans-

istor. Battery Is at lower

right.

www.americanradiohistory.com

528 Receivers and Transceivers THE RADIO

FIGURE 3

COMPONENT COLOR CODING

POWER TRANSFORMERS

PRIMARY LEADS BLACK

IF rAPPEO

COMMON BLACK

TAP

END

HIGH VOLTAGE WINDING

CENTER -TAP

BLACK /YELLOW

BLACK / RED

RED

RED /YELLOW

RECTIFIER FILAMENT WINDING- YELLOW

CENTER -TAP YELLOW /BLUE

FILAMENT WINDING N 1 GREEN

CENTER -TAP GREEN /YELLOW

FILAMENT WINDING N 2

CENTER -TAP

FILAMENT WINDING N 3

CENTER -TAP

BROWN

BROWN /YELLOW

SLATE

SLATE /YELLOW

I-F TRANSFORMERS

PLATE LEAD

0+ LEAD

GRID (OR DIODE ) LEAD

A -V -C (OR GROUND) LEAD

BLUE

RED

GREEN

BLACK

AUDIO TRANSFORMERS

PLATE LEAD (PR/.)

BY LEAD (PR /. )

GRID LEAD (SEC.)

GRID RETURN (SEC.)

BLUE OR BROWN

RED

GREEN OR YELLOW

BLACK

circuits. If possible, the wiring should be

checked by a second party as a safety mea-

sure. Some tubes can be permanently damaged

by having the wrong voltages applied to their

electrodes. Electrolytic capacitors can be

ruined by hooking them up with the wrong

voltage polarity across the capacitor terminals.

Transformer, choke and coil windings may be

damaged by incorrect wiring of the high -volt-

age leads.

The problem of meeting and overcoming

such obstacles is just part of the game. A true

radio amateur (as opposed to an amateur

broadcaster) should have adequate knowledge

of the art of communication. He should know

quite a bit about his equipment (even if pur-

chased) and, if circumstances permit, he should

build a portion of his own equipment. Those

amateurs that do such construction work are

convinced that half of the enjoyment of the

hobby may be obtained from the satisfaction

of building and operating their own receiving

and transmitting equipment.

The Transceiver A popular item of equip-

ment on "five meters"

during the late "thirties," the transceiver is

making a comeback today complete with mod-

ern tubes and circuitry. In brief, the transceiver

is a packaged radio station combining the ele-

ments of the receiver and transmitter into a

single unit having a common power supply

and audio system. The present trend toward

compact equipment and the continued growth

of single sideband techniques combine natural-

ly with the space -saving economies of the

transceiver. Various transceiver circuits for the

higher frequency amateur bands are shown in

this chapter. The experimenter can start from

these simple circuits and using modern minia-

ture tubes and components can design and

build his complete station in a cabinet no larg-

er than a pre -war receiver.

EXTERNAL

ANTENNA

10H .002 B20 100 LFD HIVOLUNE A30 100LF0

0.0.5 C o.5 + is 2 n o.S + 15

Rr BUS

Figure 4

SCHEMATIC OF TRANSISTOR BROADCAST RECEIVER

B -9 -volt transistor battery. RCA VS -300

C, -365 Nµfd. Lafayette Radio Co. MS -214, or Allied Radio Co. 61H -009

L,- Transistor "Loopstick" coil. Lafayette Radio Co. MS -166

LS -3" loudspeaker, 12 -ohm voice coil. Lafayette Radio Co. SK -39

T,-500 ohm pri., 12 ohm sec. Transistor transformer. Thordarson TR -18

www.americanradiohistory.com

527

experimenter's instinct, even in those individ-

uals owning expensive commercial receivers.

These lucky persons have the advantage of

comparing their home -built product against the

best the commercial market has to offer. Some-

times such a comparison is surprising.

When the builder has finished the wiring of

a receiver it is suggested that he check his

wiring and connections carefully for possible

errors before any voltages are applied to the

STANDARD COLOR CODE- RESISTORS AND CAPACITORS

AXIAL LEAD RESISTOR

DROWN- INSULATED

BLACK - NON -INSULATED

INSULATED FIRSTRING SECOND RING THIRD RING

UNINSULATED BODY COLOR END COLOR DOT COLOR

COLOR FIRST FIGURE SECOND FIGURE MULTIPLIER

BLACK 0 0 NONE

BROWN 1 0

RED 2 2 00

ORANGE 3 3 .000

YELLOW 4 4 0.000

GREEN S 5 00.000

BLUE IS e 000.000

VIOLET 7 7 0,000.000

GRAY e e 00,000.000

WHITE 9 9 000.000,000

DISC CERAMIC RMA CODE

5 -DOT 3-DOT

CAPACITY

I MULTIPLIER

TOLERANCE

TEMPERATURE

COEFFICIENT

-.'G1 ' l _ --

M TOLERANCE

MULTIPLIER

I STa 2ND SIGNIFICANT FIGS

WIRE-WOUND RESISTORS NAVE 1ST

DIGIT BAND DOUBLE WIDTH.

RADIAL LEAD DOT RESISTOR

¡ MULTIPLIER

5- ROTRADIAL LEAD CERAMICCAPACITOR EXTENDED

COEFF

RANGE TC CERAMIC HICAP

TOLERANCE %1I i II CURE ('' rT,

COEFF IJIISIII L'JI)aL:L4aIl1

1ST FIGURE TOLERANCE

MULTIPLIER TOLERANCE

I- MULTIPLIER

LTC MULTIPLIER

RADIAL LEAD (BAND) RESISTOR

-MULTIPLIER

BY -PASS COUPLING CERAMIC CAPACITOR

CAPACITY

AXIAL

TEMP.

LEAD CERAMIC CAPACITOR

COEFF. CAPACITY

Q1111 1.11 I IGURE OIIM1111I- V(OPT `E O I NI!

I

TOLERANCE LIST FIGURE TOLERANCE

MULTIPLIER MULTIPLIERS TOLERANCE

MOLDED MICA TYPE CAPACITORS

CURRENT STANDARD

1 ST

2ND

CODE

L SIGNIFICANT FIGURE

JAN t

/9OB RMA

CODE

RMA 3 -DOT (OBSOLETE)

RATED 300 V.D.C. ± 20 % TOL.

MULTIPLIER

BUTTON SILVER MICA CAPACITOR

CLASS

TOLERANCE ¡F A 4s,L 1ST DIGIT

MULTIPLIER , 2ND DIGIT

3RD DIGIT

WHITE (RMA)",....te

BLACK(JAN) ® o eljAULTIPLIER

CLASS ' "- TOLERANCE i____%7 SIGNIFICANT FIG.

RMA 5 -DOT

1srL sIa FIGURE

( 2ND

CODE (OBSOLETE)

WORK.

VOLT TOLERANCE

RMA 6-DOT (OBSOLETE)

- 1ST l

,qp } SIG. FIGURE

J

RMA 4 -DOT (OBSOLETE)

WORK. VOLTAGE

¡ EMULTIPLIER

FRONT MULTIPLIER

1ST ND SIG. FIG. f MULTIPLIER MULTIPLIER

WORK. VOLT.

In REAR

MULTIPLIER

I TOLERANCE L L TOLERANCE

WORKING VOLTAGE 2NDJ

157 j SIG. FIGURE

TOLERANCE WORK.

VOLT. BLANK

MOLDED PAPER TYPE CAPACITORS

TUBULAR CAPACITOR

NORMALLY STAMPED

FOR VALUE 1571

r2NDjSIGNIFICANTFIGURE

MULTIPLIER

MOLDED FLAT CAPACITOR

COMMERCIAL CODE

-WORKING VOLTS

BLACK

BOO

-MULTIPLIER

2NDJ SIGNIFICANT

1ST J FIGURE

JAN CODE CAPACITOR

SILVER 1ST SIGNIFICANT FIG.

2ND1 l

MULTIPLIER

PrI LI I

TOLERANCE SIG. VOLTAGE FIG.

1ST I

WI

A 2 -DIGIT VOLTAGE RATING INDICATES MORE THAN

900 V. ADD 2 ZEROS TO END OF 2 DIGIT NUMBER.

TOLERANCE

CHARACTERISTIC

Figure 2

STANDARD COLOR CODE FOR RESISTORS AND CAPACITORS

The standard color code provides the necessary information required to properly identify color coded

resistors and capacitors. Refer to the color code for numerical values and the number of zeros (or multi-

plier) assigned to the colors used. A fourth color band on resistors determines the tolerance rating as

follows: Gold= 5%, silver -10%. Absence of the fourth band indicates a 20% tolerance rating.

Tolerance rating of capacitors is determined by the color code. For example: Red --2 %, green =5 %, etc.

The voltage rating of capacitors is obtained by multiplying the color value by 100. For example:

Orange =3 x100, or 300 volts.

www.americanradiohistory.com

CHAPTER TWENTY -SEVEN

Receivers and Transceivers

Receiver construction has just about become

a lost art. Excellent general coverage receivers

are available on the market in many price

ranges. However, even the most modest of

these receivers is relatively expensive, and most

of the receivers are designed as a compromise

-they must suit the majority of users, and

they must be designed with an eye to the price.

It is a tribute to the receiver manufacturers

that they have done as well as they have. Even

so, the c -w man must often pay for a high -

fidelity audio system and S -meter he never

uses, and the phone man must pay for the c -w

man's crystal filter. For one amateur, the re-

ceiver has too much bandspread; for the next,

too little. For economy's sake and for ease

of alignment, low -Q coils are often found in

the r -f circuits of commercial receivers, mak-

ing the set a victim of cross -talk and over-

loading from strong local signals. Rarely does

the purchaser of a commercial receiver realize

that he could achieve the results he desires

in a home -built receiver if he left off the frills

and trivia which he does not need but which

he must pay for when he buys a commercial

product.

The ardent experimenter, however, needs

no such arguments. He builds his receiver

merely for the love of the game, and the thrill

of using a product of his own creation.

It is hoped that the receiving equipment to

be described in this chapter will awaken the

FIGURE 1

COMPONENT NOMENCLATURE

CAPACITORS:

1- VALUES BELOW 99011JFD ARE INDICATED IN UNITS.

EXAMPLE. 1SOJJmFD DESIGNATED AS IS0.

2 - VALUES ABOVE 9991J.UFD ARE INDICATED IN DECIMALS.

EXAMPLE: OOSJJFD DESIGNATED AS .00S.

3- OTHER CAPACITOR VALUES ARE AS STATED.

EXAMPLE: ,01JFD, 0.51J1JFD, ETC.

4- TYPE OF CAPACITOR IS INDICATED BENEATH THE VALUE

DESIGNATION. SM = SILVER MICA

C = CERAMIC

M' MICA

P' PAPER

EXAMPLE. 250 .01 .001

C P M

5- VOLTAGE RATING OF ELECTROLYTIC OR "FILTER

CAPACITOR IS INDICATED BELOW CAPACITY DESIGNATION.

EXAMPLE' 10 20 25

450 ' 600. 10

6- THE CURVED LINE IN CAPACITOR SYMBOL REPRESENTS

THE OUTSIDE FOIL 'GROUND OF PAPER CAPACITORS.

THE NEGATIVE ELECTRODE OF ELECTROLYTIC CAPACITORS,

OR THE ROTOR OF VARIABLE CAPACITORS.

RESISTORS

1- RESISTANCE VALUES ARE STATED IN OHMS, THOUSANDS

OF OHMS (K), AND MEGOHMS (MI.

EXAMPLE, 270 OHMS = 270

4700 OHMS = 4.7 H

33,000 OHMS = 33 H

100.000 OHMS = 100 N OR 0.1 M

33.000,000 OHMS' 33 M

2- ALL RESISTORS ARE 1 -WATT COMPOSITION TYPE UNLESS

OTHERWISE NOTED. WATTAGE NOTATION IS THEN INDICATED

BELOW RESISTANCE VALUE. 47 N

EXAMPLE: 0.5

INDUCTORS.

MICROHENRIES= JAI

MILLIHENRIES= MH

HENRIES= H

SCHEMATIC SYMBOLSI

tII OR IL -

I

-LT

I CONDUCTORS JOINED

CONDUCTORS CROSSING CHASSIS GROUNO

BUT NOT JOINED

526

www.americanradiohistory.com

HANDBOOK Noise Suppression 525

Miscellaneous There are several other poten-

tial noise sources on a pas-

senger vehicle, but they do not necessarily

give trouble and therefore require attention

only in some cases.

The heat, oil pressure, and gas gauges can

cause a rasping or scraping noise. The gas

gauge is the most likely offender. It will cause

trouble only when the car is rocked or is in

motion. The gauge units and panel indicators

should both be by- passed with the 0.1 -µfd.

paper and 0.00l-pfd. mica or ceramic combina-

tion previously described.

At high car speeds under certain atmospheric

conditions corona static may be encountered

unless means are taken to prevent it. The re-

ceiving -type auto whips which employ a plas-

tic ball tip are so provided in order to minimize

this type of noise, which is simply a discharge

of the frictional static built up on the car. A

whip which ends in a relatively sharp metal

point makes an ideal discharge point for the

static charge, and will cause corona trouble

at a much lower voltage than if the tip were

hooded with insulation. A piece of Vinylite

sleeving slipped over the top portion of the

whip and wrapped tightly with heavy thread

will prevent this type of static discharge un-

der practically all conditions. An alternative

arrangement is to wrap the top portion of the

whip with Scotch brand electrical tape.

Generally speaking it is undesirable from

the standpoint of engine performance to use

both spark -plug suppressors and a distributor

suppressor. Unless the distributor rotor clear-

ance is excessive, noise caused by sparking

of the distributor rotor will not be so bad but

what it can be handled satisfactorily by a

noise limiter. If not, it is preferable to shield

the hot lead between ignition coil and distri-

butor rather than use a distributor suppressor.

In many cases the control rods, speedometer

cable, etc., will pick up high- tension noise

under the hood and conduct it up under the

dash where it causes trouble. If so, all con-

trol rods and cables should be bonded to the

fire wall (bulkhead) where they pass through,

using a short piece of heavy flexible braid of

the type used for shielding.

In some cases it may be necessary to bond

the engine to the frame at each rubber engine

mount in a similar manner. If a rear mountéd

whip is employed the exhaust tail pipe also

should be bonded to the frame if supported by

rubber mounts.

Locating Determining the source of cer-

Noise Sources tain types of noise is made

difficult when several things

are contributing to the noise, because elimi-

nation of one source often will make little or

no apparent difference in the total noise. The

following procedure will help to isolate and

identify various types of noise.

Ignition noise will be present only when the

ignition is on, even though the engine is turn-

ing over.

Generator noise will be present when the

motor is turning over, regardless of whether

the ignition switch is on. Slipping the drive

belt off will kill it.

Gauge noise usually will be present only

when the ignition switch is on or in the "left"

position provided on some cars.

Wheel static when present will persist when

the car clutch is disengaged and the ignition

switch turned off (or to the left position), with

the car coasting.

Body noise will be noticeably worse on a

bumpy road than on a smooth road, particu-

larly at low speeds.

www.americanradiohistory.com

524 Mobile Equipment THE RADI O

of the measures may already have been taken

when the auto receiver was installed.

First either install a spark plug suppressor

on each plug, or else substitute Autolite re-

sistor plugs. The latter are more effective than

suppressors, and on some cars ignition noise

is reduced to a satisfactory level simply by

installing them. However, they may not do an

adequate job alone after they have been in use

for a while, and it is a good idea to take the

following additional measures.

Check all high tension connections for gaps,

particularly the "pinch fit" terminal connec-

tors widely used. Replace old high tension

wiring that may have become leaky.

Check to see if any of the high tension wir-

ing is cabled with low tension wiring, or run

in the same conduit. If so, reroute the low ten-

sion wiring to provide as much separation as

practicable.

By -pass to ground the 6 -volt wire from the

ignition coil to the ignition switch at each

end with a 0.1-pfd. molded case paper capaci-

tor in parallel with a .001 -µfd. mica or cer-

amic, using the shortest possible leads.

Check to see that the hood makes a good

ground contact to the car body at several

points. Special grounding contactors are avail-

able for attachment to the hood lacings on cars

that otherwise would present a grounding

problem.

If the high -tension coil is mounted on the

dash, it may be necessary to shield the high

tension wire as far as the bulkhead, unless

it already is shielded with armored conduit.

Wheel Static Wheel static is either static

electricity generated by rotation

of the tires and brake drums, or is noise gen-

erated by poor contact between the front

wheels and the axles (due to the grease in the

bearings). The latter type of noise seldom is

caused by the rear wheels, but tire static may

of course be generated by all four tires.

Wheel static can be eliminated by insertion

of grounding springs under the front hub caps,

and by inserting "tire powder" in all inner

tubes. Both items are available at radio parts

stores and from most auto radio dealers.

Voltage Regulator Certain voltage regulators

Hash generate an objectionable

amount of "hash" at the

higher frequencies, particularly in the v -h -f

range. A large by -pass will affect the operation

of the regulator and possibly damage the

points. A small by -pass can be used, however,

without causing trouble. At frequencies above

the frequency at which the hash becomes ob-

jectionable (approximately 20 Mc. or so) a

small by -pass is quite effective. A 0.001 -µfd.

mica capacitor placed from the field terminal

of the regulator to ground with the shortest

possible leads often will produce sufficient

improvement. If not, a choke consisting of a-

bout 60 turns of no. 18 d.c.c. or bell wire

wound on a h -inch form can be added. This

should be placed right at the regulator termi-

nal, and the 0.001 -µfd. by -pass placed from

the generator side of the choke to ground.

Generator Whine Generator "whine" often can

be satisfactorily suppressed

from 550 kc. to 148 Mc. simply by by- passing

the armature terminal to ground with a special

"auto radio" by -pass of 0.25 or 0.5 pfd. in

parallel with a 0.001 -µfd. mica or ceramic ca-

pacitor. The former usually is placed on the

generator when an auto radio is installed, but

must be augmented by a mica or ceramic ca-

pacitor with short leads in order to be effec-

tive at the higher frequencies as well as on the

broadcast band.

When more drastic measures are required,

special filters can be obtained which are de-

signed for the purpose. These are recommend-

ed for stubborn cases when a wide frequency

range is involved. For reception only over a

comparatively narrow band of frequencies.

such as the 10 -meter amateur band, a highly

effective filter can be improvised by connect-

ing between the previously described parallel

by -pass capacitors and the generator arma-

ture terminal a resonant choke. This may con-

sist of no. 10 enamelled wire wound on a suit-

able form and shunted with an adjustable trim-

mer capacitor to permit resonating the com-

bination to the center of the frequency band

involved. For the 10 -meter band 11 turns close

wound on a one -inch form and shunted by a

3-30 µµfd. compression -type mica trimmer is

suitable. The trimmer should be adjusted ex-

perimentally at the center frequency.

When generator whine shows up after once

being satisfactorily suppressed, the condition

of the brushes and commutator should be

checked. Unless a by -pass capacitor has

opened up, excessive whine usually indicates

that the brushes or commutator are in need of

attention in order to prevent damage to the

generator.

Body Static Loose linkages or body or frame

joints anywhere in the car are

potential static producers when the car is in

motion, particularly over a rough road. Locat-

ing the source of such noise is difficult, and

the simplest procedure is to give the car a

thorough tightening up in the hope that the

offending poor contacts will be caught by the

procedure. The use of braided bonding straps

between the various sections of the body of

the car also may prove helpful.

www.americanradiohistory.com

HANDBOOK Noise Suppression 523

660

300

60

400 100 150 200 250

OUTPUT CURRENT, MA.

PE-103A

6 V. INPUT

300

Figure 11

APPROXIMATE OUTPUT VOLTAGE VS.

LOAD CURRENT FOR A PE -103A

DYNAMOTOR

At 150 ma. or less the 12 -volt brushes will

last almost as long as the 6-volt brushes.

The reason that these particular dynamo tors

can be operated in this fashion is that there

are two 6-volt windings on the armature, and

for 12 -volt operation the two are used in series

with both commutators working. The arrange-

ment described above simply substitutes for

the regular 6-volt winding the winding and

commutator which ordinarily came into opera-

tion only on 12 -volt operation. Some operators

have reported that the regulation of the PE-

103A may be improved by operating both com-

mutators in parallel with the 6 -volt line.

The three wires now coming out of the dy-

namotor are identified as follows: The smaller

wire is the positive high voltage. The heavy

wire leaving the same grommet is positive 6

volts and negative high voltage. The single

heavy wire leaving the other grommet is nega-

tive 6 volts. Whether the car is positive or

negative ground, negative high voltage can be

taken as car -frame ground. With the negative

of the car battery grounded, the plate current

can return through the car battery and the ar-

mature winding. This simply puts the 6 volts

in series with the 500 volts and gives 6 extra

volts plate voltage.

The trunk of a car gets very warm in sum-

mer, and if the transmitter and dynamotor are

mounted in the trunk it is recommended that

the end housings be left off the dynamotor to

facilitate cooling. This is especially impor-

tant in hot climates if the dynamotor is to be

loaded to more than 200 ma.

When replacing brushes on a PE -103A check

to see if the brushes are marked negative and

positive. If so, be sure to install them accord-

ingly, because they are not of the same mater-

ial. The dynamotor will be marked to show

which holder is negative.

When using a PE -103A, or any dynamotor

for that matter, it may be necessary to devote

one set of contacts on one of the control re-

lays to breaking the plate or screen voltage

to the transmitter oscillator, If these are sup-

plied by the dynamotor, because the output of

a dynamotor takes a moment to fall to zero

when the primary power is removed.

26 -5 Vehicular Noise

Suppression

Satisfactory reception on frequencies above

the broadcast band usually requires greater

attention to noise suppression measures. The

required measures vary with the particular ve-

hicle and the frequency range involved.

Most of the various types of noise that may

be present in a vehicle may be broken down

into the following main categories:

(1) Ignition noise.

(2) Wheel static (tire static, brake static,

and intermittent ground via front wheel bear-

ings).

(3) "Hash" from voltage regulator con-

tacts.

(4) "Whine" from generator commutator seg-

ment make and break.

(5) Static from scraping connections be-

tween various parts of the car.

There is no need to suppress ignition noise

completely, because at the higher frequencies

ignition noise from passing vehicles makes

the use of a noise limiter mandatory anyway.

However, the limiter should not be given too

much work to do, because at high engine

speeds a noisy ignition system will tend to

mask weak signals, even though with the lim-

iter working, ignition "pops" may appear to

be completely eliminated.

Another reason for good ignition suppres-

sion at the source is that strong ignition

pulses contain enough energy when integrated

to block the a -v -c circuit of the receiver, caus-

ing the gain to drop whenever the engine is

speeded up. Since the a -v -c circuits of the

receiver obtain no benefit from a noise dip-

per, it is important that ignition noise be sup-

pressed enough at the source that the a -v -c

circuits will not be affected even when the

engine is running at high speed.

Ignition Noise The following procedure

should be found adequate for

reducing the ignition noise of practically any

passenger car to a level which the dipper can

handle satisfactorily at any engine speed at

any frequency from 500 kc. to 148 Mc. Some

www.americanradiohistory.com

522 Mobile Equipment THE RADI O

11

fYI

PRESS -TO -TALK

SWITCH

RING

TIP

SHELL

(GROUND)

OF MIRE

PLUG

FigUrr IO

STANDARD CONNECTIONS FOR THE

PUSH -TO -TALK SWITCH ON A HAND.

HELD SINGLE- BUTTON CARBON

MICROPHONE

microphones on the surplus market use these

connections.

There is an increasing tendency among mo-

bile operators toward the use of microphones

having better frequency and distortion char-

acteristics than the standard single -button

type. The high- impedance dynamic type is

probably the most popular, with the ceramic -

crystal type next in popularity. The conven-

tional crystal type is not suitable for mobile

use since the crystal unit will be destroyed

by the high temperatures which can be reached

in a closed car parked in the sun in the sum-

mer time.

The use of low -level microphones in mobile

service requires careful attention to the elimi-

nation of common -ground circuits in the micro-

phone lead. The ground connection for the

shielded cable which runs from the transmitter

to the microphone should be made at only one

point, preferably directly adjacent to the grid

of the first tube in the speech amplifier. The

use of a low -level microphone usually will re-

quire the addition of two speech stages (a pen-

tode and a triode), but these stages will take

only a milliampere or two of plate current, and

150 ma. per tube of heater current.

PE -103A Dyne- Because of its availability

motor Power Unit on the surplus market at a

low price and its suitability

for use with about as powerful a mobile trans-

mitter as can be employed in a passenger car

without resorting to auxiliary batteries or a

special generator, the PE -103A is probably

the most widely used dynamotor for amateur

work. Therefore some useful information will

be given on this unit.

The nominal rating of the unit is 500 volts

and 160 ma., but the output voltage will of

course vary with load and is slightly higher

with the generator charging. Actually the 160

ma. rating is conservative, and about 275 ma.

can be drawn intermittently without overheat-

ing, and without damage or excessive brush

or commutator wear. At this current the unit

should not be run for more than 10 minutes at

a time, and the average "on" time should not

be more than half the average "off" time.

The output voltage vs. current drain is

shown approximately in figure 11. The exact

voltage will depend somewhat upon the loss

resistance of the primary connecting cable

and whether or not the battery is on charge.

The primary current drain of the dynamotor

proper (excluding relays) is approximately 16

amperes at 100 ma., 21 amperes at 160 ma.,

26 amperes at 200 ma., and 31 amperes at 250

ma. Only a few of the components in the base

are absolutely necessary in an amateur mobile

installation, and some of them can just as well

be made an integral part of the transmitter if

desired. The base can be removed for salvage

components and hardware, or the dynamotor

may be purchased without base.

To remove the base proceed as follows:

Loosen the four thumb screws on the base

plate and remove the cover. Remove the four

screws holding the dynamotor to the base

plate. Trace the four wires coming out of the

dynamotor to their terminals and free the lugs.

Then these four wires can be pulled through

the two rubber grommets in the base plate when

the dynamotoris separated from the base plate.

It may be necessary to bend the eyelets in the

large lugs in order to force them through the

gromm et s.

Next remove the two end housings on the

dynamotor. Each is held with two screws. The

high -voltage commutator is easily identified

by its narrower segments and larger diameter.

Next to it is the 12 -volt commutator. The 6-

volt commutator is at the other end of the ar-

mature. The 12 -volt brushes should be removed

when only 6 -volt operation is planned, in order

to reduce the drag.

If the dynamotor portion of the PE-103A

power unit is a Pioneer type VS-25 or a Rus-

sell type 530- (most of them are), the wires to

the 12 -volt brush holder terminals can be cross

connected to the 6 -volt brush holder terminals

with heavy jumper wires. One of the wires dis-

connected from the 12 -volt brush terminals is

the primary 12 -volt pigtail and will come free.

The other wire should be connected to the op-

posite terminal to form one of the jumpers.

With this arrangement it is necessary only

to remove the 6 -volt brushes and replace the

12 -volt brushes in case the 6-volt commutator

becomes excessively dirty or worn or starts

throwing solder. No difference in output volt-

age will be noted, but as the 12 -volt brushes

are not as heavy as the 6 -volt brushes it is

not permissible vi draw more than about 150

ma. except for emergency use until the 6 -volt

commutator can be turned down or repaired.

www.americanradiohistory.com

HANDBOOK Control Circuits 521

Do not attempt to control too many relays,

particularly heavy duty relays with large coils,

by means of an ordinary push -to -talk switch

on a microphone. These contacts are not de-

signed for heavy work, and the inductive kick

will cause more sparking than the contacts on

the microphone switch are designed to handle.

It is better to actuate a single relay with the

push -to -talk switch and then control all other

relays, including the heavy duty contactor for

the dynamotor or vibrator pack, with this relay.

The procedure of operating only one relay

directly by the push -to -talk switch, with all

other relays being controlled by this control

relay, will eliminate the often -encountered dif-

ficulty where the shutting down of one item of

equipment will close relays in other items as

a result of the coils of relays being placed in

series with each other and with heater circuits.

A recommended general control circuit, where

one side of the main control relay is connected

to the hot 6 -volt circuit, but all other relays

have one side connected to ground, is illus-

trated in figure 9. An additional advantage

of such a circuit is that only one control wire

need be run to the coil of each additional re-

lay, the other side of the relay coils being

grounded.

The heavy -duty 6 -volt solenoid -type contac-

tor relays such as provided on the PE -103A

and used for automobile starter relays usually

draw from 1.5 to 2 amperes. While somewhat

more expensive, heavy -duty 6-volt relays of

conventional design, capable of breaking 30

amperes at 6 volts d.c., are available with

coils drawing less than 0.5 ampere.

When purchasing relays keep in mind that

the current rating of the contacts is not a fixed

value, but depends upon (1) the voltage, (2)

whether it is a.c. or d.c., and (3) whether the

circuit is purely resistive or is inductive. If

in doubt, refer to the manufacturer's recom-

mendations. Also keep in mind that a dynamo-

tor presents almost a dead short until the ar-

mature starts turning, and the starting relay

should be rated at considerably more than the

normal dynamotor current.

Microphones The most generally used micro -

and Circuits phone for mobile work is the

single -button carbon. With a

high -output -type microphone and a high -ratio

microphone transformer, it is possible when

"close talking" to drive even a pair of push -

pull 6L6's without resorting to a speech am-

plifier. However, there is a wide difference in

the output of the various type single button

microphones, and a wide difference in the a-

mount of step up obtained with different type

microphone transformers. So at least one

speech stage usually is desirable.

One of the most satisfactory single button

PUSH- TO-TALK PUSH -TO -TALK

SWITCH ON MIKE RELAY

r

ALTERNATE

CONTROL

SWITCH

MAIN POWER RECEIVER

RELAY MUTING

RELAY

ANTENNA

CHANGEOVER

RELAY

Ll

I V

rY

ANY

OTHER

RELAYS

Figure 9

RELAY CONTROL CIRCUIT

Simplified schematic of the recommended

relay control circuit for mobile transmitters.

The relatively small push -to -talk relay is

controlled by the button on the microphone

or the communications switch. Then one of

the contacts on this relay controls the other

relays of t he transmitter; one side of the

coil of all the additional relays controlled

should be grounded.

microphones is the standard Western Electric

type F -1 unit (or Automatic Electric Co. equi-

valent). This microphone has very high output

when operated at 6 volts, and good fidelity

on speech. When used without a speech am-

plifier stage the microphone transformer should

have a 50 -ohm primary (rather than 200 or 500

ohms) and a secondary of at least 150,000

ohms and preferably 250,000 ohms.

The widely available surplus type T -17 mi-

crophone has higher resistance (200 to 500

ohmsl and lower output, and usually will re-

quire a stage of speech amplification except

when used with a very low power modulator

stage.

Unless an F -1 unit is used in a standard

housing, making contact to the button presents

somewhat of a problem. No serious damage will

result from soldering to the button if the con-

nection is made to one edge and the solder-

ing is done very rapidly with but a small a-

mount of solder, so as to avoid heating the

whole button.

A sound -powered type microphone removed

from one of the chest sets available in the

surplus market will deliver almost as much

voltage to the grid of a modulator stage when

used with a high -ratio microphone transformer

as will an F -1 unit, and has the advantage of

not requiring button current or a "hash filter."

This is simply a dynamic microphone designed

for high output rather than maximum fidelity.

The standardized connections for a single -

button carbon microphone provided with push -

to -talk switch are shown in figure 10. Prac-

tically all hand -held military-type single -button

www.americanradiohistory.com

520 Mobile Equipment THE RADIO

Figure 8

PI- NETWORK ANTENNA COUPLER

The pi- network antenna coupler is particu-

larly satisfactory for mobile work since the

coupler affords some degree of harmonic re-

duction, provides o coupling variation to

meet varying load conditions caused by fre-

quency changes, and can cancel out react-

ance presented to the transmitter at the end

of the antenna transmission line.

For use of the coupler on the 3.9 -Mc. band

CI should hove a maximum capacitance of

about 250 µµfd., LI should be about 9 mi-

crohenrys (30 turns 1" dia. by 2" long), and

C2 may include a fixed and a variable ele-

ment with maximum capacitance of about 1400

µµ/d. A 100 -µµfd. variable capacitor will be

suitable at C1 for the 14 -Mc. and 28 -Mc.

bonds, with o 350 -44fd. variable at C2. In-

ductor LI should have an inductance of a-

bout 2 microhenrys (11 turns 1" dia. by 1"

long) for the 14 -Mc. band, and about 0.8 ml-

crohenry (6 turns 1" dia. by 1" long) for the

28 -Mc. band.

quency. Or, if the tapped type of coil is used,

taps are changed until the proper number of

turns for the desired operating frequency is

found. This procedure is repeated for the dif-

ferent bands of operation.

Feeding the After much experimenting it

Center- Loaded has been found that the most

Antenna satisfactory method for feed-

ing the coaxial line to the

base of a center- loaded antenna is with the

pi- network coupler. Figure 8 shows the basic

arrangement, with recommended circuit con-

stants. It will be noted that relatively large

values of capacitance are required for all bands

of operation, with values which seem particu-

larly large for the 75-meter band. But refer-

ence to the discussion of pi- network tank cir-

cuits in Chapter 13 will show that the val-

ues suggested are normal for the values of im-

pedance, impedance transformation, and oper-

ating Q which are encountered in a mobile in-

stallation of the usual type.

26 -4 Construction and

Installation of Mobile

Equipment

It is recommended that the following mea-

sures be taken when constructing mobile equip-

ment, either transmitting or receiving, to en-

sure trouble -free operation over long periods:

Use only a stiff, heavy chassis unless the

chassis is quite small.

Use lock washers or lock nuts when mount-

ing components by means of screws.

Use stranded hook -up wire except where r -f

considerations make it inadvisable (such as

for instance the plate tank circuit leads in a

v -h -f amplifier). Lace and tie leads wherever

necessary to keep them from vibrating or flop-

ping around.

Unless provided with gear drive, tuning ca-

pacitors in the large sizes will require a rotor

lock. Filamentary (quick heating) tubes should

be mounted only in a vertical position.

The larger size carbon resistors and mica

capacitors should not be supported from tube

socket pins, particularly from miniature sock-

ets. Use tie points and keep the resistor and

capacitor "pigtails" short.

Generally speaking, rubber shock mounts

are unnecessary or even undesirable with pas-

senger car installations, or at least with full

size passenger cars. The springing is suffi-

ciently "sott" that well constructed radio

equipment can be bolted directly to the vehicle

without damage from shock or vibration. Un-

less shock mounting is properly engineered

as to the stiffness and placement of the shock

mounts, mechanical- resonance "amplification"

effects may actually cause the equipment to

be shaken more than if the equipment were

bolted directly to the vehicle.

Surplus military equipment provided with

shock or vibration mounts was intended for

use in aircraft, jeeps, tanks, gun- firing Naval

craft, small boats, and similar vehicles and

craft subject to severe shock and vibration.

Also, the shock mounting of such equipment

is very carefully engineered in order to avoid

harmful resonances.

To facilitate servicing of mobile equipment,

all interconnecting cables between units

should be provided with separable connectors

on at least one end.

Control Circuits The send -receive control cir-

cuits of a mobile installation

are dictated by the design of the equipment,

and therefore will he left to the ingenuity of

the reader. However, a few generalizations

and suggestions are in order.

www.americanradiohistory.com

HANDBOOK Mobile Antennas 519

A more effective radiator and a better line

match may be obtained by making the whip

approximately 10% feet long and feeding it

with 75 -ohm coax (such as RG -11 /U) via a

series capacitor, as shown in figure 6. The

relay and series capacitor are mounted inside

the trunk, as close to the antenna feedthrou h

or base -mount insulator as possible. The 10-

foot length applies to the overall length from

the tip of the whip to the point where the lead

in passes through the car body. The leads in-

side the car (connecting the coaxial cable,

relay, series capacitor and antenna lead)

should be as short as possible. The outer con-

ductor of both coaxial cables should be ground-

ed to the car body at the relay end with short,

heavy conductors.

A 100 -µµtd. midget variable capacitor is

suitable for C1. The optimum setting should

be determined experimentally at the center of

the band. This setting then will be satisfac-

tory over the whole band.

One suitable coupling arrangement for either

a 1/4-wave or 5/16 -wave whip on 10 meters is

to use a conventional tank circuit, inductively

coupled to a "variable link" coupling loop

which feeds the coaxial line. Alternatively, a

pi- network output circuit may be used. If the

input impedance of the line is very low and

the tank circuit has a low C/L ratio, it may

be necessary to resonate the coupling loop

with series capacitance in order to obtain suf-

ficient coupling. This condition often is en-

countered with a % -wave whip when the line

length approximates an electrical half wave-

length.

If an all -band center -loaded mobile antenna

is used, the loading coil at the center of the

antenna may be shorted out for operation of

the antenna on the 10 -meter band. The usual

type of center -loaded mobile antenna will be

between 9 and 11 feet long, including the cen-

ter- loading inductance which is shorted out.

Hence such an antenna may be shortened to

an electrical quarter -wave for the 10 -meter

band by using a series capacitor as just dis-

cussed. Alternatively, if a pi- network is used

in the plate circuit of the output stage of the

mobile transmitter, any reactance presented

at the antenna terminals of the transmitter by

the antenna may be tuned out with the pi -net-

work.

The All -Bond The great majority of mobile

Center -Loaded operation on the 14 -Mc. band

Mobile Antenna and below is with center

loaded whip antennas. These

antennas use an insulated bumper or body

mount, with provision for coaxial feed from

the base of the antenna to the transmitter, as

shown in figure 7.

The center -loaded whip antenna must be

CAR BODY

UNSHIELDED

LOADING COIL

RG-56/U LINE

TO TRANSMITTER

COAXIAL LINE

GROUNDED TO

FRAME OF CAR

ADJACENT TO BASE

OF ANTENNA

Figure 7

THE CENTER -LOADED WHIP ANTENNA

The center -loaded whip antenna, when pro-

vided with a topped loading coil or a series

of coils, may be used over a wide frequency

range. The loading coil may be shorted for

use of the antenna on the 10 -meter bond.

tuned to obtain optimum operation on the de-

sired frequency of operation. These antennas

will operate at maximum efficiency over a

range of perhaps 20 kc. on the 75 -meter band,

covering a somewhat wider range on the 40-

meter band, and covering the whole 20 -meter

phone band. The procedure for tuning the an-

tennas is discussed in the instruction sheet

which is furnished with them, but basically

the procedure is as follows:

The antenna is installed, fully assembled,

with a coaxial lead of RG -58 /U from the base

of the antenna to the place where the trans-

mitter is installed. The rear deck of the car

should be closed, and the car should be parked

in a location as clear as possible of trees,

buildings, and overhead power lines. Objects

within 15 or 20 feet of the antenna can exert

a considerable detuning effect on the antenna

system due to its relatively high operating Q.

The end of the coaxial cable which will plug

into the transmitter is terminated in a link of

3 or 4 turns of wire. This link is then coupled

to a grid -dip meter and the resonant frequency

of the antenna determined by noting the fre-

quency at which the grid current fluctuates.

The coils furnished with the antennas normal-

ly are too large for the usual operating fre-

quency, since it is much easier to remove

turns than to add them. Turns then are removed,

one at a time, until the antenna resonates at

the desired frequency. If too many turns have

been removed, a length of wire may be spliced

on and soldered. Then, with a length of insu-

lating tubing slipped over the soldered joint,

turns may be added to lower the resonant fre-

www.americanradiohistory.com

518 Mobile Equipment THE RADIO

Figure 5

A CENTER LOADED 80 -METER WHIP

USING AIR WOUND COIL MAY BE USED

WITH HIGH POWERED TRANSMITTERS

An anti -corona loop is placed at the top

of the whip to reduce loss of power and

burning of tip of antenna. Number of turns

in coil is critical and adjustable, high -Q

coil is refommended. Whip may be used

over frequency range of about 15 kilo-

cycles without retuning.

26 -3 Antennas for

Mobile Work

10 -Meter Mobile The most popular mobile an-

Antennas retna for 10 -meter operation

is a rear -mounted whip approx-

imately 8 feet long, fed with coaxial line. This

is a highly satisfactory antenna, but a few

950 COAX TO

RI

7511 COAX TO XuTR

Figure 6

5/16 -WAVE WHIP RADIATOR FOR 10

METERS

If a whip antenna is made slightly longer

than one -quarter wave it acts as a slightly

better radiator than the usual quarter -wave

whip, and it can provide a better match to

the antenna transmission line if the react-

ance is tuned out by a serles capacitor close

to the base of the antenna. Capacitor C1 may

be a 100 -µµid. midget variable.

remarks are in order on the subject of feed

and coupling systems.

The feed point resistance of a resonant quar-

ter -wave rear -mounted whip is approximately

20 to 25 ohms. While the standing -wave ratio

when using 50 -ohm coaxial line will not be

much greater than 2 to 1, it is nevertheless

desirable to make the line to the transmitter

exactly one quarter wavelength long electri-

cally at the center of the band. This procedure

will minimize variations in loading over the

band. The physical length of RG -8 /U cable,

from antenna base to antenna coupling coil,

should be approximately 5 feet 3 inches. The

antenna changeover relay preferably should be

located either at the antenna end or the trans-

mitter end of the line, but if it is more con-

venient physically the line may be broken any-

where for insertion of the relay.

If the same rear -mounted whip is used for

broadcast -band reception, attenuation of broad-

cast -band signals by the high shunt capaci-

tance of the low impedance feed line can be

reduced by locating the changeover relay right

at the antenna lead in, and by running 95 -ohm

coax (instead of 50 or 75 ohm coax) from the

relay to the converter. Ordinarily this will pro-

duce negligible effect upon the operation of

the .converter, but usually will make a worth-

while improvement in the strength of broadcast -

band signals.

www.americanradiohistory.com

HANDBOOK One -Tube Converter 517

auto -set combination has not proven very satis-

factory. The primary reason for this is the fact

that the relatively sharp i -f channel of the auto

set imposes too severe a limitation on the sta-

bility of the high- frequency oscillator in the

converter. And if a crystal- controlled beating

oscillator is used in the converter, only a por-

tion of the band may be covered by tuning the

auto set.

The most satisfactory arrangement has been

found to consist of a separately mounted i.f.,

audio, and power supply system, with the con-

verter mounted near the steering column. The

i -f system should have a bandwidth of 30 to

100 kc. and may have a center frequency of

10.7 Mc. if standard i -f transformers are to be

used. The control head may include the 144 -

Mc. r -f, mixer, and oscillator sections, and

sometimes the first i -f stage. Alternatively,

the control head may include only the h -f os-

cillator, with a broadband r -f unit included

within the main receiver assembly along with

the i.f. and audio system. Commercially manu-

factured kits and complete units using this

general lineup are available.

An alternative arrangement is to build a

converter, 10.7 -Mc. i -f channel, and second

detector unit, and then to operate this unit in

conjunction with the auto -set power supply,

audio system, and speaker. Such a system

makes economical use of space and power

drain, and can be switched to provide normal

broadcast -band auto reception or reception

through a converter for the h -f amateur bands.

A recent development has been the VHF

transceiver, typified by the Gonset Communi-

cator. Such a unit combines a crystal con-

trolled transmitter and a tunable VHF receiver

together with a common audio system and

power supply. The complete VHF station may

be packaged in a single cabinet. Various

forms of VHF transceivers are shown in the

construction chapters of this Handbook.

26-2 Mobile Transmitters

As in the case of transmitters for fixed -sta-

tion operation, there are many schools of

thought as to the type of transmitter which is

most suitable for mobile operation. One school

states that the mobile transmitter should have

very low power drain, so that no modification

of the electrical system of the automobile will

be required, and so that the equipment may be

operated without serious regard to discharging

the battery when the car is stopped, or over-

loading the generator when the car is in mo-

tion. A total transmitter power drain of about

80 watts from the car battery (6 volts at 13

amperes, or 12 volts at 7 amperes) is about

the maximum that can be allowed under these

conditions. For maximum power efficiency it

is recommended that a vibrator type of supply

be used as opposed to a dynamotor supply,

since the conversion efficiency of the vibrator

unit is high compared to that of the dynamotor.

A second school of thought states that the

mobile transmitter should be of relatively high

power to overcome the poor efficiency of the

usual mobile whip antenna. In this case, the

mobile power should be drawn from a system

that is independent from the electrical system

of the automobile. A belt driven high voltage

generator is often coupled to the automobile

engine in this type of installation.

A variation of this idea is to employ

a complete secondary power system in the

car capable of providing 115 volts a.c. Shown

in figure 4 is a Leece -Neville three phase

alternator mounted atop the engine block, and

driven with a fan belt. The voltage regulator

and selenium rectifier for charging the car

battery from the a -c system replace the usual

d -c generator. These new items are mounted

in the front of the car radiator. The alternator

provides a balanced deltaovtput circuit where-

in the line voltage is equal to the coil volt-

age, but the line current is N/3 times the coil

current. The coil voltage is a nominal 6- volts,

RMS and three 6.3 volt 25 ampere filament

transformers may be connected in delta on

the primary and secondary windings to step

the 6 -volts up to three -phase 115-volts. If

desired, a special 115 -volt, 3 -phase step -up

transformer may be wound which will occupy

less space than the three filament trans-

formers. Since the ripple frequency of a three

phase d -c power supply will be quite high, a

single 10 mfd filter capacitor will suffice.

Figure 4

LEECE -NEVILLE 3 -PHASE

ALTERNATOR IS ENGINE DRIVEN

BY AUXILIARY FAN BELT.

www.americanradiohistory.com

516 Mobile Equipment THE RADIO

converter end or the set end of the cable be-

tween the converter and receiver. This auxil-

iary trimmer should have a range of about 3

to 50 µµfd., and may be of the inexpensive

compression mica type.

with the trimmer cut out and the converter

turned off (by- passed by the "in -out" switch),

peak the regular antenna trimmer on the auto

set at about 1400 kc. Then turn on the convert-

er, with the receiver tuned to 1500 kc., switch

in the auxiliary trimmer, and peak this trimmer

for maximum background noise. The auxiliary

trimmer then can be left switched in at all

times except when receiving very weak broad-

cast band signals.

Some auto sets, particularly certain General

Motors custom receivers, employ a high -Q high -

impedance input circuit which is very critical

as to antenna capacitance. Unless the shunt

capacitance of the antenna (including cable)

approximates that of the antenna installation

for which the set was designed, the antenna

trimmer on the auto set cannot be made to hit

resonance with the converter cut out. This is

particularly true when a long antenna cable is

used to reach a whip mounted at the rear of the

car. Usually the condition can be corrected by

unsoldering the internal connection to the an-

tenna terminal connector on the auto set and

inserting in series a 100 -µµfd. mica capacitor.

Alternatively an adjustable trimmer covering

at least 50 to 150 µµfd. may be substituted for

the 100 -µµfd. fixed capacitor. Then the adjust-

ment of this trimmer and that of the regular an-

tenna trimmer can be juggled back and forth

until a condition is achieved where the input

circuit of the auto set is resonant with the con-

verter either in or out of the circuit. This will

provide maximum gain and image rejection

under all conditions of use.

Reducing Battery When the receiving installa-

Drain of the tion is used frequently, and

Receiver particularly when the receiv-

er is used with the car

parked, it is desirable to keep the battery

drain of the receiver -converter installation at

an absolute minimum. A substantial reduction

in drain can be made in many receivers, with-

out appreciably affecting their performance.

The saving of course depends upon the de-

sign of the particular receiver and upon how

much trouble and expense one is willing to go

to. Some receivers normally draw (without the

converter connected) as much as 10 amperes.

In many cases this can be cut to about 5 am-

peres by incorporating all practicable modi-

fications. Each of the following modifications

is applicable to many auto receivers.

If the receiver uses a speaker with a field

coil, replace the speaker with an equivalent

PM type.

Practically all 0.3- ampere r -f and a -f volt-

age amplifier tubes have 0.15- ampere equiva-

lents. In many cases it is not even necessary

to change the socket wiring. However, when

substituting i -f tubes it is recommended that

the i -f trimmer adjustments be checked. Gen-

erally speaking it is not wise to attempt to

substitute for the converter tube or a -f power

output tube.

If the a -f output tube employs conventional

cathode bias, substitute a cathode resistor of

twice the value originally employed, or add

an identical resistor in series with the one

already in the set. This will reduce the B

drain of the receiver appreciably without ser-

iously reducing the maximum undistorted out-

put. Because the vibrator power supply is much

less than 100 per cent efficient, a saving of

one watt of B drain results in a saving of near-

ly 2 watts of battery drain. This also mini-

mizes the overload on the B supply when the

converter is switched in, assuming that the

converter uses B voltage from the auto set.

If the receiver uses push -pull output and if

one is willing to- accept a slight reduction in

the maximum volume obtainable without dis-

tortion, changing over to a single ended stage

is simple if the receiver employs conventional

cathode bias. Just pull out one tube, double

the value of cathode bias resistance, and add

a 25 -{ád. by -pass capacitor across the cathode

resistor if not already by- passed. In some

cases it may be possible to remove a phase

inverter tube along with one of the a -f output

tubes.

If the receiver uses a motor driven station

selector with a control tube (d -c amplifier),

usually the tube can be removed without up-

setting the operation of the receiver. One then

must of course use manual tuning.

While the changeover is somewhat expen-

sive, the 0.6 ampere drawn by a 6X4 or 6X5

rectifier can be eliminated by substituting six

115-volt r -m -s 50 -ma. selenium rectifiers (such

as Federal type 402D3200). Three in series

are substituted for each half of the full -wave

rectifier tube. Be sure to observe the correct

polarity. The selenium rectifiers also make a

good substitution for an OZ4 or OZ4 -GT which

is causing hash difficulties when using the

converter.

Offsetting the total cost of nearly $4.00 is

the fact that these rectifiers probably will last

for the entire life of the auto set. Before pur-

chasing the rectifiers, make sure that there is

room available for mounting them. While these

units are small, most of the newer auto sets

employ very compact construction.

Two -Meter Reception For reception on the 144 -

Mc. amateur band, and

those higher in frequency, the simple converter-

www.americanradiohistory.com

HANDBOOK Mobile Receiver Installation 515

KEEP THESE LEADS SHORT

OR SHIELD THEM. \ I VIBRATOR

SPD T.

6v DC

6 V VIA CONTROL

RELAY IN XMTR.

TO HOT SIDE OF

VOICE COIL WINDING

Figure 3

METHOD OF ELIMINATING THE BATTERY

DRAIN OF THE RECEIVER VIBRATOR

PACK DURING TRANSMISSION

If the receiver chassis has room for o mid-

get s.p.d.t. relay, the above arrangement not

only silences the receiver on transmit but

saves several amperes battery drain.

If the normally open contact on the relay is

connected to the hot side of the voice coil

winding as shown in figure 3 (assuming one

side of the voice coil is grounded in accord-

ance with usual practice), the receiver will

be killed instantly when switching from re-

ceive to transmit, in spite of the fact that the

power supply filter in the receiver takes a

moment to discharge. However, if a "slow

start" power supply (such as a dynamotor or

a vibrator pack with a large filter) is used with

the transmitter, shorting the voice coil prob-

ably will not be required.

Using the Receiver An alternative and high -

Plate Supply ly recommended proce-

On Transmit dure is to make use of

the receiver B supply on

transmit, instead of disabling it. One disad-

vantage of the popular PE -103A dynamotor is

the fact that its 450 -500 volt output is too high

for the low power r -f and speech stages of the

transmitter. Dropping this voltage to a more

suitable value of approximately 250 volts by

means of dropping resistors is wasteful of

power, besides causing the plate voltage on

the oscillator and any buffer stages to vary

widely with tuning. By means of a midget 6-

volt s.p.d.t. relay mounted in the receiver, con-

nected as shown in figure 2, the B supply of

the auto set is used to power the oscillator

and other low power stages (and possibly

screen voltage on the modulator). On transmit

the B voltage is removed from the receiver

and converter, automatically silencing the re-

ceiver. When switching to receive the trans-

mitter oscillator is killed instantly, thus avoid-

ing trouble from dynamotor "carry over."

The efficiency of this arrangement is good

because the current drain on the main high

voltage supply for the modulated amplifier and

modulator plate(s) is reduced by the amount

of current borrowed from the receiver. At least

80 ma. can be drawn from practically all auto

sets, at least for a short period, without dam-

age. It will be noted that with the arrangement of

figure 2, plate voltage is supplied to the audio

output stage at all times. However, when the

screen voltage is removed, the plate current

drops practically to zero.

The 200 -ohm resistor in series with the nor-

mally open contact is to prevent excessive

sparking when the contacts close. If the relay

feeds directly into a filter choke or large ca-

pacitor there will be excessive sparking at

the contacts. Even with the arrangement shown,

there will be considerable sparking at the con-

tacts; but relay contacts can stand such spark-

ing quite a while, even on d.c., before becom-

ing worn or pitted enough to require attention.

The 200 -ohm resistor also serves to increase

the effectiveness of the .01 -fifd. r -f by -pass

capacitor.

Auxiliary Antenna One other modification of

Trimmer the auto receiver which may

or may not be desirable de-

pending upon the circumstances is the addi-

tion of an auxiliary antenna trimmer capacitor.

If the converter uses an untuned output circuit

and the antenna trimmer on the auto set is

peaked with the converter cut in, then it is

quite likely that the trimmer adjustment will

not be optimum for broadcast -band reception

when the converter is cut out. For reception

of strong broadcast band signals this usually

will not be serious, but where reception of

weak broadcast signals is desired the loss in

gain often cannot be tolerated, especially in

view of the fact that the additional length of

antenna cable required for the converter in-

stallation tends to reduce the strength of

broadcast band signals.

If the converter has considerable reserve

gain, it may be practicable to peak the antenna

trimmer on the auto set for broadcast -band re-

ception rather than resonating it to the con-

verter output circuit. But oftentimes this re-

sults in insufficient converter gain, excessive

image troubles from loud local amateur sta-

tions, or both.

The difficulty can be circumvented by in-

corporation of an auxiliary antenna trimmer

connected from the "hot" antenna lead on the

auto r e c ei v e r to ground, with a switch in

series for cutting it in or out. This capacitor

and switch can be connected across either the

www.americanradiohistory.com

514 Mobile Equipment THE RADIO

6V6 OR 6AQ5

( OPTIONAL

OVC

1

Br TO REST

OF SET

6X4

30 ..UF

-÷7°'

200

2w

Br r0

CONVERTER

REGULAR

R -C FILTER

=OLF

B+ 200 TO 250 V.

TO xMTR

I 0 HY - - - 1

I

I Br FROM ((}}

""((}} O Br TO LOW POWERI

I RECEIVER STAGES SPEECH

Ti . I

I T

i 1

_ I

I

L XMTR J

6 V. VIA CONTROL

RELAY IN XMTR

Figure 2

USING THE RECEIVER PLATE SUPPLY

FOR THE TRANSMITTER

This circuit silences the receiver on trans-

mit, and in addition makes it possible to use

the receiver plate supply for feeding the ex.

citer and speech amplifier stages In the

transmitter.

is to mount a small receptacle on the receiver

cabinet or chassis, making connection via a

matching plug. An Amphenol type 77 -26 recep-

tacle is compact enough to fit in a very small

space and allows four connections (including

ground for the shield braid). The matching plug

is a type 70 -26.

To avoid the possibility of vibrator hash

being fed into the converter via the heater and

plate voltage supply leads, it is important that

the heater and plate voltages be taken from

points well removed from the power supply por-

tion of the auto receiver. If a single -ended

audio output stage is employed, a safe place

to obtain these voltages is at this tube socket,

the high voltage for the converter being taken

from the screen. In the case of a push -pull out-

put stage, however, the screens sometimes are

fed from the input side of the power supply

filter. The ripple at this point, while suffi-

ciently low for a push -pull audio output stage,

is not adequate for a converter without addi-

tional filtering. If the schematic shows that

the screens of a push -pull stage are connected

to the input side instead of the output side of

the power supply filter (usually two electroly-

tics straddling a resistor in an R -C filter), then

follow the output of the filter over into the r -f

portion of the set and pick it up there at a con-

venient point, before it goes through any addi-

tional series dropping or isolating resistors,

as shown in figure 2.

The voltage at the output of the filter usual-

ly runs from 200 to 250 volts with typical con-

verter drain and the motor not running. This

will increase perhaps 10 per cent when the

generator is charging. The converter drain will

drop the B voltage slightly at the output of the

filter, perhaps 15 to 25 volts, but this reduc-

tion is not enough to have a noticeable effect

upon the operation of the receiver. If the B

voltage is higher than desirable or necessary

for proper operation of the converter, a 2 -watt

carbon resistor of suitable resistance should

be inserted in series with the plate voltage

lead to the power receptacle. Usually some-

thing between 2200 and 4700 ohms will be

found about right.

Receiver Disabling When the battery drain is

on Transmit high on transmit, as is the

case when a PE -103A dy-

namotor is run at maximum rating and other

drains such as the transmitter heaters and auto

headlights must be considered, it is desirable

to disable the vibrator power supply in the re-

ceiver during transmissions. The vibrator

power supply usually draws several amperes,

and as the receiver must be disabled in some

manner anyhow during transmissions, opening

the 6 -volt supply to the vibrator serves both

purposes. It has the further advantage of intro-

ducing a slight delay in the receiver recovery,

due to the inertia of the power supply filter,

thus avoiding the possibility of a feedback

"yoop" when switching from transmit to re-

ceive.

To avoid troubles from vibrator hash, it is

best to open the ground lead from the vibrator

by means of a midget s.p.d.t. 6 -volt relay and

thus isolate the vibrator circuit from the ex-

ternal control and switching circuit wires. The

relay is hooked up as shown in figure 3: Stand-

ard 8- ampere contacts will be adequate for

this application.

The relay should be mounted as close to

the vibrator as practicable. Ground one of the

coil terminals and run a shielded wire from

the other coil terminal to one of the power re-

ceptacle connections, grounding the shield at

both ends. By -pass each end of this wire to

ground with .01 pfd., using the shortest pos-

sible leads. A lead is run from the correspond-

ing terminal on the mating plug to the control

circuits, to be discussed later.

www.americanradiohistory.com

HANDBOOK Mobile Receivers 513

shown. !Multi- position tone controls tied in

with the second detector circuit often permit

excessive "leak through." Hence it is recom-

mended that the tone control components be

completely removed unless they are confined

to the grid of the a -f output stage. If removed,

the highs can be attenuated any desired amount

by connecting a mica capacitor from plate to

screen on the output stage. Ordinarily from

.005 to .01 tfd. wilt r:ovide a good compro-

mise between fidelity and reduction of back-

ground hiss on weak signals.

Usually the s wit c h SW will have to be

mounted some distance from the noise limiter

components. If the leads to the switch are over

approximately 1 ii in c h e s long, a piece of

shield braid should be slipped over them and

grounded. The same applies to the "hot" leads

to the volume control if not already shielded.

Closing the switch disables the limiter. This

may be desirable for reducing distortion on

broadcast reception or when checking the in-

tensity of ignition noise to determine the ef-

fectiveness of suppression measures taken on

the car. The switch also permits one to check

the effectiveness of the noise clipper.

The 22,000 -ohm decoupling resistor at the

bottom end of the i -f transformer secondary is

not critical, and if some other value already

is incorporated inside the shield can it may be

left alone so long as it is not over 47,000

ohms, a common value. Higher values must be

replaced with a lower value even if it requires

a can opener, because anything over 47,000

ohms will result in excessive loss in gain.

There is some loss in a -f gain inherent in this

type of limiter anyhow (slightly over 6 db),

and it is important to minimize any additional

loss. It is important that the total amount of ca-

pacitance in the RC decoupling (r -f) filter not

exceed about 100 µµfd. With a value much

greater than this "pulse stretching" will occur

and the effectiveness of the noise clipper will

be reduced. Excessive capacitance will reduce

the amplitude and increase the duration of the

ignition pulses before they reach the clipper.

The reduction in pulse amplitude accomplishes

no good since the pulses are fed to the clipper

anyhow, but the greater duration of the length-

ened pulses increases the audibility and the

blanking interval associated with each pulse.

If a shielded wire to an external dipper is em-

ployed, the r -f by -pass on the "low" side of

the RC filter may be eliminated since the ca-

pacitance of a few feet of shielded wire will

accomplish the same result as the by -pass

capacitor.

The switch SW is connected in such a man-

ner that there is practically no change in gain

with the limiter in or out. If the auto set does

not have any reserve gain and more gain is

needed on weak broadcast signals, the switch

can be connected from the hot side of the vol-

ume control to the j unction of the 22,000,

270,000 and 1 megohm resistors instead of as

shown. This will provide approximately 6 db

more gain when the clipper is switched out.

Many late model receivers are provided with

an internal r -f gain control in the cathode of

the r -f and /or i -f stage. This control should

be advanced full on to provide better noise

limiter action and make up for the loss in audio

gain introduced by the noise clipper.

Installation of the noise clipper often de-

tunes the secondary of the last i -f transformer.

This should be repeaked before the set is per-

manently replaced in the car unless the trim-

mer is accessible with the set mounted in

place.

Additional clipper circuits will be found in

the receiver chapter of this llandbook.

Selectivity While not of serious concern on

10 meters, the lack of selectivity

exhibited by a typical auto receiver will result

in QRM difficulty on 20 and 75 meters. A typi-

cal auto set has only two i -f transformers of

relatively low -Q design, and the second one

is loaded by the diode detector. The skirt se-

lectivity often is so poor that a strong local

will depress the a.v.c. when listening to a

weak station as much as 15 kc. different in

frequency.

One solution is to add an outboard i -f stage

employing two good quality double -tuned trans-

formers (not the midget variety) connected

"back -to- back" through a small coupling ca-

pacitance. The amplifier tube (such as a 6BA6)

should be biased to the point where the gain

of the outboard unit is relatively small (1 or

2), assuming that the receiver already has ade-

quate gain. If additional gain is needed, it may

be provided by the outboard unit. Low- capaci-

tance shielded cable should be used to couple

into and out of the outboard unit, and the unit

itself should be thoroughly shielded.

Such an outboard unit will sharpen the nose

selectivity slightly and the skirt selectivity

greatly. Operation then will be comparable to

a home -station communications receiver,

though selectivity will not be as good as a

receiver employing a 50 -kc. or 85 -kc. "Q5'er."

Obtaining Power While the set is on the

for the Converter bench for installation of

the noise clipper, provi-

sion should be made for obtaining filament and

plate voltage for the converter, and for the ex-

citer and speech amplifier of the transmitter,

if such an arrangement is to be used. To per-

mit removal of either the converter or the auto

set from the car without removing the other, a

connector should be provided. The best method

www.americanradiohistory.com

512 Mobile Equipment THE RADIO

sary, and it is recommended that a noise clip-

per be installed without confirming the neces-

sity therefor. It has been found that quiet re-

ception sometimes may be obtained on 75 me-

ters simply by the use of resistor type plugs,

but after a few thousand miles these plugs

often become less effective and no longer do

a fully adequate job. Also, a noise clipper in-

sures against ignition noise from passing

trucks and "un- suppressed" cars. On 10 me-

ters a noise clipper is a "must" in any case.

Modifying the There are certain things that

Auto Receiver should be done to the auto set

when it is to be used with a

converter, and they might as well be done all

at the same time, because "dropping" an auto

receiver and getting into the chassis to work

on it takes quite a little time.

First, however, check the circuit of the auto

receiver to see whether it is one of the few

receivers which employ circuits which com-

plicate connection of a noise clipper or a con-

verter. If the receiver is yet to be purchased,

it is well to investigate these points ahead of

time. If the receiver uses a negative B resistor

strip for bias (as evidenced by the cathode of

the audio output stage being grounded), then

the additional plate current drain of the con-

verter will upset the bias voltages on the var-

ious stages and probably cause trouble. Be-

cause the converter is not on all the time, it

is not practical simply to alter the resistance

of the bias strip, and major modification of the

receiver probably will be required.

The best type of receiver for attachment of

a converter and noise clipper uses an r -f stage;

permeability tuning; single unit construction

(except possibly for the speaker); push button

tuning rather than a tuning motor; a high vacu-

um rectifier such as a 6X4 (rather than an OZ4

or a synchronous rectifier); a 6SQ7 (or minia-

ture or Loctal equivalent) with grounded cath-

ode as second detector, first audio, and a.v.c.;

power supply negative grounded directly (no

common bias strip); a PM speaker (to minimize

battery drain); and an internal r -f gain control

(indicating plenty of built -in reserve gain

which may be called upon if necessary). Many

current model auto radios have all of the fore-

going features, and numerous models have most

of them, something to keep in mind if the set

is yet to be purchased.

Noise Limiters A noise limiter either may be

built into the set or purchased

as a commercially manufactured unit for "out-

board" connection via shielded wires. If the

receiver employs a 6SQ7 (or Loctal or minia-

ture equivalent) in a conventional circuit, it is

a simple matter to build in a noise clipper by

I F TRANS

TO A F.

POWER

AMP.

Figure 1

SERIES -GATE NOISE LIMITER FOR AUTO

RECEIVER

Auto receivers using a 6SQ7, 786, 7X7, or

6A T6 as second detector and a.v.c. can be

converted to the above circuit with but few

wiring changes. The circuit hos the advan-

tage of not requiring an additional tube sock-

et for the limiter diode.

substituting a 6S8 octal, 7X7 Loctal, or a 6T8

9 -pin miniature as shown in figure 1. When

substituting a 6T8 for a 6ÁT6 or similar 7 -pin

miniature, the socket must be changed to a

9 -pin miniature type. This requires reaming

the socket hole slightly.

If the receiver employs cathode bias on the

6SQ7 (or equivalent), and perhaps delayed

a.v.c., the circuit usually can be changed to

the grounded -cathode circuit of figure 1 with-

out encountering trouble.

Some receivers take the r -f excitation for

the a -v -c diode from the plate of the i -f stage.

In this case, leave the a.v.c. alone and ignore

the a -v -c buss connection shown in figure 1

(eliminating the 1- megohm decoupling resistor).

If the set uses a separate a -v -c diode which

receives r -f excitation via a small capacitor

connected to the detector diode, then simply

change the circuit to correspond to figure 1.

In case anyone might be considering the use

of a crystal diode as a noise limiter in con-

junction with the tube already in the set, it

might be well to point out that crystal diodes

perform quite poorly in series -gate noise clip-

pers of the type shown.

It will be observed that no tone control is

www.americanradiohistory.com

CHAPTER TWENTY -SIX

Mobile Equipment

Design and Installation

Mobile operation is permitted on all amateur

bands. Tremendous impetus to this phase of

the hobby was given by the suitable design of

compact mobile equipment. Complete mobile

installations may be purchased as packaged

units, or the whole mobile station may be home

built, according to the whim of the operator.

The problems involved in achieving a satis-

factory two -way installation vary somewhat

with the band, but many of the problems are

common to all bands. For instance, ignition

noise is more troublesome on 10 meters than

on 75 meters, but on the other hand an effi-

cient antenna system is much more easily ac-

complished on 10 meters than on 75 meters.

Also, obtaining a worthwhile amount of trans-

mitter output without excessive battery drain

is a problem on all bands.

26 -1 Mobile Reception

When a broadcast receiver is in the car, the

most practical receiving arrangement involves

a converter feeding into the auto set. The ad-

vantages of good selectivity with good image

rejection obtainable from a double conversion

superheterodyne are achieved in most cases

without excessive "birdie" troubles, a com-

511

mon difficulty with a double conversion super-

heterodyne constructed as an integral receiver

in one cabinet. However, it is important that

the b-c receiver employ an r -f stage in order to

provide adequate isolation between the con-

verter and the high frequency oscillator in the

b -c receiver. The r -f stage also is desirable

from the standpoint of image rejection if the

converter does not employ a tuned output cir-

cuit (tuned to the frequency of the auto set,

usually about 1500 kc.). A few of the late

model auto receivers, even in the better makes,

do not employ an r -f stage.

The usual procedure is to obtain converter

plate voltage from the auto receiver. Experi-

ence has shown that if the converter does not

draw more than about 15 or at most 20 ma. tot-

al plate current no damage to the auto set or

loss in performance will occur other than a

slight reduction in vibrator life. The converter

drain can be minimized by avoiding a voltage

regulator tube on the converter h -f oscillator.

On 10 meters and lower frequencies it is pos-

sible to design an oscillator with sufficient

stability so that no voltage regulator is re-

quired in the converter.

With some cars satisfactory 75-meter opera-

tion can be obtained without a noise clipper

if resistor type spark plugs (such as those

made by Autolite) are employed. However, a

noise clipper is helpful if not absolutely neces-

www.americanradiohistory.com

510 Rotary Beams

25 -10 Indication of Direction

The most satisfactory method for indicating

the direction of transmission of a rotatable

array is that which uses Selsyns or synchros

for the transmission of the data from the ro-

tating structure to the indicating pointer at the

operating position. A number of synchros and

Selsyns of various types are available on the

surplus market. Some of them are designed for

operation on 115 volts at 60 cycles, some are

designed for operation on 60 cycles but at a

lowered voltage, and some are designed for

operation from 400 -cycle or 800 -cycle energy.

This latter type of high- frequency synchro is

the most generally available type, and the

high- frequency units are smaller and lighter

than the 60 -cycle units. Since the indicating

synchro must deliver an almost negligible

amount of power to the pointer which it drives,

the high- frequency types will operate quite

satisfactorily from 60 -cycle power if the volt-

age on them is reduced to somewhere between

6.3 and 20 volts. In the case of many of the

units available, a connection sheet is provided

along with a recommendation in regard to the

operating voltage when they are run on 60 cy-

cles. In any event the operating voltage should

be held as low as it may be and still give sat-

isfactory transmission of data from the anten-

na to the operating position. Certainly it should

not be necessary to run such a voltage on the

units that they become overheated.

A suitable Selsyn indicating system is shown

in figure 21.

Systems using a potentiometer capable of

continuous rotation and a milliammeter, along

with a battery or other source of direct current,

may also be used for the indication of direc-

tion. A commercially -available potentiometer

(Ohmite RB -2) may be used in conjunction

with a 0 -1 d -c milliammeter having a hand -

calibrated scale for direction indication.

25 -11 "Three -Band" Beams

A popular form of beam antenna introduced

during the past few years is the so- called

three -band beam. An array of this type is de-

signed to operate on three adjacent amateur

bands, such as the ten, fifteen, and twenty

meter group. The principle of operation of

this form of antenna is to employ parallel

tuned circuits placed at critical positions in

the elements of the beam which serve to elec-

trically connect and disconnect the outer sec-

tions of the elements as the frequency of ex-

citation of the antenna is changed. A typical

'three -band" element is shown in figure 22.

At the lowest operating frequency, the tuned

traps exert a minimum influence upon the ele-

ment and it resonates at a frequency determined

by the electrical length of the configuration,

plus a slight degree of loading contributed by

the traps. At some higher frequency (generally

about 1.5 times the lowest operating frequency)

the outer set of traps are in a parallel resonant

condition, placing a high impedance between

the element and the tips beyond the traps.

Thus, the element resonates at a frequency

1.5 times higher than that determined by the

overall length of the element. As the frequency

of operation is raised to approximately 2.0

times the lowest operating frequency, the inner

set of traps become resonant, effectively dis-

connecting a larger portion of the element from

the driven section. The length of the center

section is resonant at the highest frequency

of operation. The center section, plus the two

adjacent inner sections are resonant at the

intermediate frequency of operation, and the

complete element is resonant at the lowest

frequency of operation.

The efficiency of such a system is deter-

mined by the accuracy of tuning of both the

element sections and the isolating traps. In

addition the combined dielectric losses of the

traps affect the overall antenna efficiency.

As with all multi -purpose devices, some com-

promise between operating convenience and

efficiency must be made with antennas designed

to operate over more than one narrow band of

frequencies. It is a tribute to the designers of

the better multi -band beams that they perform

as well as they do, taking into account the

theoretical difficulties that must be overcome.

ISOLATING TRAPS -

r'

FEED POINT

it RESO NANT-

T

AT NIGNEST FREQUENCY

RESONANT 2 AT

INTERMEDIATE FREQUENCY

RESONANT ; AT LOWEST FREQUENCY

Figure 22

TRAP -TYPE "THREE BAND"

ELEMENT

Isolating traps permit dipole to be

self -resonant at three widely different

frequencies.

www.americanradiohistory.com

HANDBOOK Antenna Control Systems 509

- _ _ CONTROL BOX

S.P D.T. SOCKET

RELAY PLUG

ANTENNA ROTATOR

r

SOCKET 1 PLUG

fin

1 e R

IS-CONTACT JONES PLUGS SOCKETS

ROTARY BEAM CONTROL

D.P.D.T. TOGGLE SWITCH

PILOT

LIGHT

ITO 115 V.A.C.

i TOGGLE

SWITCH SOCKET j PLUG

L

r

TO PROP MOTOR 1

SYNCHRO.

GENERATOR

INDICATOR

SYNCHRO.

J

SOCKET L PLUG

DIRECTION INDICATOR

Figure 21

SCHEMATIC OF A COMPLETE ANTENNA CONTROL SYSTEM

1

J

ating position as possible. However, on a par-

ticular installation the positions of the current

minimums on the transmission line near the

transmitter may be checked with the array in

the air, and then the array may be lowered to

ascertain whether or not the positions of these

points have moved. If they have not, and in

most cases if the feeder line is strung out back

and forth well above ground as the antenna is

lowered they will not change, the positions of

the last few toward the antenna itself may be

determined. Then the calculation of the match-

ing quarter -wave section may be made, the sec-

tion installed, the standing -wave ratio again

checked, and the antenna re- installed in its

final location.

25 -9 Antenna Rotation Systems

Structures for the rotation of antenna arrays

may be divided into two general classes: the

rotating mast and the rotating platform. The

rotating mast is especially suitable where the

transmitting equipment is installed in the gar-

age or some structure away from the main

house. Such an installation is shown in figure

19. A very satisfactory rotation mechanism is

obtained by the use of a large steering wheel

located on the bottom pipe of the rotating mast,

with the thrust bearing for the structure lo-

cated above the roof.

If the rotating mast is located a distance

from the operating position, a system of pul-

leys and drive rope may be used to turn the an-

tenna, or a slow speed electric motor may be

employed.

The rotating platform system is best if a

tower or telephone pole is to be used for an-

tenna support. A number of excellent rotating

platform devices are available on the market

for varying prices. The larger and more expen-

sive rotating devices are suitable for the rdta-

of a rather sizeable array for the 14-Mc. band

while the smaller structures, such as those

designed for rotating a TV antenna are design-

ed for less load and should be used only with

a 28 -Mc. or 50 -Mc. array. Most common prac-

tice is to install the rotating device atop a

platform built at the top of a telephone pole

or on the top of a lattice mast of sizeable cross

section so that the mast will be self- support-

ing and capable of withstanding the torque im-

posed upon it by the rotating platform.

A heavy duty TV rotator may be employed

for rotation of 6 and 10 -meter arrays. Fifteen

and twenty meter arrays should use rotators

designed for amateur use such as the Cornell -

Dubilier HAM -I unit shown in figure 20.

www.americanradiohistory.com

508 Rotary B e a m s T H E R A D I O

has been determined previously, and the

Antennascope control is turned for a null

readingon the meter of the Antennascope.

The impedance presented to the Antenna -

scope by the matching device may be

read directly on the calibrated dial of

the Antennascope.

3. Adjustments should be made to the

matching device to present the desired

impedance transformation to the Antenna -

scope. If a folded dipole is used as the

driven element, the transformation ratio

of the dipole must be varied as explained

previously in this chapter to provide a

more exact match. If a T -match or gamma

match system is used, the length of the

matching rod may be changed to effect

a proper match. If the Antennascope ohm-

ic reading is lower than the desired read-

ing, the length of the matching rod should

be increased. If the Antennascope read-

ing is higher than the desired reading,

the length of the matching rod should be

decreased. After each change in length

of the matching rod, the series capacitor

in the matching system should be re-

resonated for best null on the meter of

the Antennascope.

Raising and A practical problem always pres-

Lowering ent when tuning up and matching

the Array an array is the physical location

of the structure. If the array is

atop the mast it is inaccessible for adjustment,

and if it is located on stepladders where it can

be adjusted easily it cannot be rotated. One

encouraging factor in this situation is the fact

that experience has shown that if the array is

placed 8 or 10 feet above ground on some step-

ladders for the preliminary tuning process, the

raising of the system to its full height will not

produce a serious change in the adjustments.

So it is usually possible to make preliminary

adjustments with the system located slightly

greater than head height above ground, and

then to raise the antenna to a position where

it may be rotated for final adjustments. If the

position of the sliding sections as determined

near the ground is marked so that the adjust-

ments will not be lost, the array may be raised

to rotatable height and the fastening clamps

left loose enough so that the elements may be

slid in by means of a long bamboo pole. After

a series of trials a satisfactory set of lengths

can be obtained. But the end results usually

come so close to the figures given in figure 5

that a subsequent array is usually cut to the

dimensions given and installed as -is.

The matching process does not require ro-

tation, but it does require that the antenna

proper be located at as nearly its normal oper-

P'

4041,,

Figure 20

HEAVY DUTY ROTATOR SUITABLE

FOR AMATEUR BEAMS

The new Cornell -Dubilier type HAM -1

rotor has extra heavy motor and gear-

ing system to withstand weight and

inertia of amateur array under the

buffeting of heavy winds. Steel spur

gears and rotor lock prevent "pin -

wheeling" of antenna.

www.americanradiohistory.com

HANDBOOK Tuning the Array 507

cess of tuning the array is made a substan-

tially separate process as just described. Alter

the tuning operation is complete, the resonant

frequency of the driven element of the antenna

should be checked, directly at the center of

the driven element if practicable, with a grid -

dip meter. It is important that the resonant fre-

quency of the antenna be at the center of the

frequency band to be covered. If the resonant

frequency is found to be much different from

the desired frequency, the length of the driven

element of the array should be altered until

this condition exists. A relatively small change

in the length of the driven element will have

only a second order effect on the tuning of the

parasitic elements of the array. Hence, a mod-

erate change in the length of the driven ele-

ment may be made without repeating the tuning

process for the parasitic elements.

When the resonant frequency of the antenna

system is correct, the antenna transmission

line, with impedance -matching device or net-

work between the line and antenna feed point,

is then attached to the array and coupled to a

low -power exciter unit or transmitter. Then,

preferably, a standing -wave meter is connected

in series with the antenna transmission line

at a point relatively much more close to the

transmitter than to the antenna. However, for

best indication there should be 10 to 15 feet

of line between the transmitter and the stand-

ing -wave meter. If a standing -wave meter is

not available the standing -wave ratio may be

checked approximately by means of a neon

lamp or a short fluorescent tube if twin trans-

mission line is being used, or it may be check-

ed with a thermomilliammeter and a loop, a

neon lamp, or an r -f ammeter and a pair of

clips spaced a fixed distance for clipping onto

one wire of a two -wire open line.

If the standing -wave ratio is below 1.5 to 1

it is satisfactory to leave the installation as

it is. If the ratio is greater than this range it

will be best when twin line or coaxial line is

being used, and advisable with open -wire line,

to attempt to decrease the s.w.r.

It must be remembered that no adjustments

made at the transmitter end of the transmission

line will alter the SWR on the line. All adjust-

ments to better the SWR must be made at the

antenna end of the line and to the device which

performs the impedance transformation neces-

sary to match the characteristic impedance of

the antenna to that of the transmission line.

Before any adjustments to the matching sys-

tem are made, the resonant frequency of the

driven element must be ascertained, as ex-

plained previously. If all adjustments to cor-

rect impedance mismatch are made at this fre-

quency, the problem of reactance termination

of the transmission line is eliminated, greatly

simplifying the problem. The following steps

should be taken to adjust the impedance trans-

formation:

1. The output impedance of the matching

device should be measured. An Antenna -

scope and a grid -dip oscillator are re-

quired for this step. The Antennascope

is connected to the output terminals of

the matching device. If the driven element

is a folded dipole, the Antennascope

connects directly to the split section of

the dipole. If a gamma match or T -match

are used, the Antennascope connects to

the transmission -line end of the device.

If a Q- section is used, the Antennascope

connects to the bottom end of the sec-

tion. The grid -dip oscillator is coupled

to the input terminals of the Antenna -

scope as shown in figure 18.

2. The grid -dip oscillator is tuned to the

resonant frequency of the antenna, which

Figure 19

ALL -PIPE ROTATING MAST STRUCTURE FOR ROOF INSTALLATION

An installation suitable for a building with a pitched roof is shown at (A). At (B) is shown a

similar installation for a flat or shed roof. The arrangement as shown is strong enough to sup-

port a lightweight 3- element 28 -Mc. array and a light 3- element 50 -Mc. array above the 28 -Mc.

array on the end of a 4 -foot length of I/2-inch pipe.

The lengths of pipe shown were chosen so that when the system is in the lowered position one

can stand on a household ladder and put the beam in position atop the rotating pipe. The lengths

may safely be revised upward somewhat if the array is of a particularly lightweight design with

low wind resistance.

Just before the mast is installed It is a good idea to give the rotating pipe a good smearing of

cup grease or waterproof pump grease. To get the lip of the top of the stationary section of 1 %4-

inch pipe to project above the flange plate, it will be necessary to have a plumbing shop cut a

slightly deeper thread inside the flange plate, as well as cutting an unusually long thread on

the end of the 1% -inch pipe. It is relatively easy to waterproof this assembly through the roof

since the I 1/4-inch pipe is stationary at all times. Be sure to use pipe compound on all the joints

and then really tighten these joints with a pair of pipe wrenches.

www.americanradiohistory.com

506 Rotary Beams THE RADIO

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www.americanradiohistory.com

HANDBOOK Tuning the Array 505

driven onto a wooden dowel, as shown in

figure 178. The element may then be mounted

upon an aluminum support plate by means of

four ceramic insulators. Metal based insulators,

such as the Johnson 135 -67 are recommended,

since the all- ceramic types may break at the

mounting holes when the array is subjected

to heavy winds.

25 -8 Tuning the Array

Although satisfactory results may be ob-

tained by pre -cutting the antenna array to the

dimensions given earlier in this chapter, the

occasion might arise when it is desired to

make a check on the operation of the antenna

before calling the job complete.

The process of tuning an array may fairly

satisfactorily be divided into two more or less

distinct steps: the actual tuning of the array

for best front -to -back ratio or for maximum for-

ward gain, and the project of obtaining the

best possible impedance match between the

antenna transmission line and the feed point

of the array.

Tuning the The actual tuning of the array

Array Proper for best front -to -back ratio or

maximum forward gain may best

be accomplished with the aid of a low -power

transmitter feeding a dipole antenna (polarized

the same as the array being tuned) at least

four or five wavelengths away from the anten-

na being tuned and located at the same eleva-

tion as that of the antenna under test. A cali-

brated field- strength meter of the remote -indi-

cating type is then coupled to the feed point

of the antenna array being tuned. The trans-

missions from the portable transmitter should

be made as short as possible and the call sign

of the station making the test should be trans-

mitted at least every ten minutes.

It is, of course, possible to tune an array

with the receiver connected to it and with a

station a mile or two away making transmis-

sions on your request. But this method is more

cumbersome and is not likely to give complete

satisfaction. It is also possible to carry out

the tuning process with the transmitter con-

nected to the array and with the field- strength

meter connected to the remote dipole antenna.

In this event the indicating instrument of the

remote- indicating field - strength meter should

be visible from the position where the elements

are being tuned. However, when the array is

being tuned with the transmitter connected to

it there is always the problem of making con-

tinual adjustments to the transmitter so that a

constant amount of power will be fed to the

array under test. Also, if you use this system,

use very low power (5 or 10 watts of power is

usually sufficient) and make sure that the an-

tenna transmission line is effectively grounded

as far as d -c plate voltage is concerned. The

use of the method described in the previous

paragraph of course eliminates these problems.

One satisfactory method for tuning the array

proper, assuming that it is a system with sev-

eral parasitic elements, is to set the directors

to the dimensions given in figure 5 and then

to adjust the reflector for maximum forward

signal. Then the first director should be varied

in length until maximum forward signal is ob-

tained, and so on if additional directors are

used. Then the array may be reversed in direc-

tion and the reflector adjusted for best front -

to -back ratio. Subsequent small adjustments

may then be made in both the directors and the

reflector for best forward signal with a reason-

able ratio of front -to -back signal. The adjust-

ments in the directors and the reflector will

be found to be interdependent to a certain de-

gree, but if small adjustments are made after

the preliminary tuning process a satisfactory

set of adjustments for maximum performance

will be obtained. It is usually best to make

the end sections of the elements smaller in

diameter so that they will slip inside the larger

tubing sections. The smaller sliding sections

may be clamped inside the larger main sec-

tions.

In making the adjustments described, it is

best to have the rectifying element of the re-

mote- indicating field- strength meter directly

at the feed point of the array, with a resistor

at the feed point of the estimated value of

feed -point impedance for the array.

Matching to the

Antenna Trans-

mission Line

The problem of matching the

impedance of the antenna

transmission line to the array

is much simplified if the pro-

DRIVEN ELEMENT

ANTENNASCOK RESONATING

CAPACITOR

RIDOIR MITER

Figure 18

ADJUSTMENT OF GAMMA MATCH BY USE

OF ANTENNASCOPE AND GRID -DIP

METER

www.americanradiohistory.com

504 Rotary Beams THE RADIO

LINE OF

ELEMENTS

J - -

ALUMINUM PLATE

APPROX. S' X 12"

BOOM, MADE

OF SECTIONS

- OF STEEL TV

MAST OR OF

ALUMINUM

IRRIGATION

TUBING

- ELEMENT HELD TO PLATE WITH U- BOLTS,

(2 REO'O) OR MUFFLER CLAMPS.

SHIM JOINT WITH THIN RADIATOR

STRIPS OF ALUMINUM HOSE CLAMP

IF NECESSARY

h \ J2' CENTER S_ECJI ON

ADJUSTABLE SLIT CENTER SECTION TUBE ADJUSTABLE

TIP SAT EACH END. TIP

TYPICAL ELEMENT

Figure 16

3- ELEMENT "PLUMBER'S

DELIGHT" ANTENNA ARRAY

All-metal con figuration permits rugged,

light assembly. Joints are made with

U -bolts and metal plates for maximum

rigidity.

"Plumber's Delight" It is characteristic of

the conventional type of

multi -element parasitic

array such as discussed previously and out-

lined that the centers of all the elements are at

zero r -f potential with respect to ground. It is

therefore possible to use a metallic structure

without insulators for supporting the various

elements of the array. A typical three element

array of this type is shown in figure 16. In this

particular array, U -bolts and metal plates have

been employed to fasten the elements to the

boom. The elements are made of telescoping

sections of aluminum tubing. The tips of the

inner sections of tubing are split, and a tubing

clamp is slipped over the joint, as shown in

the drawing. Before assembly of the joint, the

mating pieces of aluminum are given a thin

coat of Penetrox -A compound. (This anti -

oxidizing paste is manufactured by l3urndy Co.,

Norwalk, Conn. and is distributed by the

General Electric Supply Co.) When the tubes

are telesooped and the clamp is tightened, an

air -tight seal is produced, reducing corrosion

to a minimum.

The boom of the parasitic array may be made

from two or three sections of steel TV mast,

or it may be made of a single section of alumi-

num irrigation pipe. This pipe is made by

Reynolds Aluminum Co., and others, and may

often be purchased via the Sears, Roebuck Co.

mail -order department. Three inch pipe may be

Construction

LINE OF

ELEMENT

O

ELEMENT CLAMP

2 PIECES

U- BOLT

LINE OF BOOM

ELEMENT HELD TO 2%4

BY 2 TV- TYPE U -BOLTS

OXEN -YORE CLAMP

BOOM CLAMP

2 PIECES

LADDER

2 X BOLTED TO LADDER BY

2 PIECES OF ANGLE IRON STOCK

Figure 17

(A) OXEN -YOKE CLAMP IS DESIGNED FOR

ALL METAL ASSEMBLY

(B) ALTERNATIVE WOODEN SUPPORTING

ARRANGEMENT

A wooden ladder may be used to support a

70 or 15 meter array.

used for the 10 and 15 meter antennas, and the

huskier four inch pipe should be used for a

20 meter beam.

Automobile muffler clamps can often be

used to affix the elements to the support

plates. Larger clamps of this type will fasten

the plates to the boom. In most cases, the

muffler clamps are untreated, and they should

be given one or two coats of rust -proof paint

to protect them from inclement weather. All

bolts, nuts, and washers used in the assembly

of the array should be of the plated variety to

reduce corrosion and rust.

An alternative assembly is to employ the

`Oxen Yoke" type of clamps, shown in figure

17. These light- weight aluminum fittings are

obtainable from the Continental Electronics

and Sound Co., Dayton, 27, Ohio, and are

available in a wide range of sizes.

If it is desired to use a split driven element

for a balanced feed system, it is necessary to

insulate the element from the supporting struc-

ture of the antenna. The element should be

severed at the center, and the two halves

www.americanradiohistory.com

HANDBOOK Bi- directional Arrays 503

Figure 15

TWO GENERAL TYPES

OF BI- DIRECTIONAL

ARRAYS

Average gain figures are giv-

en for both the flat -top beam

type of array and for the

broadside- eel¡neor array with

different numbers of elements.

STUB FLAT -TOP BEAM FOR

ROTATABLE ARRAY

OPON -WIRE LINE

RAD!AL LOAD

BEARING 45A D. FEEDERS

N12 WIRE SPACED 2'

GAIN TO B DB

O

"TWO OVER TWO OVER TWO*

TYPE OF ARRAY

GAIN TOTAL NUMBER

OF ELEMENTS

I.B De

5.0 DB

7.5 DB

9.0 DB

10.0 DB

GUY WIRES

ROPES TO !NG POSITION

A.-THRUST

BEARING

z

10

availability of certain types of constructional

materials. But in any event be sure that sound

mechanical engineering principles are used in

the design of the supporting structure. There

are few things quite as discouraging as the

picking up of pieces, repairing of the roof, etc.,

when a newly constructed rotary comes down

in the first strong wind. If the principles of

mechanical engineering are understood it is

wise to calculate the loads and torques which

will exist in the various members of the struc-

ture with the highest wind velocity which may

be expected in the locality of the installation.

If this is not possible it will usually be worth

the time and effort to look up a friend who

understands these principles.

Radiating One thing more or less standard

Elements about the construction of rotatable

antenna arrays is the use of durai

tubing for the self- supporting elements. Other

materials may be used but an alloy known as

24ST has proven over a period of time to be

quite satisfactory. Copper tubing is too heavy

for a given strength, and steel tubing, unless

copper plated, is likely to add an undesirably

large loss resistance to the array. Also, steel

tubing, even when plated, is not likely to

withstand salt atmosphere such as encountered

along the seashore for a satisfactory period

of time. Do not use a soft aluminum alloy for

the elements unless they will be quite short;

24ST is a hard alloy and is best although there

are several other alloys ending in "ST" which

will be found to be satisfactory. Do not use

an alloy ending in "SO" or "S" in a position

in the array where structural strength is im-

portant, since these letters designate a metal

which has not been heat treated for strength

and rigidity. However, these softer alloys, and

aluminum electrical conduit, may be used for

short radiating elements such as would be

used for the 50 -Mc. band or as interconnecting

conductors in a stacked array.

www.americanradiohistory.com

502 Rotary Beams T H E R A D I O

O

"LAZY H" WITH REFLECTOR

GAIN APPROX. II De

BROADSIDE HALF -WAVES

WITH REFLECTORS

GAIN APPROX. 7 De

TWO OVER TWO OVER TWO

WITH REFLECTORS

GAIN APPROX. II.5 De

Figure 14

BROADSIDE ARRAYS

WITH PARASITIC

REFLECTORS

The apparent gain of the ar-

rays illustrated will be great-

er than the values given due

to concentration of the radi-

ated signal at the lower ele-

vation ongles.

If six or more elements are used in the type

of array shown in figure 15B no matching sec-

tion will be required between the antenna trans-

mission line and the feed point of the antenna.

When only four elements are used the antenna

is the familiar "lazy H" and a quarter -wave

stub should be used for feeding from the an-

tenna transmission line to the feed point of

the antenna system.

If desired, and if mechanical considerations

permit, the gain of the arrays shown in figure

15B may be increased by 3 db by placing a

half -wave reflector behind each of the ele-

ments at a spacing of one -quarter wave. The

array then becomes essentially the same as

that shown in figure 14C and the same con-

siderations in regard to reflector spacing and

tuning will apply. However, the factor that a

bi- directional array need be rotated through an

angle of less than 180° should be considered

in this connection.

25-7 Construction of

Rotatable Arrays

A considerable amount of ingenuity may be

exercised in the construction of the supporting

structure for a rotatable array. Every person

has his own ideas as to the best method of

construction. Often the most practicable meth-

od of construction will be dictated by the

www.americanradiohistory.com

HANDBOOK Driven Arrays 501

linear system which will give approximately

the same gain as the system of figure 13A, but

which requires less boom length and greater

total element length. Figure 13C illustrates

the familiar lazy -H with driven reflectors (or

directors, depending upon the point of view)

in a combination which will show wide band-

width with a considerable amount of forward

gain and good front -to -back ratio over the en-

tire frequency coverage.

Unidirectional Stacked Three practicable

Broadside Arrays types of unidirectional

stacked broadside ar-

rays are shown in figure 14. The first type,

shown at figure 14A, is the simple "lazy H"

type of antenna with parasitic reflectors for

each element. (B) shows a simpler antenna ar-

ray with a pair of folded dipoles spaced one -

half wave vertically, operating with reflectors.

In figure 14C is shown a more complex array

with six half waves and six reflectors which

will give a very worthwhile amount of gain.

In all three of the antenna arrays shown the

spacing between the driven elements and the

reflectors has been shown as one -quarter wave-

length. This has been done to eliminate the

requirement for tuning of the reflector, as a

result of the fact that a half -wave element

spaced exactly one - quarter wave from a driven

element will make a unidirectional array when

both elements are the same length. Using this

procedure will give a gain of 3 db with the re-

flectors over the gain without the reflectors,

with only a moderate decrease in the radiation

resistance of the driven element. Actually,

the radiation resistance of a half -wave dipole

goes down from 73 ohms to 60 ohms when an

identical half -wave element is placed one -

quarter wave behind it.

A very slight increase in gain for the entire

array (about 1 db) may be obtained at the ex-

pense of lowered radiation resistance, the ne-

cessity for tuning the reflectors, and decreased

bandwidth by placing the reflectors 0.15 wave-

length behind the driven elements and making

them somewhat longer than the driven elements.

The radiation resistance of each element will

drop approximately to one -half the value ob-

tained with untunedhalf -wave reflectors spaced

one -quarter wave behind the driven elements.

Antenna arrays of the type shown in figure

14 require the use of some sort of lattice work

for the supporting structure since the arrays

occupy appreciable distance in space in all

three planes.

Feed Methods The requirements for the feed

systems for antenna arrays of

the type shown in figure 14 are less critical

than those for the close- spaced parasitic ar-

rays shown in the previous section. This is a

natural result of the fact that a larger number

of the radiating elements are directly fed with

energy, and of the fact that the effective radia-

tion resistance of each of the driven elements

of the array is much higher than the feed -point

resistance of a parasitic array. As a conse-

quence of this fact, arrays of the type shown

in figure 14 can be expected to cover a some-

what greater frequency band for a specified

value of standing -wave ratio than the parasitic

type of array.

In most cases a simple open -wire line may

be coupled to the feed point of the array with-

out any matching system. The standing -wave

ratio with such a system of feed will often be

less than 2 -to -1. However, if a more accurate

match between the antenna transmission line

and the array is desired a conventional quar-

ter -wave stub, or a quarter-wave matching

transformer of appropriate impedance, may be

used to obtain a low standing -wave ratio.

25 -6 Bi- Directional

Rotatable Arrays

The bi- directional type of array is sometimes

used on the 28 -Mc. and 50 -Mc. bands where

signals are likely to be coming from only one

general direction at a time. Hence the sacri-

fice of discrimination against signals arriving

from the opposite direction is likely to be of

little disadvantage. Figure 15 shows two gen-

eral types of bi- directional arrays. The flat-

top beam, which has been described in detail

earlier, is well adapted to installation atop a

rotating structure. When self -supporting ele-

ments are used in the flat -top beam the prob-

lem of losses due to insulators at the ends of

the elements is somewhat reduced. With a

single -section flat -top beam a gain of approx-

imately 4 db can be expected, and with two

sections a gain of approximately 6 db can be

obtained.

Another type of bi- directional array which

has seen less use than it deserves is shown

in figure 15B. This type of antenna system has

a relatively broad azimuth or horizontal beam,

being capable of receiving signals with little

diminution in strength over approximately 40 °,

but it has a quite sharp elevation pattern since

substantially all radiation is concentrated at

the lower angles of radiation if more than a

total of four elements is used in the antenna

system. Figure 15B gives the approximate gain

over a half -wave dipole at the height of the

center of the array which can be expected. Al-

so shown in this figure is a type of "rotating

mast" structure which is well suited to rota-

tion of this type of array.

www.americanradiohistory.com

500 Rotary Beams THE RADIO

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Figure 12

SHORTED STUB LENGTH AND POSITION

CHART

From the standing wave ratio and current or

voltage null position it is possible to deter-

mine the theoretically correct length and

position of a shorted stub. In actual prac-

tice a slight discrepancy usually will be

found between the theoretical and the ex-

perimentally optimized dimensions; therefore

it may be necessary to "touch up" the di-

mensions after using the above data as a

starting point.

K

110 ó

50

4o u

soJ

20 `

10 Ñ

D

N

has been decided upon for the stub, and also

to determine the SWR.

Stub adjustment becomes more critical as

the SWR increases, and under conditions of

high SWR the current and voltage nulls are

more sharply defined than the current and volt-

age maxima, or loops. Therefore, it is best to

locate either a current null or voltage null, de-

pending upon whether a current indicating de-

vice or a voltage indicating device is used to

check the standing wave pattern.

The SWR is determined by means of a "di-

rectional coupler," or by noting the ratio of

Ema. to Emin or La. to Imin as read on an

indicating device.

It is assumed that the characteristic imped-

ance of the section of line used as a stub is

the same as that of the transmission line prop-

er. It is preferable to have the stub section

identical to the line physically as well as

electrically.

25 -5 Unidirectional

Driven Arrays

Three types of unidirectional driven arrays

are illustrated in figure 13. The array shown

in figure 13A is an end -fire system which may

DIRECTIONAL

1 + 4

OGAIN ABOUT e De FEED LINE

DIRECTIONAL

GAIN ABOUT e DB

FEED LINE

d4 DIRECTIONAL

GAIN ABOUT 10 De

FEED LINE

Figure 13

UNIDIRECTIONAL ALL -DRIVEN ARRAYS

A unidirectional all -driven end -fire array is

shown at (A). (B) shows an array with two

half waves in phase with driven reflectors.

A Lazy -H array with driven reflectors is

shown at (C). Note that the directivity is

through the elements with the greatest total

feed -line length in arrays such as shown at

(B) and (C).

be used in place of a parasitic array of similar

dimensions when greater frequency coverage

than is available with the yagi type is desired.

Figure 13B is a combination end -fire and co-

www.americanradiohistory.com

HANDBOOK The Gamma Match 499

Figure 10

THE GAMMA MATCHING SYSTEM

See text for details of resonating capacitor

-FLAT' LINE RESONANT

SWR 1.O SECTION

TO TRANSMITTER SIMPLE OR CONVEX

MATCHING STUB

Figure 11

IMPEDANCE MATCHING WITH A CLOSED

STUB ON A TWO WIRE TRANSMISSION

LINE

piing rings are 10 inches in diameter and are

usually constructed of % -inch copper tubing

supported one from the rotating structure and

one from the fixed structure by means of stand-

off insulators. The capacitor C in figure 9D is

adjusted, after the antenna has been tuned, for

minimum standing -wave ratio on the antenna

transmission line. The dimensions shown will

allow operation with either 14 -Mc. or 28 -Mc.

elements, with appropriate adjustment of the

capacitor C. The rings must of course be paral-

lel and must lie in a plane normal to the axis

of rotation of the rotating structure.

The Gamma Match The use of coaxial cable

to feed the driven element

of a yagi array is becoming increasingly popu-

lar. One reason for this increased popularity

lies in the fact that the TVI- reduction problem

is simplified when coaxial feed line is used

from the transmitter to the antenna system.

Radiation from the feed line is minimized when

coaxial cable is used, since the outer conduc-

tor of the line may be grounded at several

points throughout its length and since the in-

tense field is entirely confined within the out-

er conductor of the coaxial cable. Other ad-

vantages of coaxial cable as the antenna feed

line lie in the fact that coaxial cable may be

run within the structure of a building without

danger, or the cable may be run underground

without disturbing its operation. Also, trans-

mitting -type low -pass filters for 52 ohm imped-

ance are more widely available and are less

expensive than equivalent filters for two -wire

line. The gamma -match is illustrated in figure 10,

and may be looked upon as one -half of a T-

match. One resonating capacitor is used,

placed in series with the gamma rod. The ca-

pacitor should have a capacity of 7 µµfd. per

meter of wavelength. For 15-meter operation

the capacitor should have a maximum capacity

of 105 µµfd. The length of the gamma rod deter-

mines the impedance transformation between

the transmission line and the driven element

of the array, and the gamma capacitor tunes

out the inductance of the gamma rod. By ad-

justment of the length of the gamma rod, and

the setting of the gamma capacitor, the SWR

on the coaxial line may be brought to a very

low value at the chosen operating frequency.

The use of an Antennascope, described in the

Test Equipment chapter is recommended for

precise adjustment of the gamma match.

The Matching Stub If an open-wire line is

used to feed a low imped-

ance radiator, a section of the transmission

line may be employed as a matching stub as

shown in figure 11. The matching stub can

transform any complex impedance to the char-

acteristic impedance of the transmission line.

While it is possible to obtain a perfect match

and good performance with either an open stub

or a shorted one by observing appropriate di-

mensions, a shorted stub is much more readily

adjusted. Therefore, the following discussion

will be confined to the problem of using a

closed stub to match a low impedance load to

a high impedance transmission line.

If the transmission line is so elevated that

adjustment of a "fundamental" shorted stub

cannot be accomplished easily from the ground,

then the stub length may be increased by ex-

actly one or two electrical half wavelengths,

without appreciably affecting its operation.

While the correct position of the shorting

bar and the point of attachment of the stub to

the line can be determined entirely by experi-

mental methods, the fact that the two adjust-

ments are interdependent or interlocking makes

such a cut- and -try procedure a tedious one.

Much time can be saved by determining the ap-

proximate adjustments required by reference to

a chart such as figure 12 and using them as a

starter. Usually only a slight "touching up"

will produce a perfect match and flat line.

In order to utilize figure 12, it is first neces-

sary to locate accurately a voltage node or

current node on the line in the vicinity that

www.americanradiohistory.com

498 R o t a r y Beams T H E R A D I O

52 a. COAXIAL CABLE

75 A TWIN LINE

450 -000 A. LINE

OA DIRECT FEED WITH

COAXIAL CABLE

0 QUARTER -WAVE

TRANSFORMER FEED

MATCHING SYSTEM

20 MC. - 4 TURNS 2" DIA., 2" LONG

ANT. 1 TURN EACH SIDE

14 MC. - 0 TURNS 2" DIA., 2" LONG

ANT. TAPPED 2 TURNS EACH SIDE

COIL SPACED COILS 10-

APPROX. 0.5" DIAMETER

C

450 -0000. LINE

0 ROTARY LINK

COUPLING

1 TURN LINKS ARE PARALLEL

C IS 200 LUPD VARIABLE

Figure 9

ALTERNATE FEED

METHODS WHERE THE

DRIVEN ELEMENT MAY

BE BROKEN IN THE

CENTER

These capacitors should be tuned for minimum

SWR on the transmission line. The adjustment

of these capacitors should be made at the same

time the correct setting of the T -match rods is

made as the two adjustments tend to be inter-

locking. The use of the standing wave meter

(described in Test Equipment chapter) is

recommended for making these adjustments to

the T- match.

Feed Systems Using Four methods of exciting

a Driven Element the driven element of a

with Center Feed parasitic array are shown

in figure 9. The system

shown at (A) has proven to be quite satisfac-

tory in the case of an antenna -reflector two -

element array or in the case of a three -element

array with 0.2 to 0.25 wavelength spacing be-

tween the elements of the antenna system. The

feed -point impedance of the center of the driven

element is close enough to the characteristic

impedance of the 52 -ohm coaxial cable so that

the standing -wave ratio on the 52 -ohm coaxial

cable is less than 2-to-1.(B) shows an arrange-

ment for feeding an array with a broken driven

element from an open -wire line with the aid of

a quarter -wave matching transformer. With 465 -

ohm line from the transmitter to the antenna

this system will give a close match to a 12-

ohm impedance at the center of the driven ele-

ment. (C) shows an arrangement which uses an

untuned transformer with lumped inductance

for matching the transmission line to the cen-

ter impedance of the driven element.

Rotary Link In many cases it is desirable to

Coupling be able to allow the antenna ar-

ray to rotate continuously without

regard to snarling of the feed line. If this is to

be done some sort of slip rings or rotary joint

must be made in the feed line. One relatively

simple method of allowing unrestrained rotation

of the antenna is to use the method of rotary

link coupling shown in figure 9D. The two cou-

www.americanradiohistory.com

HANDBOOK Matching Systems 497

Figure 8

AVERAGE DIMENSIONS

FOR THE DELTA AND

"T" MATCH

L

frT-u7r. L L-.j

OA DELTA MATCH

L

1416 L

DIMENSIONS SHOWN GIVE

APPROX. MATCH TO SOOD

AIN - SPACED LINE

C -T MATCH

L

zoo n Dn 300 n

TWIN LINE

DI3Dz

In many cases it will be desired to use the

folded -element or yoke matching system with

different sizes of conductors or different spac-

ings than those shown in figure 7. Note, then,

that the impedance transformation ratio of

these types of matching systems is dependent

both upon the ratio of conductor diameters and

upon their spacing. The following equation

has been given by Roberts (11CA Review, June,

1947) for the determination of the impedance

transformation when using different diameters

in the two sections of a folded element:

Z,

Transformation ratio = (1 + - \ Z,

In this equation Z, is the characteristic im-

pedance of a line made up of the smaller of

the two conductor diameters spaced the center -

to- center distance of the two conductors in the

antenna, and Z, is the characteristic imped-

ance of a line made up of two conductors the

size of the larger of the two. This assumes

that the feed line will be connected in series

with the smaller of the two conductors so that

an impedance step up of greater than four will

be obtained. If an impedance step up of less

than four is desired, the feed line is connected

in series with the larger of the two conductors

and Z, in the above equation becomes the im-

pedance of a hypothetical line made up of the

larger of the two conductors and Z2 is made

up of the smaller. The folded v -h -f unipole is

an example where the transmission line is con-

nected in series with the 1 a r g e r of the two

conductors.

z

The conventional 3 -wire match to give an

impedance 'multiplication of 9 and the 5 -wire

match to give a ratio of approximately 25 are

shown in figures 7C and 7D. The 4 -wire match,

not shown, will give an impedance transforma-

tion ratio of approximately 16.

The Delta Match The Delta match and the

and T -Match T -match are shown in figure

8. The delta match has been

largely superseded by the newer T- match, how-

ever both these systems can be adjusted to

give a low value of SWR on 50 to 600 -ohm bal-

anced transmission lines. In the case of the

systems shown it will be necessary to make

adjustments in the tapping distance along the

driven radiator until minimum standing waves

on the antenna transmission line are obtained.

Since it is sometimes impracticable to elim-

inate completely the standing waves from the

antenna transmission line when using these

matching systems, it is common practice to

cut the feed line, after standing waves have

been reduced to a minimum, to a length which

will give satisfactory loading of the transmitter

over the desired frequency range of operation.

The inherent reactance of the T -match is

tuned out by the use of two identical resonat-

ing capacitors in series with each leg of the

T -rod. These capacitors should each have a

maximum capacity of 8 littfd. per meter of wave-

length. Thus for 20 meters, each capacitor

should have a maximum capacity of at least

160 µµEd. For power up to a kilowatt, 1000

volt spacing of the capacitors is adequate.

www.americanradiohistory.com

496 Rotary B e a m s T H E R A D I O

RADIATION Di FOR Di-Da R.4

FOR Di- 1-

2 Dzo. Rs'8.9

5 1.s

FOR D11

D2.z5iegAre=10.5

S1.5

roa DI1

D22.2s- 16

5 1 RAD.

D FOR D = 1-

5 3- =11

RE

*12 WI

roua D= 1-

S 2- - 14

*12 WIRE RAD.

FOR D= 1-

5 . 18

w 12 WI R1E

5-

FOLDED -ELEMENT

MATCH

s',RrecD

roa D. 1-

S= 1- =24

a0 WIRE

POR D. I

S 1 32

12 WIRE RAD.

5 -WIRE MATCH 31M12. APPROX. 25

R RAD.

Figure 7

DATA FOR

FOLDED -ELEMENT

MATCHING SYSTEMS

In all normal applications of

the data given the main ele-

ment as shown is the driven

element of a multi -element

parasitic array. Directors and

reflectors have not been

shown for the sake of clarity.

small strips of polystyrene which have been

drilled for both the main element and the small

wire and threaded on the main element.

The Folded -Element The calculation of the

Match Calculations operating conditions of

the folded - element

matching system and the yoke match, as shown

in figures 7A and 7B is relatively simple. A

selected group of operating. conditions has

been shown on the drawing of figure 7. In ap-

plying the system it is only necessary to mul-

tiply the ratio of feed to radiation resistance

(given in the figures to the right of the sug-

gested operating dimensions in figure 7) by

the radiation resistance of the antenna system

to obtain the impedance of the cable to be

used in feeding the array. Approximate values

of radiation resistance for a number of com-

monly used parasitic -element arrays are given

in figure 5.

As an example, suppose a 3- element array

with 0.15D -0.15R spacing between elements is

to be fed by m e an s of a 465 -ohm line con-

structed of no. 12 wire spaced 2 inches. The

approximate radiation resistance of such an

antenna array will be 20 ohms. Hence we need

a ratio of impedance step up of 23 to obtain

a match between the characteristic impedance

of the transmission line and the radiation re-

sistance of the driven element of the antenna

array. Inspection of the ratios given in figure

7 shows that the fourth set of dimensions

given under figure 7B will give a 24 -to -1 step

up, which is sufficiently close. So it is merely

necessary to use a 1 -inch diameter driven ele-

ment with a no.8 wire spaced on 1 inch centers

(% inch below the outside wall of the 1 -inch

tubing) below the 1 -inch element. The no. 8

wire is broken and a 2 -inch insulator placed

in the center. The feed line then carries from

this insulator down to the transmitter. The

center insulator should be supported rigidly

from the 1 -inch tube so that the spacing be-

tween the piece of tubing and the no. 8 wire

will be accurately maintained.

www.americanradiohistory.com

HANDBOOK Stacked Yagi Arrays 495

H-0.2 A -wi.-13.2 A --4

OGAIN ABOUT 12 DB

WITH 2 SECTIONS

DIRECTIONAL

I

WITH 3 SECTIONS

H 0 2 A -+- O. 2 A--+- O. 2 A-.F O 2 7.-.1

501

F Mc

DIRECTIONAL

A

FEEDER LINE

O AIN ABOUT 17 DR

Figure 6

STACKED YAGI ARRAYS

It is possible to attain a relatively large amount of gain over a limited bandwidth with stacked

yogi arrays. The two -section array at (A) will give a gain of about 12 db, while adding a third

section will bring the gain up to about 15 db. Adding two additional parasitic directors to each

section, as at (C) will bring the gain up to about 17 db.

higher where the additional section of tubing Mc. and 14 -Mc. bands since it is only neces-

may be supported below the main radiator ele- sary to suspend a wire below the driven ele-

ment without undue difficulty. The yoke -match ment proper. The wire may be spaced below

is more satisfactory mechanically on the 28- the self -supporting element by means of several

www.americanradiohistory.com

494 Rotary Beams THE RADIO

TYPE DRIVEN ELEMENT

LENGTH REFLECTOR

LENGTH IST DIRECTOR

LENGTH 2ND DIRECTOR

LENGTH 390 DIRECTOR

LENGTH SPACING BET-

WEENELEMENTS APPROX. GAIN

DO APPRO %.RADIATION

RESISTANCE (A I

3-ELEMENT 473

F(MC) SOI

F(MC) 445

.F (MC) .IS -IS 7.5 20

- -

3- ELEMENT 741?-41C) F) - .2S -.2S 9.S 35

4-ELEMENT 01(1°Z) S(1SO .2 -.2 -.2 9.S 20

.1C

5-ELEMENT F(MC) F4(L9C) .2 -.2 -.2-.2 Io.o IS

Figure 5

DESIGN CHART FOR PARASITIC ARRAYS (DIMENSIONS GIVEN IN FEET)

More Than A small amount of additional

Three Elements gain may be obtained through

use of more than two parasitic

elements, at the expense of reduced feed -point

impedance and lessened bandwidth. One addi-

tional director will add about 1 db, and a sec-

ond additional director (making a total of five

elements including the driven element) will

add s l i g ht l y less than one db more. In the

v -h -f range, where the additional elements may

be added without much difficulty, and where

required bandwidths are small, the use of more

than two parasitic elements is quite practic-

able.

Stacking of Parasitic arrays (yagis) may

Yogi Arrays be stacked to provide addition-

al gain in the same manner that

dipoles may be stacked. Thus if an array of

six dipoles would give a gain of 10 db. the

substitution of yagi arrays for each of the di-

poles would add the gain of one yagi array to

the gain obtained with the dipoles. However,

the yagi arrays must be more widely spaced

than the dipoles to obtain this theoretical im-

provement. As an example, if six 5- element

yagi arrays having a gain of about 10 db were

substituted for the dipoles, with appropriate

increase in the spacing between the arrays,

the gain of the whole system would approach

the sum of the two gains, or 20 db. A group of

arrays of yagi antennas, with recommended

spacing and approximate gains, are illus-

trated in figure 6.

25 -4 Feed Systems for

Parasitic (Yogi) Arrays

The table of figure 5 gives, in addition to

other information, the approximate radiation

resistance referred to the center of the driven

element of multi- element parasitic arrays. It is

obvious, from these low values of radiation

resistance, that especial care must be taken

in materials used and in the construction of

the elements of the array to insure that ohmic

losses in the conductors will not be an appre-

ciable percentage of the radiation resistance.

It is also obvious that some method of iglped-

ance transformation must be used in many

cases to match the low radiation resistance

of these antenna arrays to the normal range of

characteristic impedance used for antenna

transmission lines.

A group of possible methods of impedance

matching is shown in figures 7, 8, 9 and 10.

All these methods have been used but certain

of them offer advantages over some of the

other methods. Generally speaking it is not

mechanically desirable to break the center of

the driven element of an array for feeding the

system. Breaking the driven element rules out

the practicability of building an all -metal or

"plumber's delight" type of array, and im-

poses mechanical limitations with any type of

construction. However, when continuous rota-

tion is desired, an arrangement such as shown

in figure 9D utilizing a broken driven element

with a rotatable transformer for coupling from

the antenna transmission line to the driven

element has proven to be quite satisfactory.

In fact the method shown in figure 9D is prob-

ably the most practicable method of feeding

the driven element when continuous rotation

of the antenna array is required.

The feed systems shown in figure 7 will,

under normal conditions, show the lowest loss-

es of any type of feed system since the cur-

rents flowing in the matching network are the

lowest of all the systems commonly used. The

"Folded Element" match shown in figure 7A

and the "Yoke" match shown in figure 7B are

the most satisfactory electrically of all stand-

ard feed methods. However, both methods re-

quire the extension of an additional conductor

out to the end of the driven element as a por-

tion of the matching system. The folded -ele-

ment match is best on the 50 -Mc. band and

www.americanradiohistory.com

HANDBOOK Parasitic Arrays 493

0.2 wavelength between elements becomes

possible. Four -element arrays are quite com-

mon on the 28 -Mc. and 50 -Mc. bands, and five

elements are sometimes used for increased

gain and discrimination. As the number of ele-

ments is increased the gain and front -to -back

ratio increases but the radiation resistance de-

creases and the bandwidth or frequency range

over which the antenna will operate without

reduction in effectiveness is decreased.

Material for While the elements may consist

Elements of wire supported on a wood

framework, self -supporting ele-

ments of tubing are much to be preferred. The

latter type array is easier to construct, looks

better, is no more expensive, and avoids the

problem of getting sufficiently good insulation

at the ends of the elements. The voltages

reach such high values towards the ends of

the elements that losses will be excessive,

unless the insulation is excellent.

The elements may be fabricated of thin -

walled steel conduit, or hard drawn thin -walled

copper tubing, but durai tubing is much better.

Or, if you prefer, you may purchase tapered

copper -plated steel tubing elements designed

especially for the purpose. Kits are available

complete with rotating mechanism and direction

indicator, for those who desire to purchase

the whole system ready to put up.

Element Spacing The optimum spacing for a

two -element array is, as has

been mentioned be fore, approximately 0.11

wavelength for a director and 0.13 wavelength

for a reflector. However, when both a director

and a reflector are combined with the driven

element to make up a three -element array the

optimum spacing is established by the band-

width which the antenna will be required to

cover. Wide spacing (of the order of 0.25 wave-

length between elements) will result in greater

bandwidth for a specified maximum standing -

wave ratio on the antenna transmission line.

Smaller spacings may be used when boom

length is an important consideration, but for a

specified standing -wave ratio and forward gain

the frequency coverage will be smaller. Thus

the Q of the antenna system will be increased

as the spacing between the elements is de-

creased, resulting in smaller frequency cover-

age, and at the same time the feed -point im-

pedance of the driven element will be de-

creased.

For broad -band coverage, such as the range

from 26.96 to 29.7 Mc. or from 50 to 54 Mc.,

0.2 wavelength spacing from the driven ele-

ment to each of the parasitic elements is rec-

ommended. For narrower bandwidth, such as

would be adequate for the 14.0 to 14.4 Mc.

band or the 144 to 148 Mc. band, the radiator

to parasitic element spacing may be reduced

to 0.12 wavelength, while still maintaining

adequate array bandwidth for the amateur band

in question.

Length of the Experience has shown that

Parasitic Elements it is practical to cut the

prarsitic elements of a

three -element parasitic array to a predetermined

length before the installation of such an an-

tenna. A pre -tuned antenna such as this will

give good signal gain, adequate front -to -back

ratio, and good bandwidth factor. By carefully

tuning the array after it is in position the gain

may be increased by a fraction of a db, and

the front -to -back ratio by several db. However

the slight improvement in performance is us-

ually not worth the effort expended in tuning

time. The closer the lengths of the parasitic ele-

ments are to the resonant length of the driven

element, the lower will be the feed -point resist-

ance of the driven element, and the smaller

will be the bandwidth of the array. Hence, for

wide frequency coverage the director should

be considerably shorter, and the reflector con-

siderably longer than the driven element. For

example, the director should still be less than

a resonant half wave at the upper frequency

limit of the range wherein the antenna is to be

operated, and the reflector should still be long

enough to act as a reflector at the lower fre-

quency limit. Another way of stating the same

thing is to say, in the case of an array to cover

a wide frequency range such as the amateur

range from 26.96 to 29.7 Mc. or the width of a

low -band TV channel, that the director should

be cut for the upper end of the band and

the reflector for the lower end of the band. In

the case of the 26.96 to 29.7 Mc. range this

means that the director should be about 8 per

cent shorter than the driven element and the

reflector should be about 8 per cent longer.

Such an antenna will show a relatively con-

stant gain of about 6 db over its range of cov-

erage, and the pattern will not reverse at any

point in the range.

Where the frequency range to be covered is

somewhat less, such as a high -band TV chan-

nel, the 14.0 to 14.4 Mc. amateur band, or the

lower half of the amateur 28 -Mc. phone band,

the reflector should be about 5 per cent longer

than the driven element, and the director about

5 per cent shorter. Such an antenna will per-

form well over its rated frequency band, will

not reverse its pattern over this band, and will

show a signal gain of 7 to 8 db. See figure 5

for design figures for 3-element arrays.

www.americanradiohistory.com

492 Rotary Beams THE RADIO

wavelength may be employed for greater front -

to -back ratios, but the radiation resistance of

the array becomes quite low, the bandwidth

of the array becomes very narrow, and the tun-

ing becomes quite critical. Thus the Q of the

antenna system will be increased as the spac-

ing between the elements is decreased, and

smaller optimum f r e q u e n c y coverage will

result.

Element Lengths When the parasitic element

of a two -element array is

used as a director, the following formulas may

be used to determine the lengths of the driven

element and the parasitic director, assuming

an element diameter -to- length ratio of 200 to

400:

476

Driven element length (feet) = - Fmc.

450

Director length (feet) = - FAlc.

Element spacing (feet) = l

11c.

Figure 4

FIVE ELEMENT 28 MC BEAM

ANTENNA AT W6SAI

Antenna boom is made of twenty foot

length of Sears, Roebuck Co. three -

inch aluminum irrigation pipe. Spacing

between elements is five feet. Ele-

ments are made of twelve foot lengths

of 7/8 -inch aluminum tubing, with ex-

tension tips made of 3/4 -inch tubing.

Gamma matching device, element

clamps, and 'Oxen Yoke" element -to-

boom clamps are made by Continental

Electronics 8 Sound Co., Dayton 27,

Ohio. Beam dimensions are taken from

figure 5.

z o

0 I o.IS oz 0.2S

ELEMENT SPACING (X)

(PARASITIC ELEMENT TUNED FOR MAXIMUM GAIN)

Figure 3

FRONT -TO -BACK RATIO AS A FUNCTION

OF ELEMENT SPACING FOR A TWO -ELE-

MENT PARASITIC ARRAY

The effective bandwidth taken between the

1.5/1 standing wave points of an array cut to

the above dimensions is about 2.5% of the

operating frequency. This means that an array

pre -cut to a frequency of 14,150 kilocycles

would have a bandwidth of 350 kilocycles (plus

or minus 175 kilocycles of the center frequen-

cy), and therefore would be effective over the

whole 20 meter band. In like fashion, a 15

meter array should be pre -cut to 21,200 kilo-

cycles.

A beam designed for use on the 10 -meter

band would have an effective bandwidth of

some 700 kilocycles. Since the 10 -meter band

is 1700 kilocycles in width, the array should

either be cut to 28,500 kilocycles for opera-

tion in the low frequency portion of the band,

or to 29,200 kilocycles for operation in the

high frequency portion of the band. Operation

of the antenna outside the effective bandwidth

will increase the SWR on the transmission

line, and noticeably degrade both the gain and

front -to -back ratio performance. The height

above ground also influences the F/B ratio.

25 -3 The Three -Element Array

The three -element array using a director,

driven element, and reflector will exhibit as

much as 30 db front -to -back ratio and 20 db

front -to -side ratio for low -angle radiation. The

theoretical gain is about 9 db over a dipole in

free space. In actual practice, the array will

often show 7 to 10 db apparent gain over a

horizontal dipole placed the same height above

ground (at 28 and 14 Mc.).

The use of more than three elements is de-

sirable when the length of the supporting struc-

ture is such that spacings of approximately

www.americanradiohistory.com

Parasitic Arrays 491

0 1 0.15 0 2

ELEMENT SPACING (X)

Figure 1

GAIN VS ELEMENT SPACING FOR A TWO -

ELEMENT CLOSE- SPACED PARASITIC

BEAM ANTENNA WITH PARASITIC ELE-

MENT OPERATING AS A DIRECTOR OR

REFLECTOR

ais

SO

45

40 k

t

35 r ..

30 .

zs

20 .....

»

Io

s

0 1 0.15 0 2

ELEMENT SPACING (X)

Figure 2

RADIATION RESISTANCE AS A FUNCTION

OF ELEMENT SPACING FOR A TWO -ELE-

MENT PARASITIC ARRAY

Such an antenna is capable of a signal gain

of 5 db over a dipole, with a front -to -back ratio

of 7 db to 15 db, depending upon the adjust-

ment of the parasitic element. The parasitic

element may be used either as a director or

as a reflector.

The optimum spacing for a reflector in a

two -element array is approximately 0.13 wave-

length and with optimum adjustment of the

length of the reflector a gain of approximately

5 db will be obtained, with a feed -point resist-

ance of about 25 ohms.

If the parasitic element is to be used as a

director the optimum spacing between it and

the driven element is 0.11 wavelength. The

gain will theoretically be slightly greater than

with the optimum adjustment for a reflector

(about 5.5 db) and the radiation resistance

will be in the vicinity of 17 ohms.

The general characteristics of a two -element

parasitic array may be seen in figures 1, 2 and

3. The gain characteristics of a two -element

array when the parasitic element is used as a

director or as a reflector are shown. It can be

seen that the director provides a maximum of

5.3 db gain at a spacing of slightly greater

than 0.1 wavelength from the antenna. In the

interests of greatest power gain and size con-

servation, therefore, the choice of a parasitic

director would be wiser than the choice of a

parasitic reflector, although the gain differ-

ence between the two is small.

Figure 2 shows the relationship between

the element spacing and the radiation resist-

ance for the two element parasitic array for

both the reflector and the director case. Since

the optimum antenna -director spacing for maxi-

mum gain results in an antenna radiation re-

sistance of about 17 ohms, and the optimum

antenna- reflector spacing for maximum gain

results in an antenna radiation resistance of

about 25 ohms, it may be of advantage in some

instances to choose the antenna with the high-

er radiation resistance, assuming other fac-

tors to be equal.

Figure 3 shows the front -to -back ratio for

the two element parasitic array for both the

reflector and director cases. To produce these

curves, the elements were tuned for maximum

gain of the array. Better front -to -back ratios

may be obtained at the expense of array gain,

if desired, but the general shape of the curves

remains the same. It can be readily observed

that operation of the parasitic element as a

reflector produces relatively poor front -to-

back ratios except when the element spacing

is greater than 0.15 wavelength. However, at

this element spacing, the gain of the array be-

gins to suffer.

Since a radiation resistance of 17 ohms is

not unduly hard to match, it can be argued that

the best all- around performance may be ob-

tained from a two element parasitic beam em-

ploying 0.11 element spacing, with the para-

sitic element tuned to operate as a director.

This antenna will provide a forward gain of

5.3 db, with a front -to -back ratio of 10 db, or

slightly greater. Closer spacing than 0.11

www.americanradiohistory.com

CHAPTER TWENTY -FIVE

Rotary Beams

The rotatable antenna array has become al-

most standard equipment for operation on the

28 -Mc. and 50 -Mc. bands and is commonly used

on the 14 -Mc. and 21 -Mc. bands and on those

frequencies above 144 Mc. The rotatable array

offers many advantages for both military and

amateur use. The directivity of the antenna

types commonly employed, particularly the

unidirectional arrays, offers a worthwhile re-

duction in interference from undesired direc-

tions. Also, the increase in the ratio of low -

angle radiation plus the theoretical gain of

such arrays results in a relatively large in-

crease in both the transmitted signal and the

signal intensity from a station being received.

A significant advantage of a rotatable an-

tenna array in the case of the normal station is

that a relatively small amount of space is re-

quired for erection of the antenna system. In

fact, one of the best types of installation uses

a single telephone pole with the rotating struc-

ture holding the antenna mounted atop the pole.

To obtain results in all azimuth directions

from fixed arrays comparable to the gain and

directivity of a single rotatable three- element

parasitic beam would require several acres of

surface.

There are two normal configurations of radi-

ating elements which, when horizontally polar-

ized, will contribute to obtaining a low angle

of radiation. These configurations are the end -

fire array and the broadside array. The con-

ventional three- or four -element rotary beam

may properly be called a unidirectional para-

sitic end -fire array, and is actually a type of

yagi array. The flat -top beam is a type of bi-

directional end -lire array. The broadside type

of array is also quite effective in obtaining

low -angle radiation, and although widely used

in FM and TV broadcasting has seen little use

by amateur stations in rotatable arrays.

25 -1 Unidirectional

Parasitic End -Fire Arrays

(Yogi Type)

If a single parasitic element is placed on

one side of a driven dipole at a distance of

from 0.1 to 0.25 wavelength the parasitic ele-

ment can be tuned to make the array substan-

tially unidirectional.

This simple array is termed a two element

parasitic beam.

25 -2 The Two Element Beam

The two element parasitic beam provides

the greatest amount of gain per unit size of

any array commonly used by radio amateurs.

490

www.americanradiohistory.com

HANDBOOK VHF Parasitic Arrays 489

DRIVEN ELEMENT

ELEMENT DIMENSIONS 2 METER BAND

ELEMENT

(DIA M. I /6 -)

LENGTH

144 MC. 145 MC. 146 MC. 147 MC.

REFLECTOR 41^ 4ot 4°4' 404--

DIRECTORS 36 ^ 367 36a

DRIVEN ELEMENT

36.5

e WIRE FOR 300 (1

MATCH.

*10 WIRE FOR 4500

MATCH

INSULATING

PLATE BLATTEN

TUBING

AT ENDS.

DRILL HOLES THROUGH BOOM AND

PASS ELEMENTS THROUGH HOLES

BOOM LENGTH = 24 . DIAM. If

GAIN= 16.1 DB

SPACING

FROM

DIPOLE

19

D1= 7

02= 14.5

D3= z2^

D4= 36

DS= 70.

De= loz

D7= 134

De= lee.

D9= 19e

D10=230"

D11=242"

Figure 19

DESIGN DIMENSIONS FOR A 2-METER "LONG YAGI" ANTENNA

On the other hand, if a Yagi array of the same

approximate size and weight as another an-

tenna type is built, it will provide a higher

order of power gain and directivity than that

of the other antenna.

The power gain of a Yagi antenna increases

directly with the physical length of the array.

The maximum practical length is entirely a

mechanical problem of physically supporting

the long series of director elements, although

when the array exceeds a few wavelengths in

length the element lengths, spacings, and

Q's become more and more critical. The ef-

fectiveness of the array depends upon a proper

combination of the mutual coupling loops

between adjacent directors and between the

first director and the driven element.

Practically all work on Yagi antennas with

more than three or four elements has been on

an experimental, cut- and -try basis. Figure 19

provides dimensions for a typical Long Yagi

antenna for the 2 -meter VHF band. Note that

all directors have the same physical length.

If the long Yagi is designed so that the di-

rectors gradually decrease in length as they

progress from the dipole bandwidth will be

increased, and both side lobes and forward

gain will be reduced. One advantage gained

from staggered director length is that the

array can be shortened and lengthened by

adding or taking away directors without the

need for retuning the remaining group of para-

sitic elements. When all directors are the

same length, they must be all shortened en

masse as the array is lengthened, and vice -

versa when the array is shortened.

A full discussion of Long Yagi antennas,

including complete design and construction

information may be had in the VHF Handbook,

available through Radio Publications, Inc.,

Wilton., Conn.

www.americanradiohistory.com

488 V -H -F and U -H -F Antenn as THE RADIO

WOODBLOCK

Figure 16

THE MOUNTING BLOCK FOR EACH SET

OF ELEMENTS

These tubes are welded onto the center tube

of each group of three horizontal bracing tubes,

and are so located to support the horizontal di-

pole at its exact center. The dipoles are at-

tached to the supporting rods by means of

small phenolic insulating blocks, as shown in

figure 16. The radiators are therefore insulated

from the screen reflector. The inner tips of

the radiators are held by small polystyrene

blocks for rigidity, and are cross connected to

each other by a transposed length of TV -type

400 ohm open wire line. The entire array is

fed at the point A -A, illustrated in figure 15.

The matching system for the beam is mounted

behind the reflector screen, and is shown in

figure 17. A quarter -wave transformer (B) drops

the relatively high impedance of the antenna

array to a suitable value for the low imped-

ance balun (D). An adjustable matching stub

(C) and two variable capacitors (C, and C2)

are employed for impedance matching. The

two variable capacitors are mounted in a

Figure 18

HORIZONTAL RADIATION PATTERN OF

THE PE1PL ARRAY. THE FRONT -TO-

BACK RATIO IS ABOUT 28 db IN AMPLI-

TUDE, AND THE FORWARD GAIN AP-

PROXIMATELY 15 db.

-BRASS TUBING

A

STUB

C

APPROX

r

TRANSFORMER - B

SHORTING BAR -C

I C1&C2 =SO LUF

WATERTIGHT

COMPARTMENT

BALUN- D

-- SNORTING BAR - D

COPPER TUBING

72 R COAX CABLE

Figure 17

THE MATCHING UNIT IN DETAIL FOR

THE PE1PL BEAM DESIGN, WHICH AL-

LOWS THE USE OF 72 -OHM COAX

watertight box, with the balun and matching

stubs entering the bottom and top of the box,

respectively.

The matching procedure is carried out by

the use of a standing wave meter (SWR bridge).

A few watts of power are fed to the array

through the SWR meter, and the setting of the

shorting stub on C and the setting of the two

variable capacitors are adjusted for lowest

SüR at the chosen operating frequency. The

capacity settings of the two variable capaci-

tors should be equal. The final adjustment is

to set the shorting stub of the balun (D) to re-

move any residual reactance that might appear

on the transmission line. üith proper adjust-

ment, the VSWR of the array may be held to

less than 1.5 to 1 over a 2 megacycle range

of the 2 -meter band.

The horizontal radiation pattern of this array

is shown in figure 18.

Long Yogi

Antennas For a given power gain, the

Yagi antenna can be built

lighter, more compact, and

with less wind resistance than any other type.

www.americanradiohistory.com

HANDBOOK VHF Parasitic Arrays 487

The ends of the folded dipoles are made in

the following manner: Drive a length of dowel

into the short connecting lengths of aluminum

tubing. Then drill down the center of the dowel

with a clearance hole for the connecting screw.

Then shape the ends of the connecting pieces

to fit the sides of the element ends. After as-

sembly the junctions may be dressed with a

file and sandpaper until a smooth fit is ob-

tained.

The mast used for supporting the array is a

30 -foot spliced 2 by 2. A large discarded ball

bearing is used as the radial load bearing and

guy -wire termination. Enough of the upper -mast

corners were removed with a draw -knife to per-

mit sliding the ball bearing down about 9 feet

from the top of the mast. The bearing then was

encircled by an assembly of three pieces of

dural ribbon to form a clamp, with ears for

tightening screws and attachment of the guy

wires. The bearing then was greased and cov-

ered with a piece of auto inner tube to serve

as protection from the weather. Another junk -

box bearing was used at the bottom of the mast

as a thrust bearing.

The main boom s were made from 34-inch

aluminum electrical conduit. Any size of small

tubing will serve for making the elements.

Note that the main boom is mounted at the bal-

ance center and not necessarily at the physi-

cal center. The pivot bolt in the offset head

should be tightened sufficiently that there will

be adequate friction to hold the array in posi-

tion. Then an additional nut should be placed

on the pivot bolt as a lock.

In connecting the phasing sections between

the T- junction and the centers of the folded

dipoles, it is important that the center con-

ductors of the phasing sections be connected

to the same side of the driven elements of the

antennas. In other words, when the antenna is

oriented for horizontal polarization and the

center of the coaxial phasing section goes to

the left side of the top antenna, the center

conductor of the other coaxial phasing section

should go to the left side of the bottom an-

tenn a.

The "Screen Beam" This highly effective ro-

for 2 Meters tary array for the 144 Mc.

amateur band was de-

signed by the staff of the Experimental Phy-

sics Laboratory, The Hague, Netherlands for

use at the 2 meter experimental station PEIPL.

The array consists of 10 half wave radiators

fed in phase, and arranged in two stacked rows

of five radiators. 0.2 wavelength behind this

plane of radiators is a reflector screen, meas-

uring approximately 15' x 9' in size. The an-

tenna provides a power gain of 15 db, and a

front to back ratio of approximately 28 db.

ALL JOINTS WELDED

Figure 15

DETAIL OF LAYOUT AND DIMENSIONS

OF BEAM ASSEMBLY OF PEIPL

The 10 dipoles are fed in phase by means

of a length of balanced transmission line, a

quarter -wave matching transformer, and a ba-

lun. A 72 -ohm coaxial line couples the array

to the transmitter. A drawing of the array is

shown in figure 15.

The reflecting screen measures 14' 9" high

by 8' 4" wide, and is made of welded %" dia-

meter steel tubing. Three steel reinforcing

bars are welded horizontally across the frame-

work directly behind each pair of horizontal

dipoles. The intervening spaces are filled

with lengths of no. 12 enamel- coated copper

wire to complete the screen. The spacing be-

tween the wires is 2 ". Four cross braces are

welded to the corners of the frame for addi-

tional bracing, and a single vertical %" rod

runs up the middle of the frame. The complete,

welded frame is shown in figure 15. The no.

12 screening wires are run between 6 -32 bolts

placed in holes drilled in each outside verti-

cal member of the frame.

The antenna assembly is supported away

from the reflector screen by means of ten

lengths of % " steel tubing, each l' 3%4" long.

www.americanradiohistory.com

486 V -H -F and U -H -F Antennas THE RADIO

Figure 14

THE EIGHTELEMENT 144 -MC. ARRAY IN A HORIZONTAL POSITION

appropriate cord. Hence, the operation is based

on the offset head sketched in figure 13. Al-

though a wood mast has been used, the same

system may be used with a pipe mast.

The 40 -inch lengths of RG -59/U cable (elec-

trically 3i4 wavelength) running from the center

of each folded dipole driven element to the

coaxial T- junction allow enough slack to per-

mit free movement of the main boom when

changing polarity. Type RG -8 /U cable is run

from the T- junction to the operating position.

Measured standing -wave ratio was less than

2:1 over the 144 to 148 Mc. band, with the

lengths and spacings given in figure 13.

Construeion of Most of the constructional

the Array aspects of the antenna array

are self- evident from figure

13. However, the pointers given in the follow-

ing paragraphs will be of assistance to those

wishing to reproduce the array.

The drilling of holes for the small elements

should be done carefully on accurately marked

centers. A small angular error in the drilling

of these holes will result in a considerable

misalignment of the elements after the array is

assembled. The same consideration is true of

the filing out of the rounded notches in the

ends of the main boom for the fitting of the

two antenna booms.

Short lengths of wood dowel are used freely

in the construction of the array. The ends of

the small elements are plugged with an inch

or so of dowel, and the ends of the antenna

booms are similarly treated with larger discs

pressed into place.

www.americanradiohistory.com

HANDBOOK VHF Parasitic Arrays 485

16 w- -,6

I

REFLECTOR

40"

RADIATOR

35"

___1ST DIRECTOR

36"

2ND

DIRECTOR

35 5"

5

RING BOLT ß_4B BOOM

SHAPE ENDS OF SHORT PIECES

TO FIT CONTOUR

FILE END TO FIT

60- BOOM

MAIN BOOMS- S- APPROX. O.D.

ELEMENTS -- -

RG -59 /U CABLES

EACH 40 LONG

RG -B /U CABLE

TO ' T. COAXIAL

FITTING

INSULATING ROO. ENDS

CUT DOWN TO GO INTO TUBING

ABOU

ENDS OF TUBING TERMINALS

WOOD DOWELS IN-

SIDE FOR STRENGTH -

RG -13/U CA

TO RIG

AS SHOWN. ANTENNA IS

HORIZONTALLY POLARIZED

PULL TO SWING MAIN BOOM 90

FOR VERTICAL POLARITY.

CONTROL CORDS

WOOD ' 2X2

ROTATABLE MAST

RADIAL BEARING

Figure 13

CONSTRUCTIONAL DRAWING OF AN EIGHT- ELEMENT TIPPABLE 144 -MC. ARRAY

quency range. Although polarization has been

loosely standardized in various areas of the

country, exceptions are frequent enough so

that it is desirable that the polarization of an-

tenna radiation be easily changeable from hori-

zontal to vertical.

The antenna illustrated has shown a signal

gain of about 11 db, representing a power gain

of about 13. Although the signal gain of the

antenna is the same whether it is oriented for

vertical or horizontal polarization, the hori-

zontal beam width is smaller when the antenna

is oriented for vertical polarization. Conver-

sely, the vertical pattern is the sharper when

the antenna system is oriented for horizontal

polarization.

The changeover from one polarization to the

other is accomplished simply by pulling on the

www.americanradiohistory.com

484 V -H -F and U -H -F Antennas THE RADIO

sistors in series. If 2 -watt resistors are em-

ployed, this termination also is suitable for

transmitter outputs of 10 watts or less. For

higher powers, however, resistors having great-

er dissipation with negligible reactance in the

upper v -h -f range are not readily available.

For powers up to several hundred watts a

suitable termination consists of a "lossy"

line consisting of stainless steel wire (corres-

ponding to no. 24 or 26 B &S gauge) spaced 2

inches, which in turn is terminated by two

390 -ohm 2 -watt carbon resistors. The dissi-

pative line should be at least 6 wavelengths

long.

24 -8 Multi- Element V -H -F

Beam Antennas

The rotary multi -element beam is undoubted-

ly the most popular type of v -h -f antenna in

use. In general, the design, assembly and tun-

ing of these antennas follows a pattern similar

to the iarger types of rotary beam antennas

used on the lower frequency amateur bands.

The characteristics of these low frequency

beam antennas are discussed in the next chap-

ter of this Handbook, and the information con-

tained in that chapter applies in general to the

v -h -f beam antennas discussed herewith.

A Simple Three The simplest v -h -f beam for

Element Beam the beginner is the three -ele-

Antenna ment Yagi array illustrated in

figure 12. Dimensions are

given for Yagis cut for the 2 -meter and IS-

meter bands. The supporting boom for the Yagi

may be made from a smoothed piece of 1" x 2"

wood. The wood should be reasonably dry and

should be painted to prevent warpage from ex-

posure to sun and rain. The director and re-

flector are cut from lengths of %" copper tub-

ing, obtainable from any appliance store that

does service work on refrigerators. They should

be cut to length as noted in figure 12. The ele-

ments should then be given a coat of aluminum

paint. Two small holes are drilled at the center

of the reflector and director and these elements

are bolted to the wood boom by means of two

111 wood screws. These screws should be of

the plated, or rust -proof variety.

The driven element is made of a 78" length

of ia" copper tubing, the ends bent back upon

each other to form a folded dipole. If the tub-

ing is packed with fine sand and the bending

points heated over a torch, no trouble will be

had in the bending process. If the tubing does

collapse when it is bent, the break may be re-

paired with a heavy -duty soldering iron. The

FOR 1 ¡ METERS

D=22^ D-A-9

A-23}

R-y MWÚS2-

BEND RADIUS

1B

-

R- REFLECTOR

40 LONG D

A-DRIVEN ELEMENT

'LONG

MIP-D DIRECTOR

36" LONG

1X2 WELL-SEASONED

WOOD 34 LONG f L

Y

_

GAIN 7.5 DB (20. LONG FOR 11- METERS) N3ULATING_BLbC5 .il 1)

FEED LINE

TNRU HOLE,

TAIL

FLATTEN ENDS OF

TUBING AND DRILL

FOR 6/32 SCREWS WOOD BOOM

75

TV LINE

Figure 12

SIMPLE 3- ELEMENT BEAM FOR 2 AND

1'/ METERS

driven element is next attached to the center

of the wood boom, mounted atop a small in-

sulating plate made of bakelite, micarta or

some other non -conducting material. It is held

in place in the same manner as the parasitic

elements. The two free ends of the folded di-

pole are hammered flat and drilled for a 6 -32

bolt. These bolts pass through both the insu-

lating block and the boom, and hold the free

tips of the element in place.

A length of 75 -ohm Twin -Lead TV -type line

should be used with this beam antenna. It is

connected to each of the free ends of the folded

dipole. If the.antenna is mounted in the verti-

cal plane, the 75-ohm line should be brought

away from the antenna for a distance of four

to six feet before it drops down the tower to

lessen interaction between the antenna ele-

ments and the feed line. The complete antenna

is light enough to be turned by a TV rotator.

A simple Yagi antenna of this type will pro-

vide a gain of 7 db over the entire 2 -meter or

IS-meter band, and is highly recommended as

an "easy -to- build" beam for the novice or

beginner.

An 8- Element Figures 13 and 14 illus-

"Tippoble" Array trate an 8- element rotary

for 144 Mc. array for use on the 144 -

Mc. amateur band. This

array is "tippable" to obtain either horizontal

or vertical polarization. It is necessary that

the transmitting and receiving station use the

same polarization for the ground -wave signal

propagation which is characteristic of this fre-

www.americanradiohistory.com

HANDBOOK VHF Rhombic 483

4). 131. 6A

SIDE LENGTH, S

Figure 10

V -H -F RHOMBIC ANTENNA DESIGN

CHART

The optimum tilt angle (see figure 11) for

zero-angle" radiation depends upon the

length of the sides.

1OA

10 to 16 db gain with a simpler construction

than does a phased dipole array, and has the

further advantage of being useful over a wide

frequency range.

Except at the upper end of the v -h -f range

a rhombic array having a worthwhile gain is

too large to be rotated. However, in locations

75 to 150 miles from a large metropolitan area

a rhombic array is ideally suited for working

into the city on extended (horizontally polar-

ized) ground -wave while at the same time mak-

ing an ideal antenna for TV reception.

The useful frequency range of a v -h -f rhom-

bic array is about 2 to I, or about plus 40% and

minus 30% from the design frequency. This

coverage is somewhat less than that of a high -

frequency rhombic used for sky -wave communi-

cation. For ground -wave transmission or recep-

tion the only effective vertical angle is that

of the horizon, and a frequency range greater

than 2 to I cannot be covered with a rhombic

array without an excessive change in the ver-

tical angle of maximum radiation or response.

The dimensions of a v -h -f rhombic array are

determined from the design frequency and fig-

ure 10, which shows the proper tilt angle (see

figure 11) for a given leg length. The gain of

a rhombic array increases with leg length.

There is not much point in constructing a v -h -f

rhombic array with legs shorter than about 4

wavelengths, and the beam width begins to be-

come excessively sharp for leg lengths greater

than about 8 wavelengths. A leg length of 6

wavelengths is a good compromise between

beam width and gain.

The tilt angle given in figure 10 is based

upon a wave angle of zero degrees. For leg

lengths of 4 wavelengths or longer, it will be

TOP VIEW

0' TILT ANGLE

h

RI, R22390 OHMS EACH

NON -INDUCTIVE

Figure 11

V -H -F RHOMBIC ANTENNA

CONSTRUCTION

necessary to elongate the array a few per cent

(pulling in the sides slightly) if the horizon

elevation exceeds about 3 degrees.

Table I gives dimensions for two dual pur-

pose rhombic arrays. One covers the 6 -meter

amateur band and the "low" television band.

The other covers the 2 -meter amateur band,

the "high" television band, and the 1%4-meter

amateur band. The gain is approximately 12

db over a matched half wave dipole and the

beam width is about 6 degrees.

The Feed Line The recommended feed line

is an open -wire line having a

surge impedance between 450 and 600 ohms.

With such a line the VSWR will be less than

2 to 1. A line with two -inch spacing is suit-

able for frequencies below 100 Mc., but one -

inch spacing (such as used in the Gonset Line

for TV installations) is recommended for high-

er frequencies.

The Termination If the array is to be used only

for reception, a suitable ter-

mination consists of two 390 -ohm carbon re-

6 METERS 2 METERS, NIGH

AND LOW BAND BAND TV, AND

TV 11Q METERS

S

(side) 90. 32'

L

(length) 166' 10" 59' 4"

W

(Width) 67' 4" 23' 11"

S -6 warelenths at design

Tilt ongle 6B0 frequency

TABLE I.

www.americanradiohistory.com

482 V -H -F and U -H -F Antennas THE RADIO

pA UHF HORN ANTENNA

430-OHM LINE

OB VHF HORIZONTALLY POLARIZED HORN

Figure 8

TWO TYPES OF HORN ANTENNAS

The "two sided horn" of Figure BB may be

fed by means of on open -wire transmission

line.

Copper screen may also be used for the re-

flecting planes.

The values of spacing given in the corner -

reflector chart have been chosen such that the

center impedance of the driven element would

be approximately 70 ohms. This means that

the element may be fed directly with 70 -ohm

coaxial line, or a quarter -wave matching trans-

former such as a "Q" section may be used to

provide an impedance match between the cen-

ter- impedance of the element and a 460 -ohm

line constructed of no. 12 wire spaced 2 inches.

In many v -h -f antenna systems, waveguide

transmission lines are terminated by pyramidal

horn antennas. These horn antennas (figure

8A) will transmit and receive either horizon-

tally or vertically polarized waves. The use of

waveguides at 144 Mc. and 235 Mc., however,

is out of the question because of the relatively

large dimensions needed for a waveguide oper-

ating at these low frequencies.

A modified type of horn antenna may still be

used on these frequencies, since only one par-

ticular plane of polarization is of interest to

the amateur. In this case, the horn antenna

can be simplified to two triangular sides of

the pyramidal horn. When these two sides are

insulated from each other, direct excitation at

the apex of the horn by a two -wire transmission

line is possible.

In a normal pyramidal horn, all four triangu-

lar sides are covered with conducting material,

but when horizontal polarization alone is of

interest (as in amateur work) only the vertical

areas of the horn need be used. If vertical po-

larization is required, only the horizontal areas

A

450 -ONM TV LINE

ANGLE BETWEEN

SIDES OF MORN '"606

D

A

2a

ZA-A GAIN (DB)

400 3

20 9

390 1S

i

TWO SIDES MADE

OF WIRE MESH

Figure 9

THE 60° HORN ANTENNA FOR USE ON

FREQUENCIES ABOVE 144 MC.

of the horn are employed. In either case, the

system is unidirectional, away from the apex

of the horn. A typical horn of this type is shown

in figure 8B. The two metallic sides of the

horn are insulated from each other, and the

sides of the horn are made of small mesh

"chicken wire" or copper window screening.

A pyramidal horn is essentially a high -pass

device whose low frequency cut -off is reached

when a side of the horn is % wavelength. It

will work up to infinitely high frequencies,

the gain of the horn increasing by 6 db every

time the operating frequency is doubled. The

power gain of such a horn compared to a 1/2

wave dipole at frequencies higher than cut-

off is:

8.4 A2

Power gain (db)

A2

where A is the frontal area of the mouth of the

horn. For the 60 degree horn shown in figure

8B the formula simplifies to:

Power gain (db) = 8.4 D2, when D is ex-

pressed in terms of wavelength

When D is equal to one wavelength, the pow-

er gain of the horn is approximately 9 db. The

gain and feed point impedance of the 60 de-

gree horn are shown in figure 9. A 450 ohm

open wire TV -type line may be used to feed

the horn.

24 -7 VHF Horizontal

Rhombic Antenna

For v -h -f transmission and reception in a

fixed direction, a horizontal rhombic permits

www.americanradiohistory.com

HANDBOOK Helical Beam Antenna 481

D 22 in.

S .16%2 in.

G 53 in.

Tubing o.d 1 in.

The D and S dimensions are to the center of

the tubing. These dimensions must be held

rather closely, since the range from 144 through

225 Mc. represents just about the practical

limit of coverage of this type of antenna sys-

tem.

High -Band Note that an array constructed

TV Coverage with the above dimensions will

give unusually good high -band

TV reception in addition to covering the 144 -

Mc. and 220 -etc. amateur bands and the taxi

and police services.

On the 144 -Mc. band the beam width is ap-

proximately 60 degrees to the half -power

points, while the power gain is approximately

11 db over a non -directional circularly polar-

ized antenna. For high -band TV coverage the

gain will be 12 to 14 db, with a beam width

of about 50 degrees, and on the 220 -Mc. ama-

teur band the beam width will be about 40 de-

grees with a power gain of approximately 15 db.

The antenna system will receive vertically

polarized or horizontally polarized signals

with equal gain over its entire frequency range.

Conversely, it will transmit signals over the

same range, which then can be received with

equal strength on either horizontally polarized

or vertically polarized receiving antennas.

The standing -wave ratio will be very low over

the complete frequency range if RG -63/U co-

axial feed line is used.

24 -6 The Corner -Reflector

and Horn -Type Antennas

The corner -reflector antenna is a good direc-

tional radiator for the v -h -f and u -h -f region.

The antenna may be used with the radiating

element vertical, in which case the directivity

is in the horizontal or azimuth plane, or the

system may be used with the driven element

DRIVEN DIPOLE

SUPPORTING

ME H

Figure 7

CONSTRUCTION OF THE "CORNER

REFLECTOR" ANTENNA

Such an antenna is capable of giving high

gain with a minimum of complexity in the

radiating system. It may be used either with

horizontal or vertical polarization. Design

data for the antenna is given in the Corner-

Reflector Design Table.

horizontal in which case the radiation is hori-

zontally polarized and most of the directivity

is in the vertical plane. With the antenna used

as a horizontally polarized radiating system

the array is a very good low -angle beam array

although the nose of the horizontal pattern is

still quite sharp. When the radiator is oriented

vertically the corner reflector operates very

satisfactorily as a direction -finding antenna.

Design data for the corner -reflector antenna

is given in figure 7 and in the chart Cosner -

Re /lector Design Data. The planes which make

up the reflecting corner may be made of solid

sheets of copper or aluminum for the u -h -f

bands, although spaced wires with the ends

soldered together at top and bottom may be

used as the reflector on the lower frequencies.

CORNER- REFLECTOR DESIGN DATA

Corner

Angle Freq.

Band, Mc. R S H A L G Feed

Imped. Approx.

Gain, db

90 SO 110" 82" 140" 200" 230" 18" 72 10

60 50 110" 115" 140" 230" 230" 18" 70 12

60 144 38" 40" 48" 100" 100" 5 " 70 12

60 220 24.5" 25" 30" 72" 72" 3 " 70 12

60 420 13" 14" 18" 36" 36" 70 12

NOTE: Refer to figure 7 for construction of corner- reflector an

www.americanradiohistory.com

480 V -H -F and U -H -F Antennas THE RADIO

T

G

t

,,,,-ROUND OR SQUARE

GROUND SCREEN

L

TRANSMIT

RECEIVE

/\/\/\/\/\/\

COAX FEED POINT (RG -63/U)

AT CENTER OF

GROUND SCREEN

D =+ 5= á G =oer.

CONDUCTOR DIA APPROX O.t1A

%= WAVELENGTH IN FREE SPACE

L. i. A

Figure 6

THE "HELICAL BEAM" ANTENNA

This type of directional antenna system

gives excellent performance over o frequency

range of 1.7 to 1.8 to 1. Its dimensions are

such that it ordinarily is not practicable,

however, for use as a rotatable array on fre-

quencies below about 100 Mc. The center

conductor of the feed line should pass

through the ground screen for connection to

the feed point. The outer conductor of the

coaxial line should be grounded to the

ground screen.

the time of writing, there has been no stand-

ardization of the "twist" for general amateur

work.

Perhaps the simplest antenna configuration

for a directional beam antenna having circular

polarization is the helical beam popularized

by Dr. John Kraus, W8JK. The antenna con-

sists simply of a helix working against a

ground plane and fed with coaxial line. In the

u -h -f and the upper v -h -f range the physical

dimensions are sufficiently small to permit

construction of a rotatable structure without

much difficulty.

When the dimensions are optimized, the

characteristics of the helical beam antenna

are such as to qualify it as a broad band an-

tenna. An optimized helical beam shows little

variation in the pattern of the main lobe and

a fairly uniform feed point impedance averag-

ing approximately 125 ohms over a frequency

range of as much as 1.7 to 1. The direction of

"electrical twist" (right or left handed) de-

pends upon the direction in which the helix is

wound.

A six -turn helical beam is shown schemati-

cally in figure 6. The dimensions shown will

give good performance over a frequency range

of plus or minus 20 per cent of the design fre-

quency. This means that the dimensions are

not especially critical when the array is to be

used at a single frequency or over a narrow

band of frequencies, such as an amateur band.

At the design frequency the beam width is

about 50 degrees and the power gain about 12

db,referred to a non -directional circularly po-

larized antenna.

The Ground Screen For the frequency range

100 to 500 Mc. a suitable

ground screen can be made from "chicken

wire" poultry netting of 1 -inch mesh, fastened

to a round or square frame of either metal or

wood. The netting should be of the type that

is galvanized after weaving. A small, sheet

metal ground plate of diameter equal to ap-

proximately D/2 should be centered on the

screen and soldered to it. Tin, galvanized

iron, or sheet copper. is suitable. The outer

conductor of the RG -63/U (125 ohm) coax is

connected to this plate, and the inner conduc-

tor contacts the helix through a hole in the

center of the plate. The end of the coax should

be taped with Scotch electrical tape to keep

water out.

The Helix It should be noted that the beam

proper consists of six full turns.

The start of the helix is spaced a distance of

S/2 from the ground screen, and the conductor

goes directly from the center of the ground

screen to the start of the helix.

Aluminum tubing in the "SO" (soft) grade

is suitable for the helix. Alternatively, lengths

of the relatively soft aluminum electrical con-

duit may be used. In the v -h -f range it will be

necessary to support the helix on either two

or four wooden longerons in order to achieve

sufficient strength. The longerons should be

of as small cross section as will provide suf-

ficient rigidity, and should be given several

coats of varnish. The ground plane butts

against the longerons and the whole assembly

is supported from the balance point if it is to

be rotated.

Aluminum tubing in the larger diameters ordi-

narily is not readily available in lengths great-

er than 12 feet. In this case several lengths

can be spliced by means of short telescoping

sections and sheet metal screws.

The tubing is close wound on a drum and

then spaced to give the specified pitch. Note

that the length of one comp 1 e t e turn when

spaced is somewhat greater than the circumfer-

ence of a circle having the diameter D.

Broad -Band A highly useful v -h -f helical

144 to 225 Mc. beam which will receive sig-

Helical Beam nals with good gain over the

complete frequency range from

144 through 225 Mc. may be constructed by

using the following dimensions (180 Mc. de-

sign center):

www.americanradiohistory.com

HANDBOOK Discone Antenna 479

0.1 D

50n. COAX

(PIG-4/U, ETC.)

Figure SA

THE ''DISCONE " BROAD -BAND

RADIATOR

This antenna system radiates a vertically

polarized wave over a very wide frequency

range. The "disc" may be made of solid

met al sheet, a group of radials, or wire

screen; the "cone" may best be constructed

by forming a sheet of thin aluminum. A sin-

gle antenna may be used for operation on the

50, 144, and 220 Mc. amdteur bands. The

dimension D is determined by the lowest fre-

quency to be employed, and is given in the

chart of figure 58.

VSXRof less than 1.5 will be obtained through-

out the operating range of the antenna.

The Discone antenna may be considered

as a cross between an electromagnetic horn

and an inverted ground plane unipole antenna.

It looks to the feed line like a properly termi-

nated high -pass filter.

Construction Details The top disk and the

conical skirt may be

fabricated either from sheet metal, screen (such

as "hardware cloth "), or 12 or more "spine"

radials. If screen is used a supporting frame-

work of rod or tubing will be necessary for

mechanical strength except at the higher fre-

quencies.. If spines are used, they should be

terminated on a stiff ring for mechanical

strength except at the higher frequencies.

The top disk is supported by means of three

insulating pillars fastened to the skirt. Either

polystyrene or low -loss ceramic is suitable for

the purpose. The apex of the conical skirt is

grounded to the supporting mast and to the

outer conductor of the coaxial line. The line

is run down through the supporting mast. An

alternative arrangement, one suitable for cer-

tain mobile applications, is to fasten the base

400i

300

200

160

140

120'

110

MO'

90

so

To

60

50 O.! t 0 15 2 2.5 3

DIN FEET

4 6

Figure 5B

DESIGN CHART FOR THE "DISCONE"

ANT ENN A

of the skirt directly to an effective ground

plane such as the top of an automobile.

24 -5 Helical Beam

Antennas

Most v -h -f and u -h -f antennas are either ver-

tically polarized or horizontally polarized

(plane polarization). However, circularly po-

larized antennas have interesting characteris-

tics which may be useful for certain applica-

tions. The installation of such an antenna can

effectivèly solve the problem of horizontal vs.

vertical polarization.

A circularly polarized wave has its energy

divided equally between a vertically polarized

component and a horizontally polarized com-

ponent, the two being 90 degrees out of phase.

The circularly polarized wave may be either

"left handed" or "right handed," depending

upon whether the vertically polarized compo-

nent leads or lags the horizontal component.

A circularly polarized antenna will respond

to any plane polarized wave whether horizon-

tally polarized, vertically polarized, or diag-

onally polarized. Also, a circular polarized

wave can be received on a plane polarized an-

tenna, regardless of the polarization of the

latter. When using circularly polarized anten-

nas at both ends of the circuit, however, both

must be left handed or both must be right

handed. This offers some interesting possi-

bilities with regard to reduction of QRM. At

www.americanradiohistory.com

478 V -H -F and U -H -F Antennas THE RADIO

TOP APEX CONNECTS TO

INNER CONNECTOR

LOWER APEX CONNECTS

TO OUTER CONDUCTOR APICES FORMED

~ -OF SHEET METAL

RD-B /U CABLE

Figure 3

THE DOUBLE SKELETON CONE

ANTENNA

A skeleton cone has been substituted for the

single element radiator of figure 2C. This

greatly increases the bandwidth. If at least

10 elements are used for each skeleton cone

and the angle of revolution and element

length are optimized, a low SWR con be ob-

tained over o frequency range of at least two

octaves. To obtain this order of bandwidth,

the element length L should be approximate-

ly 0.2 wavelength at the lower frequency end

of the band, and the angle of revolution opti-

mized for the lowest maximum VSWR within

the frequency range to be covered. A greater

improvement in the impedance -frequency

characteristic can be achieved by adding

elements than by increasing the diameter of

the elements. With only 3 elements per

"cone'. and a much smaller angle of revo-

lution a low SWR can be obtained over a fre-

quency range of approximately 1.3 to 1.0

when the element lengths are optimized.

work over several octaves, the gain varying

only slightly over a very wide frequency range.

Commercial versions of the Discone anten-

na for various applications are manufactured

by the Federal Telephone and Radio Corpora-

tion. A Discone type antenna for amateur work

can be fabricated from inexpensive materials

with ordinary hand tools.

A Discone antenna suitable for multi -band

amateur work in the v- h /u -h -f range is shown

schematically in figure 5A. The distance D

should be made approximately equal to a free -

space quarter wavelength at the lowest oper-

1 f jALUMINUM TUBING

36"

TYP. 2X2 191 IR'

TYP.

w 19"

. E/r,

T = ak- ..UU/MUMCM7SS-

I BAB TIGHTENS IT UP.

S.I.

220 MC.

300-OHM

FEEDLIN

300-OHM

TUBULAR

TWIN LEAD

20'

300-OHM

FEEDLINE

Figure 4

NONDIRECTIONAL ARRAYS FOR 144 MC.

AND 235 MC.

On right is shown two band installation. The

whole system may easily be dissembled and

carried on a ski -rock atop a car for portable

use.

acing frequency. The antenna then will per-

form well over a frequency range of at least

8 to 1. At certain frequencies within this

range the vertical pattern will tend to "lift"

slightly, causing a slight reduction in gain at

zero angular elevation, but the reduction is

very slight.

Below the frequency at which the slant

height of the conical skirt is equal to a free -

space quarter wavelength the standing -wave

ratio starts to climb, and below a frequency

approximately 20 per cent lower than this the

standing -wave ratio climbs very rapidly. This

is termed the cut off frequency of the antenna.

By making the slant height approximately equal

to a free -space quarter wavelength at the low-

est frequency employed (refer to chart), a

www.americanradiohistory.com

HANDBOOK Vertically Polarized Arrays 477

CLOSED

I

r ~OPEN

Figure 2

THREE VERTICALLY -POLARIZED

LOW -ANGLE RADIATORS

Shown at (A) is the "sleeve" or "'hypoder-

mic" type of radiator. At (©) is shown the

ground -plane vertical, and (C) shows a modi-

fication of this antenna system which in-

creases the feed -point impedance to a value

such that the system may be fed directly

from o coaxial line with no standing waves

on the feed line.

matching transformer, and a good match is

obtained.

In actual practice the antenna would con-

sist of a quarter -wave rod, mounted by means

of insulators atop a pole or pipe mast. Elab-

orate insulation is not required, as the voltage

at the lower end of the quarter -wave radiator

is very low. Self- supporting rods from 0.25 to

0.28 wavelength would be extended out, as in

the illustration, and connected together. As

the point of connection is effectively at ground

potential, no insulation is required; the hori-

zontal rods may be bolted directly to the sup-

porting pole or mast, even if of metal. The co-

axial line should be of the low loss type es-

pecially designed for v -h -f use. The outside

connects to the junction of the radials, and

the inside to the bottom end of the vertical

radiator. An antenna of this type is moderately

simple to construct and will give a good ac-

count of itself when fed at the lower end of the

radiator directly by the 52 -ohm RG -8 /U co-

axial cable. Theoretically the standing -wave

ratio will be approximately 1.5 -to -1 but in

practice this moderate s -w -r produces no

deleterious effects, even on coaxial cable.

The modification shown in figure 2C permits

matching to a standard 50- or 70 -ohm flexible

coaxial cable without a linear transformer. If

the lower rods hug the line and supporting mast

rather closely, the feed -point impedance is

about 70 ohms. If they are bent out to form an

angle of about 30° with the support pipe the

impedance is about 50 ohms.

The number of radial legs used in a ground -

plane antenna of either type has an important

effect on the feed -point impedance and upon

the radiation characteristics of the antenna

system. Experiment has shown that three radi-

als is the minimum number that should be

used, and that increasing the number of radi-

als above six adds substantially nothing to the

effectiveness of the antenna and has no effect

on the feed -point impedance. Experiment has

shown, however, that the radials should be

slightly longer than one -quarter wave for best

results. A length of 0.28 wavelength has been

shown to be the optimum value. This means

that the radials for a 50 -Mc. ground -plane ver-

tical antenna should be 65" in length.

Double Skeleton The bandwidth of the anten-

Cone Antenna na of figure 2C can be in-

creased considerably by sub-

stituting several space -tapered rods for the

single radiating element, so that the "radia-

tor" and skirt are similar. If a sufficient num-

ber of rods are used in the skeleton cones and

the angle of revolution is optimized for the

particular type of feed line used, this antenna

exhibits a very low SWR over a 2 to 1 frequen-

cy range. Such an arrangement is illustrated

schematically in figure 3.

A Nondirectional Half -wave elements may be

Vertical Array stacked in the vertical plane

to provide a non -directional

pattern with good horizontal gain. An array

made up of four half -wave vertical elements

is shown in figure 4A. This antenna provides

a circular pattern with a gain of about 4.5 db

over a vertical dipole. It may be fed with

300 -ohm TV -type line. The feedline should be

conducted in such a way that the vertical por-

tion of the line is at least one -half wavelength

away from the vertical antenna elements. A

suitable mechanical assembly is shown in fig-

ure 4B for the 144 -Mc. and 235 -Mc. amateur

bands.

24 -4 The Discone Antenna

The Discone antenna is a vertically polar-

ized omnidirectional radiator which has very

broad band characteristics and permits a sim-

ple, rugged structure. This antenna presents a

substantially uniform feed -point impedance,

suitable for direct connection of a coaxial

line, over a range of several octaves. Alsg,

the vertical pattern is suitable for ground -wave

www.americanradiohistory.com

476 V -H -F and U -H -F Antennas THE RADIO

2=70n

TO XMTR

VECTOR SUM OF

2 PATTERNS

1

COAXIAL LINE LOW Z

TO TRANSMITTER TRANSMISSION LINE

O ©

Figure 1

THREE NONDIRECTIONAL, HORIZONTALLY POLARIZED ANTENNAS

radiation at the very low elevation angles are

not recommended for v -h -f and u -h -f work. It is

for this reason that the horizontal dipole and

horizontally- disposed colinear arrays are gen-

erally unsuitable for work on these frequen-

cies. Arrays using broadside or end -fire ele-

ments do concentrate radiation at low eleva-

tion angles and are recommended for v -h -f

work. Arrays such as the lazy -H, Sterba cur-

tain, flat -top beam, and arrays with parasiti-

cally excited elements are recommended for

this work. Dimensions for the first three types

of arrays may be determined from the data

given in the previous chapter, and reference

may be made to the Table of Wavelengths given

in this chapter.

Arrays using vertically- stacked horizontal

dipoles, such as are used by commercial tele-

vision and FM stations, are capable of giving

high gain without a sharp horizontal radiation

pattern. If sets of crossed dipoles, as shown

in figure 1A, are fed 90° out of phase the re-

sulting system is called a turnstile antenna.

The 90° phase difference between sets of di-

poles may be obtained by feeding one set of

dipoles with a feed line which is one -quarter

wave longer than the feed line to the other

set of dipoles. The field strength broadside to

one of the dipoles is equal to the field from

that dipole alone. The field strength at a point

at any other angle is equal to the vector sum

of the fields from the two dipoles at that an-

gle. A nearly circular horizontal pattern is

produced by this antenna.

A second antenna producing a uniform, hori-

zontally polarized pattern is shown in figure

1B. This antenna employs three dipoles bent

to form a circle. All dipoles are excited in

phase, and are center fed. A bazooka is in-

cluded in the system to prevent unbalance in

the coaxial feed system.

A third nondirectional antenna is shown in

figure IC. This simple antenna is made of two

half -wave elements, of which the end quarter -

wavelength of each is bent back 90 degrees.

The pattern from this antenna is very much

like that of the turnstile antenna. The field

from the two quarter -wave sections that are

bent back are additive because they are 180

degrees out of phase and are a half wave-

length apart. The advantage of this antenna is

the simplicity of its feed system and con-

struction.

24 -3 Simple Vertical -Polarized

Antennas

For general coverage with a single antenna,

a single vertical radiator is commonly em-

ployed. A two -wire open transmission line is

not suitable for use with this type antenna,

and coaxial polyethylene feed line such as

RG -8 /U is to be recommended. Three practical

methods of feeding the radiator with concen-

tric line, with a minimum of current induced

in the outside of the line, are shown in figure

2. Antenna (A) is known as the sleeve anten-

na, the lower half of the radiator being a large

piece of pipe up through which the concentric

feed line is run. At (B) is shown the ground -

plane vertical, and at (C) a modification of

this latter antenna.

The radiation resistance of the ground -

plane vertical is approximately 30 ohms, which

is not a standard impedance for coaxial line.

To obtain a good match, the first quarter wave-

length of feeder may be of 52 ohms surge im-

pedance, and the remainder of the line of ap-

proximately 75 ohms impedance. Thus, the

first quarter -wave section of line is used as a

www.americanradiohistory.com

HANDBOOK Antenna Polarization 475

Radiator Cross There is no point in using

Section copper tubing for an antenna

on the medium frequencies.

The reason is that considerable tubing would

be required, and the cross section still would

not be a sufficiently large fraction of a wave-

length to improve the antenna bandwidth char-

acteristics. At very high and ultra high fre-

quencies, however, the radiator length is so

short that the expense of large diameter con-

ductor is relatively small, even though copper

pipe of 1 inch cross section is used. With such

conductors, the antenna will tune much more

broadly, and often a broad resonance charac-

teristic is desirable. This is particularly true

when an antenna or array is to be used over

an entire amateur band.

It should be kept in mind that with such

large cross section radiators, the resonant

length of the radiator will be somewhat shorter,

being only slightly greater than 0.90 of a half

wavelength for a dipole when heavy copper

pipe is used above 100 Mc.

Insulation . The matter of insulation is of

prime importance at very high fre-

quencies. Many insulators that have very low

losses as high as 30 Mc. show up rather poor-

ly at frequencies above 100 Mc. Even the low

loss ceramics are none too good where the r -f

voltage is high. One of the best and most prac-

tical insulators for use at this frequency is

polystyrene. It has one disadvantage, however,

in that it is subject to fracture and to deforma-

tion in the presence of heat.

It is common practice to design v -h -f and

u -h -f antenna systems so that the various rad-

iators are supported only at points of relatively

low voltage; the best insulation, obviously, is

air. The voltages on properly operated untuned

feed lines are not high, and the question of

insulation is not quite so important, though in-

sulation still should be of good grade.

Antenna Commercial broadcasting in the

Polarization U.S.A. for both FM and tele-

vision in the v -h -f range has

been standarized on horizontal polarization.

One of the main reasons for this standardiza-

tion is the fact that ignition interference is

reduced through the use of a horizontally po-

larized receiving antenna. Amateur practice,

however, is divided between horizontal and

vertical polarization in the v -h -f and u -h -f

range. Mobile stations are invariably vertical -

cally polarized due to the physical limitations

imposed by the automobile antenna installa-

tion. Most of the stations doing intermittent

or occasional work on these frequencies use a

simple ground -plane vertical antenna for both

transmission and reception. However, those

TABLE OF WAVELENGTHS

Fra. t/4 Wove 1/4 Wave 1/2 Wave 1/2 Wane

quency Free An- Free An-

in Mc. Space renna Space renna

50.0 S9.1 55.5 118.1 111.0

50.5 58.5 55.0 116.9 109.9

51.0 57.9 54.4 115.9 108.8

51.5 57.4 53.9 114.7 107.8

52.0 56.8 53.4 113.5 106.7

52.5 56.3 5 2. 8 112.5 105.7

53.0 55.7 52 4 1 1 1 .5 104.7

54.0 54.7 51 .4 109.5 102.8

144 20.5 19.2 41.0 38.5

145 20.4 19.1 40.8 38.3

146 20.2 18.9 40.4 38.0

147 20.0 18.8 40.0 37.6

148 19.9 18.6 39.9 37.2

235 12.6 11.8 25.2 23.6

236 12.5 11.8 25.1 23.5

237 12.5 11.7 25.0 23.5

238 12.4 11.7 24.9 23.4

239 12.4 11.6 24.8 23.3

240 12.3 11.6 24.6 23.2

420 7.05 6.63 14.1 13.25

425 6.95 6.55 13.9 13.1

430 6.88 6.48 13.8 12.95

All dimensions ore in inches. Lengths hove in

most cases been rounded off to three significant

figures. "1/2 -Wave Free -Space' column shown

above should be used with Lecher wires for fre-

quency measurement.

stations doing serious work and striving for

maximum -range contacts on the 50 -Mc. and

144 -Mc. bands almost invariably use horizon-

tal polarization.

Experience has shown that there is a great

attenuation in signal strength when using

crossed polarization (transmitting antenna

with one polarization and receiving antenna

with the other) for all normal ground -wave con-

tacts on these bands. When contacts are be-

ing made through sporadic -E reflection, how-

ever, the use of crossed polarization seems to

make no discernible difference in signal

strength. So the operator of a station doing

v -h -f work (particularly on the 50 -Mc. band)

is faced with a problem: If contacts are to be

made with all stations doing work on the same

band, provision must be made for operation on

both horizontal and vertical polarization. This

problem has been solved in many cases through

the construction of an antenna array that may

be revolved in the plane of polarization in ad-

dition to being capable of .rotation in the azi-

muth plane.

An alternate solution to the problem which

involves less mechanical construction is sim-

ply to install a good ground -plane vertical an-

tenna for all vertically- polarized work, and

then to use a multi -element horizontally- polar-

ized array for dx work.

24 -2 Simple Horizontally -

Polarized Antennas

Antenna systems which do not concentrate

www.americanradiohistory.com

474 V -H -F and U -X -F Antennas THE RADIO

that both are directed at the station being re-

ceived. Many instances have been reported

where a frequency band sounded completely

dead with a simple dipole receiving antenna

but when the receiver was switched to a three -

element or larger array a considerable amount

of activity from 80 to 160 miles distant was

heard.

Angle of The useful portion of the signal

Radiation in the v -h -f and u -h -f range for

short or medium distance communi-

cation is that which is radiated at a very low

angle with respect to the surface of the earth;

essentially it is that signal which is radiated

parallel to the surface of the earth. A vertical

antenna transmits a portion of its radiation at

a very low angle and is effective for this rea-

son; its radiation is not necessarily effective

simply because it is vertically polarized. A

simple horizontal dipole radiates very little

low -angle energy and hence is not a satisfac-

tory v -h -f or u -h -f radiator. Directive arrays

which concentrate a major portion of the radi-

ated signal at a low radiation angle will prove

to be effective radiators whether their signal

is horizontally or vertically polarized.

In all cases, the radiating system for v -h -f

and u -h -f work should be as high and in the

clear as possible. Increasing the height of the

antenna system will produce a very marked

improvement in the number and strength of the

signals heard, regardless of the actual type

of antenna used.

Transmission Transmission lines to v -h -f and

Lines u -h -f antenna systems may be

either of the parallel- conductor

or coaxial conductor type. Coaxial line is rec-

ommended for short runs and closely spaced

open -wire line for longer runs. Wave guides

may be used under certain conditions for fre-

quencies greater than perhaps 1500 Mc. but

their dimensions become excessively great for

frequencies much below this value. Non- reson-

ant transmission lines will be found to be con-

siderably more efficient on these frequencies

than those of the resonant type. It is wise to

to use the very minimum length of transmission

line possible since transmission line losses

at frequencies above about 100 Mc. mount very

rapidly.

Open sines should preferably be spaced

closer than is common for longer wavelengths,

as 6 inches is an appreciable fraction of a

wavelength at 2 meters. Radiation from the

line will be greatly reduced if 1 -inch or 11/4-

inch spacing is used, rather than the more com-

mon 6 -inch spacing.

Ordinary TV -type 300 -ohm ribbon may be

used on the 2 -meter band for feeder lengths

of about 50 feet or less. For longer runs, either

the u -h -f or v -h -f TV open -wire lines may be

used with good overall efficiency. The v -h -f

line is satisfactory for use on the amateur

420 -Mc. band.

Antenna It is recommended that the same

Changeover antenna be used for transmitting

and receiving in the v -h -f and

u -h -f range. An ever- present problem in this

connection, however, is the antenna change-

over relay. Reflections at the antenna change-

over relay become of increasing importance

as the frequency of transmission is increased.

When coaxial cable is used as the antenna

transmission line, satisfactory coaxial anten-

na changeover relays with low reflection can

be used. One type manufactured by Advance

Electric & Relay Co., Los Angeles 26, Calif.,

will give a satisfactorily low value of re-

flection.

On the 235-Mc. and 420 -Mc. amateur bands,

the size of the antenna array becomes quite

small, and it is practical to mount two identi-

cal antennas side by side. One of these an-

tennas is used for the transmitter, and the

other antenna for the receiver. Separate trans-

mission lines are used, and the antenna relay

may be eliminated.

Effect of Feed A vertical radiator for

System on Radiation general coverage u -h -f

Angle use should be made

either 1/4 or % wavelength

long. Longer vertical antennas do not have

their maximum radiation at right angles to the

line of the radiator (unless co- phased), and,

therefore, are not practicable for use where

greatest possible radiation parallel to the

earth is desired.

Unfortunately, a feed system which is not

perfectly balanced and does some radiating,

not only robs the antenna itself of that much

power, but distorts the radiation pattern of the

antenna. As a result, the pattern of a vertical

radiator may be so altered that the radiation

is bent upwards slightly, and the amount of

power leaving the an t e n n a parallel to the

earth is greatly reduced. A vertical half -wave

radiator fed at the bottom by a quarter -wave

stub is a good example of this; the slight

radiation from the matching section decreases

the power radiated parallel to the earth by

nearly 10 db.

The only cure is a feed system which does

not disturb the radiation pattern of the antenna

itself. This means that if a 2 -wire line is used,

the current and voltages must be exactly the

same (though 180° out of phase) at any point

on the feed line. It means that if a concentric

feed line is used, there should be no current

flowing on the outside of the outer conductor.

www.americanradiohistory.com

CHAPTER TWENTY -FOUR

V-li-F and U-li-F Antennas

The very- high -frequency or v -h -f frequency

range is defined as that range falling between

30 and 300 Mc. The ultra- high -frequency or

u -h -f range is defined as falling between 300

and 3000 Mc. This chapter will be devoted to

the design and construction of antenna sys-

tems for operation on the amateur 50 -Mc., 144 -

Mc., 235 -Mc., and 420 -Mc. bands. Although the

basic principles of antenna operation are the

same for all frequencies, the shorter physical

length of a wave in this frequency range and

the differing modes of signal propagation make

it possible and expedient to use antenna sys-

tems different in design from those used on

the range from 3 to 30 Mc.

24 -1 Antenna Requirements

Any type of antenna system useable on the

lower frequencies may be used in the v -h -f and

u -h -f bands. In fact, simple non -directive half -

wave or quarter -wave vertical antennas are

very popular for general transmission and re-

ception from all directions, especially for

short -range work. But for serious v -h -f or u -h -f

work the use of some sort of directional an-

tenna array is a necessity. In the first place,

when the transmitter power is concentrated in-

to a narrow beam the apparent transmitter pow-

er at the receiving station is increased many

times. A "billboard" array or a Sterba curtain

having a gain of 16 db will make a 25 -watt

transmitter sound like a kilowatt at the other

473

station. Even a much simpler and smaller three -

or four -element parasitic array having a gain

of 7 to 10 db will produce a marked improve-

ment in the received signal at the other sta-

tion.

11 o w e v e r, as all v -h -f and u -h -f workers

know, the most important contribution of a

high -gain antenna array is in reception. If a

remote station cannot be heard it obviously is

impossible to make contact. The limiting fac-

tor in v -h -f and u -h -f reception is in almost

every case the noise generated within the re-

ceiver itself. Atmospheric noise is almost non-

existent and ignition interference can almost

invariably be reduced to a satisfactory level

through the use of an effective noise limiter.

Even with a grounded -grid or neutralized triode

first stage in the receiver the noise contribu-

tion of the first tuned circuit in the receiver

will be relatively large. Hence it is desirable

to use an antenna system which will deliver

the greatest signal voltage to the first tuned

circuit for a given field strength at the receiv-

ing location.

Since the field intensity being produced at

the receiving location by a remote transmitting

station may be assumed to be constant, the re-

ceiving antenna which intercepts the greatest

amount of wave front, assuming that the polari-

zation and directivity of the receiving antenna

is proper, will be the antenna which gives the

best received signal -to -noise ratio. An antenna

which has two square wavelengths effective

area will pick up twice as much signal power

as one which has one square wavelength area,

assuming the same general type of antenna and

www.americanradiohistory.com

472 High Frequency Directive Antennas

Thus it is seen that, while maximum gain

occurs with two stacked dipoles at a spacing

of about 0.7 wavelength and the space direc-

tivity gain is approximately 5 db over one ele-

ment under these conditions; the case of two

flat top or parasitic arrays stacked one above

the other is another story. Maximum gain will

occur at a greater spacing, and the gain over

one array will not appreciably exceed 3 db.

When two broadside curtains are placed one

ahead of the other in end -fire relationship, the

aggregate mutual impedance between the two

curtains is such that considerable spacing is

required in order to realize a gain approaching

3 db (the required spacing being a function of

the size of the curtains). While it is true that

a space directivity gain of approximately 4 db

can be obtained by placing one, half -wave di-

pole an eighth wavelength ahead of another

and feeding them 180 degrees out of phase, a

gain of less than 1 db is obtained when the

same procedure is applied to two large broad-

side curtains. To obtain a gain of approximate-

ly 3 db and retain a bidirectional pattern, a

spacing of many wavelengths is required be-

tween two large curtains placed one ahead of

the other.

A different situation exists, however, when

one driven curtain is placed ahead of an iden-

tical one and the two are phased so as to give

a unidirectional pattern. When a unidirectional

pattern is obtained, the gain over one curtain

will be approximately 3 db regardless of the

spacing. For instance, two large curtains

placed one a quarter wavelength ahead of the

other may have a space directivity gain of only

0.5 db over one curtain when the two are driv-

en 180 degrees out of phase to give a bidirec-

tional pattern (the type of pattern obtained

with a single curtain). However, if they are

driven in phase quadrature (and with equal cur-

rents) the gain is approximately 3 db.

The directivity gain of a composite array

also can be explained upon the basis of the

directivity patterns of the component arrays

alone, but it entails a rather complicated pic-

ture. It is sufficient for the purpose of this

discussion to generalize and simplify by say-

ing that the greater the directivity of an end -

fire array, the farther an identical array must

be spaced from it in broadside relationship to

obtain optimum performance; and the greater

the directivity of a broadside array, the farther

an identical array must be spaced from it in

end -fire relationship to obtain optimum per-

formance and retain the bidirectional charac-

teristic.

It is important to note that while a bidirec-

tional end -fire pattern is obtained with two

driven dipoles when spaced anything under a

half wavelength, and while the proper phase

relationship is 180 degrees regardless of the

spacing for all spacings not exceeding one

half wavelength, the situation is different in

the case of two curtains placed in end -fire re-

lationship to give a bidirectional pattern. For

maximum gain at zero wave angle, the curtains

should be spaced an odd multiple of one half

wavelength and driven so as to be 180 degrees

out of phase, or spaced an even multiple of

one half wavelength and driven in the same

phase. The optimum spacing and phase rela-

tionship will depend upon the directivity pat-

tern of the individual curtains used alone, and

as previously noted the optimum spacing in-

creases with the size and directivity of the

component arrays.

A concrete example of a combination broad-

side and end -fire array is two Lazy H arrays

spaced along the direction of maximum radia-

tion by a distance of four wavelengths and fed

in phase. The space directivity gain of such

an arrangement is slightly less than 9 db. How-

ever, approximately the same gain can be ob-

tained by juxtaposing the two arrays side by

side or one over the other in the same plane,

so that the two combine to produce, in effect,

one broadside curtain of twice the area. It is

obvious that in most cases it will be more ex-

pedient to increase the area of a broadside

array than to resort to a combination of end -

fire and broadside directivity. One exception,

of course, is where two curtains are fed in

phase quadrature to obtain a unidirectional

pattern and space directivity gain of approxi-

mately 3 db with a spacing between curtains

as small as one quarter wavelength. Another

exception is where very low angle radiation is

desired and the maximum pole height is strict-

ly limited. The two aforementioned Lazy H

arrays when placed in end -fire relationship

will have a considerably lower radiation angle

than when placed side by side if the array ele-

vation is low, and therefore may under some

conditions exhibit appreciably more practical

signal gain.

www.americanradiohistory.com

HANDBOOK Triplex Beam 471

Figure 24

THE TRIPLEX FLATTOP BEAM

ANTENNA FOR 10, 15 AND 20

METERS

MAX. RADIATION

u- 4.S DS

'

RO.[ "

S

DIMENSIONS

MAXIMUM

RADIATION

11 4.5 OR

3000 LINE TO

TRANSMITTER

ANY LENGTH

10M. 15M. 20M MATERIAL

L 1'S 21'5' 32'2 iEL[tA[b 3'

S 5'0. 7'11. II'

D 7'2' 10'7' 14'4" 3000M RIeeON

to one -quarter wave spacing may be used on

the fundamental for the one -section types and

also the two -section center -fed, but it is not

desirable to use more than 0.15 wavelength

spacing for the other types.

Although the center -fed type of flat -top gen-

erally is to be preferred because of its sym-

metry, the end -fed type often is convenient or

desirable. For example, when a flat -top beam

is used vertically, feeding from the lower end

is in most cases more convenient.

If a multisection flat -top array is end -fed

instead of center -fed, and tuned feeders are

used, stations off the ends of the array can be

worked by tying the feeders together and work-

ing the whole affair, feeders and all, as a long -

wire harmonic antenna. A single -pole double -

throw switch can be used for changing the

feeders and directivity.

The Triplex The Triplex beam is a modified

Beam version of the W8] K antenna

which uses folded dipoles for

the half wave elements of the array. The use

of folded dipoles results in higher radiation

resistance of the array, and a high overall sys-

tem performance. Three wire dipoles are used

for the elements, and 300 -ohm Twin -Lead is

used for the two phasing sections. A recom-

mended assembly for Triplex beams for 28 Mc.,

21 Mc., and 14 Mc. is shown in figure 24. The

gain of a Triplex beam is about 4.5 db over a

dipole.

23 -8 Combination End -Fire and

Broadside Arrays

Any of the end -fire arrays previously de-

scribed may be stacked one above the other or

placed end to end (side by side) to give great-

er directivity gain while maintaining a bidi-

rectional characteristic. However, it must be

kept in mind that to realize a worthwhile in-

crease in directivity and gain while maintain-

ing a bidirectional pattern the individual ar-

rays must be spaced sufficiently to reduce the

mutual impedances to a negligible value.

When two flat top beams, for instance, are

placed one above the other or end to end, a

center spacing on the order of one wavelength

is required in order to achieve a worthwhile

increase in gain, or approximately 3 db.

www.americanradiohistory.com

470 High Frequency Directive Antennas THE RADIO

CENTER FED TO CENTER END FED

or FLAT TOP

I.-A-4

11

r It 1- SECTION 1- SECTION

M MATCHING STUB

,

A 1

{-oi L, --{ -L2 -'"i

1-IX

L,

S

2- SECTION 12M

-L2 01- L2-- STUB

CONNCCF ATCr ERS l'L3 L3I

g r 3-SECTION S

t M

Ioi I---La loi

.---L3 fol La L2--1 L4 L3-

g

r M

4- SECTION

,

2M S

4pr.- 404-- k-- 1_La

I--- L3- 1-o- -- L3- L3- L3- L3-4 _J-4- La ----1-61--1-3-.4

FIGURE 23

FLAT -TOP BEAM (8JK ARRAY) DESIGN DATA.

FREQUENCY Spar.

'os S L, L. L, L. M D A (/)

approx. A ('A)

approx. A (1/4)

approx. X

approx.

7.0-7.2 Mc. X/8 17'4' 34 60' 52'8' 44' 8'10' 4' 4' 26' 26' 60' 59' 96' 4'

7.2 -7.3 a/8 17'0' 33'6' 59' 51'8' 43'1' 8'8' 94' 4'

14.0 -14.4 )/B 8'8' 17' 30' 26'4 22' 4'S' 2' 13' 30' 48' 2'

14.0 -14.4 .15X 10'5' 17' 30' 2S'3' 20' 5'4' 2' 12' 29' 47' 2'

14.0 -14.4 .20X 13'11' 17' 30' 22'10' 7'2' 2' 10' 27' 45' 3'

14.0 -14.4 a/4 17'4' 17' 30' 20'8' 8'10' 2' 8' 25' 43' 4'

28.0-29.0 .15) 5'2' 8'6' 15' 12'7' 10' 2'8' 1'6' 7' 15' 24' 1'

28.0 -29.0 a/4 8'8' 8'6' 15' 14'6'

14'6'

10'4'

12'2' 9'8' 4'S' 1'6' 5' 13' 22' 2'

29.0 -30.0

29.0 -30.0 .15X

X/4 5'0'

8'4' 8'3'

8'3' 2'7' 1'6' 7' 15' 23' 1'

10'0' 4'4' 1'6' 5' 13' 21' 2'

Dimension chart for flat -top beam antennas. The meanings of the symbols are as fo lows:

L L. L. and L,, the lengths of the sides of the flat -top sections as shown. L, is length

of the sides of single -section center -fed, L. single- section end -fed and 2- section center -fed, L, 4- section

center -fed and end -sections of 4- section end -fed, and L, middle sections of 4- section end -fed.

S, the spacing between the flat -top wires.

M, the wire length from the outside to the center of each cross -over.

D, the spacing lengthwise between sections.

A (1/4), the approximate length for a quarter -wave stub.

A (''s), the approximate length for a half -wave stub.

A (3/4), the approximate length for a three -quarter wave stub.

X, the approximate distance above the shorting wire of the stub for the connection of a 600 -ohm

line. This distance, as given in the table, is approximately correct only for 2- section flat -tops.

For single- section types it will be smaller and for 3- and 4- section types it will be larger.

The lengths given for a half -wave stub are applicable only to single -section center -fed flat -tops. To

be certain of sufficient stub length, it is advisable to make the stub a foot or so longer than shown in

the table, especially with the end -fed types. The lengths, A, are measured from the point where the

stub connects to the flat -top.

Both the center and end -fed types may be used horizontally. However, where a vertical antenna is

desired, the flat -tops can be turned on end. In this case, the end -fed types may be more convenient,

feeding from the lower end.

www.americanradiohistory.com

HANDBOOK Endlire Arrays 469

Normally the antenna tank will be located

in the same room as the transmitter, to facil-

itate adjustment when changing frequency. In

this case it is recommended that the link cou-

pled tank be located across the room from the

transmitter if much power is used, in order to

minimize r -f feedback difficulties which might

occur as a result of the asymmetrical high im-

pedance feed. If tuning of the antenna tank

from the transmitter position is desired, flex-

ible shafting can be run from the antenna tank

condenser to a control knob at the transmitter.

The lower end of the driven element is quite

"hot" if much power is used, and the lead -in

insulator should be chosen with this in mind.

The ground connection need not have very low

resistance, as the current flowing in the

ground connection is comparatively small. A

stake or pipe driven a few feet in the ground

will suffice. However, the ground lead should

be of heavy wire and preferably the length

should 'not exceed about 10 feet at 7 Mc. or

about 20 feet at 4 Mc. in order to minimize

reactive effects due to its inductance. If it is

impossible to obtain this short a ground lead,

a piece of screen or metal sheet about four

feet square may be placed parallel to the earth

in a convenient location and used as an arti-

ficial ground. A fairly high C/L ratio ordinari-

ly will be required in the antenna tank in order

to obtain adequate coupling and loading.

23 -7 End -Fire Directivity

By spacing two half -wave dipoles, or colin-

ear arrays, at a distance of from 0.1 to 0.25

wavelength and driving the two 180° out of

phase, directivity is obtained through the two

wires at right angles to them. Hence, this type

of bidirectional array is called end fire. A bet-

ter idea of end -fire directivity can be obtained

by referring to figure 10.

Remember that end-fire refers to the radia-

tion with respect to the two wires in the array

rather than with respect to the array as a

whole.

The vertical directivity of an end -fire bi-

directional array which is oriented horizontal-

ly can be increased by placing a similar end -

fire array a half wave below it, and excited in

the same phase. Such an array is a combina-

tion broadside and end -fire affair.

Kraus Flat -Top A very effective bidirectional

Beam end -fire array is the Kraus or

8JK Hai-Top Beam. Essen-

tially, this antenna consists of two close -

spaced dipoles or colinear arrays. Because of

the close spacing, it is possible to obtain the

proper phase relationships in multi- section

flat tops by crossing the wires at the voltage

loops, rather than by resorting to phasing

stubs. This greatly simplifies the array. (See

figure 23.) Any number of sections may be

used, though the one- and two -section arrange-

ments are the most popular. Little extra gain

is obtained by using more than four sections,

and trouble from phase shift may appear.

A center -fed single- section flat -top beam

cut according to the table, can be used quite

successfully on its second harmonic, the pat-

tern being similar except that it is a little

sharper. The single- section array can also be

used on its fourth harmonic with some success,

though there then will be four cloverleaf lobes,

much the same as with a full -wave antenna.

If a flat -top beam is to be used on more than

one band, tuned feeders are necessary.

The radiation resistance of a flat -top beam

is rather low, especially when only one sec-

tion is used. This means that the voltage will

be high at the voltage loops. For this reason,

especially good insulators should be used for

best results in wet weather.

The exact lengths for the radiating elements

are not especially critical, because slight de-

viations from the correct lengths can be com-

pensated in the stub or tuned feeders. Proper

stub adjustment is covered in Chapter Twenty-

five. Suitable radiator lengths and approximate

stub dimensions are given in the accompany-

ing design table.

Figure 23 shows top views of eight types

of flat -top beam antennas. The dimensions for

using these antennas on different bands are

given in the design table. The 7- and 28 -Mc.

bands are divided into two parts, but the di-

mensions for either the low- or high -frequency

ends of these bands will be satisfactory for

use over the entire band.

In any case, the antennas are tuned to the

frequency used, by adjusting the shorting wire

on the stub, or tuning the feeders, if no stub

is used. The data in the table may be extended

to other bands or frequencies by applying the

proper factor. Thus, for 50 to 52 Mc. operation,

the values for 28 to 29 Mc. are divided by 1.8.

All of the antennas have a bidirectional hori-

zontal pattern on their fundamental frequency.

The maximum signal is broadside to the flat

top. The single- section type has this pattern

on both its fundamental frequency and second

harmonic. The other types have four main lobes

of radiation on the second and higher harmon-

ics. The nominal gains of the different types

over a half -wave comparison antenna are as

follows: single- section, 4 db; two- section, 6

db; four - section, 8 db.

The maximum spacings given make the

beams less critical in their adjustments. Up

www.americanradiohistory.com

468 High Frequency Directive Antennas THE RADIO

L

DIMENSIONS

IOM. 134. 20M.

L IT' 22,3 33.6"

D tree 22.9- 34'6'

L L

30011 RIBBON LINE

GAIN APPROX. 7.5 Da

Figure 21

THE "SIX- SHOOTER" BROADSIDE ARRAY

wire line should be employed if the antenna is

used with a high power transmitter.

To tune the reflector, the back of the an-

tenna is aimed at a nearby field -strength meter

and the reflector stub capacitor is adjusted

for minimum received signal at the operating

frequency.

This antenna provides high gain for its small

size, and is recommended for 28 -Mc. work.

The elements may be made of number 14 en-

amel wire, and the array may be built on a light

bamboo or wood framework.

The "Six- Shooter" As a good compromise be-

Broadside Array tween gain, directivity,

compactness, mechanical

simplicity, ease of adjustment, and band width

the array of figure 21 is recommended for the

10 to 30 Mc. range when the additional array

width and greater directivity are not obtain-

able. The free space directivity gain is ap-

proximately 7.5 db over one element, and the

practical dx signal gain over one element at

the same average elevation is of about the

same magnitude when the array is sufficiently

elevated. To show up to best advantage the

array should be elevated sufficiently to put

the lower elements well in the clear, and pre-

ferably at least 0.5 wavelength above ground.

The "Bobtail" Another application of ver-

Bidirectional tical orientation of the ra-

Broadside Curtain diating elements of an ar-

ray in order to obtain low -

angle radiation at the lower end of the h -f

range with low pole heights is illustrated in

figure 22. When precut to the specified dimen-

sions this single pattern array will perform

well over the 7 -Mc. amateur band or the 4 -Mc.

amateur phone band. For the 4 -Mc. band the

required two poles need be only 70 feet high,

and the array will provide a practical signal

DI DI

D2 3

saw END-LINK COIL

TO TUNE FREQUENCY

32 a

COAXIAL LINE

C. 100 LUF DIMENSIONS

40M. 60M.

DI 126,

Dz 33, 60,

D3 30.7036 34'TOM'

Figure 22

"BOBTAIL" BIDIRECTIONAL BROAD-

SIDE CURTAIN FOR THE 7 -MC. OR THE

4.0 -MC. AMATEUR BANDS

This simple vertically polarized array pro-

vides low angle radiation and response with

comparatively low pole heights, and is very

effective for dx work on the 7 -Mc. band or

the 4.0 -Mc. phone band. Because of the

phase relationships, radiation from the hori-

zontal portion of the antenna is effectively

suppressed. Very little current flows in the

ground lead to the coupling tank; so an elab-

orate ground system is not required, and the

length of the ground lead is not critical so

long os it uses heavy wire and is reason-

ably short.

gain averaging from 7 to 10 db over a horizon-

tal half -wave dipole utilizing the same pole

height when the path length exceeds 2500

miles.

The horizontal directivity is only moderate,

the beam width at the half power points being

slightly greater than that obtained from three

cophased vertical radiators fed with equal cur-

rents. This is explained by the fact that the

current in each of the two outer radiators of

this array carries only about half as much cur-

rent as the center, driven element. While this

"binomial" current distribution suppresses

the end -fire lobe that occurs when an odd num-

ber of parallel radiators with half -wave spac-

ing are fed equal currents, the array still ex-

hibits some high -angle radiation and response

off the ends as a result of imperfect cancella-

tion in the flat top portion. This is not suffi-

cient to affect the power gain appreciably, but

does degrade the discrimination somewhat.

A moderate amount of sag can be tolerated

at the center of the flat top, where it connects

to the driven vertical element. The poles and

antenna tank should be so located with respect

to each other that the driven vertical element

drops approximately straight down from the

flat top.

www.americanradiohistory.com

HANDBOOK Broadside Arrays 467

D

DIMENSIONS

101.4 ISM. 50M.

B'B

B'S 2YY

12.57

SS' O SECTION

151E. WIRE

SPACED 4-

GAIN APPROX. DB

150.11 LINE TO TRANSMITTER

Figure 19

THE "BI- SQUARE" BROADSIDE ARRAY

This bidirectional array Is related to the

"Lazy H," and in spite of the oblique ele-

ments, is horizontally polarized. It has slight-

ly less gain and directivity than the Lazy

H, the free space directivity gain being ap-

proximately 4 db. Its chief advantage is the

fact that only a single pole is required for

support, and two such arrays may be sup-

ported from a single pole without interaction

if the planes of the elements are at right

angles. A 600 -ohm line may be substituted

for the Twin -Lead, and either operated os a

resonant line, or made non -resonant by the

incorporation of a matching stub.

still worthwhile, being approximately 4 db over

a half -wave horizontal dipole at the same aver-

age elevation.

When two Bi -Square arrays are suspended

at right angles to each other (for general cov-

erage) from a single pole, the Q sections

should be well separated or else symmetrically

arranged in the form of a square (the diagonal

conductors forming one Q section) in order to

minimize coupling between them. The same

applies to the line if open construction is used

instead of Twin -Lead, but if Twin -Lead is

used the coupling can be made negligible sim-

ply by separating the two Twin -Lead lines by

at least two inches and twisting one Twin -

Lead so as to effect a transposition every foot

or so.

sT EACH

SIDE

EACN SIOE

REFLECTOR 'RADIATOR

CDR

CAIN

C' MIN LLF I I

PUNK FOR M RM ON BALANCED

CED LINE

PICKUP REAR 5% 8.107 TUNING UNIT OR

OF BEI M. Id TRANSMITTER.

SPACED )^

NOTE s,oe LENCTN- rI' B FOR 2I MC

IT T FOR 74 MC.

ELEMENT SPACING PB- FOR LACS BAND.

STUD LENCT PPROX IS' FOR 21 MC.

20 FOR 14 MC.

Figure 20

THE CUBICAL -QUAD ANTENNA FOR

THE 10 -METER BAND

When tuned feeders are employed, the Bi-

Square array can be used on half frequency as

an end -fire vertically polarized array, giving

a slight practical dx signal gain over a verti-

cal half -wave dipole at the same height.

A second Bi -Square serving as a reflector

may be placed 0.15 wavelength behind this an-

tenna to provide an overall gain of 8.5 db. The

reflector may be tuned by means of a quarter -

wave stub which has a moveable shorting bar

at the bottom end. The stub is used as a sub-

stitute for the Q- section, since the reflector

employs no feed line.

The "Cubical- A smaller version of the Bi-

Quad" Antenna Square antenna is the Cubi-

cal -Quad antenna. Two half -

waves of wire are folded into a square that is

one -quarter wavelength on a side, as shown in

figure 20. The arraÿ radiates a horizontally

polarized signal. A reflector placed 0.15 wave-

length behind the antenna provides an overall

gain of some 6 db. A shorted stub with a paral-

leled tuning capacitor is used to resonate the

reflector. The Cubical -Quad is fed with a 150-ohm

line, and should employ some sort of antenna

tuner at the transmitter end of the line if a pi-

network type transmitter is used. There is a

small standing wave on the line, and an open

www.americanradiohistory.com

466 High Frequency Directive Antennas THE RADIO

L D + L

DIMENSIONS

10M. I5M. 20M.

L 13'3- 22' 32'10.

S 20' 30' 40'

P 14.2 21'3 213'4

D 3'Y 7.0-

GAIN APPRO*. 3 DB

7511 TRANSMISSION LINE

Figure 17

THE X -ARRAY FOR 28 MC., 21 MC.,

OR 14 MC.

The entire array (with the exception of the

75 -ohm feed line) is constructed of 300 -ohm

ribbon line. Be sure phasing lines (P) are

poled correctly, as shown.

in a vertical plane and properly phased, a

simplified form of in -phase curtain is formed,

providing an overall gain of about 6 db. Such

an array is shown in figure 17. In this X- array,

the four dipoles are all in phase, and are fed

by four sections of 300 -ohm line, each one -

half wavelength long, the free ends of all four

lines being connected in parallel. The feed

impedance at the junction of these four lines

is about 75 ohms, and a length of 75 -ohm

Twin -Lead may be used for the feedline to

the array.

An array of this type is quite small for the

28 -Mc. band, and is not out of the question

for the 21 -Mc. band. For best results, the bot-

tom section of the array should be one -half

wavelength above ground.

The Double -Bruce The Bruce Beam consists

Array of a long wire folded so

that vertical elements

carry in -phase currents while the horizontal

elements carry out of phase currents. Radia-

tion from the horizontal sections is low since

only a small current flows in this part of the

wire, and it is largely phased -out. Since the

height of the Bruce Beam is only one -quarter

wavelength, the gain per linear foot of array

is quite low. Two Bruce Beams may be com-

bined as shown in figure 18 to produce the

Double Bruce array. A four section Double

Bruce will give a vertically polarized emis-

sion, with a power gain of 5 db over a simple

L

'

' - L

1111111111

9004 LINE

DIMENSION L

10M I5M. 20 /IOM.

13'9- 113' 17.3 ANTENNA

TUNER

GAIN APPROX. 5 DB

TRANSMITTER

Figure 18

THE DOUBLE -BRUCE ARRAY FOR 10,

15, AND 20 METERS

If a 600 -ohm feed line is used, the 20 -meter

array will also perform on 10 meters as o

Sterba curtain, with an approximate gain of

9 db.

dipole, and is a very simple beam to construct.

This antenna, like other so- called "broadside"

arrays, radiates maximum power at right angles

to the plane of the array.

The feed impedance of the Double Bruce is

about 750 ohms. The array may be fed with a

one -quarter wave stub made of 300 -ohm ribbon

line and a feedline made of 150 -ohm ribbon

line. Alternatively, the array may be fed di-

rectly with a wide- spaced 600 -ohm transmis-

sion line (figure 18). The feedline should

be brought away from the Double Bruce for a

short distance before it drops downward, to

prevent interaction between the feedline and

the lower part of the center phasing section of

the array. For best results, the bottom sec-

tions of the array should be one -half wave-

length above ground.

Arrays such as the X -array and the Double

Bruce are essentially high impedance devices,

and exhibit relatively broad -band characteris-

tics. They are less critical of adjustment than

a parasitic array, and they work well over a

wide frequency range such as is encountered

on the 28 -29.7 Mc. band.

The "Bi- Square" Illustrated in figure 19 is a

Broadside Array simple method of feeding a

small broadside array first

described by W6BCX several years ago as a

practical method of suspending an effective

array from a single pole. As two arrays of this

type can be supported at right angles from a

single pole without interaction, it offers a

solution to the problem of suspending two ar-

rays in a restricted space with a minimum of

erection work. The free space directivity gain

is slightly less than that of a Lazy I1, but is

www.americanradiohistory.com

HANDBOOK Broadside Arrays 465

GAIN APPROX. 6 DB

NON - RESONANT FELDER

GAIN APPROX. 8 DB

Figure 16

THE STERBA CURTAIN ARRAY

Approximate directive gains along with alter-

native feed methods are shown.

sent points of maximum current. All arrows

should point in the same direction in each por-

tion of the radiating sections of an antenna in

order to provide a field in phase for broadside

radiation. This condition is satisfied for the

arrays illustrated in figure 16. Figures 16A

and 16C show simple methods of feeding

a short Sterba curtain, while an alternative

method of feed is shown in the higher gain an-

tenna of figure 16B.

In the case of each of the arrays of figure

16, and also the "Lazy H" of figure 15, the

array may be made unidirectional and the gain

increased by 3 db if an exactly similar array

is constructed and placed approximately 14

wave behind the driven array. A screen or mesh

of wires slightly greater in area than the an-

tenna array may be used instead of an addi-

tional array as a reflector to obtain a unidirec-

tional system. The spacing between the re-

flecting wires may vary from 0.05 to 0.1 wave-

length with the spacing between the reflecting

wires the smallest directly behind the driven

elements. The wires in the untuned reflecting

system should be parallel to the radiating ele-

ments of the array, and the spacing of the com-

plete reflector system should be approximately

0.2 to 0.25 wavelength behind the driven ele-

ments.

On frequencies below perhaps 100 Mc. it

normally will be impracticable to use a wire -

screen reflector behind an antenna array such

as a Sterba curtain or a "Lazy H." Parasitic

elements may be used as reflectors or direc-

tors, but parasitic elements have the disadvan-

tage that their operation is selective with re-

spect to relatively small changes in frequency.

Nevertheless, parasitic reflectors for such ar-

rays are quite widely used.

The X -Array In section 23 -5 it was shown

how two dipoles may be arranged

in phase to provide a power gain of (some) 3

db. If two such pairs of dipoles are stacked

LAZY -H AND STERBA

(STACKED DIPOLE) DESIGN TABLE

FREQUENCY

IN MC. L, L. L,

7.0 68'2" 70' 35'

7.3 65'10" 67'6" 33'9"

14.0 34'1" 35' 17'6"

14.2 33'8" 34.7" 17'3"

14.4 33'4" 34'2" 17'

21.0 22'9" 23'3" 11'B"

21.5 22'3" 22'9" 11'5"

27.3 17'7" 17'10" 8'11"

28.0 17' 17'7" 8'9"

29.0 16'6" 17' 8'6"

50.0 9'7" 9'10" 4'11"

52.0 9'3" 9'S" 4'8"

54.0 8'10" 9'1" 4'6"

144.0 39.8" 40.5" 20.3'

146.0 39" 40" 20"

148.0 38.4" 39.5" 19.8"

www.americanradiohistory.com

464 High Frequency Directive Antennas THE RADIO

of a colinear antenna is proportional to the

overall length, whether the individual radiating

elements are 1/4 wave, 1/2 wave or 1/4 wave in

length.

Spaced Half The gain of two colinear half

Wave Antennas waves may be increased by

increasing the physical spac-

ing between the elements, up to a maximum of

about one half wavelength. If the half wave

elements are fed with equal lengths of trans-

mission line, poled correctly, a gain of about

3.3 db is produced. Such an antenna is shown

in figure 13. By means of a phase reversing

switch, the two elements may be operated out

of phase, producing a cloverleaf pattern with

slightly less maximum gain.

A three element "precut" array for 40 meter

operation is shown in figure 14. It is fed di-

rectly with 300 ohm "ribbon line," and may

be matched to a 52 ohm coaxial output trans-

mitter by means of a Balun, such as the Barker

& illiamson 3975. The antenna has a gain of

about 3.2 db, and a beam width at half -power

points of 40 degrees.

23 -6 Broadside Arrays

Colinear elements may be stacked above or

below another string of colinear elements to

produce what is commonly called a broadside

array. Such an array, when horizontal elements

are used, possesses vertical directivity in

proportion to the number of broadsided (ver-

tically stacked) sections which have been

used. Since broadside arrays do have good ver-

tical directivity their use is recommended on

the 14 -Mc. band and on those higher in fre-

quency. One of the most popular of simple

broadside arrays is the "Lazy 11" array of fig-

ure 15. Horizontal colinear elements stacked

two above two make up this antenna system

which is highly recommended for work on fre-

quencies above perhaps 14 -Mc. when moderate

gain without too much directivity is desired.

It has high radiation resistance and a gain of

approximately 5.5 db. The high radiation re-

sistance results in low voltages and a broad

resonance curve, which permits use of inex-

pensive insulators and enables the array to be

used over a fairly wide range in frequency.

For dimensions, see the stacked dipole design

table.

Stacked Vertical stacking may be applied

Dipoles to strings of colinear elements

longer than two half waves. In

such arrays, the end quarter wave of each

string of radiators usually is bent in to meet

LI Li

NON -RESONANT

FEED LIN CAIN APPROX. 5.5 DR

2

OUARTER-WAVE STUR

RESONANT FEED LINE

Figure 15

THE "LAZY H" ANTENNA SYSTEM

Stacking the colinear pairs gives both hori-

zontal and vertical directivity. As shown, the

array will give about 5.5 db gain. Note that

the array may be fed either at the center of

the phasing section or at the bottom; if fed

at the bottom the phasing section must be

twisted through 180 °.

a similar bent quarter wave from the opposite

end radiator. This provides better balance and

better coupling between the upper and lower

elements when the array is current -fed. Arrays

of this type are shown in figure 16, and are

commonly known as curtain arrays.

Correct length for the elements and stubs

can be determined for any stacked dipole array

from the Stacked -Dipole Design Table.

In the sketches of figure 16 the arrowheads

represent the direction of current flow at any

given instant. The dots on the radiators repre-

www.americanradiohistory.com

HANDBOOK Colinear Arrays 463

COLINEAR ANTENNA DESIGN CHART

FREQUENCY

IN MC. Li La L3

26.5 16'8' 17' e'6

21.2 22'e" 23'3 1+'e

14.2 33'e 34.7 17'3

7.15 e7' 66'6" 34'4

.0 120' 123' 61'6

3.e 133' 136'5- 68'2

Colinear The simple colinear antenna array

Arrays is a very effective radiating system

for the 3.5 -Mc. and 7.0 -Mc. bands,

but its use is not recommended on higher fre-

quencies since such arrays do not possess

any vertical directivity. The elevation radia-

tion pattern for such an array is essentially

the same as for a half -wave dipole. This con-

sideration applies whether the elements are of

normal length or are extended.

The colinear antenna consists of two or

more radiating sections from 0.5 to 0.65 wave-

lengths long, with the current in phase in each

section. The necessary phase reversal between

sections is obtained through the use of reso-

nant tuning stubs as illustrated in figure 11.

The gain of a colinear array using half -wave

elements (in decibels) is approximately equal

to the number of elements in the array. The

exact figures are as follows:

Number of Elements 2 3 4 5 6

Gain in Decibels 1.8 3.3 4.5 5.3 6.2

As additional in -phase colinear elements

are added to a doublet, the radiation resistance

goes up much faster than when additional half

waves are added out of phase (harmonic oper-

ated antenna).

For a colinear array of from 2 to 6 elements,

h~- e'(,`1 i' F(1AC)

PHASE -REVERSING SWITCH

FOR CLOVERLEAF PATTERN

Figure 13

TWO COLINEAR HALF -WAVE ANTENNAS

IN PHASE PRODUCE A 3 DB GAIN WHEN

SEPARATED ONE -HALF WAVELENGTH

FMC) RuCI

s

F(I)

A B

A-B =15011 FEED POINT GAINAPPROX. 3D6

Figure 12

DOUBLE EXTENDED ZEPP ANTENNA

For best results, antenna should be tuned to

operating frequency by means of griddip

oscillator.

the terminal radiation resistance in ohms at

any current loop is approximately 100 times

the number of elements.

It should be borne in mind that the gain from

a colinear antenna depends upon the sharpness

of the horizontal directivity since no vertical

directivity is provided. An array with several

colinear elements will give considerable gain,

but will have a sharp horizontal radiation

pattern.

Double Extended The gain of a conventional

Zepp two- element Franklin colin-

ear antenna can be increased

to a value approaching that obtained from a

three -element Franklin, simply by making the

two radiating elements 230° long instead of

180° long. The phasing stub is shortened cor-

respondingly to maintain the whole array in

resonance. Thus, instead of having 0.5 -wave-

length elements and 0.25-wavelength stub, the

elements are made 0.64 wavelength long and

the s tub is approximately 0.11 wavelength

long. Dimensions for the double extended Zepp

are given in figure 12.

The vertical directivity of a colinear anten-

na having 230° elements is the same as for

one having 180° elements. There is little ad-

vantage in using extended sections when the

total length of the array is to be greater than

about 1.5 wavelength overall since the gain

r'-- 65 e -Ai - 65 6 -H 65 e-+{

MID

32'9 -

MAKE STUBS OFl 14 E.

WIRE,SPACED TOe

32'9'

e ms

"-3OOR RIBBON TO

TRANSMITTER, ANY LENGTH

GAIN APPROX. 3 DB

Figure 14

PRE -CUT LINEAR ARRAY FOR 40 -METER

OPERATION

www.americanradiohistory.com

462 High Frequency Directive Antennas THE RADIO

f 4-

PLANE OF WIRES

END VIEW

14)

5= =

ISO. OUT OF PHASE I IN PHASE

(FLAT -TOP BEAM, ETC.) / (LAZY H, SIERRA CURTAIN)

Figure 10

RADIATION PATTERNS OF A PAIR OF

DIPOLES OPERATING WITH IN -PHASE

EXCITATION, AND WITH EXCITATION

180° OUT OF PHASE

If the dipoles are oriented horizontally most

of the directivity will be in the vertical

plane; if they are oriented vertically most

of the directivity will be in the horizontal

plane.

and 180° (45 °, 90 °, and 135° for instance),

the pattern is unsymmetrical, the radiation be-

ing greater in one direction than in the oppo-

site direction.

With spacings of more than 0.8 wavelength,

more than two main lobes appear for all phas-

ing combinations; hence, such spacings are

seldom used.

In -Phase With the dipoles driven so as to

Spacing be in phase, the most effective

spacing is between 0.5 and 0.7

wavelength. The latter provides greater gain,

but minor lobes are present which do not ap-

pear at 0.5- wavelength spacing. The radiation

is broadside to the plane of the wires, and the

gain is slightly greater than can be obtained

from two dipoles out of phase. The gain falls

off rapidly for spacings less than 0.375 wave-

length, and there is little point in using spac-

ing of 0.25 wavelength or less with in -phase

dipoles, except where it is desirable to in-

crease the radiation resistance. (See Multi -

Wire Doublet.)

Out of Phase When the dipoles are fed 180°

Spacing out of phase, the directivity is

through the plane of the wires,

and is greatest with close spacing, though

there is but little difference in the pattern

after the spacing is made less than 0.125

wavelength. The radiation resistance becomes

so low for spacings of less than 0.1 wave-

length that such spacings are not practicable.

Lt

---- L2

NON- RESONANT

FEED LIN

L2 Lt

- _ - -----e--

L! L3

QUARTER-WAVE STUBS

GAIN APPROS 4 S DB

Figure 11

THE FRANKLIN OR COLINEAR

ANTENNA ARRAY

An antenna of this type, regardless of the

number of elements, attains all of its direc-

tivity through sharpening of the horizontal

or azimuth radiation pattern; no vertical di-

rectivity is provided. Hence a long antenna

of this type has an extremely sharp azimuth

pattern, but no vertical directivity.

In the three foregoing examples, most of the

directivity provided is in a plane at a right

angle to the wires, though when out of phase,

the directivity is in a line through the wires,

and when in phase, the directivity is broadside

to them. Thus, if the wires are oriented verti-

cally, mostly horizontal directivity will be

provided. If the wires are oriented horizontally,

most of the directivity obtained will be verti-

cal directivity.

To increase the sharpness of the directivity

in all planes that include one of the wires,

additional identical elements are added in the

line of the wires, and fed so as to be in phase.

The familiar H array is one array utilizing

both types of directivity in the manner pre-

scribed. The two -section Kraus flat -top beam

is another.

These two antennas in their various forms

are directional in a horizontal plane, in addi-

tion to being low -angle radiators, and are per-

haps the most practicable of the bidirectional

stacked -dipole arrays for amateur use. More

phased elements can be used to provide great-

er directivity in planes including one of the

radiating elements. The fl then becomes a

Sterba- curtain array.

For unidirectional work the most practicable

stacked -dipole arrays for amateur -band use

are parasitically- excited systems using rela-

tively close spacing between the reflectors

and the directors. Antennas of this type are

described in detail in a later chapter. The

next most practicable unidirectional array is

an H or a Sterba curtain with a similar system

placed approximately one -quarter wave behind.

The use of a reflector system in conjunction

with any type of stacked -dipole broadside ar-

ray will increase the gain by 3 db.

www.americanradiohistory.com

HANDBOOK The Rhombic Antenna 461

Figure 8

TYPICAL RHOMBIC

ANTENNA DESIGN

The antenna system illus-

trated above may be used

over the frequency ronge from

7 to 29 Mc. without change.

The directivity of the system

may be reversed by the sys-

tem discussed in the text.

J,

LINE TO TX

N14 SPACED e'

SPACING BETWEEN SIDES S. 214 FEET

TOTAL LENGTH 5112 FEET

H50

TERMINATING LINE

OF 250' OF N 26

NICHROME SPACED 6"

AND B00 -OHM 16 -WATT

CARBON RESISTOR AT

END 6 2-WATT 100-OHM

RESISTORS IN SERIES

A considerable amount of directivity is lost

when the terminating resistor is left off the

end and the system is operated as a resonant

antenna. If it is desired to reverse the direc-

tion of the antenna it is much better practice

to run transmission lines to both ends of the

antenna, and then run the terminating line to

the operating position. Then with the aid of

two d -p -d -t switches it will be possible to con-

nect either feeder to the antenna changeover

switch and the other feeder to the terminating

line, thus reversing the direction of the array

and maintaining the same termination for

either direction of operation.

Figure 7 gives curves for optimum- design

rhombic antennas by both the maximum -out-

put method and the alignment method. The

alignment method is about 1.5 db down from

the maximum output method but requires only

about 0.74 as much leg length. The height and

tilt angle is the same in either case. Figure

8 gives construction data for a recommended

rhombic antenna for the 7.0 through 29.7 Mc.

bands. This antenna will give about 11 db

gain in the 14.0 -Mc. band. The approximate

gain of a rhombic antenna over a dipole, both

above normal soil, is given in figure 9.

23 -5 Stacked -Dipole

Arrays

The characteristics of a half -wave dipole

already have been described. When another

dipole is placed in the vicinity and excited

either directly or parasitically, the resultant

radiation pattern will depend upon the spacing

and phase differential, as well as the relative

magnitude of the currents. With spacings less

than 0.65 wavelength, the radiation is mainly

broadside to the two wires (bidirectional) when

there is no phase difference, and through the

wires (end fire) when the wires are 180° out

of phase. With phase differences between 0°

Figure 9

RHOMBIC ANTENNA GAIN

Showing the theoretical gain of a rhombic

antenna, in terms of the side length, over a

half -wave antenna mounted at the same

height above the same type of soil.

ILI ie

J 215

p 14

w 1

312

LL 11

J1

= 3

CC

w

Ó

CO e

z 3

Z

30

...........mm%_

..

../NM

........1,,.. .....

.......,...........

....MM............

.... III ..............

.r...._...

WI ..... ..........

..01.........

. / ........ .........

. ..................

II N........

M. .. . .

2 3 4 5 6 7 6 S 10 11 R 13 14 15 le Ti 16 /f 20

Il "LENGTH OF EACH LEG OF RHOMBIC IN WAVELENGTHS

www.americanradiohistory.com

460 High Frequency Directive Antennas THE RADIO

23 -4 The Rhombic

Antenna

The terminated rhombic or diamond is prob-

ably the most effective directional antenna

that is practical for amateur communication.

This antenna is non -resonant, with the result

that it can be used on three amateur bands,

such as 10, 20, and 40 meters. When the an-

tenna is non -resonant, i.e., properly termi-

nated, the system is undirectional, and the

wire dimensions are not critical.

Rhombic When the free end is terminated

Termination with a resistance of a value

between 700 and 800 ohms the

backwave is eliminated, the forward gain is

increased, and the antenna can be used on

several bands without changes. The terminat-

ing resistance should be capable of dissipat-

ing one -third the power output of the trans-

mitter, and should have very little reactance.

For medium or low power transmitters, the

non -inductive plaque resistors will serve as

a satisfactory termination. Several manufactur-

ers offer special resistors suitable for termi-

nating a rhombic antenna. The terminating de-

vice should, for technical reasons, present a

small amount of inductive reactance at the

point of termination.

A compromise terminating device commonly

used consists of a terminated 250 -foot or

longer length of line, made of resistance wire

which does not have too much resistance per

unit length. If the latter qualification is not

met, the reactance of the line will be exces-

sive. A 250 -foot line consisting of no. 25

nichrome wire, spaced 6 inches and terminated

with 800 ohms, will serve satisfactorily. Be-

cause of the attenuation of the line, the

lumped resistance at the end of the line need

dissipate but a few watts even when high pow-

er is used. A half -dozen 5000 -ohm 2 -watt car-

bon resistors in parallel will serve for all ex-

cept very high power. The attenuating line

may be folded back on itself to take up less

room.

The determination of the best value of termi-

nating resistor may be made while receiving,

if the input impedance of the receiver is ap-

proximately 800 ohms. The value of resistor

which gives the best directivity on reception

will not give the most gain when transmitting,

but there will be little difference between the

two conditions.

The input resistance of the rhombic which

is reflected into the transmission line that

feeds it is always somewhat less than the

terminating resistance, and is around 700 to

750 ohms when the terminating resistor is 800

ohms.

o

.... H u a no

c..,ww.cwa .a.n

.... . . .. . .. .. .

.

r r M Ir n.

WAVE ANGLE A ar sr

Figure 7

RHOMBIC ANTENNA DESIGN TABLE

Design data is given in terms of the wave

angle (vertical angle of transmission and re-

ception) of the antenna. The lengths I are

for the "maximum output" design; the shorter

lengths I' are for the "alignment" method

which gives approximately 1.5 db less gain

with o considerable reduction in the space

required for the antenna. The values of side

length, tilt angle, and height for a given

wave angle are obtained by drawing o ver-

tical line upward from the desired wave

angle.

Zr

The antenna should be fed with a non -reso-

nant line having a characteristic impedance

of 650 to 700 ohms. The four corners of the

rhombic should be at least one -half wave-

length above ground for the lowest frequency

of operation. For three -band operation the

proper tilt angle ,;4 for the center band should

be observed.

The rhombic antenna transmits a horizon-

tally- polarized wave at a relatively low angle

above the horizon. The angle of radiation

(wave angle) decreases as the height above

ground is increased in the same manner as

with a dipole antenna. The rhombic should

not be tilted in any plane. In other words, the

poles should all be of the same height and

the plane of the antenna should be parallel

with the ground.

www.americanradiohistory.com

HANDBOOK The V Antenna 459

TRAN3MIT

b ' RECEIVE

Figure 5

TYPICAL "V" BEAM ANTENNA

for a long wire. The reaction of one upon the

other removes two of the four main lobes, and

increases the other two in such a way as to

form two lobes of still greater magnitude.

The correct wire lengths and the degree of

the angle b are listed in the V- Antenna Design

Table for various frequencies in the 10 -, 20-

and 40 -meter amateur bands. Apex angles for

all side lengths are given in figure 4. The

gain of a "V" beam in terms of the side length

when optimum apex angle is used is given in

figure 6.

The legs of a very long V antenna are usual-

ly so arranged that the included angle is twice

the angle of the major lobe from a single wire

if used alone. This arrangement concentrates

the radiation of each wire along the bisector

of the angle, and permits part of the other

lobes to cancel each other.

üith legs shorter than 3 wavelengths, the

best directivity and gain are obtained with a

somewhat smaller angle than that determined

by the lobes. Optimum directivity for a one -

wave V is obtained when the angle is 90°

13

1:/

i 1P.i

>lo11SL

6/ 3 L w. / t 7 I yl M16

o / Jul 3=1111/411011111

M . u 411(Ii

p l Z - 2 I z If

RIMIIIIIME111101111111.11MMIEll

oo 2 2 4 3 6 7

LENGTH OF SIDE "L" 10 11 12

Figure 6

DIRECTIVE GAIN OF A "V" BEAM

This curve shows the approximate directive

gain of a V beam with respect to a half -wave

antenna located the same distance above

ground, in terms of the side length L.

rather than 180 °, as determined by the ground

pattern alone.

If very long wires are used in the V, the

angle between the wires is almost unchanged

when the length of the wires in wavelengths

is altered. However, an error of a few degrees

causes a much larger loss in directivity and

gain in the case of the longer V than in the

shorter one.

The vertical angle at which the wave is

best transmitted or received from a horizontal

V antenna depends largely upon the included

angle. The sides of the V antenna should be

at least a half wavelength above ground; com-

mercial practice dictates a height of approxi-

mately a full wavelength above ground.

V- ANTENNA DESIGN TABLE

FREQUENCY

IN KILOCYCLES L= ñ

6'so L =2r

6° 70 L =4a

6 =52 L =BT

6F39

26000 34'6" 69'6 140' 260'

29000 33.6 67.3" 135' 271'

21100 45'9" 91.9" 163' 366'

21300 45'4" 91'4" 162'6 365'

14050 69' 139' 279' 356'

14150 66'6" 136' 277' 555'

14250 66'2' 137' 275' 552'

7020 136'2' 276' 556' 1120'

7100 136.6' 275' 552' 1106'

7200 134.10" 271' 545' 1060'

www.americanradiohistory.com

458 High Frequency Directive Antennas THE RADIO

LONG- ANTENNA DESIGN TABLE

APPROXIMATE LENGTH IN FEET-END-FED ANTENNAS

FREQUENCY

IN M 1A IZA 2X 2 +A 3A 3 X 4A 4ZA

29 33 50 67 84 101 118 135 152

26 34 52 69 67 104 122 140 157

21.4 45 66 911/2 114 1/2 136 1/2 160 1/2 165 1/2 2091/2

21.2 45 1M 66 1/4 91 3/4 114 3/4 136 3.41 160 3.4 163 3/4 209 3/4

21.0 451/2 66 1/2 92 115 137 161 166 210

14.2 67 1/2 102 137 171 206 240 275 310

14.0 88 1/2 103 1/2 I39 174 209 244 279 31

7.3 138 206 276 346 418 66 555 625

7.15 136 1/2 207 277 347 17 467 557 627

7.0 137 207 1/2 277 1/2 348 416 488 356 628

.0 240 362 465 616 730 633 977 1100

3.6 232 361 511 60 770 900 1030 1160

3.6 268 403 540 676 612 950 1090 1220

3.5 274 414 555 696 633 977 1120

2.0 480 725 972 1230 1473

1.9 304 763 1020 1280

1.6 532 605 1060

One of the most practical methods of feed-

ing a long -wire antenna is to bring one end of

it into the radio room for direct connection to

a tuned antenna circuit which is link- coupled

through a harmonic- attenuating filter to the

transmitter. The antenna can be tuned effec-

tively to resonance for operation on any har-

monic by means of the tuned circuit which is

connected to the end of the antenna. A ground

is sometimes connected to the center of the

tuned coil.

If desired, the antenna can be opened and

current -fed at a point of maximum current b'

means cf low- impedance ribbon line, or by a

quarter -wave matching section and open line.

23 -3 The V Antenna

If two long -wire antennas are built in the

form of a V, it is possible to make two of the

maximum lobes of one leg shoot in the same

direction as two of the maximum lobes of the

other leg of the V. The resulting antenna is

bidirectional (two opposite directions) for the

main lobes of radiation. Each side of the V

can be made any odd or even number of quarter

wavelengths, depending on the method of feed-

ing the apex of the V. The complete system

must be a multiple of half waves. If each leg

is an even number of quarter waves long, the

antenna must be voltage -fed at the apex; if an

odd number of quarter waves long, current feed

must be used.

By choosing the proper apex angle, figure

4 and figure 5, the lobes of radiation from the

two long -wire antennas aid each other to form

a bidirectional beam. Each wire by itself

would have a radiation pattern similar to that

ISO

140

120

40

20

o o 4 6

LENGTH IN "L' WAVELENGTHS

10 12

Figure 4

INCLUDED ANGLE FOR A

"V" BEAM

Showing the included angle be-

tween the legs of a V beam for

various leg lengths. For opti-

mum alignment of the radiation

lobe at the correct vertical

angle with leg lengths less thon

three wavelengths, the optimum

Included angle is shown by the

dashed curve.

www.americanradiohistory.com

HANDBOOK Long Wire Radiators 457

Figure 3

DIRECTIVE GAIN OF

LONG -WIRE ANTENNAS

2

°o

LONG STRAIGHT WIRE ANTENNAS

2 3 4 3 7 e s

DB POWER RATIO OF MAIN LOBE TO A DIPOLE

10

Types of There is an enormous vari-

Directive Arrays ety of directive antenna ar-

rays that can give a substan-

tial power gain in the desired direction of trans-

mission or reception. However, some are more

effective than others requiring the same space.

In general it may be stated that long -wire an-

tennas of various types, such as the single

long wire, the V beam, and the rhombic, are

less effective for a given space than arrays

composed of resonant elements, but the long -

wire arrays have the significant advantage

that they may be used over a relatively large

frequency range while resonant arrays are usa-

ble only over a quite narrow frequency band.

23 -2 Long Wire Radiators

Harmonically operated long wires radiate

better in certain directions than others, but

cannot be considered as having appreciable

directivity unless several wavelengths long.

The current in adjoining half -wave elements

flows in opposite directions at any instant,

and thus, the radiation from the various ele-

ments adds in certain directions and cancels

in others.

A half -wave do u b l e t in free space has a

"doughnut" of radiation surrounding it. A full

wave has 2 lobes, 3 half waves 3, etc. When

the radiator is made more than 4 half wave-

lengths long, the end lobes (cones of radia-

tion) begin to show noticeable power gain over

a half -wave doublet, while the broadside lobes

get smaller and smaller in amplitude, even

though numerous (figure 2).

The horizontal radiation pattern of such an-

tennas depends upon the vertical angle of radi-

ation being considered. If the wire is more

than 4 wavelengths long, the maximum radia-

tion at vertical angles of 15° to 20° (useful

for dx) is in line with the wire, being slightly

greater a few degrees either side of the wire

than directly off the ends. The directivity of

the main lobes of radiation is not particularly

sharp, and the minor lobes fill in between the

main lobes to permit working stations in near-

ly all directions, though the power radiated

broadside to the radiator will not be great if

the radiator is more than a few wavelengths

long. The directive gain of long -wire antennas,

in terms of the wire length in wavelengths is

given in figure 3.

To maintain the out -of -phase condition in

adjoining half -wave elements throughout the

length of the radiator, it is necessary that a

harmonic antenna be fed either at one end or

at a current loop. If fed at a voltage loop, the

adjacent sections will be fed in phase, and a

different radiation pattern will result.

The directivity of a long wire does not in-

crease very much as the length is increased

beyond about 15 wavelengths. This is due to

the fact that all long -wire antennas are ad-

versely affected by the r -f resistance of the

wire, and because the current amplitude be-

gins to become unequal at different current

loops, as a result of attenuation along the wire

caused by radiation and losses. As the length

is increased, the tuning of the antenna be-

comes quite broad. In fact, a long wire about

15 waves long is practically aperiodic, and

works almost equally well over a wide range

of frequencies.

www.americanradiohistory.com

456 High Frequency Directive Antennas THE RADIO

7,

,,.

,°.

,O. S0.

20

,O

n.

M11Ì11111

0. 111111

- 1111

1111

DOUBLE HOP

U1111III . 11111 111111

1111111 h. 1101 111111

__ 11111 111111

1111h HID

_eÌ

SINGLE HOP 1 1111.,11

1111111 1 1 UMW

30 SO 100 300 500 ,000 3000

GREAT CIRCLE DISTANCE IN MILES

ro 000

Figure 1

OPTIMUM ANGLE OF RADIATION

WITH RESPECT TO DISTANCES

Shown above is o plot of the optimum angle

of radiation for one -hop and two -hop com-

munication. An operating frequency close to

the optimum working frequency for the com-

munication distance is assumed.

use a directive antenna than to increase trans-

mitter power, if more than a few watts of power

is being used.

Directive antennas for the high- frequency

range have been designed and used commer-

cially with gains as high as 23 db over a sim-

ple dipole radiator. Gains as high as 35 db are

common in direct -ray microwave communication

and radar systems. A gain of 23 db represents

a power gain of 200 times and a gain of 35 db

represents a power gain of almost 3500 times.

However, an antenna with a gain of only 15 to

20 db is so sharp in its radiation pattern that

it is usable to full advantage only for point -

to -point work.

The increase in radiated power in the de-

sired direction is obtained at the expense of

radiation in the undesired directions. Power

gains of 3 to 12 db seem to be most practi-

cable for amateur communication, since the

width of a beam with this order of power gain

is wide enough to sweep a fairly large area.

Gains of 3 to 12 db represent effective trans-

mitter power increases from 2 to 16 times.

Horizontal Pattern There is a certain optimum

vs. Vertical Angle vertical angle of radiation

for sky -wave communica-

tion, this angle being dependent upon distance,

frequency, time of day, etc. Energy radiated at

an angle much lower than this optimum angle

is largely lost, while radiation at angles much

W1111'14 ir

..®:4

1p v v 1 I

,.',, 1 I ,.,

'.I.iiat- ' l j --

1 í ÿ s

et* q `,.

ihoir , `'1 ..I 711G.++..-,

HALT WAVE ANT. -- FULL WAVE ANT. 2 WAVES ANT.

HORIZONTAL ANTENNAS IN FREE SPACE

Figure 2

FREE -SPACE FIELD PATTERNS OF

LONG -WIRE ANTENNAS

The presence of the earth distorts the field

pattern in such a manner that the azimuth

pattern becomes a function of the elevation

angle.

higher than this optimum angle oftentimes is

not nearly so effective.

For this reason, the horizontal directivity

pattern as measured on the ground is of no im-

port when dealing with frequencies and dis-

tances dependent upon sky -wave propagation.

It is the horizontal directivity (or gain or dis-

crimination) measured at the most useful ver-

tical angles of radiation that is of conse-

quence. The horizontal radiation pattern, as

measured on the ground, is considerably differ-

ent from the pattern obtained at a vertical

angle of 15 °, and still more different from a

pattern obtained at a vertical angle of 30 °.

In general, the energy which is radiated at

angles higher than approximately 30° above

the earth is effective at any frequency only for

local work.

For operation at frequencies in the vicinity

of 14 Mc., the most effective angle of radiation

is usually about 15° above the horizon, from

any kind of antenna. The most effective angles

for 10 -meter operation are those in the vicinity

of 10 °. Figure 1 is a chart giving the optimum

vertical angle of radiation for sky -wave propa-

gation in terms of the great -circle distance be-

tween the transmitting and receiving antennas.

www.americanradiohistory.com

CHAPTER TWENTY -THREE

High Frequency Antenna Arrays

It is becoming of increasing importance in

most types of radio communication to be cap-

able of concentrating the radiated signal from

the transmitter in a certain desired direction

and to be able to discriminate at the receiver

against reception from directions other than

the desired one. Such capabilities involve the

use of directive antenna arrays.

Few simple antennas, except the single ver-

tical element, radiate energy equally well in

all azimuth (horizontal or compass) directions.

All horizontal antennas, except those specifi-

cally designed to give an omnidirectional azi-

muth radiation pattern such as the turnstile,

have some directive properties. These proper-

ties depend upon the length of the antenna in

wavelengths, the height above ground, and the

slope of the radiator.

The various forms of the half -wave horizon-

tal antenna produce maximum radiation at right

angles to the wire, but the directional effect

is not great. Nearby objects also minimize the

directivity of a dipole radiator, so that it hard-

ly seems worth while to go to the trouble to

rotate a simple half -wave dipole in an attempt

to improve transmission and reception in any

direction.

The half -wave doublet, folded dipole, zepp,

single- wire -fed, matched impedance, and John-

son Q antennas all have practically the same

radiation pattern when properly built and ad-

455

justed. They all are dipoles, and the feeder

system, if it does not radiate in itself, will

have no effect on the radiation pattern.

23 -1 Directive Antennas

When a multiplicity of radiating elements is

located and phased so as to reinforce the radi-

ation in certain desired directions and to neu-

tralize radiation in other directions, a direc-

tive antenna array is formed.

The function of a directive antenna when

used for transmitting is to give an increase in

signal strength in some direction at the ex-

pense of radiation in other directions. For re-

ception, one might find useful an antenna giv-

ing little or no gain in the direction from which

it is desired to receive signals if the antenna

is able to discriminate against interfering sig-

nals and static arriving from other directions.

A good directive transmitting antenna, however,

can also be used to good advantage for recep-

tion. If radiation can be confined to a narrow

beam, the signal intensity can be increased a

great many times in the desired direction of

transmission. This is equivalent to increasing

the power output of the transmitter. On the

higher frequencies, it is more economical to

www.americanradiohistory.com

Figure 45

REAR VIEW OF

TUNER SHOWING

PLACEMENT OF

MAJOR COMPON-

ENTS

Rotary inductor is driv-

n by Johnson 116.208-

4 counter dial. Coaxial

Input receptacle JI

Is mounted directly be-

low rotary inductor.

ohm termination. The transmitter is turned on

(preferably at reduced input) and resonance is

established in the amplifier tank circuit. The

sensitivity control of the tuner is adjusted to

provide near full scale deflection on the bridge

meter. Various settings of Si, L2, and Cl

should be tried to obtain a reduction of bridge

reading. As tuner resonance is approached,

the meter reading will decrease and the sensi-

tivity control should be advanced. When the

system is in resonance, the meter will read

zero. All loading adjustments may then be

made with the transmitter controls. The tuner

should be readjusted whenever the frequency

of the transmitter is varied by an appreciable

amount.

Figure 46

CLOSE -UP OF SWR BRIDGE

Simple SWR bridge is mounted

below the chassis of the tuner.

Carbon resistors are mounted to

two copper rings to form low

inductance one -ohm resistor.

Bridge capacitors form triangular

configuration for lowest lead

inductance. Balancing capacitor

C2 is at lower right.

www.americanradiohistory.com

HANDBOOK Single -wire Antenna Tuner 453

Figure 43

ANTENNA TUNER IS HOUSED

IN METAL CABINET 7' x 8" IN

SIZE.

Inductance switch SI and sensi-

tivity control are at left with

counter dial for L2 at center.

Output tuning capacitor CI is at

right. SWR meter is mounted

above SI.

52 a INPUT

FROM XMTR R1 n

5

250

C2 25 1Q Si

SINGLE

WIRE

L2 ANT

5 010 V

MICA

SENSITIVITY

0 -1

L1- 35 TURNS e 18, 2- DIA.,

3.9- LONG (A /R -DL/.e)

TAP AT 15 T., 27 T.,

FROM POINT A

L2- JOHNSON 229 -207 VARIABL

INDUCTOR (10 NH)

TUNE

C1 'll

330 =

2RV.

CI- JOHNSON 350E20

C2- CENTRALA8 TYPE 822

J1 -TYPE SO -239 RECEPTACLE

E R1-TEN 10-OHM 1 -WATT CAR-

BON RESISTORS IN PARA-

LLEL. INC TYPE LTA

Figure 44

SCHEMATIC, SINGLE -WIRE

ANTENNA TUNER

mum (clockwise) position. The bridge is

balanced when the input impedance of the

tuner is 52 ohms resistive. This is the con-

dition for maximum energy transfer between

transmission line and antenna. The meter is

graduated in arbitrary units, since actual SWR

value is not required.

Tuner Major parts placement in

Construction the tuner is shown in

figures 43 and 45. Tapped

coil L1 is mounted upon 1-inch ceramic in-

sulators, and all major components are mounted

above deck with the exception of the SWR

bridge (figure 46). The components of the

bridge are placed below deck, adjacent to

the coaxial input plug mounted on the rear

apron of the chassis. The ten 10 -ohm resistors

are soldered to two 1 -inch rings made of copper

wire as shown in the photograph. The bridge

capacitors are attached to this assembly with

extremely short leads.The 1N56 crystal mounts

at right angles to the resistors to insure mini-

mum amount of capacitive coupling between the

resistors and the detector. The output lead

from the bridge passes through a ceramic feed -

thru insulator to the top side of the chassis.

Connection to the antenna is made by means

of a large feedthru insulator mounted on the

back of the tuner cabinet. This insulator is

not visible in the photographs.

Bridge The SWR bridge must be

Calibration calibrated for 52 ohm ser-

vice. This can be done by

temporarily disconnecting the lead between the

bridge and the antenna tuner and connecting a

2 -watt, 52 ohm carbon resistor to the junction

of R1 and the negative terminal of the 1N56

diode. The opposite lead of the carbon resistor

is grounded to the chassis of the bridge. A

small amount of r -f energy is fed to the input

of the bridge until a reading is obtained on

the r -f voltmeter. The 25 mmfd bridge balancing

capacitor C2 (see figure 46) is then adjusted

with a fibre -blade screwdriver until a zero

reading is obtained on the meter. The sensi-

tivity control is advanced as the meter null

grows, in order to obtain the exact point of

bridge balance. When this point is found, the

carbon resistor should be removed and the

bridge attached to the antenna tuner. The

bridge capacitor is sealed with a drop of nail

polish to prevent misadjustment.

Tuner All tuning adjustments are

Adjustments made to obtain proper

transmitter loading with a

balanced (zero meter reading) bridge condition.

The tuner is connected to the transmitter

through a random length of 52 ohm coaxial

line, and the single wire antenna is attached

to the output terminal of the tuner. Transmitter

loading controls are set to approximate a 52

www.americanradiohistory.com

452 Antennas and Antenna Matching THE RADIO

PARALLEL -WIRE TO

40-e0 M. ANTENNA

r

L

TO COAX. LINES TO

RECEIVER IOM. ANT 20M ANT

1

.001 CERAMICS

TO

TRANSMITTER THROUGH

HARMONIC FILTER

Figure 41

ALTERNATIVE COAXIAL ANTENNA

COUPLER

This circuit is recommended not only as be-

ing most desirable when coaxial lines with

low s.w.r. are being used to feed antenna

systems such as rotatable beams, but when

It also Is desired to feed through open -wire

line to some sort of multi -band antenna for

the lower frequency ranges. The tuned cir-

cuit of the antenna coupler is operative only

when using the open -wire feed, and then It

is In operation both for transmit and receive.

in such an application will be found to be ade-

quate, since harmonic attenuation has been

accomplished ahead of the antenna coupler.

However, the circuit will be easier to tune,

although it will not have as great a bandwidth,

if the operating Q is made higher.

An alternative arrangement shown in figure

41 utilizes the antenna coupling tank circuit

only when feeding the coaxial output of the

transmitter to the open -wire feed line (or simi-

lar multi -band antenna) of the 40- 80 meter an-

tenna. The coaxial lines to the 10 -meter beam

and to the 20 -meter beam would be fed directly

from the output of the coaxial antenna change-

over relay through switch S.

22 -12 A Single -Wire Antenna Tuner

One of the simplest and least expensive

antennas for transmission and reception is

the single wire, end -fed Hertz antenna. When

used over a wide range of frequencies, this

type of antenna exhibits a very great range of

input impedance. At the low frequency end of

the spectrum such an antenna may present a

resistive load of less than one ohm to the

transmitter, combined with a large positive or

negative value of reactance. As the frequency

of operation is raised, the resistive load may

55 IL

COAXIAL

LINE

FROM

XMTR

SWR

INDICATOR

SINGLE WIRE

ANTENNA

Figure 42

ANTENNA TUNER AND SWR

INDICATOR FOR RANDOM

LENGTH HERTZ ANTENNA

rise to several thousand ohms (near half -wave

resonance) and the reactive component of the

load can rapidly change from positive to nega-

tive values, or vice -versa.

It is possible to match a 52 -ohm trans-

mission line to such an antenna at almost any

frequency between 1.8 me and 30 me with the

use of a simple tuner of the type shown in

figure 42. A variable series inductor L, and a

variable shunt capacitor Cl permit circuit

resonance and impedance transformation to

be established for most antenna lengths.

Switch S1 permits the selection of series

capacitor C for those instances when the

single wire antenna exhibits large values of

positive reactance.

To provide indication for the tuning of the

network, a radio frequency bridge (SWR meter)

is included to indicate the degree of mis-

match (standing wave ratio) existing at the

input to the tuner. All adjustments to the

tuner are made with the purpose of reaching

unity standing wave ratio on the coaxial feed

system between the tuner and the transmitter.

A Practical A simple antenna tuner for

Antenna Tuner use with transmitters of

250 watts power or less

is shown in figures 43 through 46. A SWR

bridge circuit is used to indicate tuner re-

sonance. The resistive arm of the bridge con-

sists of ten 10 -ohm, 1 -watt carbon resistors

connected in parallel to form a 1 -ohm resistor

(R1). The other pair of bridge arms are ca-

pacitive rather than resistive. The bridge

detector is a simple r -f voltmeter employing a

1N56 crystal diode and a 0 -1 d.c. milliammeter.

A sensitivity control is incorporated to prevent

overloading the meter when power is first

applied to the tuner. Final adjustments are

made with the sensitivity control at its maxi-

www.americanradiohistory.com

HANDBOOK Antenna Couplers 451

ter, assuming that the antenna feed line is be-

ing operated with a low standing -wave ratio.

However, there are many cases where it is de-

sirable to feed a multi -band antenna from the

output of the harmonic filter, where a tuned

line is being used to feed the antenna, or

where a long wire without a separate feed line

is to be fed from the output of the harmonic

filter. In such cases an antenna coupler is re-

quired.

Some harmonic attenuation will be provided

by the antenna coupler, particularly if it is

well shielded. In certain cases when a pi net-

work is being used at the output of the trans-

mitter, the addition of a shielded antenna cou-

pler will provide sufficient harmonic attenua-

tion. But in all normal cases it will be neces-

sary to include a harmonic filter between the

output of the transmitter and the antenna cou-

pler. When an adequate harmonic filter is be-

ing used, it will not be necessary in normal

cases to shield the antenna coupler, except

from the standpoint of safety or convenience.

Function of an The function of the antenna

Antenna Coupler coupler is, basically, to

transform the impedance of

the antenna system being used to the correct

value of resistive impedance for the harmonic

filter, and hence for the transmitter. Thus the

antenna coupler may be used to resonate the

feeders or the radiating portion of the antenna

system, in addition to its function of imped-

ance transformation.

It is important to remember that there is

nothing that can be done at the antenna cou-

pler which will eliminate standing waves on

the antenna transmission line. Standing waves

are the result of reflection from the antenna,

and the coupler can do nothing about this con-

dition. However, the antenna coupler can reso-

nate the feed line (by introducing a conjugate

impedance) in addition to providing an imped-

ance transformation. Thus, a resistive imped-

ance of the correct value can be presented to

the harmonic filter, as in figure 36, regardless

of any reasonable value of standing -wave ratio

on the antenna transmission line.

Types of All usual types of antenna

Antenna Couplers couplers fall into two clas-

sifications: (1) inductively

coupled resonant systems as exemplified by

those shown in figure 39, and (2) conductively

coupled pi- network systems such as shown in

figure 40. The inductively -coupled system is

much more commonly used, since it is conven-

ient for feeding a balanced line from the co-

axial output of the usual harmonic filter. The

pi- network system is most useful for feeding

a length of wire from the output of a trans-

mitter.

TRANSMITTER HARMONIC

FILTER

COAX TO

RECEIVER

COAX ANT.

CHANGEOVER

RELAY

SINGLE WIRE

ANTENNA

Figure 40

PI- NETWORK

ANTENNA COUPLER

An arrangement such as illustrated

above is convenient for feeding an

end -fed Hertz antenna, or a random

length of wire for portable or emer-

gency operation, from the nominal

value of impedance of the harmonic

filter.

Several general methods for using the induc-

tively- coupled resonant type of antenna cou-

pler are illustrated in figure 39. The coupling

between the link coil L and the main tuned cir-

cuit need not be variable; in fact it is prefer-

able that the correct link size and placement

be determined for the tank coil which will be

used for each band, and then that the link be

made a portion of the plug -in coil. Capacitor

C then can be adjusted to a pre- determined

value for each band such that it will resonate

with the link coil for that band. The reactance

of the link coil (and hence the reactance of the

capacitor setting which will resonate the coil)

should be about 3 or 4 times the impedance of

the transmission line between the antenna cou-

pler and the harmonic filter, so that the link

coupling circuit will have an operating Q of

3 or 4. The use of capacitor C to resonate with

the inductance of the link coil L will make it

easier to provide a low standing -wave ratio

to the output of the harmonic filter, simply by

adjustment of the antenna- coupler tank circuit

to resonance. If this capacitor is not included,

the system still will operate satisfactorily, but

the tank circuit will have to be detuned slight-

ly from resonance so as to cancel the induc-

tive reactance of the coupling link and thus

provide a resistive load to the output of the

harmonic filter. Variations in the loading of

the final amplifier should be made by the cou-

pling adjustment at the final amplifier, not at

the antenna coupler.

The pi- network type of antenna coupler, as

shown in figure 40 is useful for certain appli-

cations, but is primarily useful in feeding a

single -wire antenna from a low- impedance

transmission line. In such an application the

operating Q of the pi network may be somewhat

lower than that of a pi network in the plate cir-

cuit of the final amplifier of a transmitter, as

shown in figure 38. An operating Q of 3 or 4

www.americanradiohistory.com

450 Antennas and Antenna Matching THE RADIO

U

COAX. TO

RECEIVER

HARMONIC LJ COAX ANT.

TRANSMITTER FILTER ¡ CNANGEOVEq

RELAY

CEPP

FEEDERS

O

©

L PARALLEL-WIRE

LINE TO ANTENNA

SINGLE-WIRE

ANTENNA OR

FEEDLINE

O

SINGLE -WIRE

HERTZ ANTENNA

OR FEEDER

COAX. LINE

TO ANTENNA

Figure 39

ALTERNATIVE ANTENNA -COUPLER CIRCUITS

Plug -in coils, one or two variable capacitors of the split -stator variety, and a system of

switches or plugs and jacks may be used in the antenna coupler to accomplish the feeding of

different types of antennas and antenna transmission lines from the coaxial input line from the

transmitter or from the antenna changeover relay. Link L should be resonated with capacitor

C at the operating frequency of the transmitter so that the harmonic filter will operate into o

resistive load impedance of the correct nominal value.

ended output stage down to the 50 -ohm imped-

ance of the usual harmonic filter and its sub -

sequent load.

In a pi network of this type the harmonic

attenuation of the section will be adequate

when the correct value of C, and L are being

used and when the r e s on a n t dip in C, is

sharp. If the dip in C, is broad, or if the plate

current persists in being too high with C2 at

maximum setting, it means that a greater value

of capacitance is required at C2, assuming that

the values of C, and L are correct.

22 -11 Antenna Couplers

As stated in the previous section, an anten-

na coupler is not required when the impedance

of the antenna transmission line is the same

as the nominal impedance of the harmonic fil-

www.americanradiohistory.com

HANDBOOK Pi- Network Coupling Systems 449

SHIELO

COAX. TO

RECEIVER

i

C 1 HARMONIC 1 COAX ANT. I TO ANTENNA

oAK---- ATTENUATING CHAANGEOVER - - - - FEEDLINE OR

1 1

CO ANTENNA

COUPLER

Figure 37

TUNED -LINK OUTPUT CIRCUIT

Capacitor C should be adjusted so os to tune out the inductive reactance of the coupling link,

L. Loading of the amplifier then is varied by physically varying the coupling between the plate

tank of the final amplifier and the antenna coupling link,

Pi- Network The pi- network coupling system

Coupling offers two advantages: (1) a me-

chanical coupling variation is

not required to vary the loading of the final am-

plifier, and (2) the pi network (if used with an

operating Q of about 15) offers within itself a

harmonic attenuation of 40 db or more, in ad-

dition to the harmonic attenuation provided by

the additional harmonic attenuating filter. Some

commercial equipments (such as the Collins

amateur transmitters) incorporate an L network

in addition to the pi network, for accomplish-

ing the impedance transformation in two steps

and to provide additional harmonic attenuation.

Tuning the Tuning of a pi- network

Pi- Section Coupler coupling circuit such as

illustrated in figure 38 is

accomplished in the following manner: First

remove the connection between the output of

the amplifier and the harmonic filter (load).

Tune C2 to a capacitance which is large for

the band in use, adding suitable additional ca-

pacitance by switch S if operation is to be on

one of the lower frequency bands. Apply re-

duced plate voltage to the stage and dip to

resonance with C,. It may be necessary to vary

the inductance in coil L, but in any event reso-

nance should be reached with a setting of C,

which is approximately correct for the desired

value of operating Q of the pi network.

Next, couple the load to t h e amplifier

(through the harmonic filter), apply reduced

plate voltage again and dip to resonance with

C,. If the plate current dip with load is too low

(taking into consideration the reduced plate

voltage), decrease the capacitance of C2 and

again dip to resonance, repeating the proce-

dure until the correct value of plate current is

obtained with full plate voltage on the stage.

There should be a relatively small change re-

quired in the setting of C, (from the original

setting of C, without load) if the operating Q

of the network is correct and if a large value

of impedance transformation is being em-

ployed-as would be the case when transform-

ing from the plate impedance of a single-

COAX TO

RECEIVER

HARMONIC COAXIAL

_ATTENUATING = ANTENNA

FILTER r ,CHANGEOVER

RELAY

TO FCCDLINE

-MOR ANTENNA

COUPLER

Figure 38

PI- NETWORK ANTENNA COUPLER

The design of pi-network output circuits is discussed in Chapter Thirteen. The additional output -

end shunting capacitors selected by switch S are for use on the lower frequency ranges. Induc-

tor L may be selected by a tap switch, it may be continuously variable, or plug -in inductors

may be used.

www.americanradiohistory.com

448 Antennas and Antenna Matching THE RADIO

I EXCITER

PORTION rI NAL

AMPLIFIE COUPLING

RDJUSTMENT

AT TRANSMITTER

HARMONIC

ATTENUATING

SYSTEM

ANTENNA

COUPLER

AT ANTENNA

IMPEDANCE

MATCHING

TRANSMISSION TANTENNA

LIN[

RADIATING

SYSTEM

Figure 36

ANTENNA COUPLING SYSTEM

The antenna coupling system illustrated above is for use when the antenna transmission line

does not have the same characteristic impedance as the TVI filter, and when the standing -wave

ratio on the antenna transmission line may or may not be low.

within or adjacent to the antenna. The feed

line coming down from the an t e n n a system

should have a characteristic impedance equal

to the nominal impedance of the harmonic fil-

ter, and the impedance matching at the anten-

na should be such that the standing -wave ratio

on the antenna feed line is less than 2 to 1

over the range of frequency to be fed to the

antenna. Such an arrangement may be used

with open -wire line, ribbon or tubular line, or

with coaxial cable. The use of coaxial cable

is to be recommended, but in any event the

impedance of the antenna transmission line

should be the same as the nominal impedance

of the harmonic filter. The arrangement of fig-

ure 35 is more or less standard for commercial-

ly manufactured equipment for amateur and

commercial use in the h -f and v -h -f range.

The arrangement of figure 36 merely adds

an antenna coupler between the output of the

harmonic attenuating filter and the antenna

transmission line. The antenna coupler will

have some harmonic- attenuating action, but its

main function is to transform the impedance

at the station end of the antenna transmission

line to the nominal value of the harmonic filter.

Hence the arrangement of figure 36 is more

general than the figure 35 system, since the

inclusion of the antenna coupler allows the

system to feed an antenna transmission line

of any reasonable impedance value, and also

without regard to the standing -wave ratio

which might exist on the antenna transmission

line. Antenna couplers are discussed in a fol-

lowing section.

Output Coupling It will be noticed by refer-

Adjustment ence to both figure 35 and

figure 36 that a box labeled

Coupling Adjustment is included in the block

diagram. Such an element is necessary in the

complete system to afford an adjustment in the

value of load impedance presented to the tubes

in the final amplifier stage of the transmitter.

The impedance at the input terminal of the

harmonic filter is established by the antenna,

through its matching system and the antenna

coupler, if used. In any event the impedance

at the input terminal of the harmonic filter

should be very close to the nominal impedance

of the filter. Then the Coupling Adjustment

provides means for transforming this imped-

ance value to the correct operating value of

load impedance which should be presented to

the final amplifier stage.

There are two common ways for accomplish-

ing the antenna coupling adjustment, as illus-

trated in figures 37 and 38. Figure 37 shows

the variable -link arrangement most commonly

used in home -constructed equipment, while the

pi- netowrk coup ling arrangement commonly

used in commercial equipment is illustrated in

figure 38. Either method may be used, and each

has its advantages.

Variable -Link The variable -link method il-

Coupling lustrated in figure 37 has the

advantage that standard man-

ufactured components m a y be used with no

changes. However, for greatest bandwidth of

operation of the coupling circuit, the reactance

of the link coil, L, and the reactance of the

link tuning capacitor, C, should both be be-

tween 3 and 4 times the nominal load imped-

ance of the harmonic filter. This is to say that

the inductive reactance of the coupling link L

should be tuned out or resonated by capacitor

C, and the operating Q of the L -C link circuit

should be between 3 and 4. If the link coil is

not variable with respect to the tank coil of

the final amplifier, capacitor C may be used

as a loading control; however, this system is

not recommended since its use will require

adjustment of C whenever a frequency change

is made at the transmitter. If L and C are made

resonant at the center of a band, with a link

circuit Q of 3 to 4, and coupling adjustment is

made by physical adjustment of L with respect

to the final amplifier tank coil, it usually will

be possible to operate over an entire amateur

band without change in the coupling system.

Capacitor C normally may have a low voltage

rating, even with a high power transmitter, due

to the low Q and low impedance of the coupling

circuit.

www.americanradiohistory.com

HANDBOOK Antenna Coupling Systems 447

When insulators are subject to very high r -f

voltages, they should be cleaned occasionally

if in the vicinity of sea water or smoke. Salt

scum and soot are not readily dislodged by

rain, and when the coating becomes heavy

enough, the efficiency of the insulators is

greatly impaired.

If a very pretentious installation is to be

made, it is wise to check up on both under-

writer's rules and local ordinances which

might be applicable. If you live anywhere near

an airport, and are contemplating a tall pole,

it is best to investigate possible regulations

and ordinances pertaining to towers in the dis-

trict, before starting construction.

22 -10 Coupling to the

Antenna System

When coupling an antenna feed system to a

transmitter the most important considerations

are as follows: (1) means should be provided

for varying the load on the amplifier; (2) the

two tubes in a push -pull amplifier should be

equally loaded; (3) the load presented to the

final amplifier should be resistive (non -reac-

tive) in character; and (4) means should be

provided to reduce harmonic coupling between

the final amplifier plate tank circuit and the

antenna or antenna transmission line to an ex-

tremely low value.

The Transmitter- The problem of coupling the

Loading Problem power output of a high -fre-

quency or v -h -f transmitter

to the radiating portion of the antenna system

has been materially complicated by the virtual

necessity for eliminating interference to TV re-

ception. However, the TVI- elimination portion

of the problem may always be accomplished

by adequate shielding of the transmitter, by

filtering of the control and power leads which

enter the transmitter enclosure, and by the in-

clusion of a harmonic -attenuating filter be-

tween the output of the transmitter and the an-

tenna system.

Although TVI may be eliminated through in-

clusion of a filter between the output of a

shielded transmitter and the antenna system,

the fact that such a filter must be included in

the link between transmitter and antenna makes

it necessary that the transmitter -loading prob-

lem be re- evaluated in terms of the necessity

for inclusion of such a filter.

Harmonic- attenuating filters must be oper-

ated at an impedance level which is close to

their design value; therefore they must operate

into a resistive termination substantially equal

to the characteristic impedance of the filter.

If such filters are operated into an impedance

which is not resistive and approximately equal

to their characteristic impedance: (1) the ca-

pacitors used in the filter sections will be

subjected to high peak voltages and may be

damaged, (2) the harmonic- attenuating proper-

ties of the filter will be decreased, and (3) the

impedance at the input end of the filter will

be different from that seen by the filter at the

load end (except in the case of the half -wave

type of filter). It is therefore important that

the filter be included in the transmitter-to-an -

tenna circuit at a point where the impedance

is close to the nominal value of the filter, and

at a point where this impedance is likely to

remain fairly constant with variations in fre-

quency.

Block Diagrams of There are two basic

Transmitter -to- Antenna arrangements which

Coupling Systems include all the provi-

sions required in the

transmitter -to- antenna coupling system, and

which permit the harmonic -attenuating filter to

be placed at a position in the coupling system

where it can be operated at an impedance

level close to its nominal value. These ar-

rangements are illustrated in block -diagram

form in figures 35 and 36.

The arrangement of figure 35 is recommend-

ed for use with a single -band antenna system,

such as a dipole or a rotatable array, wherein

an impedance matching system is included

r-

E%CITER

PORTION

1NIELD

FINAL

AMPLIFIER

AT TRANSMITTER

HARMONIC

COUPLING _ TTENUATI

Ap1U5TMENT SYSTEM TRANSMISSION

LINE

AT ANTENNA

I MPE DANCED RADIATING

MATCHING SYSTEM

T ANTENNAn

Figure 35

ANTENNA COUPLING SYSTEM

The harmonic suppressing antenna coupling system illustrated above is for use when the anten-

na transmission line has a low standing -wave ratio, and when the characteristic impedance of

the antenna transmission line is the same as the nominal impedance of the low -pass harmonic -

attenuating filter.

www.americanradiohistory.com

4 4 6 Antennas and Antenna M atching THE RADIO

waiting for it to show excessive wear or de-

terioration.

It is an excellent idea to tie both ends of

the halyard line together in the manner of a

flag -pole line. Then the antenna is tied onto

the place where the two ends of the halyard

are joined. This procedure of making the hal-

yard into a loop prevents losing the top end

of the halyard should the antenna break near

the end, and it also prevents losing the hal-

yard completely should the end of the halyard

carelessly be allowed to go free and be pulled

through the pulley at the top of the mast by

the antenna load. A somewhat longer piece

of line is required but the insurance is well

worth the cost of the additional length of rope.

Trees as Often a tall tree can be called up-

Supports on to support one end of an anten-

na, but one should not attempt to

attach anything to the top, as the swaying of

the top of the tree during a heavy wind will

complicate matters.

If a tree is utilized for support, provision

should be made for keeping the antenna taut

without submitting it to the possibility of be-

ing severed during a heavy wind. This can be

done by the simple expedient of using a pul-

ley and halyard, with weights attached to the

lower end of the halyard to keep the antenna

taut. Only enough weight to avoid excessive

sag in the antenna should be tied to the hal-

yard, as the continual swaying of the tree sub-

mits the pulley and halyard to considerable

wear. Galvanized iron pipe, or steel -tube conduit,

is often used as a vertical radiator, and is

quite satisfactory for the purpose. However,

when used for supporting antennas, it should

be remembered that the grounded supporting

poles will distort the field pattern of a verti-

cally polarized antenna unless spaced some

distance from the radiating portion.

Painting The life of a wood mast or pole can

be increased several hundred per

cent by protecting it from the elements with a

coat or two of paint. And, of course, the ap-

pearance is greatly enhanced. The wood should

first be given a primer coat of flat white out-

side house paint, which can be thinned down

a bit to advantage with second -grade linseed

oil. For the second coat, which should not be

applied until the first is thoroughly dry, alumi-

num paint is not only the best from a preserva-

tive standpoint, but looks very well. This type

of paint, when purchased in quantities, is con-

siderably cheaper than might be gathered from

the price asked for quarter -pint cans.

Portions of posts or poles below the surface

of the soil can be protected from termites and

moisture by painting with creosote. While not

so strong initially, redwood will deteriorate

much more slowly when buried than will the

white woods, such as pine.

Antenna Wire The antenna or array itself

presents no especial problem.

A few considerations should be borne in mind,

however. For instance, soft -drawn copper

should not be used, as even a short span will

stretch several per cent after whipping around

in the wind a few weeks, thus affecting the

resonant frequency. Enameled -copper wire,

as ordinarily available at radio stores, is us-

ally soft drawn, but by tying one end to some

object such as a telephone pole and the other

to the frame of an auto, a few husky tugs can

be given and the wire, after stretching a bit,

is equivalent to hard drawn.

Where a long span of wire is required, or

where heavy insulators in the center of the

span result in considerable tension, copper -

clad steel wire is somewhat better than hard -

drawn copper. It is a bit more expensive,

though the cost is far from prohibitive. The

use of such wire, in conjunction with strain

insulators, is advisable, where the antenna

would endanger persons or property should it

break.

For transmission l i n e s and tuning stubs

steel -core or hard -drawn wire will prove awk-

ward to handle, and soft -drawn copper should,

therefore, be used. If the line is Ion g, the

strain can be eased by supporting it at several

points.

More important from an electrical standpoint

than the actual size of wire used is the sol-

dering of joints, especially at current loops

in an antenna of low radiation resistance. In

fact, it is good practice to solder all joints,

thus insuring quiet operation when the anten-

na is used for receiving.

Insulation A question that often arises is

that of insulation. It depends, of

course, upon the r -f voltage at the point at

which the insulator is placed. The r -f voltage,

in turn, depends upon the distance from a cur-

rent node, and the radiation resistance of the

antenna. Radiators having low radiation re-

sistance have very high voltage at the voltage

loops; consequently, better than usual insula-

tion is advisable at those points.

Open -wire lines operated as non -resonant

lines have little voltage across them; hence

the most inexpensive ceramic types are suffi-

ciently good electrically. With tuned lines, the

voltage depends upon the amplitude of the

standing waves. If they are very great, the

voltage will reach high values at the voltage

loops, and the best spacers available are none

too good. At the current loops the voltage is

quite low, and almost anything will suffice.

www.americanradiohistory.com

HANDBOOK Antenna Construction 445

Figure 34

TWO SIMPLE WOOD MASTS

Shown at (A) is the method of as-

sembly, and at (B) is the completed

structure, of the conventional "A-

frame" antenna most. At (C) is

shown a structure which is heavier

but more stable than the A -frame

for heights above about 40 feet.

2X4

SAWHORSES /-\\ 2X2

CROSSPIECES

Iv-S GROUND LEVEL i I

CONCRETE éd s'

'9JJéi

©

if a gin pole about 20 feet high is installed

about 30 or 40 feet to the rear of the direction

in which the antenna is to be raised. A line

from a pulley on the top of the gin pole is then

run to the top of the pole to be raised. The

gin pole comes into play when the center of

the mast has been raised 10 to 20 feet above

the ground and an additional elevated pull is

required to keep the top of the mast coming

up as the center is raised further above

ground.

Using TV Masts Steel tubing masts of the

telescoping variety are wide-

ly available at a moderate price for use in sup-

porting television antenna arrays. These masts

usually consist of several 10 -foot lengths of

electrical metal tubing (EMT) of sizes such

that the sections will telescope. The 30 -foot

and 40 -foot lengths are well suited as masts

for supporting antennas and arrays of the

types used on the amateur bands. The masts

are constructed in such a manner that the bot-

tom 10 -foot length may be guyed permanently

before the other sections are raised. Then the

upper sections may be extended, beginning

with the top -mast section, until the mast is at

full length(provided a strong wind is not blow-

ing) following which all the guys may be an-

chored. It is important that there be no load

on the top of the mast when the "vertical"

raising method is to be employed.

Guy Wires Guy wires should never be pulled

taut; a small amount of slack is

desirable. Galvanized wire, somewhat heavier

than seems sufficient for the job, should be

used. The heavier wire is a little harder to

handle, but costs only a little more and takes

longer to rust through. Care should be taken

to make sure that no kinks exist when the pole

or tower is ready for erection, as the wire will

be greatly weakened at such points if a kink

is pulled tight, even if it is later straightened.

If "dead men" are used for the guy wire

terminations, the wire or rod reaching from the

dead men to the surface should be of non -rust-

ing material, such as brass, or given a heavy

coating of asphalt or other protective sub-

stance to prevent destructive action by the

damp soil. Galvanized iron wire will last only

a short time when buried in moist soil.

Only strain -type (compression) insulators

should be used for guy wires. Regular ones

might be sufficiently strong for the job, but it

is not worth taking chances, and egg -type

strain halyard insulators are no more ex-

pensive.

Only a brass or bronze p u l l e y should be

used for the halyard, as a high pole with a

rusted pulley is truly a sad affair. The bear-

ing of the pulley should be given a few drops

of heavy machine oil before the pole or tower

is raised. The halyard itself should be of good

material, preferably water -proofed. Hemp rope

of good quality is better than window sash

cord from several standpoints, and is less ex-

pensive. Soaking it thoroughly in engine oil of

medium viscosity, and then wiping it off with

a rag, will not only extend its life but minim-

ize shrinkage in wet weather. Because of the

difficulty of replacing a broken halyard it is

a good idea to replace it periodically, without

www.americanradiohistory.com

444 Antennas and Antenna Matching THE RADIO

ance of 123 ohms. Z, is one -quarter wavelength

long at the mid -frequency and has an imped-

ance of 224 ohms. 4 is the balanced line to

be matched(in this case 300 ohms) and may be

any length.

Other system parameters for different output

and input impedances may be calculated from

the following:

Transformation ratio (r) for each section is:

N Z

r= Zrn

Where N is the number of sections. In the

above case,

3 Z 5

r Z

Impedance between sections, as Z,_ is r

times the preceding section. Z=_, = r X Z and

= r X Z,_,

Mid -frequency (m):

m= F, + F,

2

7 +30

For 40 -20 -10 meters = - 18.5 Mc.

2

and one -quarter wavelength = 12 feet.

For 20 -10 -6 meters - 14 + 54

- - 34 Mc.

2

and one -quarter wavelength = 5.5 feet.

The impedances of the sections are:

Z, =V Z, X Z,_,

Z, = V Z:-, X Z3-4

Z4 -7- \7Z74.727

Zo= XZ,

Generally, the larger number of taper sec-

tions the greater will be the bandwidth of the

system.

22 -9 Antenna

Construction

The foregoing portion of this chapter has

been concerned primarily with the electrical

characteristics and considerations of anten-

nas. Some of the physical aspects and mech-

anical problems incident to the actual erec-

tion of antennas and arrays will be discussed

in the following section.

Up to 60 feet, there is little point in using

mast -type antenna supports unless guy wires

either must be eliminated or kept to a mini-

mum. while a little more difficult to erect, be-

cause of their floppy nature, fabricated wood

poles of the type to be described will be just

as satisfactory as more rigid types, provided

many guy wires are used.

Rather expensive when purchased through

the regular channels, 40- and 50 -foot tele-

phone poles sometimes can be obtained quite

reasonably. In the latter case, they are hard

to beat, inasmuch as they require no guying

if set in the ground six feet (standard depth),

and the resultant pull in any lateral direction

is not in excess of a hundred pounds or so.

For heights of 80 to 100 feet, either three -

sided or four -sided lattice type masts are most

practicable. They can be made self- support-

ing, but a few guys will enable one to use a

smaller cross section without danger from

high winds. The torque exerted on the base

of a high self- supporting mast is terrific dur-

ing a strong wind.

The "A- Frame" Figures 34A and 34B show

Most the standard method of con-

struction of the A -frame

type of mast. This type of mast is quite fre-

quently used since there is only a moderate

amount of work involved in the construction

of the assembly and since the material cost is

relatively small. The three pieces of selected

2 by 2 are first set up on three sawhorses or

boxes and the holes drilled for the three 1/4-

inch bolts through the center of the assembly.

Then the base legs are spread out to about 6

feet and the bottom braces installed. Then the

upper b r aces and the cross pieces are in-

stalled and the assembly given several coats

of good -quality paint as a protection against

weathering.

Figure 34C shows another common type of

mast which is made up of sections of 2 by 4

placed end -to -end with stiffening sections of

1 by 6 bolted to the edge of the 2 by 4 sec-

tions. Both types of masts will require a set

of top guys and another set of guys about one -

third of the way down from the top. Two guys

spaced about 90 to 100 degrees and pulling

against the load of the antenna will normally

be adequate for the top guys. Three guys are

usually used at the lower level, with one di-

rectly behind the load of the antenna and two

more spaced 120 degrees from the rear guy.

The raising of the mast is made much easier

www.americanradiohistory.com

HANDBOOK Matching Systems 443

- L FEET 46B

Ì` =

F Mc

TUBING Q MATCHING SECTION

_ 234

L FIMCI Zo= Z2

So

UNTUNED LINE

ANY LENGTH

z

Figure 32

HALF -WAVE RADIATOR FED

BY "Q BARS"

The Q matching section is simply o quarter -

wave transformer whose impedance is equal

to the geometric mean between the imped-

ance at the center of the antenna and the

impedance of the transmission line to be

used to feed the bottom of the transformer.

The transformer may be made up of parallel

tubing, a four -wire line, or any other type

of transmission line which has the correct

value of impedance.

ally can be obtained by either designing or

adjusting the matching section for a dipole to

have a surge impedance that is the geometric

mean between the line impedance and 72 ohms,

the latter being the theoretical radiation re-

sistance of a half -wave doublet either infi-

nitely high or a half wave above a perfect

ground.

Though the radiation resistance may depart

somewhat from 72 ohms under actual condi-

tions, satisfactory results will be obtained

with this assumed value,so long as the dipole

radiator is more than a quarter wave above

effective earth, and reasonably in the clear.

The small degree of standing waves intro-

duced by a slight mismatch will not increase

the line losses appreciably, and any small

amount of reactance present can be tuned out

at the transmitter termination with no bad ef-

fects. If the reactance is objectionable, it may

be minimized by making the untuned line an

integral number of quarter waves long.

A Q- matched system can be adjusted pre-

cisely, if desired, by constructing a matching

section to the calculated dimensions with pro-

vision for varying the spacing of the Q sec-

tion conductors slightly, after the untuned

line has been checked for standing waves.

Center to Impedance Impedance

Center in Ohms in Ohms

Spacing for %z" for I/4 "

in Inches Diameters Diameters

1 170 250

1.25 188 277

1.5 207 298

1.75 225 318

2 248 335

PARALLEL TUBING SURGE IMPEDANCE FOR

MATCHING SECTIONS

The Collins The advantage of unbal-

Transmission Line anced output networks

Matching System fo r transmitters a r e nu-

merous; however this out-

put system becomes awkward when it is de-

sired to feed an antenna system utilizing a

balanced i n p u t. For some time the Collins

Radio Co. has been experimenting with a bal-

un and tapered line system for matching a co-

axial output transmitter to an open -wire bal-

anced transmission line. Considerable success

has been obtained and matching systems good

over a frequency range as great as four to one

have been developed. Illustrated in figure 33

is one type of matching system which is prov-

ing satisfactory over this range. Z, is the

transmitter end of the system and may be any

length of 52 -ohm coaxial cable. Z2 is one -

quarter wavelength long at the mid -frequency

of the range to be covered and is made of 75

ohm coaxial cable. ZA is a quarter- wavelength

shorted section of cable at the mid -frequency.

ZD (ZA and Z2) forms a 200 -ohm quarter -wave

section. The ZA section is formed of a con-

ductor of the same diameter as Z2. The differ-

ence in length between ZA and Z2 is accounted

for by the fact that Z2 is a coaxial conductor

with a solid dielectric, whereas the dielectric

for ZD is air. Z2 is one -quarter wavelength

long at the mid -frequency and has an imped-

Zo ZJ Z4

If__,_,,./____n____ ZS

ZI Zz I

---- v 2J=123R Z= 22411 Z5=30011

(ANY LENGTH)

ZA }

INNER L OUTER CONDUCTORS

SHORTED AT EACH END

Figure 33

COLLINS TRANSMISSION LINE MATCHING

SYSTEM

A wide -band system for matching a 52 -ohm

coaxial line to a balanced 300 -ohm line over

a 4:7 wide frequency range.

www.americanradiohistory.com

442 Antennas and Antenna Matching THE RADIO

The open stub should be resonated in the

same manner as the shorted stub before at-

taching the transmission line; however, in

this case, it is necessary to prune the stub

to resonance, as there is no shorting bar.

Sometimes it is handy to have a stub hang

from the radiator to a point that can be reached

from the ground, in order to facilitate adjust-

ment of the position of the transmission -line

attachment. For this reason, a quarter -wave

stub is sometimes made three -quarters wave-

length long at the higher frequencies, in order

to bring the bottom nearer the ground. Opera-

tion with any odd number of quarter waves is

the same as for a quarter -wave stub.

Any number of half waves can be added to

either a quarter -wave stub or a half -wave stub

without disturbing the operation, though losses

and frequency sensitivity will be lowest if

the shortest usable stub is employed. See fig-

ure 31.

Stub Length

(Electrical) Current -Fed

Radiator Voltage -Fed

Radiator

1/4- % -1 t %.etc.

wavelengths

Y2 -1 -1 i5.2 -etc .

wavelength s

Open Shorted

Stub Stub

Shorted Open

Stub Stub

Linear R -F A resonant quarter -wave line

Transformers has the unusual property of

acting much as a transformer.

Let us take, for example, a section consisting

of no. 12 wire spaced 6 inches, which happens

to have a surge impedance of 600 ohms. Let

the far end be terminated with a pure resist-

ance, and let the near end be fed with radio -

frequency energy at the frequency for which

the line is a quarter wavelength long. If an

impedance measuring set is used to measure

the impedance at the near end while the im-

pedance at the far end is varied, an interest-

ing relationship between the 600 -ohm charac-

teristic surge impedance of this particular

quarter -wave matching line, and the imped-

ance at the ends will be discovered.

When the impedance at the far end of the

line is the same as the characteristic surge

impedance of the line itself (600 ohms), the

impedance measured at the near end of the

quarter -wave line will also be found to be

600 ohms.

Under these conditions, the line would not

have any standing waves on it, since it is

terminated in its characteristic impedance.

Now, let the resistance at the far end of the

line be doubled, or changed to 1200 ohms.

The impedance measured at the near end of

the line will be found to have been cut in

half, to 300 ohms. If the resistance at the far

end is made half the original value of 600

ohms or 300 ohms, the impedance at the near

end doubles the original value of 600 ohms,

and becomes 1200 ohms. As one resistance

goes up, the other goes down proportionately.

It will always be found that the character-

istic surge impedance of the quarter -wave

matching line is the geometric mean between

the impedance at both ends. This relationship

is shown by the following formula:

ZMS = ZA ZL

where

ZMS = Impedance of matching section.

ZA = Antenna resistance.

ZL = Line impedance.

Quarter -Wave

Matching

Transformers

The impedance inverting char-

acteristic of a quarter -wave

section of transmission line is

widely used by making such a

section of line act as a quarter -wave trans-

former. The Johnson Q feed system is a wide-

ly known application of the quarter -wave trans-

former to the feeding of a dipole antenna and

array consisting of two dipoles. However, the

quarter -wave transformer may be used in a

wide number of applications wherever a trans-

former is required to match two impedances

whose geometric mean is somewhere between

perhaps 25 and 750 ohms when transmission

line sections can be used. Paralleled coaxial

lines may be used to obtain the lowest im-

pedance mentioned, and open -wire lines com-

posed of small conductors spaced a moderate

distance may be used to obtain the higher im-

pedance. A short list of impedances, which

may be matched by quarter -wave sections of

transmission line having specified imped-

ances, is given below.

Load or Ant. 1

Impedance 300 480 600 r Feed -Line

Impedance

20 77 98 110 Quarter -

30 95 120 134 Wave

50 110 139 155 Transformer

75 150 190 212 Impedance

100 173 220 245

Johnson -Q The standard form of Johnson -

Feed System n feed to a doublet is shown

in figure 32. An impedance

match is obtained by utilizing a matching sec-

tion, the surge impedance of which is the geo-

metric mean between the transmission line

surge impedance and the radiation resistance

of the radiator. A sufficiently good match usu-

www.americanradiohistory.com

HANDBOOK Matching Systems 441

section. If they are not used, the T- section

will detune the dipole when the T- section is

attached to it. The two capacitors may be

ganged together, and once adjusted for mini-

mum detuning action, they may be locked. A

suitable housing should be devised to protect

these capacitors from the weather. Additional

information on the adjustment of the T -match

is given in the chapter covering rotary beam

antennas.

The "Gamma" Match An unbalanced version of

the T -match may be used

to feed a dipole from an unbalanced coaxial

line. Such a device is called a Gamma Match,

and is illustrated in figure 30.

The length of the Gamma rod and the spac-

ing of it from the dipole determine the imped-

ance level at the transmission line end of the

rod. The series capacitor is used to tune out

the reactance introduced into the system by

the Gamma rod. The adjustment of the Gamma

Match is discussed in the chapter covering

rotary beam antennas.

Matching Stubs By connecting a resonant

section of transmission line

(called a matching stub) to either a voltage or

current loop and attaching parallel -wire non -

resonant feeders to the resonant stub at a

suitable voltage (impedance) point, standing

waves on the line may be virtually eliminated.

The stub is made to serve as an auto- trans-

former. Stubs are particularly adapted to

matching an open line to certain directional

arrays, as will be described later.

Voltage Feed When the stub attaches to the

antenna at a voltage loop, the

stub should be a quarter wavelength long

electrically, and be shorted at the bottom end.

The stub can be resonated by sliding the

shorting bar up and down before the non -reso-

nant feeders are attached to the stub, the an-

tenna being shock -excited from a separate

radiator during the process. Slight errors in

the length of the radiator can be compensated

for by adjustment of the stub if both sides of

the stub are connected to the radiator in a

symmetrical manner. Where only one side of

the stub connects to the radiating system, as

in the Zepp and in certain antenna arrays, the

radiator length must be exactly right in order

to prevent excessive unbalance in the untuned

line. A dial lamp may be placed in the center of

the shorting stub to act as an r -f indicator.

Current Feed When a stub is used to current -

feed a radiator, the stub should

either be left open at the bottom end instead

of shorted, or else made a half wave long.

SHORTING BAR

ANTENNA

11 STUB

SHORTING BAR NON-RESONANT

FEEDERS

ANTENNA

NON- RESONANT

FEEDERS STUB

FEEDER TAPS NEAR

END OF STUB

OPEN

t7

STUB

NON- RESONANTSHORTING BAR

FEEDER

Figure 31

MATCHING -STUB APPLICATIONS

An end -fed half -wave antenna with a quarter -

wave shorted stub is shown at (A). (B) shows

the use of a half -wave shorted stub to feed

a relatively low impedance point such as the

center of the driven element of a parasitic

array, or the center of a half -wave dipole.

The use of an open -ended quarter -wave stub

to feed a low impedance is illustrated at

(C). (D) shows the conventional use of a

shorted quarter -wave stub to voltage feed

two half -wave antennas with a 180. phase

difference.

www.americanradiohistory.com

440 Antennas and Antenna Matching THE RADIO

-.:7°-LE.D- EL.T-T0. 0. I

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ON

Figure 29

FOLDED -ELEMENT MATCHING SYSTEMS

Drawing (A) above shows a half -wave made

up to two parallel wires. If one of the wires

is broken as in (B) and the feeder connected,

the feed -point impedance is multiplied by

four; such an antenna is commonly called a

"folded doublet." The feed -point impedance

for a simple half -wave doublet fed in this

manner is approximately 300 ohms, depend-

ing upon antenna height. Drawing (C) shows

how the feed -point impedance can be multi-

plied by a factor greater than four by making

the half of the element that is broken smaller

in diameter than the unbroken half. An ex-

tension of the principles of (B) and (C) is

the arrangement shown at (D) where the sec-

tion into which the feeders are connected is

considerably shorter than the driven element.

This system is most convenient when the

driven element is too long (such as for o

28 -Mc. or 14 -Mc. array) for a convenient

mechanical arrangement of the system shown

at (C).

wire of such a radiator, as shown in figure 29,

the effective feed -point resistance of the an-

tenna or array will be increased by a factor

of N2 where N is equal to the number of con-

ductors, all in parallel, of the same diameter

in the array. Thus if there are two conductors

of the same diameter in the driven element or

the antenna the feed -point resistance will be

multiplied by 22 or 4. If the antenna has a ra-

diation resistance of 75 ohms its feed -point

resistance will be 300 ohms, this is the case

DRIVEN ELEMENT

MOVEABLE CLAMP

GAMMA ROD

RESONATING CONDENSER

50 -70 OHM COAXIAL FEED LINE

Figure 30

THE GAMMA MATCH FOR CONNECTING

AN UNBALANCED COAXIAL LINE TO A

BALANCED DRIVEN ELEMENT

of the conventional folded- dipole as shown in

figure 29B.

If three wires are used in the driven radia-

tor the feed -point resistance is increased by

a factor of 9; if four wires are used the im-

pedance is increased by a factor of 16, and

so on. In certain cases when feeding a para-

sitic array it is desirable to have an imped-

ance step up different from the value of 4:1

obtained with two elements of the same dia-

meter and 9:1 with three elements of the same

diameter. Intermediate values of impedance

step up may be obtained by using two elements

of different diameter for the complete driven

element as shown in figure 29C. If the con-

ductor that is broken for the feeder is of small-

er diameter than the other conductor of the

radiator, the impedance step up will be greater

than 4:1. On the other hand if the larger of the

two elements is broken for the feeder the im-

pedance step up will be less than 4:1.

The "T" Match A method of matching a bal-

anced low- impedance trans-

mission line to the driven element of a para-

sitic array is the T match illustrated in figure

29D. This method is an adaptation of the

multi -wire doublet principle which is more

practicable for lower -frequency parasitic ax-

rays such as those for use on the 14 -Mc. and

28 -Mc. bands. In the system a section of tub-

ing of approximately one -half the diameter of

the driven element is spaced about four in-

ches below the driven element by means of

clamps which hold the T- section mechanically

and which make electrical connection to the

driven element. The length of the T- section

is normally between 15 and 30 inches each

side of the center of the dipole for transmis-

sion lines of 300 to 600 ohms impedance, as-

suming 28 -Mc. operation. In series with each

leg of the T- section and the transmission line

is a series resonating capacitor. These two

capacitors tune out the reactance of the T-

www.americanradiohistory.com

HANDBOOK Matching Systems 439

D -I

NON-RESONANT

LINE

MATCHING SECTION

Figure 28

THE DELTA- MATCHED DIPOLE

ANTENNA

The dimensions for the portions of the an.

tenno ore given in the text.

between three types of transmission line: (1)

Ribbon or tubular molded 300 -ohm line is

widely used up to moderate power levels (the

"transmitting" type is useable up to the kilo-

watt level). (2) Open -wire 400 to 600 ohm line

is most commonly used when the antenna is

some distance from the transmitter, because of

the low attenuation of this type of line. (3) Co-

axial line (usually RG -8 /U with a 52 -ohm

characteristic impedance) is widely used in

v -h -f work and also on the lower frequencies

where the feed line must run underground or

through the walls of a building. Coaxial line

also is of assistance in TVI reduction since

the r -f field is entirely enclosed within the

line. Molded 75 -ohm line is sometimes used

to feed a doublet antenna, but the doublet has

been largely superseded by the folded -dipole

antenna fed by 300 -ohm ribbon or tubular line

when an antenna for a single band is required.

Standing Waves As was discussed earlier,

standing waves on the anten-

na transmission line, in the transmitting case,

are a result of reflection from the point where

the feed line joins the antenna system. The

magnitude of the standing waves is deter-

mined by the degree of mismatch between the

characteristic impedance of the transmission

line and the input impedance of the antenna

system. When the feed -point impedance of the

antenna is resistive and of the same value as

the characteristic impedance of the feed line,

standing waves will not exist on the feeder.

It may be well to repeat at this time that there

is no adjustment which can be made at the

transmitter end of the feed line which will

change the magnitude of the standing waves

on the antenna transmission line.

Delta- Marched The delta type matched -im-

pedance antenna system is

shown in figure 28. The im-

pedance of the transmission line is trans-

formed gradually into a higher value by the

fanned -out Y portion of the feeders, and the

Y portion is tapped on the antenna at points

where the antenna impedance is a compromise

between the impedance at the ends of the Y

and the impedance of the unfanned portion of

the line.

The constants of the system are rather criti-

cal, and the antenna must resonate at the

operating frequency in order to minimize stand-

ing waves on the line. Some slight readjust-

ment of the taps on the antenna is desirable,

if appreciable standing waves persist in ap-

pearing on the line.

The constants for a doublet are determined

by the following formulas:

Antenna System

L feet -

Dfeet

467.4

F megacycles

175

F megacycles

147.6

Efeet

F megacycles

Where L is antenna length; D is the distance in

from each end at which the Y taps on; E is the

height of the Y section.

Since these constants are correct only for a

600 -ohm transmission line, the spacing S of

the line must be approximately 75 times the

diameter of the wire used in the transmission

line. For no. 14 B & S wire, the spacing will

be slightly less than 5 inches. This system

should never be used on either its even or odd

harmonics, as entirely different constants are

required when more than a single half wave-

length appears on the radiating portion of the

system.

Multi -Wire Doublets When a doublet antenna

or the driven element in

an array consists of more than one wire or

tubing conductor the radiation resistance of

the antenna or array is increased slightly as a

result of the increase in the effective diameter

of the element. Further, if we split just one

www.americanradiohistory.com

438 Antennas and Antenna Matching THE RADIO

!CwLtILMIw ro

wo wwL*oRa

TOS MSC -TOP VIA

APE, Z' HIGH

a2 Om. cODUL L«[

JOA« ALL wiAEST SASE L.wE

Figure 26

MECHANICAL CONSTRUCTION OF 20-

METER DISCONE

pending upon the wire size and the point of

attachment to the antenna. The earth losses

are comparatively low over ground of good con-

ductivity. Since the single wire feeder radiates,

it is necessary to bring it away from the an-

tenna at right angle s to the antenna wire

for at least one -half the length of the an-

tenna.

The correct point for best impedance match

on the fundamental frequency is not suitable

for harmonic operation of the antenna. In addi-

tion, the correct length of the antenna for

fundamental operation is not correct for har-

monic operation. Consequently, a compromise

VOLTAGE CURVES

40 MET R5 20v METERS IO METERS

\%

8 AN

33.5 FT.

87 FT.

CENTER

ANTENNA WIRE

FEEDER

Figure 27

SINGLE- WIRE -FED ANTENNA FOR ALL -

BAND OPERATION

An antenna of this type for 40 -, 20- and 10-

meter operation would have a radiator 67

feet long, with the feeder tapped 11 feet off

center. The feeder can be 33, 66 or 99 feet

long. The some type of antenna for 80 -, 40 -,

20- and 10 -meter operation would have a

radiator 134 feet long, with the feeder topped

22 feet off center. The feeder can be either

66 or 132 feet long. This system should be

used only with those coupling methods which

provide good harmonic suppression.

must be made in antenna length and point of

feeder connection to enable the single -wire-

fed antenna to operate on more than one band.

Such a compromise introduces additional re-

actance into the single wire feeder, and might

cause loading difficulties with pi- network

transmitters. To minimize this trouble, the

single wire feeder should be made a multiple

of 33 feet long.

Two typical single -wire -fed antenna sys-

tems are shown in figure 27 with dimensions

for multi -band operation.

22 -8 Matching Non -Resonant

Lines to the Antenna

Present practice in regard to the use of

transmission lines for feeding antenna systems

on the amateur bands is about equally divided

www.americanradiohistory.com

HANDBOOK Low Frequency Discone 437

i S

D

H

II

I I

II

I I

II

I

II

20,15.11,10,4 METERS

D. 12 L. 1e

S=10- R. is,

H.15' 7.

R

32 OHM COAXIAL

FEED LINE

DIMENSIONS

15, 1 1.10,6 METERS

D =e L= 12'

S. e. R =12,

H =10.5

11,10.6,2 METERS

D= e L=Ye-

S = R=ee

H=e'3^

Figure 24

DIMENSIONS OF LOW- FREQUENCY DIS-

CONE ANTENNA FOR LOW FREQUENCY

CUTOFF AT 13.2 MC., 20.1 MC., AND

26 MC.

The Discone is a vertically polarized radia-

tor, producing an omnidirectional pattern

similar to a ground plane. Operation on sev-

eral amateur bands with low SWR on the co-

axial feed line is possible. Additional in-

formation on L -F Discone by W2RYI in July,

1950 CQ magazine.

of the radials may be reduced to 25 feet. As

with all multi -band antennas that employ no

lumped tuned circuits, this antenna offers no

attenuation to harmonics of the transmitter.

When operating on the lower frequency band,

it would be wise to check the transmitter for

second harmonic emission, since this antenna

will effectively radiate this harmonic.

The Low - Frequency The discone antenna is

Discone widely used on the v -h -f

bands, but until recently

it has not been put to any great use on the

lower frequency bands. Since the discone is a

broad -band device, it may be used on several

harmonically related amateur bands. Size is

the limiting factor in the use of a discone, and

the 20 meter band is about the lowest practi-

cal frequency for a discone of reasonable di-

mensions. A discone designed for 20 meter

operation may be used on 20, 15, 11, 10 and

S`4.0

F

3.3

30

O z 25

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I

. ...........11

i x.o...

..1....

M.^¡E

e

CC l.s

m Io 14 le 22 26 30 34 3e 42 4e 50 54 5e

FREQUENCY (Mc)

Figure 25

SWR CURVE FOR A 13.2 MC. DISCONE

ANTENNA. SWR IS BELOW 1.5 TO 1 FROM

13.0 MC. TO 58 MC.

6 meters with excellent results. It affords a

good match to a 50 ohm coaxial feed system

on all of these bands. A practical discone an-

tenna is shown in figure 24, with a SWR curve

for its operation over the frequency range of

13 -55 Mc. shown in figure 25. The discone

antenna radiates a vertically polarized wave

and has a very low angle of radiation. For

v -h -f work the discone is constructed of sheet

metal, but for low frequency work it may be

made of copper wire and aluminum angle

stock. A suitable mechanical layout for a low

frequency di scone is shown in figure 26.

Smaller versions of this antenna may be con-

structed for 15, 11, 10 and 6 meters, or for 11,

10, 6 and 2 meters as shown in the chart of

figure 24.

For minimum wind resistance, the top "hat"

of the discone is constructed from three -quar-

ter inch aluminum angle stock, the rods being

bolted to an aluminum plate at the center of

the structure. The tips of the rods are all con-

nected together by lengths of no. 12 enamelled

copper wire. The cone elements are made of

no. 12 copper wire and act as guy wires for

the discone structure. A very rigid arrange-

ment may be made from this design; one that

will give no trouble in high winds. A 4" x 4"

post can be used to support the discone struc-

ture. The discone antenna may be fed by a length

of 50 -ohm coaxial cable directly from the trans-

mitter, with a very low SWR on all bands.

The Single -Wire- The old favorite single -wire-

Fed Antenna fed antenna system is quite

satisfactory for an impromp-

tu all band antenna system. It is widely used

for portable installations and "Field Day"

contests where a simple, multi -band antenna

is required. A single wire feeder has a char-

acteristic impedance of some 500 ohms, de-

www.americanradiohistory.com

436 Antennas and Antenna Matching THE RADIO

of transmission line of any characteristic im-

pedance into a feeder system such as this and

the impedanc' at the far end of the line will

be exactly the same value of impedance which

the half -was e line sees at its termination.

Hence this has been done in the antenna sys-

tem shown in figure 22; an electrical half

wave of line has been inserted between the

feed point of the antenna and the 300 -ohm

transmission line to the transmitter.

The characteristic impedance of this addi-

tional half -wave section of transmission line

has been made about 715 ohms (no. 20 wire

spaced 6 inches), but since it is an electrical

half wave long at 7 Mc. and operates into a

load of 300 ohms at the antenna the 300 -ohm

Twin -Lead at the bottom of the half -wave sec-

tion still sees an impedance of 300 ohms. The

additional half -wave section of transmission

line introduces a negligible amount of loss

since the current flowing in the section of line

is the same which would flow in a 300 -ohm

line at each end of the half -wave section, and

at all other points it is less than the current

which would flow in a 300 -ohm line since the

effective impedance is greater than 300 ohms

in the center of the half -wave section. This

means that the loss is less than it would be in

an equivalent length of 300 -ohm TwinLead

since this type of manufactured transmission

line is made up of conductors which are equiv-

alent to no. 20 wire.

So we see that the added section of 715 -ohm

line has substantially no effect on the opera-

tion of the antenna system on the 7 -Mc. band.

However, when the flat top of the antenna is

operated on the 3.5-Mc. band the feed -point

impedance of the flat top is approximately

3500 ohms. Since the section of 715 -ohm trans-

mission line is an electrical quarter -wave in

length on the 3.5-Mc. band, this section of

line will have the effect of transforming the

approximately 3500 ohms feed -point imped-

ance of the antenna down to an impedance of

about 150 ohms which will result in a 2:1

standing -wave ratio on the 300 -ohm Twin -Lead

transmission line from the transmitter to the

antenna system.

The antenna system of figure 22 operates

with very low standing waves over the entire

7 -Mc. band, and it will operate with moderate

standing waves from 3500 to 3800 kc. in the

3.5-Mc. band and with sufficiently low stand-

ing -wave ratio so that it is quite usable over

the entire 3.5 -Mc. band.

This antenna system, as well as all other

types of multi -band antenna systems, must be

used in conjunction with some type of har-

monic- reducing antenna tuning network even

though the system does present a convenient

impedance value on both bands.

L

I6Á -e0 METERS

300 OHM OPEN -WIRE L. TO'

TV TYPE LINE

5 OHM COAX IA INE

V=52'

60 -40 METERS

L =33'

V =2s'

/*P-6 RADIALS

Figure 23

THE MULTEE TWO -BAND ANTENNA

This compact antenna can be used with ex-

cellent results on 160/80 and 80/40 meters.

The feedline should be held as vertical as

possible, since it radiates when the antenna

is operated on its fundamental frequency.

The "Multee" An antenna that works well

Antenna on 160 and 80 meters, or 80

and 40 meters and is suffi-

ciently compact to permit erection on the aver-

age city lot is the W68CX Multee antenna,

illustrated in figure 23. The antenna evolves

from a vertical two wire radiator, fed on one

leg only. On the low frequency band the top

portion does little radiating, so it is folded

down to form a radiator for the higher frequen-

cy band. On the lower frequency band, the an-

tenna acts as a top loaded vertical radiator,

while on the higher frequency band, the flat-

top does the radiating rather than the vertical

portion. The vertical portion acts as a quarter -

wave linear transformer, matching the 6000

ohm antenna impedance to the 50 ohm imped-

ance of the coaxial transmission line.

The earth below a vertical radiator must be

of good conductivity not only to provide a low

resistance ground connection, but also to pro-

vide a good reflecting surface for the waves

radiated downward towards the ground. For

best results, a radial system should be in-

stalled beneath the antenna. For 160 -80 me-

ter operation, six radials 50 feet in length,

made of no. 16 copper wire should be buried

just below the surface of the ground. While an

ordinary water pipe ground system with no

radials may be used, a system of radials will

provide a worthwhile increase in signal

strength. For 80 -40 meter operation, the length

www.americanradiohistory.com

HANDBOOK Multi -band Antennas 435

144

33' OR NV LONG- 400 01.163 OPEN -WIRE

TV TYPE LINE

ANTENNA TUNER

OR

MATCH NOX SMAL

LINE

Figure 21

MULTI - BAND ANTENNA USING FAN -

DIPOLE TO LIMIT IMPEDANCE EXCUR-

SIONS ON HARMONIC FREQUENCIES

14 wire spaced 4 to 6 inches the antenna sys-

tem is sometimes called a center -fed zepp.

With this type of feeder the impedance at the

transmitter end of the feeder varies from about

70 ohms to approximately 5000 ohms, the same

as is encountered in an end -fed zepp antenna.

This great impedance ratio requires provision

for either series or parallel tuning of the feed-

ers at the transmitter, and involves quite high

r -f voltages at various points along the feed

line. If the feed line between the transmitter and

the antenna is made to have a characteristic

impedance of approximately 300 ohms the ex-

cursions in end -of- feeder impedance are great-

ly reduced. In fact the impedance then varies

from approximately 75 ohms to 1200 ohms.

With this much lowered impedance variation

it is usually possible to use series tuning on

all bands, or merely to couple the antenna di-

rectly to the output tank circuit or the har-

monic reduction circuit without any separate

feeder tuning provision.

There are several practicable types of trans-

mission line which can give an impedance of

approximately 300 ohms. The first is, obvious-

ly, 300 -ohm Twin -Lead. Twin -Lead of the re-

ceiving type may be used as a resonant feed

line in this case, but its use is not recom-

mended with power levels greater than perhaps

150 watts, and it should not be used when

lowest loss in the transmission line is desired.

For power levels up to 250 watts or so, the

transmitting type tubular 300 -ohm line may be

used, or the open -wire 300 -ohm TV line may

be employed. For power levels higher than

this, a 4- wire transmission line, or a line

built of one -quarter inch tubing should be

used.

Figure 22

FOLDED -TOP DUAL -BAND ANTENNA

Even when a 300 -ohm transmission line is

used, the end -of- feeder impedance may reach

a high value, particularly on the second har-

monic of the antenna. To limit the impedance

excursions,, a two -wire flat -top may be em-

ployed for the radiator, as shown in figure 21.

The use of such a radiator will limit the im-

pedance excursions on the harmonic frequen-

cies of the antenna and make the operation of

the antenna matching unit much less critical.

The use of a two -wire radiator is highly recom-

mended for any center -fed multi -band antenna.

Folded Flot -Top As has been mentioned

Dual -Band Antenna earlier, there is an increas-

ing tendency among ama-

teur operators to utilize rotary or fixed arrays

for the 14 -Mc. band and those higher in fre-

quency. In order to afford complete coverage

of the amateur bands it is then desirable to

have an additional system which will operate

with equal effectiveness on the 3.5 -Mc. and

7 -Mc. bands, but this low- frequency antenna

system will not be required to operate on any

bands higher in frequency than the 7 -Mc. band.

The antenna system shown in figure 22 has

been developed to fill this need.

This system consists essentially of an

open -line folded dipole for the 7 -Mc. band with

a special feed system which allows the an-

tenna to be fed with minimum standing waves

on the feed line on both the 7 -Mc. and 3.5 -Mc.

bands. The feed -point impedance of a folded

dipole on its fundamental frequency is approxi-

mately 300 ohms. Hence the 300 -ohm Twin -

Lead shown in figure 22 can be connected di-

rectly into the center of the system for opera-

tion only on the 7 -Mc. band and standing waves

on the feeder will be very small. However, it

is possible to insert an electrical half -wave

www.americanradiohistory.com

4 34 Antennas and Antenna Matching THE RADIO

250 LIMP

-lao -

L.90' FOR /0 -40 IMETES OPERATION

Figure 19

A TWO -BAND MARCONI ANTENNA FOR

160 -80 METER OPERATION

Since this antenna type is an unbalanced radi-

ating system, its use is not recommended with

high -power transmitters where interference to

broadcast listeners is likely to be encountered.

The r -f voltages encountered at the end of

zepp feeders and at points an electrical half

wave from the end are likely to be quite high.

Hence the feeders should be supported an ade-

quate distance from surrounding objects and

sufficiently in the clear so that a chance en-

counter between a passerby and the feeder is

unlikely.

The coupling coil at the transmitter end of

the feeder system should be link coupled to

the output of the low -pass TVI filter in order

to reduce harmonic radiation.

The Two -Band A three- eighths wavelength

Marconi Antenna Marconi antenna may be

operated on its harmonic

frequency, providing good two band perform-

ance from a simple wire. Such an arrangement

for operation on 160 -80 meters, and 80 -40 me-

ters is shown in figure 19. On the fundamental

(lowest) frequency, the antenna acts as a

three- eighths wavelength series -tuned Marconi.

On the second harmonic, the antenna is a cur-

rent -fed three -quarter wavelength antenna oper-

ating against ground. For proper operation,

the antenna should be resonated on its second

harmonic by means of a grid -dip oscillator to

the operating frequency most used on this par-

ticular band. The Q of the antenna is relatively

low, and the antenna will perform well over a

frequency range of several hundred kilocycles.

The overall length of the antenna may be

varied slightly to place its self- resonant fre-

quency in the desired region. Bends or turns

in the antenna tend to make it resonate higher

in frequency, and it may be necessary to

lengthen it a bit to resonate it at the chosen

frequency. For fundamental operation, the

series condenser is inserted in the circuit, and

the antenna may be resonated to any point in

the lower frequency band. As with any Marconi

r

BANDS LI

orHc

La TOPE OF

TuNINO

11. SS

PARALLEL

PALACELI

105

n5

n

stain

MRCS

PARALLEL

PA

PARALLEL

7 Mc

!di MC SI S]

RS Mc

5ERIEs

PA

T Mc

14 MC SI 100

211 MC

PARALLEL

iP fOOR 1551014 LIN[ If

VIED FOR 1-2 /MI IMPEDANCE AT

TNt TRANSMITTER END OF TOE

LiNt if AP02. L2

1200 ONMf 50

1200 ONMf

1100 OHMS

1100 Ow.

1100 ON25

M

0.045

Tf 00uf w

taw 0005

1200 0021

1200 o02í A

75 OHMS

/200 OHMS

1200 ONMs 55

1200 OHMS

Haw Ms

:00 Ms

CENTER-FED ANTENNA

Figure 20

DIMENSIONS FOR CENTER -FED MULTI -

BAND ANTENNA

type antenna, the use of a good ground is es-

sential. This antenna works well with trans-

mitters employing coaxial antenna feed, since

its transmitting impedance on both bands is in

the neighborhood of 40 to 60 ohms. It may be

attached directly to the output terminal of such

transmitters as the Collins 32V and the Viking

H. The use of a low -pass TVI filter is of

course recommended.

The Center -Fed For multi -band operation,

Multi -Band Antenna the center fed antenna is

without doubt the best

compromise. It is a balanced system on all

bands, it requires no ground return, and when

properly tuned has good rejection properties

for the higher harmonics generated in the trans-

mitter. It is well suited for use with the various

multi -band 150 -watt transmitters that are cur-

rently so popular. For proper operation with

these transmitters, an antenna tuning unit

must be used with the center -fed antenna. In

fact, some sort of tuning unit is necessary for

any type of efficient, multi -band antenna. The

use of such questionable antennas as the "off -

center fed15 doublet is an invitation to TVI

troubles and improper operation of the trans-

mitter. A properly balanced antenna is the

best solution to multi -band operation. When

used in conjunction with an antenna tuning

unit, it will perform with top efficiency on all

of the major amateur bands.

Several types of center -fed antenna systems

are shown in figure 20. If the feed line is made

up in the conventional manner of no. 12 or no.

www.americanradiohistory.com

HANDBOOK Multi -band Antennas 433

`[ rant+ NH... a

100 OMM N*

O '"ITEM

L I11 'OR 3310 NC AND 7114 NC

L N' rpm 7I0O NC AND 1310 NC.

L 411' rpA I4100 NC. AND a= MC

Figure 15

THE THREE -QUARTER WAVE FOLDED

DOUBLET

This antenna arrangement will give very

satisfactory operation with a 600 -ohm feed

line for operation with the switch open on

the fundamental frequency and with the

switch closed on twice frequency.

effective radiator on the second harmonic but

the pattern of radiation will be different from

that on the fundamental, and the standing -wave

ratio on the feed line will be greater. The flat

top of the antenna must be made of open wire

rather than ribbon or tubular line.

For greater operating convenience, the short-

ing switch may be replaced with a section of

transmission line. If this transmission line is

made one -quarter wavelength long for the fun-

damental frequency, and the free end of the

line is shorted, it will act as an open circuit

across the center insulator. At the second har-

monic, the transmission line is one -half wave-

length long, and reflects the low impedance

of the shorted end across the center insulator.

Thus the switching action is automatic as the

frequency of operation is changed. Such an

installation is shown in figure 16.

The End -Fed The end -fed Hertz antenna

Hertz shown in figure 17 is not as

effective a radiating system as

3.3, 7, 14 AND 26 MC.

3.5, 7 AND 14 MC.

3.5 AND 7 MC.

3.9 MC. AND 26 MC.

LINK

FROM

xMTR.

Ll36

L= 137'

L136

LO 20'

Figure 17

RECOMMENDED LENGTHS FOR THE END -

FED HERTZ

6' FEEDER SPREADERS

600 A LINE

SHORTED END 600 OHM LINE

TO TRANSMITTER

L. 67 FT WHEN ANTENNA IS 195 FT.

L' 33 FT - - - 96 Fr.

L ' 1 6 . S F T - 496 FT

Figure 16

AUTOMATIC BANDSWITCHING STUB FOR

THE THREE -QUARTER WAVE FOLDED

DOUBLET

The antenna of Figure 15 may be used with

a shorted stub line in place of the switch

normally used for second harmonic operation.

many other antenna types, but it is particular-

ly convenient when it is desired to install an

antenna in a hurry for a test, or for field -day

work. The flat top of the radiator should be

as high and in the clear as possible. In any

event at least three quarters of the total wire

length should be in the clear. Dimensions for

optimum operation on various amateur bands

are given in addition in figure 17.

The End -Fed The end- f ed Zepp has long

Zepp been a favorite for multi -band

operation. It is shown in fig-

ure 18 along with recommended dimensions

for operation on various amateur band groups.

- LI r01÷ .1ta SILO.- - -- - -

OR1

Sr1IAD[RS

BANDS L1 TTn Dr

TIMING

a.1 MC

SS MC 1n a $10,01

MRALL[L

3,5 11C

7 MC

1 MC

all MC

137 n 110610/

M1ALL11L

/ARALLIL

M1

$1 14 MIES

END-FED ZEPP

FIGURE 18

www.americanradiohistory.com

432 Antennas and Antenna Matching THE RADIO

3

3.5 3.1 3.7 3

FREQUENCY (MC)

30

Figure 13

SWR CURVE OF 80 -METER BROAD -BAND

DIPOLE

4.0

ohms. The ground losses are now reduced by

a factor of 4. In addition, the antenna may be

directly fed from a 50 -ohm coaxial line, or di-

rectly from the unbalanced output of a pi- net-

work transmitter.

Since a certain amount of power may still

be lost in the ground connection, it is still of

greatest importance that a good, low resist-

ance ground be used with this antenna.

The Collins Shown in figures 11 and 12

Brood -bond are broad -band dipoles for

Dipole System the 40 and 80 meter amateur

bands, designed by Collins

Radio Co. for use with the Collins 32V -3 and

KW -1 transmitters. These fan -type dipoles

have excellent broad -band response, and are

designed to be fed with a 52 -ohm unbalanced

coaxial line, making them suitable for use with

many of the other modem transmitters, such

as the Barker and Williamson 5100, Johnson

Ranger, and Viking. The antenna system con-

sists of a fan -type dipole, a balun matching

section, and a suitable coaxial feedline. The

Q of the half -wave 80 meter doublet is low-

ered by decreasing the effective length -to-

diameter ratio. The frequency range of opera-

tion of the doublet is increased considerably

by this change. A typical SWR curve for the

80 meter doublet is shown in figure 13.

The balanced doublet is matched to the un-

balanced coaxial line by the one -quarter wave

balun. If desired, a shortened balun may be

used (figure 14). The short balun is capacity

loaded at the junction between the balun and

the broad -band dipole.

22 -7 Multi -Band Antennas

The availability of a multi -band antenna is

a great operating convenience to an amateur

station. In most cases it will be found best to

install an antenna which is optimum for the

band which is used for the majority of the

PNENOUC !LOCKS

SEE rIC.12

ANTENNA -

INNUR CONDUCTOR NOT USED

eO METER7

L13s-

C 4001lUr

40 M

L7'3-

C200YLr

SEE FIG.12 FOR CONNECTION

52 OHM COAXIAL LINE

Figure 14

SHORT BALUN FOR 40 AND 80 METERS

available operating time, and then to have an

additional multi -band antenna which may be

pressed into service for operation on another

band when propagation conditions on the most

frequently used band are not suitable. Most

amateurs use, or plan to install, at least one

directive array for one of the higher- frequency

bands, but find that an additional antenna

which may be used on the 3.5 -Mc. and 7.0 -Mc.

band, or even up through the 28 -Mc. band is

almost indispensable.

The choice of a multi -band antenna depends

upon a number of factors such as the amount

of space available, the band which is to be

used for the majority of operating with the an-

tenna, the radiation efficiency which is de-

sired, and the type of antenna tuning network

to be used at the transmitter. A number of

recommended types are shown in the next

pages.

The 3í -Wave Figure 15 shows an antenna

Folded Doublet type which will be found to

be very effective when a

moderate amount of space is available, when

most of the operating will be done on one band

with occasional operation on the second har-

monic. The system is quite satisfactory for

use with high -power transmitters since a 600 -

ohm non -resonant line is used from the anten-

na to the transmitter and since the antenna

system is balanced with respect to ground.

With operation on the fundamental frequency

of the antenna where the flat top is % wave

long the switch SW is left open. The system

affords a very close match between the 600 -

ohm line and the feed point of the antenna.

Kraus has reported a standing -wave ratio of

approximately 1.2 to 1 over the 14 -Mc. band

when the antenna was located approximately

one -half wave above ground.

For operation on the second harmonic the

switch SW is closed. The antenna is still an

www.americanradiohistory.com

HANDBOOK Space Conserving Antennas 431

r

FOR DETAIL SEE FIG. A

PHENOLIC BLOCK 2XI.5XC +j//

WRAP CABLES AND BLOCK /

WITH SCOTCH ELECTRICALTAPE /%////

///

SPACE BLOCKS 0' APART / / /// //

ALONG BALUN

44.9

1ldllllllr

'- 11.11.IuIrI

OS-

FIGURE A

CUT OFF SHIELD AND OUTER

JACKET AS SHOWN. ALLOW

DIELECTRIC TO E %TEND PART

WAY TO OTHER CABLE. COVER

ALL EXPOSED SHIELD AND

DIELECTRIC ON BOTH CABLES

WITH A CONTINUOUS WRAP-

PING OF SCOTCH ELECTRICAL

TAPE TO EXCLUDE MOISTURE.

rI.Y

KEEP BALUN AT LEAST B CLEAR

OF GROUND AND OTHER OBJECTS.

FOR DETAIL SEE FIGURE B

52 OHM RG-8/U, ANY LENGTH

114

FIGURE B

REMOVE OUTER JACKET

FROM A SHORT LENGTH OF

CABLE AS SHOWN HERE.

UNBRAID THE SHIELD OF

COAX CUT OFF THE DI-

ELECTRIC AND INNER CON-

DUCTOR FLUSH WITH THE

OUTER JACKET. DO HOT CUT

THE SHIELD. WRA SHIELD

OF COAX C AROUND SHIELD

OF COAX D. SOLDER THE

CONNECTION. BEING VERY

CAREFUL NOT TO DAMAGE

THE DIELECTRIC MATERIAL.

HOLD CABLE O STRAIGHT

WHILE SOLDERING. COVER

THE AREA WITH A CONTIN-

UOUS WRAPPING OF SCOTCH

ELECTRICAL TAPE. NO CON-

NECTION TO INNER CONDUC-

TORS.

DIMENSIONS SHOWN NERE ARE FOR THE 40 METER BAND. THIS ANT-

ENNA MAY BE BUILT FOR OTHER BANDS BY US/Ni DIMENSIONS THAT

ARE MULTIPLES OR SUBMUL TIPLES OF THE DIMENSIONS SHOWN.

BALUN SPACING IS S. ON ALL BANDS.

Figure 11

HALF -WAVE ANTENNA WITH QUARTER -

WAVE UNBALANCED TO BALANCED

TRANSFORMER (BALUN) FEED SYSTEM

FOR 40 -METER OPERATION

FOR DETAIL SEE FIG A

ENOLIC BLOCK 2 X 1.3 X 0.5

WRAP CABLES AND BLOCK

ITN SCOTCH ELECTRICAL T

SPACE BLOCKS B'APART

ALONG BALUN

iW.N.I

0.5W

LS-

FIGURE A

CUT OFF SHIELD AND OUTER

JACKET AS SHOWN. ALLOW

DIELECTRIC TO EXTEND PART

WAY TO OTHER CABLE. COVER

ALL EXPOSED SHIELD AND

DIELECTRIC ON BOTH CABLES

WRAP-

PING OF SCOTCH ELECTRICAL

TAPE TO EXCLUDE MOISTURE.

110'

11.10.10c - 1 411111110

i

THE TWO W IRES MAY BE

SPREAD EITHER HORIZ-

ONTALLYOR VERTICALLY.

KEEP BALUN AT LEAST B- LEAR

OF GROUND AND OTHER OBJECTS.

FOR DETAIL SEE FIGURE B

52 OHM RD -B /U, ANY LENGTH

FIGURE B

REMOVE OUTER JACKET

FROM A SHORT LENGTH OF

CABLE AS SHOWN HERE.

UNBRAID THE SHIELD OF

COAX C CUTOFF THE DI-

ELECTI(IC AND INNER CON -

DUCTOR FLUSH WITH THE

OUTER JACKET. DO NOT CUT

THE SHIELD. WRAP SHIELD

OF COAX C AROUND SHIELD

OF COAX D. SOLDER THE

CONNECTION. BEING VERY

CAREFUL NOT TO DAMAGE

THE DIELECTRIC MATERIAL.

HOLD CABLE D STRAIGHT

WHILE SOLDERING. COVER

THE AREA WITHACONTIN-

UOUS WRAPPING OF SCOTCH

ELECTRICAL TAPE. N0 CON-

NEC T ION TO INNER CONDUC-

TORS.

DIMENSIONS SHOWN HERE ARE FOR THE b METER BAND. THIS ANT-

ENNA MAY BE BUILT FOR OTHER BANDS BY USINE DIMENSIONS THAT

ARE MULTIPLES OR SUBMUL TIPLES OF THE DIMENSIONS SHOWN.

BALUN SPACING IS /.5. ON ALL BANDS.

Figure 12

BROADBAND ANTENNA WITH QUARTER -

WAVE UNBALANCED TO BALANCED

TRANSFORMER (BALUN) FEED SYSTEM

FOR 80 -METER OPERATION

sions in terms of frequency are given on the

drawing. An antenna of this type is 93 feet

long for operation on 3600 kc. and 86 feet long

for operation on 3900 kc. This type of antenna

has the additional advantage that it may be

operated on the 7 -Mc. and 14 -Mc. bands, when

the flat top has been cut for the 3.5 -Mc. band,

simply by changing the position of the short-

ing bar and the feeder line on the stub.

A sacrifice which must be made when using

a shortened radiating system, as for example

the types shown in figure 9, is in the band-

width of the radiating system. The frequency

range which may be covered by a shortened

antenna system is approximately in proportion

to the amount of shortening which has been

employed. For example, the antenna system

shown in figure 9C may be operated over the

range from 3800 kc. to 4000 kc. without ser-

ious standing waves on the feed line. If the

antenna had been made full length it would

be possible to cover about half again as much

frequency range for the same amount of mis-

match on the extremes of the frequency range.

The Twin -Lead Much of the power loss in

Marconi Antenna the Marconi antenna is a re-

sult of low radiation resist-

ance and high ground resistance. In some

cases, the ground resistance may even be

be higher than the radiation resistance, caus-

ing a loss of 50 per cent or more of the trans-

mitter power output. If the radiation resistance

of the Marconi antenna is raised, the amount

of power lost in the ground resistance is pro-

portionately less. If a Marconi antenna is made

out of 300 ohm TV -type ribbon line, as shown

in figure 10, the radiation resistance of the

antenna is raised from a low value of 10 or 15

ohms to a more reasonable value of 40 to 60

www.americanradiohistory.com

430 Antennas and Antenna Matching THE RADIO

.. .,. AT Lo..,T /MM.,

YrLCGaa

areeoaXa

Figure 9

THREE EFFECTIVE SPACE CONSERVING

ANTENNAS

The arrangements shown at (A) and (B) are

satisfactory where resonant feed line can be

used. However, non- resonant 75 -ohm feed

line may be used in the arrangement at (A)

when the dimensions in wavelengths are as

shown. In the arrangement shown at (B) low

standing waves will be obtained on the feed

line when the overall length of the antenna

is a half wave. The arrangement shown at

(C) may be tuned for any reasonable length

of flat top to give a minimum of standing

waves on the transmission line.

quarter wavelength can be lengthened elec-

trically by means of a series loading coil, and

used as a quarter -wave Marconi. However, if

the wire is made shorter than approximately

one -eighth wavelength, the radiation resist-

ance will be quite low. This is a special prob-

lem in mobile work below about 20 -Mc.

22 -6 Space -Conserving

Antennas

In many cases it is desired to undertake a

considerable amount of operation on the 80-

meter or 40 -meter band, but sufficient space

is simply not available for the installation of

a half -wave radiator for the desired frequency

of operation. This is a common experience of

apartment dwellers. The shortened Marconi

antenna operated against a good ground can

be used under certain conditions, but the short-

ened Marconi is notorious for the production

of broadcast interference, and a good ground

connection is usually completely unobtainable

in an apartment house.

52A COAXIAL

TEED LINE

300 OHM -RIBBON- LINE

WIRES SHORTED TO-

GETHER AT END

Figure 10

TWIN -LEAD MARCONI ANTENNA FOR THE

80 AND 160 METER BANDS

Essentially, the problem in producing an

antenna for lower frequency operation in re-

stricted space is to erect a short radiator

which is balanced with respect to ground and

which is therefore independent of ground for

its operation. Several antenna types meeting

this set of conditions are shown in figure 9.

Figure 9A shows a conventional center -fed

doublet with bent -down ends. This type of an-

tenna can be fed with 75-ohm Twin -Lead in the

center, or it may be fed with a resonant line

for operation on several bands. The overall

length of the radiating wire will be a few per

cent greater than the normal length for such

an antenna since the wire is bent at a posi-

tion intermediate between a current loop and

a voltage loop. The actual length will have to

be determined by the cut -and -try process be-

cause of the increased effect of interfering ob-

jects on the effective electrical length of an

antenna of this type.

Figure 9B shows a method for using a two -

wire doublet on one half of its normal operat-

ing frequency. It is recommended that spaced

open conductor be used both for the radiating

portion of the folded dipole and for the feed

line. The reason for this recommendation lies

in the fact that the two wires of the flat top

are not at the same potential throughout their

length when the antenna is operated on one -

half frequency. Twin -Lead may be used for

the feed line if operation on the frequency

where the flat top is one -half wave in length

is most common, and operation on one -half fre-

quency is infrequent. However, if the antenna

is to be used primarily on one -half frequency

as shown, it should be fed by means of an

open -wire line. If it is desired to feed the an-

tenna with a non -resonant line, a quarter -wave

stub may be connected to the antenna at the

points X, X in figure 9B. The stub should be

tuned and the transmission line connected to

it in the normal manner.

The antenna system shown in figure 9C may

be used when not quite enough length is avail-

able for a full half -wave radiator. The dimen-

www.americanradiohistory.com

HANDBOOK Marconi Antenn a 429

Figure 8

LOADING THE

MARCONI ANTENNA

The various loading systems

are discussed in the accom-

panying text.

a 7

LOADING

COILS `MAT

O © © 0 0 0

current flows through a r e s i s t o r, or if the

ground itself presents some resistance, there

will be a power loss in the form of heat. Im-

proving the ground connection, therefore, pro-

vides a definite means of reducing this power

loss, and thus increasing the radiated power.

The best possible ground consists of as

many wires as possible, each at least a quar-

ter wave long, buried just below the surface

of the earth, and extending out from a common

point in the form of radials. Copper wire of

any size larger than no. 16 is satisfactory,

though the larger sizes will take longer to dis-

integrate. In fact, the radials need not even

be buried; they may be supported just above

the earth, and insulated from it. This arrange-

ment is called a counterpoise, and operates

by virtue of its high capacitance to ground.

If the antenna is physically shorter than a

quarter wavelength, the antenna current is

higher, due to lower radiation resistance. Con-

sequently, the power lost in resistive soil is

greater. The importance of a good ground with

short, inductive -loaded Marconi radiators is,

therefore, quite obvious. With a good ground

system, even very short (one- eighth wave-

length) antennas can be expected to give a

high percentage of the efficiency of a quarter -

wave antenna used with the same ground sys-

tem. This is especially true when the short

radiator is top loaded with a high Q (low loss)

coil.

Water -Pipe Water pipe, because of its corn -

Grounds paratively large surface and cross

section, has a relatively low r -f

resistance. If it is possible to attach to a

junction of several water pipes (where they

branch in several directions and run for some

distance under ground), a satisfactory ground

connection will be obtained. If one of the

pipes attaches to a lawn or garden sprinkler

system in the immediate vicinity of the anten-

na, the effectiveness of the system will ap-

proach that of buried copper radials.

The main objection to water -pipe grounds

is the possibility of high resistance joints in

the pipe, due to the "dope" put on the cou-

pling threads. By attaching the ground wire

to a junction with three or more legs, the pos-

sibility of requiring the main portion of the

r -f current to flow through a high resistance

connection is greatly reduced.

The presence of water in the pipe adds

nothing to the conductivity; therefore it does

not relieve the problem of high resistance

joints. Bonding the joints is the best insur-

ance, but this is, of course, impracticable

where the pipe is buried. Bonding together

with copper wire the various water faucets

above the surface of the ground will improve

the effectiveness of a water -pipe ground sys-

tem hampered by high -resistance pipe cou-

plings.

Marconi A Marconi antenna is an odd

Dimensions number of electrical quarter

waves long (usually only one

quarter wave in length), and is always reso-

nated to the operating frequency. The correct

loading of the final amplifier is accomplished

by varying the coupling, rather than by detun-

ing the antenna from resonance.

Physically, a quarter -wave Marconi may be

made anywhere from one - eighth to three -eighths

wavelength overall, meaning the total length of

the antenna wire and ground lead from the end

of the antenna to the point where the ground

lead attaches to the junction of the radials or

counterpoise wires, or where the water pipe

enters the ground. The longer the antenna

is made physically, the lower will be the cur-

rent flowing in the ground connection, and the

greater will be the overall radiation efficiency.

However, when the antenna length exceeds

three -eighths wavelength, the antenna be-

comes difficult to resonate by means of a

series capacitor, and it begins to take shape

as an end -fed Hertz, requiring a method of

feed such as a pi network.

A radiator physically much shorter than a

www.americanradiohistory.com

428 Antennas and Antenna Matching THE RADIO

used for the radiator. Such an ant e nn a is

shown in figure 6. The loaded ground -plane

tends to have a rather high operating Q and

operates only over a narrow band of frequen-

cies. An operating range of about 100 kilo-

cycles with a low SWR is possible on 80 me-

ters. Operation over a larger frequency range

is possible if a.higher standing wave ratio is

tolerated on the transmission line. The radia-

tion resistance of a loaded 80 -meter ground -

plane is about 15 ohms. A quarter wavelength

(45 feet) of 52 -ohm coaxial line will act as an

efficient feed line, presenting a load of ap-

proximately 180 ohms to the transmitter.

22 -5 The Marconi

Antenna

A grounded quarter -wave Marconi antenna,

widely used on frequencies below 3 Mc., is

sometimes used on the 3.5-Mc. band, and is

also used in v -h -f mobile services where a

compact antenna is required. The Marconi type

antenna allows the use of half the length of

wire that would be required for a half -wave

Hertz radiator. The ground acts as a mirror,

in effect, and takes the place of the additional

quarter -wave of wire that would be required

to reach resonance if the end of the wire were

not returned to ground.

The fundamental practical form of the Mar-

coni antenna system is shown in figure 7.

Other Marconi antennas differ from this type

primarily in regard to the method of feeding

the energy to the radiator. The feed method

shown in figure 7B can often be used to advan-

tage, particularly in mobile work.

Variations on the basic Marconi antenna

are shown in the illustrations of figure 8. Fig-

ures 8B and 8C show the "L" -type and "T "-

type Marconi antennas. These arrangements

have been more or less superseded by the top -

loaded forms of the Marconi antenna shown in

figures 8D, 8E, and 8F. In each of these lat-

ter three figures an antenna somewhat less

than one quarter wave in length has been

loaded to increase its effective length by the

insertion of a loading coil at or near the top

of the radiator. The arrangement shown at fig-

ure 8D gives the least loading but is the most

practical mechanically. The system shown at

figure 8E gives an intermediate amount of

loading, while that shown at figure 8F, utiliz-

ing a "hat" just above the loading coil, gives

the greatest amount of loading. The object of

all the top -loading methods shown is to pro-

duce an increase in the effective length of

the radiator, and thus to raise the point of

maximum current in the radiator as far as pos-

COAX. PROM TRANS.

Figure 7

FEEDING A QUARTER -WAVE MARCONI

ANTENNA

When an open -wire line is to be used, it may

be link coupled to o series- resonant circuit

between the bottom end of the Marconi and

ground, as of (A). Alternatively, a reason-

ably good impedance match may be obtained

between 52 -ohm coaxial line and the bottom

of a resonant quarter -wave antenna, as illus-

trated at (B) above.

sible above ground. Raising the maximum -cur-

rent point in the radiator above ground has

two desirable results: The percentage of low -

angle radiation is increased and the amount of

ground current at the base of the radiator is

reduced, thus reducing the ground losses.

To estimate whether a loading coil will

probably be required, it is necessary only to

note if the length of the antenna wire and

ground lead is over a quarter wavelength; if

so, no loading coil is needed, provided the

series tuning capacitor has a high maximum

capacitance.

Amateurs primarily interested in the higher

frequency bands, but who like to work 80 me-

ters occasionally, can usually manage to reso-

nate one of their antennas as a Marconi by

working the whole system, feeders and all,

against a water pipe ground, and resorting to

a loading coil if necessary. A high- frequency-

rotary, zepp, doublet, or single- wire -fed an-

tenna will make quite a good 80 -meter Marconi

if high and in the clear, with a rather Long

feed line to act as a radiator on 80 meters.

Where two-wire feeders are used, the feeders

should be tied together for Marconi operation.

Importance of With a quarter -wave anten-

Ground Connection na and a ground, the an-

tenna current generally is

measured with a meter placed in the antenna

circuit close to the ground connection. If this

www.americanradiohistory.com

HANDBOOK Vertical Antennas 427

RADIALS EACH la

52 OHM COAXIAL LINE,

CENTER CONDUCTOR CONNECTS

TO VERTICAL WHIP

Figure 5

THE LOW -FREQUENCY GROUND PLANE

ANTENNA

The radials o f the ground plane antenna

should lie in a horizontal plane, although

slight departures from this caused by nearby

objects is allowable. The whip may be

mounted on a short post, or on the roof of a

building. The wire radials may slope down-

wards towards their tips, acting as guy

wires for the installation.

ground is an effective transmitting antenna for

low -angle radiation, where ground conditions

in the vicinity of the antenna are good. Such

an antenna is not good for short -range sky -

wave communication, such as is the normal

usage of the 3.5 -Mc. amateur band, but is ex-

cellent for short -range ground -wave communi-

cation such as on the standard broadcast band

and on the amateur 1.8 -Mc. band. The vertical

antenna normally will cause greater BO than

an equivalent horizontal antenna, due to the

much greater ground -wave field intensity. Al-

so, the vertical antenna is poor for receiving

under conditions where man -made interference

is severe, since such interference is predomi-

nantly vertically polarized.

Three ways of feeding a half -wave vertical

antenna from an untuned transmission line are

illustrated in figure 4. The J -fed system shown

in figure 4A is obviously not practicable ex-

cept on the higher frequencies where the ex-

tra length for the stub may easily be obtained.

However, in the normal case the ground -plane

vertical antenna is to be recommended over

the J -fed system for high frequency work.

22 -4 The Ground Plane Antenna

An effective low angle radiator for any ama-

LOADING COIL

APROXI MAYFLY D! TURNS

RIZ WIRE, .S" DIAMETER

AND I FOOT LONG

RADIALS EACH

52 OHM COAXIAL LINE

45 FEET LONG

Figure 6

80 METER LOADED GROUND PLANE

ANTENNA

Number of turns in loading coil to be adjusted

until antenna system resonates at desired

frequency in 80 meter band.

teur band is the ground -plane antenna, shown

in figure 5. So called because of the radial

ground wires, the ground -plane antenna is not

affected by soil conditions in its vicinity due

to the creation of an artificial ground system

by the radial wires. The base impedance of

the ground plane is of the order of 30 to 35

ohms, and it may be fed with 52 -ohm coaxial

line with only a slight impedance mis -match.

For a more exact match, the ground -plane an-

tenna may be fed with a 72 -ohm coaxial line

and a quarter -wave matching section made of

52 -ohm coaxial line.

The angle of radiation of the ground -plane

antenna is quite low, and the antenna will be

found less effective for contacts under 1000

miles or so on the 80 and 40 meter bands than

a high angle radiator, such as a dipole. How-

ever, for DX contacts of 1000 miles or more,

the ground -plane antenna will prove to be

highly effective.

The 80 -Meter A vertical antenna of 66 feet

Loaded in height presents quite a prob-

Ground -Plane lern on a small lot, as the sup-

porting guy wires will tend to

take up quite a large portion of the lot. Under

such conditions, it is possible to shorten the

length of the vertical radiator of the ground -

plane by the inclusion of a loading coil in the

vertical whip section. The ground -plane an-

tenna may be artificially loaded in this man-

ner so that a 25 -foot vertical whip may be

www.americanradiohistory.com

426 Antennas and Antenna Matching THE RADIO

462

FMc

tDZ

oaPaii o.A.:

I FNC. - -

1 ©

' FED STVB-FED L-C FED

VERTICAL VERTICAL VERTICAL

300 -OHM RIBBON

404

FNc

300 -ONM RIBBON

30 FMC.

Figure 3

FOLDED DIPOLE WITH SHORTING

STRAPS

The impedance match and bandwidth char-

acteristics ofa folded dipole maybe improved

by shorting the two wires of the ribbon a dis-

tance out from the center equal to the veloci-

ty factor of the ribbon times the half -length

of the dipole as shown at (A). An alternative

arrangement with bent down ends for space

conservation is illustrated at (13).

times over the radiation resistance of the ele-

ment, have both contributed to the frequent

use of the multi -wire radiator as the driven

element in a parasitic antenna array.

Delta-Matched These two types of radiat-

Doublet and ing elements are shown in

Standard Doublet figure 2L and figure 2M. The

delta- matched doublet is

described in detail in Section 22 -8 of this

chapter. The standard doublet, shown in fig-

ure 2M, is fed in the center by means of 75-

ohm Twin -Lead, either the transmitting or the

receiving type, or it may be fed by means of

twisted -pair feeder or by means of parallel -

wire lamp -cord. Any of these types of feed

line will give an approximate match to the

center impedance of the dipole, but the 75-

ohm Twin -Lead is far to be preferred over the

other types of low -impedance feeder due to

the much lower losses of the polyethylene -

dielectric transmission line.

The coaxial- cable -fed doublet shown in fig-

ure 2N is a variation on the system shown in

figure 2M. Either 52 -ohm coaxial cable or 75-

ohm coaxial cable may be used to feed the

center of the dipole, although the 75 -ohm type

Figure 4

HALF -WAVE VERTICAL ANTENNA SHOW-

ING ALTERNATIVE METHODS OF FEED

will give a somewhat better impedance match

at normal antenna heights. Due to the asym-

metry of the coaxial feed system difficulty

may be encountered with waves traveling on

the outside of the coaxial cable. For this rea-

son the use of Twin -Lead is normally to be

preferred over the use of coaxial cab 1 e for

feeding the center of a half -wave dipole.

Off- Center The system shown in figure

Fed Doublet 2(0) is sometimes used to

feed a half -wave dipole, espe-

cially when it is desired to use the same an-

tenna on a number of harmonically -related fre-

quencies. The feeder wire (no. 14 enamelled

wire should be used) is tapped a distance of

14 per cent of the total length of the antenna

either side of center. The feeder wire, operat-

ing against ground for the return current, has

an impedance of approximately 600 ohms. The

system works well over highly conducting

ground, but will introduce rather high losses

when the antenna is located above rocky or

poorly conducting soil. The off -center fed an-

tenna has a further disadvantage that it is

highly responsive to harmonics fed to it from

the transmitter.

The effectiveness of the antenna system in

radiating harmonics is of course an advantage

when operation of the antenna on a number of

frequency bands is desired. But it is neces-

sary to use a harmonic filter to insure that

only the desired frequency is fed from the

transmitter to the antenna.

22 -3 The Half -Wave

Vertical Antenna

The half -wave vertical antenna with its bot-

tom end from 0.1 to 0.2 wavelength a bo v e

www.americanradiohistory.com

THE RADIO Multi -wire Doublets 425

in series with the antenna coil or in parallel

with it. A series tuning c a p a c i tor can be

placed in series with one feeder leg without

unbalancing the system.

The tuned -doublet antenna is shown in fig-

ure 2D. The antenna is a current -fed system

when the radiating wire is a half wave long

electrically, or when the system is operated

on its odd harmonics, but becomes a voltage -

fed radiator when operated on its even har-

monics.

The antenna has a different radiation pat-

tern when operated on its harmonics, as would

be expected. The arrangement used on the

second harmonic is better known as the Frank-

lin colinear array and is described in Chapter

Twenty- three. The pattern is similar toa 1,j-wave

dipole except that it is sharper in the broad-

side direction. On higher harmonics of oper-

ation there will be multiple lobes of radiation

from the system.

Figures 2E and 2F show alternative arrange-

ments for using an untuned transmission line

between the transmitter and the tuned -doublet

radiator. In figure 2E a half -wave shorted line

is used to resonate the radiating system,

while in figure 2F a quarter -wave open line is

utilized. The adjustment of quarter -wave and

half -wave stubs is discussed in Section 19 -8.

Doublets with The average value of feed im-

Quarter -Wave pedance for a center -fed half -

Transformers wave doublet is 75 ohms. The

actual value varies with height

and is shown in Chapter Twenty -one. Other

methods of matching this rather low value of

impedance to a medium -impedance transmis-

sion line are shown in (G), (H), and (I) of fig-

ure 2. Each of these three systems uses a

quarter -wave transformer to accomplish the

impedance transformation. The only difference

between the three systems lies in the type of

transmission line used in the quarter -wave

transformer. (G) shows the Johnson Q system

whereby a line made up of 1/2-inch dural tubing

is used for the low- impedance linear trans-

former. A line made up in this manner is fre-

quently called a set of Q bars. Illustration

(H) shows the use of a four -wire line as the

linear transformer, and (I) shows the use of a

piece of 150 -ohm Twin -Lead electrically 1/2-

wave in length as the transformer between the

center of the dipole and a piece of 300 -ohm

Twin -Lead. In any case the impedance of the

quarter -wave transformer will be of the order

of 150 to 200 ohms. The use of sections of

transmission line as linear transformers is

discussed in detail in Section 22 -8.

Multi -Wire

Doublets An alternative method for increas-

ing the feed -point impedance of a

dipole so that a medium -imped-

ance transmission line may be used is shown

in figures 2J and 2K. This system utilizes

more than one wire in parallel for the radiating

element, but only one of the wires is broken

for attachment of the feeder. The most com-

mon arrangement uses two wires in the flat

top of the antenna so that an impedance multi-

plication of four is obtained.

The antenna shown in figure 2J is the so-

called Twin -Lead folded dipole which is a

commonly used antenna system on the medium -

frequency amateur bands. In this arrangement

both the antenna and the transmission line to

the transmitter are constructed of 300 -ohm

Twin -Lead. The flat top of the antenna is

made slightly less than the conventional

length (462 /FMc, instead of 468 /FMc, for a

single -wire flat top) and the two ends of the

Twin -Lead are joined together at each end.

The center of one of the conductors of the

Twin -Lead flat top is broken and the two ends

of the Twin -Lead feeder are spliced into the

flat top leads. As a protection against mois-

ture pieces of flat polyethylene taken from

another piece of 300 -ohm Twin -Lead may be

molded over the joint between conductors with

the aid of an electric iron or soldering iron.

Better bandwidth characteristics can be ob-

tained with a folded dipole made of ribbon line

if the two conductors of the ribbon line are

shorted a distance of 0.82 (the velocity factor

of ribbon line) of a free -space quarter wave-

length from the center or feed point. This pro-

cedure is illustrated in figure 3A. An alter-

native arrangement for a Twin -Lead folded

dipole is illustrated in figure 3B. This type of

half -wave antenna system is convenient for

use on the 3.5-Mc. band when the 116 to 132

foot distance required for a full half -wave is

not quite available in a straight line, since the

single -wire end pieces may be bent away or

downward from the direction of the main sec-

tion of the antenna.

Figure 2K shows the basic type of 2 -wire

doublet or folded dipole wherein the radiating

section of the system is made up of standard

antenna wire spaced by means of feeder

spreaders. The feeder again is made of 300 -

ohm Twin -Lead since the feed -point imped-

ance is approximately 300 ohms, the same as

that of the Twin -Lead folded dipole.

The folded -dipole type of antenna has the

broadest response characteristic (greatest

bandwidth) of any of the conventional half -

wave antenna systems constructed of small

wires or conductors. Hence such an antenna

may be operated over the greatest frequency

range without serious standing waves of any

common half -wave antenna type.

The increased bandwidth of the multi -wire

doublet type of radiator, and the fact that the

feed -point resistance is increased sever al

www.americanradiohistory.com

424 Antennas and Antenna Matching THE RADIO

Z EPP END -FED HERTZ STUB -FED

e O

A

300 -600

11 LINE -

END -FED TYPES

r-- 0.95 A/2-1 --0.95 A/2--of-

O TUNED DOUBLET

-0 .95 A/2 -+{

O T

0-FEO

6000 LINE

W---0.94 5/2{

O

TWINLEAD

OLOEDDIPOIE

HALF -WAVE

STUB- FED

SHORTED

300-60011 LINE

r-- 0.95 9/2

O 4

FOUR -WIRE

LINE -FED

Y1/4

3000 TWINLEAD

LOW SIDE OPENED

IN CENTER

L

60011 LINE

0.94 5/2

O

2 -WIRE DOUBLET

OR `FOLDED DIPOLE

1

300-600 O11M

LINE OPEN

QUARTER-WAVE

STUB- FED

0.95 9/2 --{

15011 TWINLEAD

0.193 OF FREE

SPACE WAVE-

LENGTH OR

0.77 OF 9/4

TWIN LEAD

FED

300 n TWINLEAD

ANY LENGTH

-0.95 5/2 -+I

2 -OR 6

FEEDER

SPREADERS

DELTA MATCHED

DOUBLET

300 OHM TWINLEAD 300 OHM TWINLEAD

ANY LENGTH

- -0.95A/2--

STANDARD

DOUBLET

750 TWINLEAD

ANY LENGTH

f~--0.95 A/2 -41

CO -AA FED

ti FOR DELTA

DIMENSIONS

SEE CHAP 19

600 OHM LINE

ANY LENGTH

r--095 A/2--.{

CENTER -FED TYPES,

0 14% OF

TOTAL LENGTH

N 14 WIRE

Figure 2

ALTERNATIVE

METHODS OF

FEEDING A

HALF -WAVE DIPOLE

www.americanradiohistory.com

Center -Fed Antennas 423

FROM TRANSMITTER

r

V

T

f I V

NIGH LOW

CAPACITANCE /1 T T CAPACITANCE

I

TI ANT NUMBER OF HALF -WAVES

ANY EVEN NUMBER OF QUARTER -WAVES

Figure 1

THE END -FED HERTZ ANTENNA

Showing the manner in which an end -fed Hertz

antenna may be fed through a low -impedance

line and low -pass filter by using a resonant

tank circuit as at (A), or through the use of

a reverse- connected pi network as at (B).

Some harmonic -attenuating provision (in addi-

tion to the usual low -pass TVI filter) must be

included in the coupling system, as an end -

fed antenna itself offers no discrimination

against harmonics, either odd or even.

The end -fed Hertz antenna has rather high

losses unless at least three -quarters of the

radiator can be placed outside the operating

room and in the clear. As there is r -f voltage

at the point where the antenna enters the

operating room, the insulation at that point

should be several times as effective as the

insulation commonly used with low- voltage

feeder systems. This antenna can be operated

on all of its higher harmonics with good effi-

ciency, and can be operated at half frequency

against ground as a quarter -wave Marconi.

As the frequency of an antenna is raised

slightly when it is bent anywhere except at a

voltage or current loop, an end -fed Hertz an-

tenna usually is a few per cent longer than a

straight half -wave doublet for the same fre-

quency, because, ordinarily, it is impractical

to bring a wire in to the transmitter without

making several bends.

The Zepp Antenna The zeppelin or zepp an-

System terma system, illustrated

in figure 2A is very con-

venient when it is desired to operate a single

radiating wire on a number of harmonically re-

lated frequencies.

The zepp antenna system is easy to tune,

and can be used on several bands by merely

retuning the feeders. The overall efficiency of

the zepp antenna system is not quite as high

for long feeder lengths as for some of the an-

tenna systems which employ non -resonant

transmission lines, but where space is limited

and where operation on more than one band

is desired, the zepp has some decided ad-

vantages.

As the radiating portion of the zepp antenna

system must always be some multiple of a

half wave long, there is always high voltage

present at the point where the live zepp feed-

er attaches to the end of the radiating portion

of the antenna. Thus, this type of zepp an-

tenna system is voltage led.

Stub -Fed Zepp- Figure 2C shows a modifica-

Type Radiator Lion of the zepp -type antenna

system to allow the use of

a non -resonant transmission line between the

radiating portion of the antenna and the trans-

mitter. The zepp portion of the antenna is

resonated as a quarter -wave stub and the non -

resonant feeders are connected to the stub at

a point where standing waves on the feeder

are minimized. The procedure for making these

adjustments is described in detail in Section

22 -8 This type of antenna system is quite

satisfactory when it is necessary physically

to end feed the antenna, but where it is neces-

sary also to use non -resonant feeder between

the transmitter and the radiating system.

22 -2 Center -Fed Half -

Wave Horizontal Antennas

The center feeding of a half -wave antenna

system is usually to be desired over an end -

fed system since the center -fed system is in-

herently balanced to ground and is therefore

less likely to be troubled by feeder radiation.

A number of center -fed systems are illustrated

in figure 2.

The Tuned The current -fed do u b l e t with

Doublet spaced feeders, sometimes

called a center -fed zepp, is an

inherently balanced system if the two legs of

the radiator are electrically equal. This fact

holds true regardless of the frequency, or of

the harmonic, on which the system is oper-

ated. The system can successfully be oper-

ated over a wide range of frequencies if the

system as a whole (both tuned feeders and the

center -fed flat top) can be resonated to the

operating frequency. It is usually possible to

tune such an antenna system to resonance

with the aid of a tapped coil and a tuning ca-

pacitor that can optionally be placed either

www.americanradiohistory.com

CHAPTER TWENTY -TWO

Antennas and Antenna Matching

Antennas for the lower frequency portion of

the h -f spectrum (perhaps from 1.8 to 7.0 Mc.),

and temporary or limited use antennas for the

upper portion of the h -f range, usually are of

a relatively simple type in which directivity

is not a prime consideration. Also, it often is

desirable, in amateur work, that a single an-

tenna system be capable of operation at least

on the 3.5 -Mc. and 7.0 -Mc. range, and prefer-

ably on other frequency ranges. Consequently,

the first portion of this chapter will be de-

voted to a discussion of such antenna sys-

tems. The latter portion of the chapter is de-

voted to the general problem of matching the

antenna transmission line to antenna systems

of the fixed type. Matching the antenna trans-

mission line to the rotatable directive array

is discussed in Chapter Twenty -five.

22 -1 End -Fed Half -Wave

Horizontal Antennas

The half -wave horizontal dipole is the most

common and the most practical antenna for the

3.5 -Mc. and 7 -Mc. amateur bands. The form of

the dipole, and the manner in which it is fed

are capable of a large number of variations.

Figure 2 shows a number of practicable forms

of the simple dipole antenna along with meth-

ods of feed.

422

Usually a high- frequency doublet is mounted

as high and as much in the clear as possible,

for obvious reasons. However, it is sometimes

justifiable to bring part of the radiating sys-

tem directly to the transmitter, feeding the an-

tenna without benefit of a transmission line.

This is permissible when (1) there is insuffi-

cient room to erect a 75- or 80 -meter horizon-

tal dipole and feed line, (2) when a long wire

is also to be operated on one of the higher

frequency bands on a harmonic. In either case,

it is usually possible to get the main portion

of the antenna in the clear because of its

length. This means that the power lost by

bringing the antenna directly to the transmitter

is relatively small.

Even so, it is not best practice to bring the

high -voltage end of an antenna into the oper-

ating room because of the increased difficulty

in eliminating BC! and TVI. For this reason

one should dispense with a feed line in con-

junction with a Hertz antenna only as a last

resort.

End -Fed The end -fed antenna has no form

Antennas of transmission line to couple it

to the transmitter, but brings the

radiating portion of the antenna right down to

the transmitter, where some form of coupling

system is used to transfer energy to the an-

tenna.

Figure 1 shows two common methods of

feeding the Fuchs antenna or end -fed Hertz.

www.americanradiohistory.com

HANDBOOK Tuned Lines 421

amplitude, in turn, depends upon the mismatch

at the line termination. A line of no. 12 wire,

spaced 6 inches with good ceramic or plastic

spreaders, has a surge impedance of approx-

imately 600 ohms, and makes an excellent

tuned feeder for feeding anything between 60

and 6000 ohms (at frequencies below 30 Mc.).

If used to feed a load of higher or lower imped-

ance than this, the standing waves become

great enough in amplitude that some loss will

occur unless the feeder is kept short. At fre-

quencies above 30 Mc., the spacing becomes

an appreciable fraction of a wavelength, and

radiation from the line no longer is negligible.

Hence, coaxial line or close- spaced parallel -

wire line is recommended for v -h -f work.

If a transmission line is not perfectly match-

ed, it should be made resonant, even though

the amplitude of the standing waves (voltage

variation) is not particularly great. This pre-

vents reactance from being coupled into the

final amplifier. A feed system having moderate

standing waves may be made to present a non -

reactive load to the amplifier either by tuning

or by pruning the feeders to approximate reso-

nance.

Usually it is preferable with tuned feeders

to have a current loop (voltage minimum) at the

transmitter end of the line. This means that

when voltage- feeding an antenna, the tuned

feeders should be made an odd number of quar-

ter wavelengths long, and when current -feeding

an antenna, the feeders should be made an

even number of quarter wavelengths long. Actu-

ally, the feeders are made about 10 per cent of

a quarter wave longer than the calculated

value (the value given in the tables) when

they are to be series tuned to resonance by

means of a capacitor, instead of being trimmed

and pruned to resonance.

When tuned feeders are used to feed an an-

tenna on more than one band, it is necessary

to compromise and make provision for both

series and parallel tuning, inasmuch as it is

impossible to cut a feeder to a length that

will be optimum for several bands. If a voltage

loop appears at the transmitter end of the line

on certain bands, parallel tuning of the feed-

ers will be required in order to get a transfer

of energy. It is impossible to transfer energy

by inductive coupling unless current is flow-

ing. This is effected at a voltage loop by the

presence of the resonant tank circuit formed

by parallel tuning of the antenna' coil.

21 -12 Line Discontinuities

In the previous discussion we have assumed

a transmission line which was uniform through-

out its length. In actual practice, this is

usually not the case.

Whenever there is any sudden change in the

characteristic impedance of the line, partial

reflection will occur at the point of discon-

tinuity. Some of the energy will be transmitted

and some reflected, which is essentially the

same as having some of the energy absorbed

and some reflected in so far as the effect upon

the line from the generator to that point is

concerned. The discontinuity can by ascribed

a reflection coefficient just as in the case of

an unmatched load.

In a simple case, such as a finite length of

uniform line having a characteristic impedance

of 500 ohms feeding into an infinite length of

uniform line having a characteristic impedance

of 100 ohms, the behavior is easily predicted.

The infinite 100 ohm lin& will have no standing

waves and will accept the same power from the

500 ohm line as would a 100 ohm resistor,

and the rest of the energy will be reflected at

the discontinuity to produce standing waves

from there back to the generator. However, in

the case of a complex discontinuity placed at

an odd distance down a line terminated in a

complex impedance, the picture becomes com-

plicated, especially when the discontinuity is

neither sudden nor gradual, but intermediate

between the two. This is the usual case with

amateur lines that must be erected around

buildings and trees.

In any case, when a discontinuity exists

somewhere on a line and is not a smooth,

gradual change embracing several wavelengths,

it is not possible to avoid standing waves

throughout the entire length of the line. If the

discontinuity is sharp enough and is great

enough to be significant, standing waves must

exist on one side of the discontinuity, and

may exist on both sides in many cases.

www.americanradiohistory.com

420 Radiation, Propagation and Lines THE RADIO

line fed by a transmitter. It is the reflection

from the antenna end which starts waves mov-

ing back toward the transmitter end. When

waves moving in both directions along a con-

ductor meet, standing waves are set up.

Semi -Resonant A well- constructed open -

Parallel -Wire Lines wire line has acceptably

low losses when its length

is less than about two wavelengths even when

the voltage standing -wave ratio is as high as

10 to 1. A transmission line constructed of

ribbon or tubular line, however, should have

the standing -wave ratio kept down to not more

than about 3 to 1 both to reduce power loss

and because the energy dissipation on the line

will be localized, causing overheating of the

line at the points of maximum current.

Because moderate standing waves can be

tolerated on open -wire lines without much loss,

a standing -wave ratio of 2/1 or 3/1 is con-

sidered acceptable with this type of line, even

when used in an untuned system. Strictly

speaking, a line is untuned, or non -resonant,

only when it is perfectly flat, with a standing -

wave ratio of 1 (no standing waves). However,

some mismatch can be tolerated with open -wire

untuned lines, so long as the reactance is not

objectionable, or is eliminated by cutting the

line to approximately resonant length.

21 -11 Tuned or

Resonant Lines

If a transmission line is terminated in its

characteristic surge impedance, there will be

no reflection at the end of the line, and the

current and voltage distribution will be uni-

form along the line. If the end of the line is

either open- circuited or short -circuited, the

reflection at the end of the line will be 100

per cent, and standing waves of very great am-

plitude will appear on the line. There will still

be practically no radiation from the line if it is

closely spaced, but voltage nodes will be

found every half wavelength, the voltage loops

corresponding to current nodes (figure 23).

If the line is terminated in some value of

resistance other than the characteristic surge

impedance, there will be some reflection, the

amount being determined by the amount of mis-

match. With reflection, there will be standing

waves (excursions of current and voltage)

along the line, though not to the same extent

as with an open- circuited or short- circuited

line. The current and voltage loops will occur

at the same points along the line as with the

open or short- circuited line, and as the ter-

minating impedance is made to approach the

characteristic impedance of the line, the cur-

1.0

o

1;)

t Zo

swR t.o ZL. Zo

SWR = 1.5 ZL. +.s on 0.e1 Zo

1.s

o SWR 3.0 ZL 3.0 OR 0.31 ZO

SWRo ZLooRo

Figure 23

STANDING WAVES ON A TRANS-

MISSION LINE

As shown at (A), the voltage and current are

constant on a transmission line which is

terminated in its characteristic impedance,

assuming that losses are small enough so

that they may be neglected. (B) shows the

variation in current or in voltage on a line

terminated in a load with a reflection co-

efficient of 0.2 so that a standing wave ratio

of 1.5 to I is set up. At (C) the reflection

coefficient has been increased to 0.5, with

the formation of a 3 to 1 standing -wove ratio

on the line. At (D) the line has been termi-

nated in a load which has a reflection co-

efficient of I.0 (short, open circuit, or a pure

reactance) so that all the energy is reflected

with the formation of an infinite standing -

wave ratio.

rent and voltage along the line will become

more uniform. The foregoing assumes, of

course, a purely resistive (non -reactive) load.

If the load is reactive, standing waves also

will be formed. But with a reactive load the

nodes will occur at different locations from

the node locations encountered with wrong -

value resistive termination.

A well built 500- to 600 -ohm transmission

line may be used as a resonant feeder for

lengths up to several hundred feet with very

low loss, so long as the amplitude of the

standing waves (ratio of maximum to minimum

voltage along the line) is not too great. The

www.americanradiohistory.com

THE RADIO Transmission Lines 419

ribbon and tubular configuration, with char-

acteristic impedance values from 75 to 300

ohms. Receiving types, and transmitting types

for power levels up to one kilowatt in the h -f

range, are listed with their pertinent char-

acteristics, in the table of figure 21.

Coaxial Line Several types of coaxial cable

have come into wide use for

feeding power to an antenna system. A cross -

sectional view of a coaxial cable (sometimes

called concentric cable or line) is shown in

figure 22.

As in the parallel -wire line, the power lost

in a properly terminated coaxial line is the

sum of the effective resistance losses along

the length of the cable and the dielectric

losses between the two conductors.

Of the two losses, the effective resistance

loss is the greater; since it is largely due to

the skin effect, the line loss (all other condi-

tions the same) will increase directly as the

square root of the frequency.

Figure 22 shows that, instead of having two

conductors running side by side, one of the

conductors is placed inside of the other. Since

the outside conductor completely shields the

inner one, no radiation takes place. The con-

ductors may both be tubes, one within the

other; the line may consist of a solid wire

within a tube, or it may consist of a stranded

or solid inner conductor with the outer con-

ductor made up of one or two wraps of copper

shielding braid.

In the type of cable most popular for mili-

tary and non -commercial use the inner con-

ductor consists of a heavy stranded wire, the

outer conductor consists of a braid of copper

wire, and the inner conductor is supported

within the outer by means of a semi -solid

dielectric of exceedingly low loss character-

istics called polyethylene. The Army -Navy

designation on one size of this cable suitable

for power levels up to one kilowatt at fre-

quencies as high as 30 Mc. is AN /RG -8 /U.

The outside diameter of this type of cable is

approximately one -half inch. The character-

istic impedance of this cable type is 52 ohms,

but other similar types of greater and smaller

power- handling capacity are available in im-

pedances of 52, 75, and 95 ohms.

When using solid dielectric coaxial cable

it is necessary that precautions be taken to

insure that moisture cannot enter the line. If

the better grade of connectors manufactured

for the line are employed as terminations, this

condition is automatically satisfied. If con-

nectors are not used, it is necessary that

some type of moisture -proof sealing compound

be applied to the end of the cable where it

will be exposed to the weather.

Nearby metallic objects cause no loss, and

coaxial cable may be run up air ducts or ele-

204

170

us

loo

70

s2

30

o 2.81 5 7

3.21

RATIO OF DIAMETERS

to iS 30

Zo7SSLOGp

COAXIAL OR

CONCENTRIC LINE

Di. INSIDE DIAMETER OF

OUTER CONDUCTOR

D= OUTSIDE DIAMETER OF

INNER CONDUCTOR

Figure 22

CHARACTERISTIC IMPEDANCE OF AIR -

FILLED COAXIAL LINES

If the filling of the line is o dielectric ma-

terial other than air, the characteristic im-

pedance of the line will be reduced by a

factor proportional to the square -root of the

dielectric constant of the material used as a

dielectric within the line.

vator shafts, inside walls, or through metal

conduit. Insulation troubles can be forgotten.

The coaxial cable may be buried in the ground

or suspended above ground.

Standing Waves Standing waves on a trans-

mission line always are the

result of the reflection of energy. The only

significant reflection which takes place in a

normal installation is that at the load end of

the line. But reflection can take place from

discontinuities in the line, such as caused by

insulators, bends, or metallic objects adjacent

to an unshielded line.

When a uniform transmission line is termi-

nated in an impedance equal to its surge im-

pedance, reflection of energy does not occur,

and no standing waves are present. When the

load termination is exactly the same as the

line impedance, it simply means that the load

takes energy from the line just as fast as the

line delivers it, no slower and no faster.

Thus, for proper operation of an untuned

line (with standing waves eliminated), some

form of impedance- matching arrangement must

be used between the transmission line and

the antenna, so that the radiation resistance

of the antenna is reflected back into the line

as a nonreactive impedance equal to the line

impedance.

The termination at the antenna end is the

only critical characteristic about the untuned

www.americanradiohistory.com

418 Radiation, Propagation and Lines THE RADIO

CHARACTERISTICS OF COMMON TRANSMISSION LINES

ATTENUATION

db/ 00 FEET

vswR =1.0 VELO-

CITYUUFD

PER

FT

REMARKS

30 Mc 100 MC

1-

300 MC

ACTOR

V

OPEN WIRE LINE, N' 12

COPPER. 0.15 0.3 0.6 O.q8 -099 - BASED UPON 4" SPACING BELOW 50 MC ; 2- SPACING ABOVE 50 MC. RADIATION

LOSSES INCLUDED. CLEAN, LOW LOSS CERAMIC INSULATION ASSUMED RADIATION

HIGH ABOVE 150 MC

RIBBON LINE, REC.TYPE,

300 OHMS.

(7/2e CONDUCTORS) 0.86 2.2 5.3 .,

0.62 Ny

6 FOR CLEAN. DRY LINE. wET WEATHER PERFORMANCE RATHER POOR BEST LINE IS

SLIGHTLY CONVEX. AVOID LINE THAT HAS CONCAVE DIELECTRIC SUITABLE FOR

LOW POWER TRANSMITTING APPLICATIONS. LOSSES INCREASE AS LINE WEATHERS.

HANDLES 400 WATTS AT 30 MC. IF VSWR IS LOW.

TUBULAR "TWIN-LEAD"

REC TYPE. 300 OHMS,

S /16.0.0., (AMPHENOL

TYPE 1-271) - - - - - CHARACTERISTICS SIMILAR TO RECEIVING TYPE RIBBON LINE EXCEPT FOR MUCH

BETTER wET WEATHER PERFORMANCE.

RIBBON LINE, TRANS.

TYPE. 300 OHMS. - - - - - CHARACTERISTICS VARY SOMEWHAT WITH MANUFACTURER. BUT APPROXIMATE

THOSE OF RECEIVING TYPE RIBBON EXCEPT FOR GREATER POWER HANDLING

CAPABILITY AND SLIGHTLY BETTER WET WEATHER PERFORMANCE.

TUBULAR "TWIN -LEAD-

TRANS. TYPE, 7/160.D.

(AMPHENOL 14 -076) 0.65 2 3 5.4 0.79 8.1 FOR USE WHERE RECEIVING TYPE TUBULAR -TWIN -LEAD DOES NOT HAVE SUM-

CIENT POWER HANDLING CAPABILITY. WILL HANDLE / KW AT 30 MC. I F VSWR

IS LOW.

RIBBON LIKE, RECEIVE

TYPE, ISO OHMS. 1 1 2 7 6 O 0 77 V' 10 USEFUL FOR QUARTER WAVE MATCHING SECTIONS. NO LONGER WIDELY USED

AS A LINE.

RIBBON LINE, RECEIVE.

TYPE, 75 OHMS. 2 O 5 O 11.0 o.BB' 19 V' USEFUL MAINLY IN THE H -F RANGE BECAUSE OF EXCESSIVE LOSSES AT V -H -F

AND U-H-F. LESS AFFECTED BY WEATHER THAN 300 OHM_RIBBON.

RIBBON LINE, TRANS.

TYPE, 75 OHMS. 1.5 3.9. 6.0 0.71Y f 6`t VERY SATISFACTORY FOR TRANSMITTING APPLICATIONS BELOW 30 MC. AT

POWERS UP TO 1 KW. NOT SIGNIFICANTLY AFFECTED BY WET WEATHER.

RG-6/U COAX (52 OHMS) 1.0 2.1 4.2 0.88 29.5 WILL HANDLE 2 KW AT 4O MC. IF VSWR IS LOW. 0.. O.D. 7/21 CONDUCTOR.

RG-11 /U COAX (75 OHMS) 0.94 I 9 3.6 0.88 20.5 WILL HANDLE 1. KW AT 30 MC. IF VSWR IS LOW. 0 4 "0.0. 7/28 CONDUCTOR.

RG -17 /U COAX (520HMS) 0.38 0.85 1.8 0.66 29.5 WILL HANDLE 7 8 KW. AT 30 MC. IF VSWR IS LOW. 087" OD. 0.19" DIA. CONDUCTOR

RG -58/U COAX (53 OHMS) 1.95 4.1 8.0 0.66 28.5 WILL HANDLE 430 WATTS AT 30 MC. IF VSWR IS LOW. 0.2000. N 20 CONDUCTOR.

RG-S9 /U COAX (73 OHMS) 1.9 3.8 7.0 0.66 21 WILL HANDLE 680 WATTS AT 30 MC. IF VSWR IS LOW. 0.24" O.D. N 22 CONDUCTOR.

TV -59 COAX (720HMS) 2.0 4.0 7.0 0.66 22 COMMERCIAL VERSION OF RG-59/U FOR LESS EXACTING APPLICATIONS. LESS

EXPENSIVE.

RG -22/U SHIELDED

PAIR (95 OHMS) 1.7 3.0 5.5 0.66 18 FOR SHIELDED, BALANCED -TO- GROUND APPLICATIONS. VERY LOW NOISE

PICK UP. 0.4" 0.D.

K -I11 SHIELDED PAIR

(300 OHMS) 2.0 3.5 6.1 - 4 DESIGNED FOR TV LEAD -IN IN NOISY LOCATIONS. LOSSES HIGHER THAN

REGULAR 300 OHM RIBBON, BUT DO NOT INCREASE AS MUCH FROM WEATHERING

0 APPROXIMATE. EXACT FIGURE VARIES SLIGHTLY WITH MANUFACTURER

FIGURE 21

2S

Z. = 276 1og10-

d

Where:

S is the exact distance between wire centers

in some convenient unit of measurement, and

d is the diameter of the wire measured in the

same units as the wire spacing, S.

2S

Since - expresses a ratio only, the units

d

of measurement may be centimeters, milli-

meters, or inches. This makes no difference

in the answer, so long as the substituted

values for S and d are in the same units.

The equation is accurate so long as the

wire spacing is relatively large as compared

to the wire diameter.

Surge impedance values of less than 200

ohms are seldom used in the open -type two -

wire line, and, even at this rather high value

of Z. the wire spacing S is uncomfortably

close, being only 2.7 times the wire diameter d.

Figure 20 gives in graphical form the surge

impedance of practicable two -wire lines. The

chart is self -explanatory, and is sufficiently

accurate for practical purposes.

Ribbon and Instead of using spacer in-

Tubular Trans- sulators placed periodically

mission Line along the transmission line

it is possible to mold the

line conductors into a ribbon or tube of flex-

ible low -loss dielectric material. Such line,

with polyethylene dielectric, is used in enor-

mous quantities as the lead -in transmission

line for FM and TV receivers. The line is

available from several manufacturers in the

www.americanradiohistory.com

HANDBOOK Transmission Lines 417

ever, mechanical or electrical considerations

often make one type of transmission line better

adapted for use to feed a particular type of

antenna than any other type.

Transmission lines for carrying r -f energy

are of two general types: non -resonant and

resonant. A non -resonant transmission line

is one on which a successful effort has been

made to eliminate reflections from the termi-

nation (the antenna in the transmitting case

and the receiver for a receiving antenna) and

hence one on which standing waves do not

exist or are relatively small in magnitude. A

resonant line, on the other hand, is a trans-

mission line on which standing waves of ap-

preciable magnitude do appear, either through

inability to match the characteristic impedance

of the line to the termination or through in-

tentional design.

The principal types of transmission line in

use or available at this time include the open -

wire line (two -wire and four -wire types), two -

wire solid -dielectric line ( "Twin- Lead" and

similar ribbon or tubular types), two -wire poly-

ethylene- filled shielded line, coaxial line of

the solid -dielectric, beaded, stub -supported,

or pressurized type, rectangular and cylindrical

wave guide, and the single -wire feeder oper-

ated against ground. The significant charac-

teristics of the more popular types of trans-

mission line available at this time are given

in the chart of figure 21.

21 -10 Non -Resonant

Transmission Lines

A non -resonant or untuned transmission line

is a line with negligible standing waves.

Hence, a non -resonant line is a line carrying

r -f power only in one direction -from the source

of energy to the load.

Physically, the line itself should be iden-

tical throughout its length. There will be a

smooth distribution of voltage and current

throughout its length, both tapering off very

slightly towards the load end of the line as a

result of line losses. The attenuation (loss)

in certain types of untuned lines can be kept

very low for line lengths up to several thou-

sand feet. In other types, particularly where

the dielectric is not air (such as in the twisted -

pair line), the losses may become excessive

at the higher frequencies, unless the line is

relatively short.

Transmission -Line

Impedance

ance. Neglecting

minor importance

All transmission lines have

distributed inductance,

capacitance and resist -

the resistance, as it is of

in short lines, it is found

1111111111ESSMINI%s's_í.s

%%M III / .--

ILW IIM/11111Ï11

/Oií%E'/_Er1111111111

iiilMEM ;m t=1 111111111

s . io is 3 s + w w

INCHES. CENTER TO CENTER

Figure 20

CHARACTERISTIC IMPEDANCE OF TYPI-

CAL TWO -WIRE OPEN LINES

u

that the inductance and capacitance per unit

length determine the characteristic or surge

impedance of the line. Thus, the surge im-

pedance depends upon the nature and spacing

of the conductors, and the dielectric sepa-

rating them.

Speaking in electrical terms, the charac-

teristic impedance of a transmission line is

simply the ratio of the voltage across the line

to the current which is flowing, the same as

is the case with a simple resistor: Z. = E /1.

Also, in a substantially loss -less line (one

whose attenuation per wavelength is small)

the energy stored in the line will be equally

divided between the capacitive field and the

inductive field which serve to propagate the

energy along the line. Hence the character-

istic impedance of a line maybe expressed as:

Z. = N/ L/C.

Two -Wire A two -wire transmission system

Open Line is easy to construct. Its surge im-

pedance can be calculated quite

easily, and when properly adjusted and bal-

anced to ground, with a conductor spacing

which is negligible in terms of the wave-

length of the signal carried, undesirable feeder

radiation is minimized; the current flow in

the adjacent wires is in opposite directions,

and the magnetic fields of the two wires are

in opposition to each other. When a two -wire

line is terminated with the equivalent of a

pure resistance equal to the characteristic

impedance of the line, the line becomes a non -

resonant line.

Expressed in physical terms, the character-

istic impedance of a two -wire open line is

equal to:

www.americanradiohistory.com

416 Radiation, Propagation and Lines THE RADIO

dent, particularly a "flutter fade" and a char-

acteristic "hollow" or echo effect.

Deviations from a great circle path are es-

pecially noticeable in the case of great circle

paths which cross or pass near the auroral

zones, because in such cases there often is

complete or nearly complete absorption of the

direct sky wave, leaving off -path scattered

reflections the only mechanism of propagation.

Under such conditions the predominant wave

will appear to arrive from a direction closer

to the equator, and the signal will be notice-

ably if not considerably weaker than a direct

sky wave which is received under favorable

conditions.

Irregular reflection of radio waves from

"scattering patches" is divided into two cate-

gories: "short scatter" and "long scatter ".

Short scatter is the scattering that occurs

when a radio wave first reaches the scattering

patches or media. Ordinarily it is of no parti-

cular benefit, as in most cases it only serves

to fill in the inner portion of the skip zone

with a weak, distorted signal.

Long scatter occurs when a wave has been

refracted from the F2 layer and strikes scatter-

ing patches or media on the way down. When

the skip distance exceeds several hundred

miles, long scatter is primarily responsible

for reception within the skip zone, particu-

larly the outer portion of the skip zone. Dis-

tortion is much less severe than in the case

of short scatter, and while the signal is like-

wise weak, i t sometimes can be utilized for

satisfactory communication.

During a severe ionosphere disturbance in

the north auroral zone, it sometimes is possible

to maintain communication between the Eastern

United States and Northern Europe by the fol-

lowing mechanism: That portion of the energy

which is radiated in the direction of the great

circle path is completely absorbed upon reach-

ing the auroral zone. However, the portion of

the wave leaving the United States in a south-

easterly direction is refracted downward from

the F2 layer and encounters scattering patches

or media on its downward trip at a distance

of approximately 2000 miles from the trans-

mitter. There it is reflected by "long scatter"

in all directions, this scattering region acting

like an isotropic radiator fed with a very small

fraction of the original transmitter power. The

great circle path from this southerly point to

northern Europe does not encounter unfavor-

able ionosphere conditions, and the wave is

propagated the rest of the trip as though it had

been radiated from the scattering region.

Another type of scatter is produced when

a sky wave strikes certain areas of the earth.

Upon striking a comparatively smooth surface

such as the sea, there is little scattering, the

wave being shot up again by what could be

considered specular or mirror reflection. But

upon striking a mountain range, for instance,

the reradiation or reflected energy is scattered,

some of it being directed back towards the

transmitter, thus providing another mechanism

for producing a signal within the skip zone.

Metéors and When a meteor strikes the earth's

"Bursts" atmosphere, a cylindrical region

of free electrons is formed at

approximately the height of the E layer. This

slender ionized column is quite long, and when

first formed is sufficiently dense to reflect

radio waves back to earth most readily, in-

cluding v -h -f waves which are not ordinarily

returned by the F= layer.

The effect of a single meteor, of normal size,

shows up as a sudden "burst" of signal of

short duration at points not ordinarily reached

by the transmitter. After a period of from 10

to 40 seconds, recombination and diffusion

have progressed to the point where the effect

of a single fairly large meteor is not percep-

tible. However, there are many small meteors

impinging upon earth's atmosphere every min-

ute, and the aggregate effect of their transient

ionized trails, including the small amount of

residual ionization that exists for several

minutes after the original flash but is too weak

and dispersed to prolong a "burst ", is be-

lieved to contribute to the existence of the

"nighttime E" layer, and perhaps also to

sporadic E patches.

While there are many of these very small

meteors striking the earth's atmosphere every

minute, meteors of normal size (sufficiently

large to produce individual "bursts ") do not

strike nearly so frequently except during some

of the comparatively rare meteor "showers ".

During one of these displays a "quivering"

ionized layer is produced which is intense

enough to return signals in the lower v -h -f

range with good strength, but with a type of

"flutter" distortion which is characteristic

of this type of propagation.

21 -9 Transmission Lines

For many reasons it is desirable to place

an antenna or radiating system as high and in

the clear as is physically possible, utilizing

some form of nonradiating transmission line

to carry energy with as little loss as possible

from the transmitter to the radiating antenna,

and conversely from the antenna to the re-

ceiver.

There are many different types of transmis-

sion lines and, generally speaking, practically

any type of transmission line or feeder system

may be used with any type of antenna. How-

www.americanradiohistory.com

HANDBOOK 11 -Year Sunspot Cycle 415

225

200

175

150

125

100

75

50

25

o

i' ,

'5 é ; t

C

ri

ó tt

; , ,-- `

I 2

3 . ,

YEAR

48 50 52 54 58 56 80 62 64 86

Figure 18

THE YEARLY TREND OF THE SUNSPOT

CYCLE. RADIO CONDITIONS IN GEN-

ERAL WILL DETERIORATE DURING 1960-

1965 AS THE CYCLE DECLINES.

zon, the farther away will the wave return to

earth, and the greater the skip distance. The

wave can be reflected back up into the iono-

sphere by the earth, and then be reflected back

down again, causing a second skip distance

area. The drawing of figure 19 shows the mul-

tiple reflections possible. When the receiver

receives signals which have traveled over

more than one path between transmitter and

receiver, the signal impulses will not all arrive

at the same instant, as they do not all travel

the same distance. When two or more signals

arrive in the same phase at the receiving an-

tenna, the resulting signal in the receiver will

be quite strong. On the other hand, if the sig-

nals arrive 180° out of phase, so they tend to

cancel each other, the received signal will

drop -perhaps to zero if perfect cancellation

occurs. This explains why high -frequency

signals are subject to fading.

Fading can be greatly reduced on the high

frequencies by using a transmitting antenna

with sharp vertical directivity, thus cutting

down the number of possible paths of signal

arrival. A receiving antenna with similar char-

acteristics (sharp vertical directivity) will

further reduce fading. It is desirable, when

using antennas with sharp vertical directivity,

to use the lowest vertical angle consistent

with good signal strength for the frequency

used.

Scattered

Reflections Scattered reflections are random,

diffused, substantially isotropic

reflections which are partly re-

TRANSMITTER

Figure 19

IONOSPHERE -REFLECTION WAVE PATHS

Showing typical ionosphere- reflection wave

paths during daylight hours when ionization

density is such that frequencies as high as

28 Mc. will he returned to earth. The dis-

tance between ground -wave range and that

range where the ionosphere -reflected wove of

a specific frequency first will be returned to

earth is called the skip distance.

sponsible for reception within the skip zone,

and for reception of signals from directions

off the great circle path.

In a heavy fog or mist, it is difficult to see

the road at night because of the bright glare

caused by scattered reflection of the head-

light beam by the minute droplets. In fact, the

road directly to the side of the car will be

weakly illuminated under these conditions,

whereas it would riot on a clear night (assum-

ing flat, open country). This is a good example

of propagation of waves by scattered reflec-

tions into a zone which otherwise would not

be illuminated.

Scattering occurs in the ionosphere at all

times, because of irregularities in the medi-

um (which result in "patches" corresponding

to the water droplets) and because of random -

phase radiation due to the collision or recom-

bination of free electrons. However, the nature

of the scattering varies widely with time, in

a random fashion. Scattering is particularly

prevalent in the f: region, but scattered re-

flections may occur at any height, even well

out beyond the virtual height of the /-'2 layer.

There is no "critical frequency" or "low-

est perforating frequency" involved in the

scattering mechanism, though the intensity

of the scattered reflections due to typical

scattering in the F. region of the ionosphere

decreases with frequency.

CChen the received signal is due primarily

to scattered reflections, as is the case in the

skip zone or where the great circle path does

not provide a direct sky wave (due to low crit-

ical or perforation frequency, or to an iono-

sphere storm) very bad distortion will be evi-

www.americanradiohistory.com

414 Radiation, Propagation and Lines T H E R A D I O

ù ï

U Z w a

rc a

10

34

32

30

2e WINTER

SUNSPOT

MAXIMUM

26

24 1 l'

22

20

III

16 SUMMER

SUNSPOT-

14 -.1,, MINIMUM

12 -

ii'

e

4

2

0 2 6 e 10 12 14 16 16 20 22 24

LOCAL TIME

Figure 17

TYPICAL CURVES SHOWING CHANGE IN

M.U.F. AT MAXIMUM AND MINIMUM

POINTS IN SUNSPOT CYCLE

The m.u.f. often drops to frequencies below

10 Mc. in the early morning hours. The high

m.u.f. in the middle of the day is brought about

by reflection from the F2 layer. M.u.f. data is

published periodically in the magazines de-

voted to amateur work, and the m.u.f. can be

calculated with the aid of Basic Radio Propa-

gation Predictions, CRPL -D, published month-

ly by the Government Printing Office, Wash-

ington, D.C.

Absorption and The optimum working fre-

Optimum Working quency for any particular

Frequency direction and distance is

usually about 15 per cent

less than the m.u.f. for contact with that par-

ticular location. The absorption by the iono-

sphere becomes greater and greater as the

operating frequency is progressively lowered

below the m.u.f. It is this condition which

causes signals to increase tremendously in

strength on the 14 -Mc. and 28 -Mc. bands just

before the signals drop completely out. At the

time when the signals are greatest in ampli-

tude the operating frequency is equal to the

m.u.f. Then as the signals drop out the m.u.f.

has become lower than the operating frequency.

Skip Distance The shortest distance from a

transmitting location at which

signals reflected from the ionosphere can be

returned to the earth is called the skip dis-

tance. As was mentioned above under Critical

Frequency there is no skip distance for a fre-

quency below the critical frequency of the

most highly ionized layer of the ionosphere

at the time of transmission. However, the skip

distance is always present on the 14 -Mc. band

and is almost always present on the 3.5 -Mc.

and 7 -Mc. bands at night. The actual measure

of the skip distance is the distance between

the point where the ground wave falls to zero

and the point where the sky wave begins to

return to earth. This distance may vary from

40 to 50 miles on the 3.5-Mc. band to thou-

sands of miles on the 28 -Mc. band.

The Sporadic -E Occasional patches of ex-

Loyer tremely high ionization den-

sity appear at intervals

throughout the year at a height approximately

equal to that of the F layer. These patches,

called the sporadic -F. layer may be very small

or may be up to several hundred miles in ex-

tent. The critical frequency of the sporadic -F

layer may be greater than twice that of the

normal ionosphere layers which exist at the

same time.

It is this sporadic -E condition which pro-

vides "short- skip" contacts from 400 to per-

haps 1200 miles on the 28 -Mc. band in the

evening. It is also the sporadic -E condition

which provides the more common type of "band

opening" experienced on the 50 -Mc. band when

very loud signals are received from stations

from 400 to 1200 miles distant.

Cycles in The ionization density of

Ionosphere Activity the ionosphere is deter-

mined by the amount of

radiation (probably ultra violet) which is be-

ing received from the sun. Consequently, iono-

sphere activity is a function of the amount of

radiation of the proper character being emitted

by the sun and is also a function of the rela-

tive aspect of the regions in the vicinity of

the location under discussion to the sun. There

are four main cycles in ionosphere activity.

These cycles are: the daily cycle which is

brought about by the rotation of the earth, the

27 -day cycle which is caused by the rotation

of the sun, the seasonal cycle which is caused

by the movement of the earth in its orbit, and

the 11 -year cycle which is a cycle in sunspot

activity. The effects of these cycles are super-

imposed insofar as ionosphere activity is con-

cerned. Also, the cycles are subject to short

term variations as a result of magnetic storms

and similar terrestrial disturbances.

The most recent minimum of the 11 -year sun-

spot cycle occurred during the winter of 1954-

1955, and we are currently moving along the

slope of a new cycle, the maximum of which

occurred during 1958. The current cycle is pic-

tured in figure 18.

Fading The lower the angle of radiation of

the wave, with respect to the hori-

www.americanradiohistory.com

HANDBOOK Ionospheric Propagation 413

200

130

too

F2

Ft MID DAY

D

E

too

50

F2

MIDNIGHT

IONIZATION DENSITY -a

Figure 16

IONIZATION DENSITY IN THE IONO-

SPHERE

Showing typical ionization density of the

ionosphere in mid -summer. Note that the Ft

and D layers disappear at night, and that the

density of the E layer falls to such a low

value that it is ineffective.

which the sky wave can undergo depends up-

on its frequency, and the amount of ionization

in the ionosphere, which is in turn dependent

upon radiation from the sun. The sun increases

the density of the ionosphere layers (figure 16)

and lowers their effective height. For this

reason, the ionosphere acts very differently

at different times of day, and at different times

of the year.

The higher the frequency of a radio wave,

the farther it penetrates the ionosphere, and

the less it tends to be bent back toward the

earth. The lower the frequency, the more easily

the waves are bent, and the less they pene-

trate the ionosphere. 160 -meter and 80 -meter

signals will usually be bent back to earth

even when sent straight up, and may be con-

sidered as being reflected rather than refract-

ed. As the frequency is raised beyond about

5,000 kc. (dependent upon the critical frequen-

cy of the ionosphere at the moment), it is

found that waves transmitted at angles higher

than a certain critical angle never return to

earth. Thus, on the higher frequencies, it is

necessary to confine radiation to low angles,

since the high angle waves simply penetrate

the ionosphere and are lost.

The F2 Layer The higher of the two major

reflection regions of the iono-

sphere is called the F, layer. This layer has

a virtual height of approximately 175 miles at

night, and in the daytime it splits up into two

layers, the upper one being called the F, layer

and the lower being called the F, layer. The

height of the F2 layer during daylight hours is

normally about 250 miles on the average and

the F, layer often has a height of as low as

140 miles. It is the F2 layer which supports

all nighttime dx communication and nearly all

daytime dx propagation.

The E Layer Below the F2 layer is another

layer, called the E layer, which

is of importance in daytime communication

over moderate distances in the frequency range

between 3 and 8 Mc. This layer has an almost

constant height at about 70 miles. Since the

re- combination time of the ions at this height

is rather short, the E layer disappears almost

completely a short time after local sunset.

The D Layer Below the E layer at a height of

about 35 miles is an absorbing

layer, called the D layer, which exists in the

middle of the day in the summertime. The layer

also exists during midday in the winter time

during periods of high solar activity, but the

layer disappears completely at night. It is this

layer which causes high absorption of signals

in the medium and high- frequency range during

the middle of the day.

Critical Frequency The critical frequency of

an ionospheric layer is the

highest frequency which will be reflected when

the wave strikes the layer at vertical inci-

dence. The critical frequency of the most high-

ly ionized layer of the ionosphere may be as

low as 2 Mc. at night and as high as 12 to 13

Mc. in the middle of the day. The critical fre-

quency is directly of interest in that a skip -

distance zone will exist on all frequencies

greater than the highest critical frequency at

that time. The critical frequency is a measure

of the density of ionization of the reflecting

layers. The higher the critical frequency the

greater the density of ionization.

Maximum Usable The maximum usable /re-

Frequency quency or m.u. /. is of great

importance in long- distance

communication since this frequency is the high-

est that can be used for communication be-

tween any two specified areas. The m.u.f. is

the highest frequency at which a wave pro-

jected into space in a certain direction will

be returned to earth in a specified region by

ionospheric reflection. The m.u.f. is highest

at noon or in the early afternoon and is high-

est in periods of greatest sunspot activity,

often going to frequencies higher than 50 Mc.

(figure 17).

www.americanradiohistory.com

412 Radiation, Propagation and Lines THE RADIO

T INVERSION

__

DUCT

T

INVERSION AND DUCT

REFRACTIVE INDEX

Figure 15

ILLUSTRATING DUCT TYPES

Showing two types of variation in refractive

index with height which will give rise to the

formation of a duct. An elevated duct is

shown at (A), and o ground -based duct is

shown at (B). Such ducts can propagate

ground -wave signals far beyond their normal

range.

may give rise to the formation of a duct which

can propagate waves with very little attenu-

ation over great distances in a manner similar

to the propagation of waves through a wave

guide. Guided propagation through a duct in

the atmosphere can give quite remarkable

transmission conditions (figure 15). However,

such ducts usually are formed only on an over -

water path. The depth of the duct over the

water 's surface may be only 20 to 50 feet, or

it may be 1000 feet deep or more. Ducts ex-

hibit a low -frequency cutoff characteristic

similar to a wave guide. The cutoff frequency

is determined by depth of the duct and by the

strength of the discontinuity in refractive in-

dex at the upper surface of the duct. The low -

est'frequency that can be propagated by such

a duct seldom goes below 50 Mc., and usually

will be greater than 100 Mc. even along the

Pacific Coast.

Stratospheric Communication by virtue of

Reflection stratospheric reflection can be

brought about during magnetic

storms, aurora borealis displays, and during

meteor showers. Dx communication during ex-

tensive meteor showers is characterized by

frequent bursts of great signal strength fol-

lowed by a rapid decline in strength of the

received signal. The motion of the meteor

forms an ionized trail of considerable extent

which can bring about effective reflection of

signals. However, the ionized region persists

only for a matter of seconds so that a shower

of meteors is necessary before communication

becomes possible.

The type of communication which is possible

during visible displays of the aurora borealis

and during magnetic storms has been called

aurora -type dx. These conditions reach a max-

imum somewhat after the sunspot cycle peak,

possibly because the spots on the sun are

nearer to its equator (and more directly in line

with the earth) in the latter part of the cycle.

Ionospheric storms generally accompany mag-

netic storms. The normal layers of the iono-

sphere may be churned or broken up, making

radio transmission over long distances diffi-

cult or impossible on high frequencies. Un-

usual conditions in the ionosphere sometimes

modulate v -h -f waves so that a definite tone or

noise modulation is noticed even on transmit-

ters located only a few miles away.

A pecularity of this type of auroral propaga-

tion of v -h -f signals in the northern hemisphere

is that directional antennas usually must be

pointed in a northerly direction for best results

for transmission or reception, regardless of

the direction of the other station being con-

tacted. Distances out to 700 or 800 miles have

been covered during magnetic storms, using

30 and 50 Mc. transmitters, with little evi-

dence of any silent zone between the stations

communicating with each other. Generally,

voice -modulated transmissions are difficult or

impossible due to the tone or noise modulation

on the signal. Most of the communication of

this type has taken place by c.w. or by tone

modulated waves with a keyed carrier.

21-8 Ionospheric

Propagation

Propagation of radio waves for communica-

tion on frequencies between perhaps 3 and

30 Mc. is normally carried out by virtue of

ionospheric reflection or refraction. Under con-

ditions of abnormally high ionization in the

ionosphere, communication has been known

to have taken place by ionospheric reflection

on frequencies higher than 50 Mc.

The ionosphere consists of layers of ionized

gas located above the stratosphere, and ex-

tending up to possibly 300 miles above the

earth. Thus we see that high- frequency radio

waves may travel over short distances in a

direct line from the transmitter to the receiver,

or they can be radiated upward into the iono-

sphere to be bent downward in an indirect ray,

returning to earth at considerable distance

from the transmitter. The wave reaching a re-

ceiver via the ionosphere route is termed a

sky wave. The wave reaching a receiver by

traveling in a direct line from the transmitting

antenna to the receiving antenna is commonly

called a ground wave.

The amount of bending at the ionosphere

www.americanradiohistory.com

HANDBOOK Ground Wave Communication 411

TRANSMITTING

ANTENNA Di DIRECT WAVES

GROUND- REFLECTED D2

WAVES

e e

RECEIVING

D3 ANTENNA

AT DIFFERENT

HEIGHTS

Figure 14

WAVE INTERFERENCE WITH HEIGHT

When the source of a horizontally -polarized

space -wave signal is above the horizon, the

received signal at a distant location will go

through a cyclic variation as the antenna

height is progressively raised. This is due

to the difference in total path length between

the direct wove and the ground -reflected

wave, and to the fact that this path length

difference changes with antenna height.

When the path length difference is such that

the two waves arrive at the receiving anten-

na with a phase difference of 3600 or some

multiple of 3600, the two waves will appear

to be in phase as for as the antenna is con-

cerned and maximum signal will be obtained.

On the other hand, when the antenna height

is such that the path length difference for

the two waves causes the waves to arrive

with a phase difference of an odd multiple

of 1800 the two waves will substantially can-

cel, and a null will be obtained at that an-

tenna height. The difference between DI

and D2 plus D3 is the path- length difference.

Note also that there is an additional 1800

phase shift in the ground -reflected wave at

the point where it is reflected from the

ground. It is this latter phase shift which

causes the space -wave field intensity of a

horizontally polarized wave to be zero with

the receiving antenna at ground level.

d is in miles and the antenna height N is in

feet. This equation must be applied separately

to the transmitting and receiving antennas and

the results added. However, refraction and

diffraction of the signal around the spherical

earth cause a smaller reduction in field strength

than would occur in the absence of such bend-

ing, so that the average radio horizon is some-

what beyond the geometrical horizon. The

equation d = 1.4 N,/ f is sometimes used for

determining the radio horizon.

Tropospheric Propagation by signal bending

Propagation in the lower atmosphere, called

tropospheric propagation, can

result in the reception of signals over a much

greater distance than would be the case if the

lower atmosphere were homogeneous. In a

homogeneous or well -mixed lower atmosphere,

called a normal or standard atmosphere, there

is a gradual and uniform decrease in index of

refraction with height. This effect is due to

the combined effects of a decrease in temper-

ature, pressure, and water -vapor content with

height.

This gradual decrease in refractive index

with height causes waves radiated at very low

angles with respect to the horizontal to be

bent downward slightly in a curved path. The

result of this effect is that such waves will be

propagated beyond the true or geometrical

horizon. In a so- called standard atmosphere

the effect of the curved path is the same as

though the radius of the earth were increased

by approximately one third. This condition ex-

tends the horizon by approximately 30 per cent

for normal propagation, and the extendedhori-

zon is known as the radio path horizon, men-

tioned before.

Conditions Leading to When the temperature,

Tropospheric pressure, or water -vapor

Stratification content of the atmos-

phere does not change

smoothly with rising altitude, the discontinuity

or stratification will result in the reflection

or refraction of incident v -h -f signals. Ordi-

narily this condition is more prevalent at night

and in the summer. In certain areas, such as

along the west coast of North America, it is

frequent enough to be considered normal. Sig-

nal strength decreases slowly with distance

and, if the favorable condition in the lower

atmosphere covers sufficient area, the range

is limited only by the transmitter power, an-

tenna gain, receiver sensitivity, and signal -to-

noise ratio. There is no skip distance. Usually,

transmission due to this condition is accom-

panied by slow fading, although fading can be

violent at a point where direct waves of about

the same strength are also received.

Bending in the troposphere, which refers to

the region from the earth's surface up to about

10 kilometers, is more likely to occur on days

when there are stratus clouds than on clear,

cool days with a deep blue sky. The tempera-

ture or humidity discontinuities may be broken

up by vertical convection currents over land

in the daytime but are more likely to continue

during the day over water. This condition is

in some degree predictable from weather infor-

mation several days in advance. It does not

depend on the sunspot cycle. Like direct com-

munication, best results require similar an-

tenna polarization or orientation at both the

transmitting and receiving ends, whereas in

transmission via reflection in the ionosphere

(that part of the atmosphere between about 50

and 500 kilometers high) it makes little dif-

ference whether antennas are similarly polar-

ized.

Duct Formation When bending conditions are

particularly favorable they

www.americanradiohistory.com

41 0 Radiation, Propayation and Lines THE RADIO

@DIRECT WAVE

©GROUND -REFLECTED

WAVE l-

@SURFACE WAVE -- ' - - - -

Figure 13

GROUND -WAVE SIGNAL PROPAGATION

The illustration above shows the three com-

ponents of the ground wave: (A), the surface

wave; (B), the direct wave; and (C), the

ground -reflected wove. The direct wave and

the ground -reflected wove combine at the

receiving antenna to make up the space

wive.

may take place as a result of the ground wave,

or as a result of the sky wave or ionospheric

wave.

The Ground Wave The term ground wave actu-

ally includes several dif-

ferent types of waves which usually are called:

(1) the surface wave, (2) the direct wave, and

(3) the ground -reflected wave. The latter two

waves combine at the receiving antenna to

form the resultant wave or the space wave.

The distinguishing characteristic of the com-

ponents of the ground wave is that all travel

along or over the surface of the earth, so that

they are affected by the conductivity and ter-

rain of the earth's surface.

The Ionospheric Wove Intense bombardment of

or Sky Wave the upper regions of

the atmosphere by radi-

ations from the sun results in the formation

of ionized layers. These ionized layers, which

form the ionosphere, have the capability of

reflecting or refracting radio waves which im-

pinge upon them. A radio wave which has been

propagated as a result of one or more reflec-

tions from the ionosphere is known as an

ionospheric wave or a sky wave. Such waves

make possible long distance radio communica-

tion. Propagation of radio signals by iono-

spheric waves is discussed in detail in Sec-

tion 21 -8.

21 -7 Ground -Wave

Communication

As stated in the preceding paragraph, the

term ground wave applies both to the surface

wave and to the space wave (the resultant

wave from the combination of the direct wave

and the ground -reflected wave) or to a com-

bination of the two. The three waves which

may combine to make up the ground wave are

illustrated in figure 13.

The Surface Wove The surface wave is that

wave which we normally

receive from a standard broadcast station. It

travels directly along the ground and termin-

ates on the earth's surface. Since the earth is

a relatively poor conductor, the surface wave

is attenuated quite rapidly. The surface wave

is attenuated less rapidly as it passes over

sea water, and the attenuation decreases for

a specific distance as the frequency is de-

creased. The rate of attenuation with distance

becomes so large as the frequency is increased

above about 3 Mc. that the surface wave be-

comes of little value for communication.

The Space Wave The resultant wave or space

wave is illustrated in figure

13 by the combination of (B) and (C). It is this

wave path, which consists of the combination

of the direct wave and the ground -reflected

wave at the receiving antenna, which is the

normal path of signal propagation for line -of-

sight or near line -of -sight communication or

FM and TV reception on frequencies above

about 40 Mc.

Below line -of -sight over plane earth or

water, when the signal source is effectively

at the horizon, the ground -reflected wave does

not exist, so that the direct wave is the only

component which goes to make up the space

wave. But when both the signal source and the

receiving antenna are elevated with respect to

the intervening terrain, the ground -reflected

wave is present and adds vectorially to the

direct wave at the receiving antenna. The vec-

torial addition of the two waves, which travel

over different path lengths (since one of the

waves has been reflected from the ground) re-

sults in an interference pattern. The interfer-

ence between the two waves brings about a

cyclic variation in signal strength as the re-

ceiving antenna is raised above the ground.

This effect is illustrated in figure 14. From

this figure it can be seen that best space -

wave reception of a v -h -f signal often will be

obtained with the receiving antenna quite close

to the ground. This subject, along with other

aspects of v -h -f signal propagation and recep-

tion, are discussed in considerable detail in

a book on fringe -area TV reception.

The distance from an elevated point to the

geometrical horizon is gitiren by the approxi-

mate equation: d = 1.221 where'the distance

"Better TV Reception," by W. W. Smith and R. L. Dow -

ley, published by Editors and Engineers, Ltd., Summer -

land, Calif.

www.americanradiohistory.com

HANDBOOK Antenna Bandwidth 409

Figure 11

COMPARATIVE VERTICAL

RADIATION PATTERNS

Showing the vertical radiation

patterns of a horizontal single -

section flat -top beam (A), an

array of two stacked horizontal

in -phase half -wave elements -

half of a "Lazy H "-(8), and

a horizontal dipole (C). In each

case the top of the antenna sys-

tem is 0.75 wavelength above

ground, as shown to the left of

the curves.

angle radiation at the expense of the useless

high -angle radiation with these simple arrays

as contrasted to the dipole is quite marked.

Figure 12 compares the patterns of a 3 ele-

ment beam and a dipole radiator at a height of

0.75 wavelength. It will be noticed that al-

though there is more energy in the lobe of the

beam as compared to the dipole, the axis of

the beam is at the same angle above the hori-

zontal. Thus, although more radiated energy

is provided by the beam at low angles, the

average angle of radiation of the beam is no

lower than the average angle of radiation of

the dipole.

21 -5 Bandwidth

The bandwidth of an antenna or an antenna

array is a function primarily of the radiation

resistance and of the shape of the conductors

which make up the antenna system. For arrays

of essentially similar construction the band-

width (or the deviation in frequency which the

system can handle without mismatch) is in-

creased with increasing radiation resistance,

and the bandwidth is increased with the use

of conductors of larger diameter (smaller ratio

of length to diameter). This is to say that if

an array of any type is constructed of large

diameter tubing or spaced wires, its bandwidth

will be greater than that of a similar array

constructed of single wires.

The radiation resistance of antenna arrays

of the types mentioned in the previous para-

graphs may be increased through the use of

wider spacing between elements. With increased

radiation resistance in such arrays the radi-

ation efficiency increases since the ohmic

losses within the conductors become a smaller

percentage of the radiation resistance, and the

bandwidth is increased proportionately.

21 -6 Propagation of

Radio Waves

The preceding sections have discussed the

manner in which an electromagnetic -wave or

radio -wave field may be set up by a radiating

system. However, for this field to be useful

for communication it must be propagated to

some distant point where it may be received,

or where it may be reflected so that it may be

received at some other point. Radio waves

may be propagated to a remote point by either

or both of two general methods. Propagation

A- DIPOLE \

B-3- ELEMENT

PARASITIC

0 1.5 2.0 2.5 3.0 3.3

GAIN IN FIELD STRENGTH

Figure 12

VERTICAL RADIATION PATTERNS

Showing vertical radiation patterns of a hori-

zontal dipole (A) and a horizontal 3- element

parasitic array (8) at a height above ground

of 0.75 wavelength. Note that the axis of the

main radiation lobes are at the some angle

above the horizontal. Note also the suppres-

sion of high angle radiation by the parasitic

array.

www.americanradiohistory.com

408 Radiation, Propagation and Lines THE RADIO

POWER OUTPUT

Figure 9

VERTICAL RADIATION

PATTERNS

Showing the vertical radiation

patterns for half -wave antennas

(or colinear half -wave or ex-

tended half -wave antennas) at

different heights above average

ground and perfect ground. Note

that such antennas one -quarter

wave above ground concentrate

most radiation at the very high

angles which are useful for com-

munication only on the lower fre-

quency bands. Antennas one -half

wave above ground are not

shown, but the elevation pattern

shows one lobe on each side at

an angle of 30. above hori-

zontal.

dipole could be increased by raising the an-

tenna higher above the ground. This is true to

an extent in the case of the horizontal dipole;

the low -angle radiation does increase slowly

after a height of 0.6 wavelength is reached

but at the expense of greatly increased high -

angle radiation and the formation of a number

of nulls in the elevation pattern. No signal

can be transmitted or received at the elevation

angles where these nulls have been formed.

Tests have shown that a center height of 0.6

wavelength for a vertical dipole (0.35 wave-

length to the bottom end) is about optimum for

this type of array.

Figure 9 shows the effect of placing a hori-

zontal dipole at various heights above ground.

It is easily seen by reference to figure 9 (and

figure 10 which shows the radiation from a di-

pole at ja wave height) that a large percentage

of the total radiation from the dipole is being

radiated at relatively high angles which are

useless for communication on the 14 -Mc. and

28 -Mc. bands. Thus we see that in order to ob-

tain a worthwhile increase in the ratio of low -

angle radiation to high -angle radiation it is

necessary to place the antenna high above

ground, and in addition it is necessary to use

additional means for suppressing high -angle

radiation.

Suppression of High -angle radiation can be

High -angle suppressed, and this radiation

Radiation can be added to that going out

at low angles, only through the

use of some sort of directive antenna system.

There are three general types of antenna ar-

rays composed of dipole elements commonly

used which concentrate radiation at the lower

more effective angles for high -frequency com-

munication. These types are: (1) The close -

spaced out -of -phase system as exemplified by

the "flat -top" beam or a8JK array. Such con-

figurations are classified as end fire arrays.

(2) The wide - spaced in -phase arrays, as exem-

plified by the "Lazy H" antenna. These con-

figurations are classified as broadside arrays.

(3) The close- spaced parasitic systems, as

exemplified by the three element rotary beam.

A comparison between the radiation from a

dipole, a "flat -top beam" and a pair of dipoles

stacked one above the other (half of a "lazy

H "), in each case with the top of the antenna

at a height of Sa wavelength is shown in figure

11. The improvement in the amplitude of low-

.5 1.0 1.5 2.0 Q.0 3.0 .0 1.0 1.5 2.0 2.0 3.0

GAIN IN FIELD STRENGTH

Figure 10

VERTICAL RADIATION

PATTERNS

Showing vertical -plane radiation

patterns of a horizontal single -

section flat -top beam with one -

eighth wave spacing (solid

curves) and a horizontal half -

wav antenna (dashed curves)

when both are 0.5 wavelength

(A) and 0.75 wavelength (B) a-

bove ground.

www.americanradiohistory.com

HANDBOOK Antenna Directivity 407

.2

.4

.3

.2

.1

0 30 22 26 24 22 20 M N 14 12 IS 2 0

WAVE ANGLE IN DEGREES

Figure 8

VERTICALPLANE DIRECTIONAL CHAR-

ACTERISTICS OF HORIZONTAL AND VER-

TICAL DOUBLETS ELEVATED 0.6 WAVE-

LENGTH AND ABOVE TWO TYPES OF

GROUND

H, represents a horizontal doublet over typi-

cal farmland. H2 over salt water. VI is a

vertical pattern of radiation from o vertical

doublet over typical farmland, V2 over salt

water. A salt water ground is the closest

approach to an extensive ideally perfect

ground that will be met in actual practice.

great -circle path, or within 2 or 3 degrees of

that path under all normal propagation condi-

tions. However, under turbulent ionosphere

conditions, or when unusual propagation con-

ditions exist, the deviation from the great -circle

path for greatest signal intensity may be as

great as 90 °. Making the array rotatable over-

comes these difficulties, but arrays having ex-

tremely high horizontal directivity become too

cumbersome to be rotated, except perhaps

when designed for operation on frequencies

above 50 Mc.

Vertical Vertical directivity is of the great -

Directivity est importance in obtaining satis-

factory communication above 14

Mc. whether or not horizontal directivity is

used. This is true simply because only the

energy radiated between certain definite eleva-

tion angles is useful for communication. Ener-

gy radiated at other elevation angles is lost

and performs no useful function.

Optimum Angle The optimum angle of radiation

of Radiation for propagation of signals be-

tween two points is dependent

upon a number of variables. Among these sig-

nificant variables are: (1) height of the iono-

sphere layer which is providing the reflection,

(2) distance between the two stations, (3) num-

ber of hops for propagation between the two

stations. For communication on the 14 -Mc.

band it is often possible for different modes

of propagation to provide signals between two

points. This means, of course, that more than

one angle of radiation can be used. If no eleva-

tion directivity is being used under this con-

dition of propagation, selective fading will

take place because of interference between the

waves arriving over the different paths.

On the 28 -Mc. band it is by far the most com-

mon condition that only one mode of propagation

will be possible between two points at any

one time. This explains, of course, the reason

why rapid fading in general and selective fad-

ing in particular are almost absent from sig-

nals heard on the 28 -Mc. band (except for fad-

ing caused by local effects).

Measurements have shown that the angles

useful for communication on the 14 -Mc. band

are from 3° to about 30 °; angles above about

15° being useful only for local work. On the

28 -Mc. band measurements have shown that

the useful angles range from about 3° to 18 °;

angles above about 12° being useful only for

local (less than 3000 miles) work. These fig-

ures assume normal propagation by virtue of

the 1:2 layer.

Angle of Radiation It now becomes of inter -

of Typical Antennos est to determine the s-

and Arrays mount of radiation avail-

able at these useful low-

er angles of radiation from commonly used an-

tennas and antenna arrays. Figure 8 shows

relative output voltage plotted against eleva-

tion angle (wave angle) in degrees above the

horizontal, for horizontal and vertical doublets

elevated 0.6 wavelength above two types of

ground. It is obvious by inspection of the

curves that a horizontal dipole mounted at this

height above ground (20 feet on the 28 -Mc.

band) is radiating only a small amount of ener-

gy at angles useful for communication on the

28 -Mc. band. Most of the energy is being radi-

ated uselessly upward. The vertical antenna

above a good reflecting surface appears much

better in this respect -and this fact has been

proven many times by actual installations.

It might immediately be thought that the a-

mount of radiation from a horizontal or vertical

www.americanradiohistory.com

406 Radiation, Propagation and Lines THE RADIO

is resistance of the wire, ground resistance

(in the case of a Marconi), corona discharge,

and insulator losses.

The approximate effective radiation effi-

ciency (expressed as a decimal) is equal to:

Nr = Ra /(Ra+ RL) where R. is equal to the

radiation resistance and RL is equal to the

effective loss resistance of the antenna. The

loss resistance will be of the order of 0.25

ohm for large- diameter tubing conductors such

as are most commonly used in multi- element

parasitic arrays, and will be of the order of

0.5 to 2.0 ohms for arrays of normal construc-

tion using copper wire.

When the radiation resistance of an antenna

or array is very low, the current at a voltage

node will be quite high for a given power. Like-

wise, the voltage at a current node will be very

high. Even with a heavy conductor and excel-

lent insulation, the losses due to the high volt-

age and current will be appreciable if the radi-

ation resistance is sufficiently low.

Usually, it is not considered desirable to

use an antenna or array with a radiation resist-

ance of less than approximately 5 ohms unless

there is sufficient directivity, compactness,

or other advantage to offset the losses result-

ing from the low radiation resistance.

Ground The radiation resistance of a Mar-

Resistance coni antenna, especially, should

be kept as high as possible. This

will reduce the antenna current for a given

power, thus minimizing loss resulting from the

series resistance offered by the earth connec-

tion. The radiation resistance can be kept high

by making the Marconi radiator somewhat longer

than a quarter wave, and shortening it by series

capacitance to an electrical quarter wave. This

reduces the current flowing in the earth con-

nection. It also should be removed from ground

as much as possible (vertical being ideal).

Methods of minimizing the resistance of the

earth connection will be found in the discus-

sion of the Marconi antenna.

21 -4 Antenna Directivity

All practical antennas radiate better in some

directions than others. This characteristic is

called directivity. The more directive an an-

tenna is, the more it concentrates the radiation

in a certain direction, or directions. The more

the radiation is concentrated in a certain direc-

tion, the greater will be the field strength pro-

duced in that direction for a given amount of

total radiated power. Thus the use of a direc-

tional antenna or array produces the same re-

sult in the favored direction as an increase in

the power of the transmitter.

The increase in radiated power in a certain

direction with respect to an antenna in free

space as a result of inherent directivity is

called the free space directivity power gain

or just space directivity gain of the antenna

(referred to a hypothetical isotropic radiator

which is assumed to radiate equally well in

all directions). Because the fictitious isotropic

radiator is a purely academic antenna, not phy-

sically realizable, it is common practice to use

as a reference antenna the simplest unground-

ed resonant radiator, the half -wave Hertz, or

resonant doublet. As a half -wave doublet has

a space directivity gain of 2.15 db over an iso-

tropic radiator, the use of a resonant dipole

as the comparison antenna reduces the gain

figure of an array by 2.15 db. However, it should

be understood that power gain can be expressed

with regard to any antenna, just so long as it

is specified.

As a matter of interest, the directivity of

an infinitesimal dipole provides a free space

directivity power gain of 1.5 (or 1.76 db) over

an isotropic radiator. This means that in the

direction of maximum radiation the infinitesi-

mal dipole will produce the same field of

strength as an isotropic radiator which is radi-

ating 1.5 times as much total power.

A half -wave resonant doublet, because of

its different current distribution and signifi-

cant length, exhibits slightly more free space

power gain as a result of directivity than does

the infinitesimal dipole, for reasons which will

be explained in a later section. The space

directivity power gain of a half -wave resonant

doublet is 1.63 (or 2.15 db) referred to an iso-

tropic radiator.

Horizontal When choosing and orienting an

Directivity antenna system, the radiation pat-

terns of the various common types

of antennas should be given careful considera-

tion. The directional characteristics are of

still greater importance when a directive an-

tenna array is used.

Horizontal directivity is always desirable

on any frequency for point -to -point work. How-

ever, it is not always attainable with reason-

able antenna dimensions on the lower fre-

quencies. Further, when it is attainable, as

on the frequencies above perhaps 7 Mc., with

reasonable antenna dimensions, operating con-

venience is greatly furthered if the maximum

lobe of the horizontal directivity is control-

lable. It is for this reason that rotatable an-

tenna arrays have come into such common

usage.

Considerable horizontal directivity can be

used to advantage when: (1) only point -to-

point work is necessary, (2) several arrays are

available so that directivity may be changed

by selecting or reversing antennas, (3) a single

rotatable array is in use. Signals follow the

www.americanradiohistory.com

HANDBOOK Antenna Impedance 405

HEIGHT IN WAVELENGTHS OF CENTER OF VERTICAL

HALF -WAVE ANTENNA ABOVE PERFECT GROUND

25 .3 .4 .5 .5 .7 .75

V 571 -MON OH7AL

w-_

0 .1 .2 .3 .4 .S 41 .7 . .5 to

HEIGHT IN WAVELENGTHS OF HORIZONTAL HALF -

WAVE ANTENNA ABOVE PERFECT GROUND

Figure 7

EFFECT OF HEIGHT ON THE RADIATION

RESISTANCE OF A DIPOLE SUSPENDED

ABOVE PERFECT GROUND

the radiation resistance to approximately 100

ohms. When a horizontal half -wave antenna is

used, the radiation resistance (and, of course,

the amount of energy radiated for a given an-

tenna current) depends on the height of the

antenna above ground, since the height deter-

mines the phase and amplitude of the wave

reflected from the ground back to the antenna.

Thus the resultant current in the antenna for

a given power is a function of antenna height.

Center -fed When a linear radiator is series fed

Feed Point at the center, the resistive and

Impedance reactive components of the driving

point impedance are dependent up-

on both the length and diameter of the radiator

in wavelengths. The manner in which the resis-

tive component varies with the physical dimen-

sions of the radiator is illustrated in figure 5.

The manner in which the reactive component

varies is illustrated in figure 6.

Several interesting things will be noted with

respect to these curves. The reactive com-

ponent disappears when the overall physical

length is slightly less than any number of half

waves long, the differential increasing with

conductor diameter. For overall lengths in the

vicinity of an odd number of half wavelengths,

the center feed point looks to the generator or

transmission line like a series -resonant lumped

circuit, while for overall lengths in the vici-

nity of an even number of half wavelengths, it

looks like a parallel- resonant or anti- resonant

lumped circuit. Both the feed point resistance

and the feed point reactance change more slow-

ly with overall radiator length (or with fre-

quency with a fixed length) as the conductor

diameter is increased, indicating that the ef-

fective "Q" is lowered as the diameter is in-

creased. However, in view of the fact that the

damping resistance is nearly all "radiation

resistance" rather than loss resistance, the

lower Q does not represent lower efficiency.

Therefore, the lower Q is desirable, because

it permits use of the radiator over a wider fre-

quency range without resorting to means for

eliminating the reactive component. Thus, the

use of a large diameter conductor makes the

overall system less frequency sensitive. If the

diameter is made sufficiently large in terms of

wavelengths, the Q will be low enough to qual-

ify the radiator as a "broad- band" antenna.

The curves of figure 7 indicate the theoreti-

cal center -point radiation resistance of a half -

wave antenna for various heights above perfect

ground. These values are of importance in

matching untuned radio -frequency feeders to

the antenna, in order to obtain a good imped-

ance match and an absence of standing waves

on the feeders.

Ground Losses Above average ground, the

actual radiation resistance

of a dipole will vary from the exact value of

figure 7 since the latter assumes a hypothet-

ical, perfect ground having no loss and perfect

reflection. Fortunately, the curves for the radi-

ation resistance over most types of earth will

correspond rather closely with those of the

chart, except that the radiation resistance for

a horizontal dipole does not fall off as rapidly

as is indicated for heights below an eighth

wavelength. However, with the antenna so

close to the ground and the soil in a strong

field, much of the radiation resistance is ac-

tually represented by ground loss; this means

that a good portion of the antenna power is

being dissipated in the earth, which, unlike

the hypothetical perfect ground, has resistance.

In this case, an appreciable portion of the

radiation resistance actually is loss resist-

ance. The type of soil also has an effect upon

the radiation pattern, especially in the vertical

plane, as will be seen later.

The radiation resistance of an antenna gen-

erally increases with length, although this in-

crease varies up and down about a constantly

increasing average. The peaks and dips are

caused by the reactance of the antenna, when

its length does not allow it to resonate at the

operating frequency.

Antenna Antennas have a certain loss re-

Efficiency sistance as well as a radiation re-

sistance. The loss resistance de-

fines the power lost in the antenna due to ohm-

www.americanradiohistory.com

404 Radiation, Propagation and Lines THE RADIO

10000

9000

8000

7000

6000

5000.

4000

3000

2000

1000

0 O.15Á 05A 1.05 1.5A 2.05

OVERALL LENGTH OF RADIATOR

DIAMETER= rka-

i l

DIAMETER-

=r' 2.55

Figure 5

FEED POINT RESISTANCE OF A CENTER

DRIVEN RADIATOR AS A FUNCTION OF

PHYSICAL LENGTH IN TERMS OF FREE

SPACE WAVELENGTH

When the antenna is resonant, and it always

should be for best results, the impedance at

the center is substantially resistive, and is

termed the radiation resistance. Radiation re-

sistance is a fictitious term; it is that value

of resistance (referred to the current loop)

which would dissipate the same amount of

power as being radiated by the antenna, when

fed with the current flowing at the current loop.

The radiation resistance depends on the

antenna length and its proximity to nearby

objects which either absorb or re- radiate pow-

er, such as the ground, other wires, etc.

The Marconi Before going too far with the

Antenna discussion of radiation resist-

ance, an explanation of the Mar-

coni (grounded quarter wave) antenna is in

order. The Marconi antenna is a special type

of Hertz antenna in which the earth acts as the

"other half" of the dipole. In other words, the

current flows into the earth instead of into a

similar quarter -wave section. Thus, the current

loop of a Marconi antenna is at the base rather

than in the center. In either case it is a quarter

wavelength from the end.

A half -wave dipole far from ground and other

reflecting objects has a radiation resistance

at the center of about 73 ohms. A Marconi an-

+9000

+5000

+4000

+3000

+2000

+ 1000

000,

2000

3000

4000

5000

0 155 055 I OA 1.25 2.0

OVERALL LENGTH OF RADIATOR

2.55

Figure 6

REACTIVE COMPONENT OF THE FEED

POINT IMPEDANCE OF A CENTER

DRIVEN RADIATOR AS A FUNCTION OF

PHYSICAL LENGTH IN TERMS OF FREE

SPACE WAVELENGTH

tenna is simply one -half of a dipole. For that

reason, the radiation resistance is roughly

half the 73-ohm impedance of the dipole or

36.5 ohms. The radiation resistance of a Mar-

coni antenna such as a mobile whip will be

lowered by the proximity of the automobile

body.

Antenna Because the power throughout the

Impedance antenna is the same, the imped-

ance of a resonant antenna at any

point along its length merely expresses the

ratio between voltage and current at that point.

Thus, the lowest impedance occurs where the

current is highest, namely, at the center of a

dipole, or a quarter wave from the end of a

Marconi. The impedance rises uniformly toward

each end, where it is about 2000 ohms for a

dipole remote from ground, and about twice as

high for a vertical Marconi.

If a vertical half -wave antenna is set up so

that its lower end is at the ground level, the

effect of the ground reflection is to increase

www.americanradiohistory.com

HANDBOOK Radiation Resistance 403

A harmonic operated antenna is somewhat

longer than the corresponding integral number

of dipoles, and for this reason, the dipole

length formula cannot be used simply by mul-

tiplying by the corresponding harmonic. The

intermediate half wave sections do not have

end effects. Also, the current distribution is

disturbed by the fact that power can reach

some of the half wave sections only by flowing

through other sections, the latter then acting

not only as radiators, but also as transmission

lines. For the latter reason, the resonant length

will be dependent to an extent upon the method

of feed, as there will be less attenuation of

the current along the antenna if it is fed at or

near the center than if fed towards or at one

end. Thus, the antenna would have to be some-

what longer if fed near one end than if fed near

the center. The difference would be small,

however, unless the antenna were many wave-

lengths long.

The length of a center fed harmonically oper-

ated doublet may be found from the formula:

L (K -.05) x 492

Freq. in Mc.

where K = number of i waves on

antenna

L = length in feet

Under conditions of severe current attenua-

tion, it is possible for some of the nodes, or

loops, actually to be slightly greater than a

physical half wavelength apart. Practice has

shown that the most practical method of reson-

ating a harmonically operated antenna accu-

rately is by cut and try, or by using a feed

system in which both the feed line and antenna

are resonated at the station end as an integral

system.

A dipole or half -wave antenna is said to

operate on its fundamental or first harmonic.

A full wave antenna, 1 wavelength long, oper-

ates on its second harmonic. An antenna with

five half- wavelengths on it would be operating

on its fifth harmonic. Observe that the fifth

harmonic antenna is 2tfs wavelengths long, not

5 wavelengths.

Antenna Most types of antennas operate

Resonance most efficiently when tuned or

resonated to the frequency of

operation. This consideration of course does

not apply to the rhombic antenna and to the

parasitic elements of arrays employing para-

sitically excited elements. However, in practi-

cally every other case it will be found that in-

creased efficiency results when the entire an-

tenna system is resonant, whether it be a sim-

ple dipole or an elaborate array. The radiation

efficiency of a resonant wire is many times

that of a wire which is not resonant.

000

Di

o.sa rv=.

Figure 4

EFFECT OF SERIES INDUCTANCE AND

CAPACITANCE ON THE LENGTH OF A

HALF -WAVE RADIATOR

The top antenna has been electrically length-

ened by placing o coil in series with the cen-

ter. In other words, an antenna with a lumped

inductance in its center can be mode shorter

for a given frequency than a plain wire radia-

tor. The bottom antenna has been capacitive-

ly shortened electrically. In other words, on

antenna with o capacitor in series with it

must be mode longer for o given frequency

since its effective electrical length os com-

pared to plain wire is shorter.

If an antenna is slightly too long, it can be

resonated by series insertion of a variable

capacitor at a high current point. If it is slight-

ly too short, it can be resonated by means of

a variable inductance. These two methods,

illustrated schematically in figure 4, are gen-

erally employed when part of the antenna is

brought into the operating room.

With an antenna array, or an antenna fed by

means of a transmission line, it is more com-

mon to cut the elements to exact resonant

length by "cut and try" procedure. Exact an-

tenna resonance is more important when the

antenna system has low radiation resistance;

an antenna with low radiation resistance has

higher Q (tunes sharper) than an antenna with

high radiation resistance. The higher Q does

not indicate greater efficiency; it simply indi-

cates a sharper resonance curve.

21 -3 Radiation Resistance

and Feed -Point Impedance

In many ways, a half -wave antenna is like a

tuned tank circuit. The main difference lies in

the fact that the elements of inductance, capac-

itance, and resistance are lumped in the tank

circuit, and are distributed throughout the

length of an antenna. The center of a half -wave

radiator is effectively at ground potential as

far as r -f voltage is concerned, although the

current is highest at that point.

www.americanradiohistory.com

402 Radiation, Propagation and Lines THE RADIO

distance in meters between adjacent peaks

or adjacent troughs of a wave train.

As a radio wave travels 300,000,000 meters

a second (speed of light), a frequency of 1

cycle per second corresponds to a wavelength

of 300,000,000 meters. So, if the frequency is

multiplied by a million, the wavelength must

be divided by a million, in order to maintain

their correct ratio.

A frequency of 1,000,000 cycles per second

(1,000 kc.) equals a wavelength of 300 meters.

Multiplying frequency by 10 and dividing wave-

length by 10, we find: a frequency of 10,000 kc.

equals a wavelength of 30 meters. Multiplying

and dividing by 10 again, we get: a frequency

of 100,000 kc. equals 3 meters wavelength.

Therefore, to change wavelength to frequency

(in kilocycles), simply divide 300,000 by the

wavelength in meters (À).

300,000

Fkc = À

300,000

À - Fkc

Now that we have a simple conversion for-

mula for converting wavelength to frequency

and vice versa, we can combine it with our

wavelength versus antenna length formula, and

we have the following:

Length of a half -wave radiator made from

wire (no. 14 to no. 10):

3.5 -11c. to 30 -Mc. bands

Length in feet = 468

Freq. in Mc.

50 -Mc. band

Length in feet -

Length in inches -

460

Freq. in Mc.

5600

Freq. in Mc.

144 -Mc. band

Length in inches = 5500

Freq. in Mc.

Length -to- Diameter When a half -wave radiator

Ratio is constructed from tubing

or rod whose diameter is

an appreciable fraction of the length of the

radiator, the resonant length of a half -wave

antenna will be shortened. The amount of

40 .40 130 100 200 300 X00 400 400,000

RATIO OF LENGTH TO DIAMETER

:000

Figure 3

CHART SHOWING SHORTENING OF A

RESONANT ELEMENT IN TERMS OF

RATIO OF LENGTH TO DIAMETER

The use of this chart is based on the basic

formula where radiator length in feet is

equal to 468 /frequency in Mc. This formula

applies to frequencies below perhaps 30 Mc.

when the radiator is made from wire. On

higher frequencies, or on 14 and 28 Mc. when

the radiator is made of large- diameter tubing,

the radiator is shortened from the value ob-

tained with the above formula by on amount

determined by the ratio of length to diameter

of the radiator. The amount of this shorten-

ing is obtainable from the chart shown above.

.00

shortening can be determined with the aid of

the chart of figure 3. In this chart the amount

of additional shortening over the values given

in the previous paragraph is plotted against

the ratio of the length to the diameter of the

half -wave radiator.

The length of a wave in free space is some-

what longer than the length of an antenna for

we same frequency. The actual free -space

half- wavelength is given by the following

expressions:

492

Half- wavelength = in feet

Freq. in Mc.

5905

Half- wavelength w in inches

Freq. in Mc.

Harmonic A wire in space can resonate at

Resonance more than one frequency. The low-

est frequency at which it resonates

is called its fundamental frequency, and at

that frequency it is approximately a half wave-

length long. A wire can have two, three, four,

five, or more standing waves on it, and thus

it resonates at approximately the integral har-

monics of its fundamental frequency. However,

the higher harmonics are not exactly integral

multiples of the lowest resonant frequency as

a result of end effects.

www.americanradiohistory.com

HANDBOOK Antenna Characteristics 401

Figure 2

ANTENNA POLARIZATION

The polarization (electric field) of

the radiation from a resonant dipole

such as shown at (A) above is paral-

lel to the length of the radiator. In

the case of o resonant slot cut in a

sheet of metal and used as a radia-

tor, the polarization (of the elec-

tric field) is perpendicular to the

length of the slot. In both cases,

however, the polarization of the

radiated field is parallel to the po-

tential gradient of the radiator; in

the case of the dipole the electric

lines of force are from end to end,

while in the case of the slot the

field is across the sides of the

slot. The metallic sheet containing

the slot may be formed into a cyl-

inder to make up the radiator shown

at (C). With this type of radiator

the radiated field will be horizon-

tally polarized even though the

radiator is mounted vertically.

ELECTRIC

FIELD O

(POLARIZATION)

VERTICAL

ELECTRIC FIELD

(POLARIZATION)

.or HORIZONTAL

FEEDERS CONNECT

TO POINTS Aas

NSIDC CYLINDER

is a graph showing the relative radiated field

intensity against azimuth angle for horizontal

directivity and field intensity against elevation

angle for vertical directivity.

The bandwidth of an antenna is a measure

of its ability to operate within specified limits

over a range of frequencies. Bandwidth can

be expressed either "operating frequency plus -

or -minus a specified per cent of operating fre-

quency" or "operating frequency plus -or -minus

a specified number of megacycles" for a cer-

tain standing- wave -ratio limit on the trans-

mission line feeding the antenna system.

The effective power gain or directive gain

of an antenna is the ratio between the power

required in the specified antenna and the power

required in a reference antenna (usually a half -

wave dipole) to attain the same field strength

in the favored direction of the antenna under

measurement. Directive gain may be expressed

either as an actual power ratio, or as is more

common, the power ratio may be expressed

in decibels.

Physical Length If the cross section of the

of a Half -Wave conductor which makes up

Antenna the antenna is kept very

small with respect to the

antenna length, an electrical half wave is a

fixed percentage shorter than a physical half -

wavelength. This percentage is approximately

5 per cent. Therefore, most linear half -wave an-

tennas are close to 95 per cent of a half wave-

length long physically. Thus, a half -wave an-

tenna resonant at exactly 80 meters would be

one -half of 0.95 times 80 meters in length. An-

other way of saying the same thing is that a

wire resonates at a wavelength of about 2.1

times its length in meters. If the diameter of

the conductor begins to be an appreciable frac-

tion of a wavelength, as when tubing is used

as a v -h -f radiator, the factor becomes slightly

less than 0.95. For the use of wire and not

tubing on frequencies below 30 Mc., however,

the figure of 0.95 may be taken as accurate.

This assumes a radiator removed from sur-

rounding objects, and with no bends.

Simple conversion into feet can be obtained

by using the factor 1.56. To find the physical

length of a half -wave 80 -meter antenna, we

multiply 80 times 1.56, and get 124.8 feet for

the length of the radiator.

It is more common to use frequency than

wavelength when indicating a specific spot in

the radio spectrum. For this reason, the rela-

tionship between wavelength and frequency

must be kept in mind. As the velocity of radio

waves through space is constant at the speed

of light, it will be seen that the more waves

that pass a point per second(higher frequency),

the closer together the peaks of those waves

must be (shorter wavelength). Therefore, the

higher the frequency, the lower will be the

wavelength.

A radio wave in space can be compared to

a wave in water. The wave, in either case, has

peaks and troughs. One peak and one trough

constitute a full wave, or one wavelength.

Frequency describes the number of wave

cycles or peaks passing a point per second.

Wavelength describes the distance the wave

travels through space during one cycle or

oscillation of the antenna current; it is the

www.americanradiohistory.com

400 Radiation, Propagation and Lines THE RADIO

` VOLTAGE

C[NTCII

. ,---t--_

me/m..1.5,-k\ ` f

.

' `{

1...-14ALW -WAVE ANTENNA i .

SHOWING NOW STANDING WAVES

CRUST ON A HORIZONTAL ANTENNA.

CURRENT IS MAXIMUM AT CENTRA.

VOLTAGE IS MAXIMUM AT C. VOLTAGE

Figure 1

STANDING WAVES ON A RESONANT

ANTENNA

transmission lines, both from single -wire lines

and from lines comprised of more than one

wire. In addition, radiation can be made to

take place in a very efficient manner from elec-

tromagnetic horns, from plastic lenses or from

electromagnetic lenses made up of spaced con-

ducting planes, from slots cut in a piece of

metal, from dielectric wires, or from the open

end of a wave guide.

Directivity of The radiation from any phys-

Radiation ically practicable radiating

system is directive to a certain

degree. The degree of directivity can be en-

hanced or altered when desirable through the

combination of radiating elements in a pre-

scribed manner, through the use of reflecting

planes or curved surfaces, or through the use

of such systems as mentioned in the preceding

paragraph. The construction of directive an-

tenna arrays is covered in detail in the chap-

ters which follow.

Polarization Like light waves, radio waves

can have a definite polarization.

In fact, while light waves ordinarily have to

be reflected or passed through a polarizing

medium before they have a definite polariza-

tion, a radio wave leaving a simple radiator

will have a definite polarization, the polar-

ization being indicated by the orientation of

the electric -field component of the wave. This,

in turn, is determined by the orientation of the

radiator itself, as the magnetic -field component

is always at right angles to a linear radiator,

and the electric -field component is always in

the same plane as the radiator. Thus we see

that an antenna that is vertical with respect

to the earth will transmit a vertically polar-

ized wave, as the electrostatic lines of force

will be vertical. Likewise, a simple horizontal

antenna will radiate horizontally polarized

waves.

Because the orientation of a simple linear

radiator is the same as the polarization of the

waves emitted by it, the radiator itself is re-

ferred to as being either vertically or horizon-

tally polarized. Thus, we say that a horizontal

antenna is horizontally polarized.

Figure 2A illustrates the fact that the polar-

ization of the electric field of the radiation

from a vertical dipole is vertical. Figure 2B,

on the other hand, shows that the polarization

of electric -field radiation from a vertical slot

radiator is horizontal. This fact has been uti-

lized in certain commercial FM antennas where

it is desired to have horizontally polarized

radiation but where it is more convenient to

use an array of vertically stacked slot arrays.

If the metallic sheet is bent into a cylinder

with the slot on one side, substantially omni-

directional horizontal coverage is obtained

with horizontally -polarized radiation when the

cylinder with the slot in one side is oriented

vertically. An arrangement of this type is shown

in figure 2C. Several such cylinders may be

stacked vertically to reduce high -angle radia-

tion and to concentrate the radiated energy

at the useful low radiation angles.

In any event the polarization of radiation

from a radiating system is parallel to the elec-

tric field as it is set up inside or in the vici-

nity of the radiating system.

21 -2 General Character-

istics of Antennas

All antennas have certain general character-

istics to be enumerated. It is the result of

differences in these general characteristics

which makes one type of antenna system most

suitable for one type of application and an-

other type best for a different application. Six

of the more important characteristics are: (1)

polarization, (2) radiation resistance, (3) hori-

zontal directivity, (4) vertical directivity,

(5) bandwidth, and (6) effective power gain.

The polarization of an antenna or radiating

system is the direction of the electric field

and has been defined in Section 21 -1.

The radiation resistance of an antenna sys-

tem is normally referred to the feed point in

an antenna fed at a current loop, or it is re-

ferred to a current loop in an antenna system

fed at another point. The radiation resistance

is that value of resistance which, if inserted

in series with the antenna at a current loop,

would dissipate the same energy as is actually

radiated by the antenna if the antenna current

at the feed point were to remain the same.

The horizontal and vertical directivity can

best be expressed as a directive pattern which

www.americanradiohistory.com

CHAPTER TWENTY -ONE

Radiation, Propagation

and Transmission Lines

Radio waves are electromagnetic waves

similar in nature but much lower in frequency

than light waves or heat waves. Such waves

represent electric energy traveling through

space. Radio waves travel in free space with

the velocity of light and can be reflected and

refracted much the same as light waves.

21 -1 Radiation from an

Antenna

Alternating current passing through a con-

ductor creates an alternating electromagnetic

field around that conductor. Energy is alter-

nately stored in the field, and then returned

to the conductor. As the frequency is raised,

more and more of the energy does not return

to the conductor, but instead is radiated off

into space in the form of electromagnetic

waves, called radio waves. Radiation from a

wire, or wires, is materially increased when-

ever there is a sudden change in the electrical

constants of the line. These sudden changes

produce reflection, which places standing

waves on the line.

When a wire in space is fed radio frequency

energy having a wavelength of approximately

2.1 times the length of the wire in meters, the

wire resonates as a half -wave dipole antenna

at that wavelength or frequency. The greatest

399

possible change in the electrical constants

of a line is that which occurs at the open end

of a wire. Therefore, a dipole has a great mis-

match at each end, producing a high degree of

reflection. We say that the ends of a dipole

are terminated in an infinite impedance.

A returning wave which has been reflected

meets the next incident wave, and the voltage

and current at any point along the antenna are

the vector sum of the two waves. At the ends

of the dipole, the voltages add, while the cur-

rents of the two waves cancel, thus producing

high voltage and low current at the ends of the

dipole or half wave section of wire. In the

same manner, it is found that the currents add

while the voltages cancel at the center of the

dipole. Thus, at the center there is high cur-

rent but low voltage.

Inspection of figure 1 will show that the

current in a dipole decreases sinusoidally

towards either end, while the voltage similarly

increases. The voltages at the two ends of the

antenna are 180° out of phase, which means

that the polarities are opposite, one being plus

while the other is minus at any instant. A

curve representing either the voltage or cur-

rent on a dipole represents a standing wave

on the wire.

Radiation from Radiation can and does take

Sources other place from sources other than

than Antennas antennas. Undesired radiation

can take place from open -wire

www.americanradiohistory.com

3 98

OSCILLATOR BUFFER i I BUFFER Sta

-120V

Figure 19

DIFFERENTIAL KEYING SYSTEM WITH

OSCILLATOR SWITCHING DIODE

+300 V.

OUTPUT

CONTROL

RI

° 100E

2w

Vi

OSCILLATOR V2

BUFFER V3

DRIVER

300 V.

V4 °

12AU7 REVER TUBE

6

22

REV

n 100 It

005

R2 R3

100 E 100E

VFO"MOLD -50V

4TE

C11`OS

100E 330E

Figure 20

DIFFERENTIAL KEYER EMPLOYED IN

"JOHNSON" TRANSMITTERS

005

conducting--and then continue operating This may be adjusted to cut off the VFO

until atter V2 and V3 have stopped con- between marks of keyed characters, thus

ducting. Potentiometer R1 adjusts the "hold" allowing rapid break -in operation.

time for VFO operation after the key is opened.

www.americanradiohistory.com

HANDBOOK Differential Keying 397

2

o

ñ

u

o

O Ó O

f VI

r

u U

ó

-Y-CUr-OFF VALUE

I AMPLIFIER

-- KEY iS DEPRESSED

DURING THIS TIME

6-TRANSMITTER IS YON THE

AIR- DURING THIS TIME

Figure 17

TIME SEQUENCE OF A

DIFFERENTIAL KEYER

CUT -OFF

\ VALUE

OSC.

on a moment before the rest of the stages are

energized, and remains on a moment longer

than the other stages. The "chirp" or frequen-

cy shift associated with abrupt switching of

the oscillator is thus removed from the emitted

signal. In addition, the differential keyer can

apply waveshaping to the amplifier section

of the transmitter, eliminating the "click"

caused by rapid keying of the latter stages.

The ideal keying system would perform as

illustrated in figure 17. When the key is

closed, the oscillator reaches maximum out-

put almost instantaneously. The following

stages reach maximum output in a fashion

determined by the waveshaping circuits of

the keyer. When the key is released, the out-

put of the amplifier stages starts to decay

in a predetermined manner, followed shortly

thereafter by cessation of the oscillator. The

overall result of these actions is to provide

relatively soft "make" and "break" to the

keyed signal, meanwhile preventing oscilla-

tor frequency shift during the keying se-

quence.

The rates of charge and decay in a typical

R -C keying circuit may be varied independent-

ly of each other by the blocking diode system

of figure 18. Each diode permits the charging

current of the timing capacitor to flow through

only one of the two variable potentiometers,

thus permitting independent adjustment of

the "make" and "break" characteristics of

the keying system.

A practical differential keying system de-

-250 V. 170 Pt

6AL5 TO CATHODE

BLOCKING DIODES CIRCUIT OF

KEYED STAGE

1

Figure 18

BLOCKING DIODES EMPLOYED

TO VARY TIME CONSTANT OF

"MAKE" AND "BREAK" CHARAC-

TERISTICS OF VACUUM TUBE

KEYER

VACUUM

TUBE

KEYER

(FIG. Po)

veloped by WIICP (Feb., 1956 QST) is shown

in figure 19. A 6AL5 switch tube turns the

oscillator on before the keying action starts,

and holds it on until after the keying se-

quence is completed. Time constant of the

keying cycle is determined by values of C

and R. When the key is open, a cut -off bias

of about -110 volts is applied to the screen

grid circuits of the keyed stages. When the

key is closed, the screen grid voltage rises

to the normal value at a rate determined by

the time constant R -C. Upon opening the key

again, the screen voltage returns to cut -off

value at the predetermined rate.

The potentiometer R1 serves as an output

control, varying the minimum internal re-

sistance of the 12BH7 keyer tube, and is a

useful device to limit power input during tune -

up periods. Excitation to the final amplifier

stage may be controlled by the screen po-

tentiometer R3 in the second buffer stage.

An external bias source of approximately -120

volts at 10 milliamperes is required for oper-

ation of the keyer, in addition to the 300 -volt

screen supply.

Blocking voltage may be removed from the

oscillator for "zeroing" purposes by closing

switch Si, rendering the diode switch in-

operative.

A second popular keying system is shown

in figure 20, and is widely used in many

Johnson transmitters. Grid block keying is

used on tubes V2 and V3. A waveshaping

filter consisting of R2, R3, and C1 is used

in the keying control circuit of V2 and V3.

To avoid chirp when the oscillator (V1) is

keyed, the keyer tube V4 allows the oscillator

to start quickly -- before V2 and V3 start

www.americanradiohistory.com

396 Transmitter Keying and Control THE RADIO

'wet

LOW POWER SUFFER

6AG7

KEYER UNIT

&LOCK /NG 64 /0 VOLTAGE

VOLTS t

TIME -

.025

470K,1 W

0--

6Ax5- GT

4

5 3

D+

AN

25K1

IOW

4.70,2W

10M.S0MA.

'n81

ce 450V

OUTPUT TO SCREEN CIS 807

+ r -

VOLTS O+-+

TIME

6116

5Y3

a v.

Io K

Io W

OAKS SKIS MTR.

12AÚ7

350-0-350

50 MA.

Figure 16

TWO -STAGE SCREEN GRID KEYER UNIT

+M.V.

POINT KEY VP KEY DOWN

A -35 340

B -Ito 0

C -no 0

D 375 375

E -275 -273

guished, removing the screen voltage from the

tetrode r -f tube. At the same time, rectified

grid bias is applied to the screen of the tetrode

through the I megohm resistor between screen

and key. This voltage effectively cuts off the

screen of the tetrode until the key is closed

again. The RC circuit in the grid of the 6L6

tube determines the keying characteristic of

the tetrode tube.

A more elaborate screen grid keyer is shown

in figures 15 and 16. This keyer is designed

to block -grid key the oscillator or a low pow-

ered buffer stage, and to screen key a medium

powered tetrode tube such as an 807, 2E26 or

6146. The unit described includes a simple

dual voltage power supply for the positive

screen voltage of the tetrode, and a negative

supply for the keyer stages. A 6K6 is used as

the screen keyer, and a 12AU7 is used as a

cathode follower and grid block keyer. As in

the W1DX keyer, this keyer turns on the ex-

citer a moment before the tetrode stage is

turned on. The tetrode stage goes off an in-

stant before the exciter does. Thus any key-

ing chirp of the oscillator is effectively re-

moved from the keyed signal.

By listening in the receiver one can hear

the exciter stop operating a fraction of a sec-

ond after the tetrode stage goes off. In fact,

during rapid keying, the exciter may be heard

as a steady signal in the receiver, as it has

appreciable time lag in the keying circuit. The

clipping effect of following stages has a defi-

nite hardening effect on this, however.

20 -8 Differential Keying Circuits

Excellent waveshaping may be obtained by

a differential keying system whereby the

master oscillator of the transmitter is turned

www.americanradiohistory.com

HANDBOOK Screen Grid Keying 395

EXC.

807. ETC

Figure 14

SINGLE -STAGE SCREEN GRID KEYER

FOR TETRODE TUBES

tetrode is keyed by this method, there is the

possibility of a considerable backwave caused

by r -f leakage through the grid -plate capacity

of the tube.

Certain hi -µ triode tubes, such as the 811 -A

and the 805, automatically block themselves

when the grid return circuit is opened. It is

merely necessary to insert a key and associ-

ated key click filter in the grid return lead of

these tubes. No blocking bias supply is need-

ed. This circuit is shown in figure 12.

A more elaborate blocked -grid keying sys-

tem has been developed by W1DX, and was

shown in the February, 1954 issue of QST

magazine. This highly recommended circuit

is shown in figure 13. Two stages are keyed,

Figure 15

TOP VIEW OF SCREEN

GRID KEYER SHOWN IN

FIGURE 16

preventing any backwave emission. The first

keyed stage may be the oscillator, or a low

powered buffer. The last keyed stage may be

the driver stage to the power amplifier, or the

amplifier itself. Since the circuit is so pro-

portioned that the lower powered stage comes

on /first and goes off last, any keying chirp in

the oscillator is not emitted on the air. Keying

lag is applied to the high powered keyed stage

only.

20.7 Screen Grid Keying

The screen circuit of a tetrode tube may be

keyed for c -w operation. Unfortunately, when

the screen grid of a tetrode tube is brought to

zero potential, the tube still delivers con-

siderable output. Thus it is necessary to place

a negative blocking voltage on the screen grid

to reduce the backwave through the tube. A

suitable keyer circuit that will achieve this

was developed by W6DTY, and was described

in the February, 1953 issue of CQ magazine.

This circuit is shown in figure 14. A 6L6 is

used as a combined clamper tube and keying

tube. When the key is closed, the 6L6 tube

has blocking bias applied to its control grid.

This bias is obtained from the rectified grid

bias of the keyed tube. Screen voltage is ap-

plied to the keyed stage through a screen drop-

ping resistor and a VR -105 regulator tube.

then the key is open, the 6L6 is no longer

cut -off, and conducts heavily. The voltage

drop across the dropping resistor caused by

the heavy plate current of the 6L6 lowers the

voltage on the VR -105 tube until it is extin-

I^N

www.americanradiohistory.com

394 Transmitter Keying and Control THE RADIO

-BLOCKING

B AS H.V

Figure 11

SIMPLE BLOCKED -GRID KEYING

SYSTEM

The blocking bias must be sufficient to cut-

off plate current to the amplifier stage in the

presence of the excitation voltage. R¡ is nor-

mal bias resistor for the tube. R2 and C1

should be adjusted for correct keying wave-

form.

HI -MU TRIODE

(61 I -A ETC.)

Figure 12

SELF -BLOCKING KEYING SYSTEM FOR

HIGH -MU TRIODE

R, and C1 adjusted for correct keying wave-

form. R, is bias resistor of tube.

recommended for general use, as considerable

voltage will be developed across the key when

it is open.

An electronic switch can take the place of

the hand key. This will remove the danger of

shock. At the same time, the opening and clos-

ing characteristics of the electronic switch

may easily be altered to suit the particular

need at hand. Such an electronic switch is

called a vacuum tube keyer. Low internal re-

sistance triode tubes such as the 45, 6A3, or

6AS7 are used in the keyer. These tubes act

as a very high resistance when sufficient

LOW POWER BUFFER

(5457 ETC.)

807, 6146, ETC.

ISO LUF

RFC

2 SUN

33K

2W

1001t

IW +M V.

VR-150

-+00 V.

6.3V. TO 6J5

REY

Figure 13

TWO -STAGE BLOCKED -GRID KEYER

A separate filament transformer must be used

for the 6J5, as its filament is at a potential

of -400 volts.

blocking bias is applied to them, and as a

very low resistance when the bias is removed.

The desired amount of lag or cushioning effect

can be obtained by employing suitable resist-

ance and capacitance values in the grid of the

keyer tube(s). Because very little spark is

produced at the key, due to the small amount

of power in the key circuit, sparking clicks

are easily suppressed.

One type 45 tube should be used for every

50 ma. of plate current. Type 6B4G or 2A3

tubes may also be used; allow one 6B4G tube

for every 80 ma. of plate current.

Because of the series resistance of the keyer

tubes, the plate voltage at the keyed tube will

be from 30 to 60 volts less than the power

supply voltage. This voltage appears as cath-

ode bias on the keyed tube, assuming the bias

return is made to ground, and should be taken

into consideration when providing bias.

Some typical cathode circuit vacuum tube

keying units are shown in figure 10.

20 -6 Grid Circuit Keying

Grid circuit, or blocked grid keying is an-

other effective method of keying a c -w trans-

mitter. A basic blocked grid keying circuit is

shown in figure 11. The time constant of the

keying is determined by the RC circuit, which

also forms part of the bias circuit of the tube.

When the key is closed, operating bias is de-

veloped by the flow of grid current through 121.

When the key is open, sufficient fixed bias is

applied to the tube to block it, preventing the

stage from functioning. If an un- neutralized

www.americanradiohistory.com

HANDBOOK Cathode Keying 393

SOMA SELE-

NIUM RECTIFIER

7. I W

TO

6Y6 CATHODE OF

IM (BREAK) I M(MAKE) KEYED STAGE

STANCOR PAB421

350.0 -350

SOMA

BO

G

2K ,2W 47014 IW IM 4704,0.1W

70 a°) RFC

4 2.SMH

I e00V

RF TUBE

45/2A3

I

45/2A3

KEY 1 "T IoODUF

Q_ooe

1 MI

TY705 t

M.Y. SUPPLY

Figure 10

VACUUM TUBE KEYERS FOR CENTER -TAP KEYING CIRCUITS

The type A keyer is suitable for keying stages running up to 1250 volts on the plate. Two 2A3

or 6A3 tubes can safely key 160 milliamperes of cathode current. The simple 6Y6 keyer in fig-

ure B Is for keying stages running up to 650 volts on the plate. A single 6Y6 can key 80 milli-

amperes. Two in parallel may be used for plate currents under 160 ma. If softer keying is de-

sired, the 500 -µofd. mica condenser should be increased to .001 pfd.

amplifier. If a low -level stage, which is fol-

lowed by a series of class C amplifiers, is

keyed, serious transients will be generated

in the output of the transmitter even though

the keyed stage is being turned on and off very

smoothly. This condition arises as a result of

pulse sharpening, which has been discussed

previously.

Third, the output from the stage should be

completely cut off when the key is up, and the

time constant of the rise and decay of the key-

ing wave should be easily controllable.

Fourth, it should be possible to make the

rise period and the decay period of the keying

wave approximately equal. This type of keying

envelope is the only one tolerable for commer-

cial work, and is equally desirable for obtain-

ing clean cut and easily readable signals in

amateur work.

Fifth, it is desirable that the keying circuit

be usable without a keying relay, even when

a high -power stage is being keyed.

Last, for the sake of simplicity and safety,

it should be possible to ground the frame of

the key, and yet the circuit should be such

that placing the fingers across the key will

not result in an electrical shock. In other words,

the keying circuit should be inherently safe.

All these requirements have been met in the

keying circuits to be described.

20-5 Cathode Keying

The lead from the cathode or center -tap con-

nection of the filament of an r -f amplifier can

be opened and closed for a keying circuit. Such

a keying system opens the plate voltage cir-

cuit and at the same time opens the grid bias

return lead. For this reason, the grid circuit

is blocked at the same time the plate circuit

is opened. This helps to reduce the backwave

that might otherwise leak through the keyed

stage.

The simplest cathode keying circuit is il-

lustrated in figure 9, where a key -click filter

is employed, and a hand key is used to break

the circuit. This simple keying circuit is not

www.americanradiohistory.com

392 Transmitter Keying and Control THE RADIO

a wide frequency band as sidebands and are

heard as clicks.

The cure for transient key clicks is rela-

tively simple, although one would not believe

it, judging from the hordes of clicky, "snappy"

signals heard on the air.

To be capable of transmitting code charac-

ters and at the same time not splitting the

eardrums of neighboring amateurs, the c -w

transmitter MUST meet two important speci-

fications.

1- It must have no parasitic oscillations

either in the stage being keyed or in any

succeeding stage.

2- It must have some device in the keying

circuit capable of shaping the leading

and trailing edge of the waveform.

Both these specifications must be met be-

/ore the transmitter is capable of c -w opera-

tion. Merely turning a transmitter on and off

by the haphazard insertion of a telegraph key

in some power lead is an invitation to trouble.

The two general methods of keying a trans-

mitter are those which control the excitation

to the keyed amplifier, and those which con-

trol the plate or screen voltage applied to the

keyed amplifier.

Key -Click Key -click elimination is accom-

Elimination plished by preventing a too -rapid

make- and -break of power to the

antenna circuit, rounding off the keying char-

acters so as to limit the sidebands to a value

which does not cause interference to adjacent

channels. Too much lag will prevent fast key-

ing, but fortunately key clicks can be prac-

tically eliminated without limiting the speed

of manual (hand) keying. Some circuits which

eliminate key clicks introduce too much time -

lag and thereby add tails to the dots. These

tails may cause the signals to be difficult to

copy at high speeds.

Location of Considerable thought should be

Keyed Stage given as to which stage in a

transmitter is the proper one to

key. If the transmitter is keyed in a stage close

to the oscillator, the change in r -f loading of

the oscillator will cause the oscillator to shift

frequency with keying. This will cause the

signal to have a distinct chirp. The chirp will

be multiplied as many times as the frequency

of the oscillator is multiplied. A chirpy oscil-

lator that would be passable on 80 meters

would be unusable on 28 Mc. c.w.

Keying the oscillator itself is an excellent

way to run into keying difficulties. If no key

click filter is used in the keying circuit, the

transmitter will have bad key clicks. If a key

click filter is used, the slow rise and decay

of oscillator voltage induced by the filter ac-

tion will cause a keying chirp. This action is

O IS

Figure 9

CENTER -TAP KEYING WITH CLICK

FILTER

The constants shown above are suggested as

starting values; considerable variation in

these values can be expected for optimum

keying of amplifiers of different operating

conditions. It is suggested that a keying re-

lay be substituted for the key in the circuit

above wherever practicable.

true of all oscillators, whether electron coupled

or crystal controlled.

The more amplifier or doubler stages that

follow the keyed stage, the more difficult it is

to hold control of the shape of the keyed wave-

form. A heavily excited doubler stage or class

C stage acts as a peak clipper, tending to

square up a rounded keying impulse, and the

cumulative effect of several such stages cas-

caded is sufficient to square up the keyed

waveform to the point where bad clicks are

reimposed on a clean signal.

A good rule of thumb is to never key back

farther than one stage removed from the final

amplifier stage, and never key closer than one

stage removed from the frequency controlling

oscillator of the transmitter. Thus there will

always be one isolating stage between the

keyed stage and the oscillator, and one isolat-

ing stage between the keyed stage and the

antenna. At this point the waveform of the

keyed signal may be most easily controlled.

Keyer Circuit In the first place it may be es-

Requirements tablished that the majority of

new design transmitters, and

many of those of older design as well, use a

medium power beam tetrode tube either as the

output stage or as the exciter for the output

stage of a high power transmitter. Thus the

transmitter usually will end up with a tube

such as type 2E26, 807, 6146, 813, 4 -65A,

4E27/257B, 4 -125A or similar, or one of these

tubes will be used as the stage just ahead of

the output stage.

Second, it may be established that it is un-

desirable to key further down in the transmitter

chain than the stage just ahead of the final

www.americanradiohistory.com

HANDBOOK Transmitter Keying 391

For 100 per cent protection, just obey the

following rule: never work on the transmitter

or reach inside any protective cover except

when the green pilots are glowing. To avoid

confusion, no other green pilots should be used

on the transmitter; if you want an indicator

jewel to show when the filaments are lighted,

use amber instead of green.

Safety Bleeders Filter capacitors of good qual-

ity hold their charge for some

time, and when the voltage is more than 1000

volts it is just about as dangerous to get across

an undischarged 4 -pfd. filter capacitor as it is

to get across a high -voltage supply that is

turned on. Most power supplies incorporate

bleeders to improve regulation, but as these

are generally wire -wound resistors, and as

wire -wound resistors occasionally open up

without apparent cause, it is desirable to in-

corporate an auxiliary safety bleeder across

each heavy -duty bleeder. Carbon resistors will

not stand much dissipation and sometimes

change in value slightly with age. However,

the chance of their opening up when run well

within their dissipation rating is very small.

To make sure that all capacitors are bled, it

is best to short each one with an insulated

screwdriver. However, this is sometimes awk-

ward and always inconvenient. One can be vir-

tually sure by connecting auxiliary carbon

bleeders across all wire -wound bleeders used

on supplies of 1000 volts or more. For every

500 volts, connect in series a 500,000 -ohm

1 -watt carbon resistor. The drain will be neg-

ligible (1 ma.) and each resistor will have to

dissipate only 0.5 watt. Under these condi-

tions the resistors will last indefinitely with

little chance of opening up. For a 1500 -volt

supply, connect three 500,000 -ohm resistors in

series. If the voltage exceeds an integral num-

ber of 500 volt divisions, assume it is the next

higher integral value; for instance, assume

1800 volts as 2000 volts and use four resistors.

Do not attempt to use fewer resistors by

using a higher value for the resistors; not over

500 volts should appear across any single

1 -watt resistor.

In the event that the regular bleeder opens

up, it will take several seconds for the auxil-

iary bleeder to drain the capacitors down to a

safe voltage, because of the very high resist-

ance. Therefore, i t is best to allow 10 or 15

seconds after turning off the plate supply be-

fore attempting to work on the transmitter.

If a 0 -1 d -c milliammeter is at hand, it may

be connected in series with the auxiliary

bleeder to act as a high voltage voltmeter.

"Hot" Adjustments Some amateurs contend

that it is almost impossible

to make certain adjustments, such as coupling

and neutralizing, unless the transmitter is run-

ning. The best.thing to do is to make all neu-

tralizing and coupling devices adjustable from

the front panel by means of flexible control

shafts which are broken with insulated cou-

plings to permit grounding of the panel bearing.

If your particular transmitter layout is such

that this is impracticable and you refuse to

throw the main switch to make an adjustment

-throw the main switch -take a reading -throw

the main switch -make an adjustment -and so

on, then protect yourself by making use of long

adjusting rods made from 1/-inch dowel sticks

which have been wiped with oil when perfectly

free from moisture.

If you are addicted to the use of pickup loop

and flashlight bulb as a resonance and neutral-

izing indicator, then fasten it to the end of a

long dowel stick and use it in that manner.

Protective Interlocks With the increasing ten-

dency toward construc-

tion of transmitters in enclosed steel cabinets

a transmitter becomes a particularly lethal de-

vice unless adequate safety provisions have

been incorporated. Even with a combined safety

signal and switch as shown in figure 8 it is

still conceivable that some person unfamiliar

with the transmitter could come in contact with

high voltage. It is therefore recommended that

the transmitter, wherever possible, be built

into a complete metal housing or cabinet and

that all doors or access covers be provided

with protective interlocks (all interlocks must

be connected in series) to remove the high

voltage whenever these doors or covers are

opened. The term "high voltage" should mean

any voltage above approximately 150 volts,

although it is still possible to obtain a serious

burn from a 150 -volt circuit under certain cir-

cumstances. The 150 -volt limit usually will

mean that grid -bias packs as well as high -

voltage packs should have their primary cir-

cuits opened when any interlock is opened.

20 -4 Transmitter Keying

The carrier from a c -w telegraph transmitter

must be broken into dots and dashes for the

transmission of code characters. The carrier

signal is of constant amplitude while the key

is closed, and is entirely removed when the

key is open. When code characters are being

transmitted, the carrier may be considered as

being modulated by the keying. If the change

from the no- output condition to full -output, or

vice versa, occurs coo rapidly, the rectangular

pulses which form the keying characters con-

tain high- frequency components which take up

www.americanradiohistory.com

390 Transmitter Keying and Control THE RADIO

sary chances. However, no one is infallible,

and chances of an accident are greatly less-

ened if certain factors are taken into consid-

eration in the design of a transmitter, in order

to protect the operator in the event of a lapse

of caution. If there are too many things one

must "watch out for" or keep in mind there is

a good chance that sooner or later there will

be a mishap; and it only takes one. When de-

signing or constructing a transmitter, the fol-

lowing safety considerations should be given

attention.

Grounds For the utmost in protection, every-

thing of metal on the front panel of

a transmitter capable of being touched by the

operator should be at ground potential. This

includes dial set screws, meter zero adjuster

screws, meter cases if of metal, meter jacks,

everything of metal protruding through the front

panel or capable of being touched or nearly

touched by the operator. This applies whether

or not the panel itself is of metal. Do not rely

upon the insulation of meter cases or tuning

knobs for protection.

The B negative or chassis of all plate power

supplies should be connected together, and to

an external ground such as a waterpipe.

Exposed Wires It is not necessary to resort

and Components to rack and panel construc-

tion in order to provide com-

plete enclosure of all components and wiring

of the transmitter. Even with metal- chassis

construction it is possible to arrange things so

as to incorporate a protective shielding hous-

ing which will not interfere with ventilation

yet will prevent contact with all wires and

components carrying high voltage d.c. or a.c.,

in addition to offering shielding action.

If everything on the front panel is at ground

potential (with respect to external ground) and

all units are effectively housed with protective

covers, then there is no danger except when

the operator must reach into the interior part

of the transmitter, as when changing coils,

neutralizing, adjusting coupling, or shooting

trouble. The latter procedure can be made safe

by making it possible for the operator to be

absolutely certain that all voltages have been

turned off and that they cannot be turned on

either by short circuit or accident. This can be

done by incorporation of the following system

of main primary switch and safety signal lights.

Combined Safety The common method of

Signal and Switch using red pilot lights to

show when a circuit is on

is useless except from an ornamental stand-

point. When the red pilot is not lit it usually

means that the circuit is turned off, but it can

6 3V TO GREEN PILOT LIGHTS ON

FRONT PANEL AND ON EACH CHASSIS

.Q FIL TRANS

MAIN 113 V. SUPPLY 0 -. -(D PDT SWITCH

11 IS V.A C TO ENTIRE TRANSMITTER

Figure 8

COMBINED MAIN SWITCH AND

SAFETY SIGNAL

When shutting down the transmitter, throw

the main switch to neutral. If work is to be

done on the transmitter, throw the switch all

the way to "pilot," thus turning on the green

pilot lights on the panel and on each chas-

sis, and insuring that no voltage can exist

on the primary of any transformer, even by

virtue of a short or accidental ground.

mean that the circuit is on but the lamp is

burned out or not making contact.

To enable you to touch the tank coils in

your transmitter with absolute assurance that

it is impossible for you to obtain a shock ex-

cept from possible undischarged filter conden-

sers (see following topic for elimination of

this hazard), it is only necessary to incorpo-

rate a device similar to that of figure 8. It is

placed near the point where the main 110 -volt

leads enter the room (preferably near the door)

and in such a position as to be inaccessible

to small children. Notice that this switch breaks

both leads; switches that open just one lead

do not afford complete protection, as it is

sometimes possible to complete a primary cir-

cuit through a short or accidental ground. Break-

ing just one side of the line may be all right

for turning the transmitter on and off, but when

you are going to place an arm inside the trans-

mitter, both 110 -volt leads should be broken.

When you are all through working your trans-

mitter for the time being, simply throw the

main switch to neutral.

When you find it necessary to work on the

transmitter or change coils, throw the switch

so that the green pilots light up. These can be

ordinary 6.3 -volt pilot lamps behind green

bezels or dipped in green lacquer. One should

be placed on the front panel of the transmitter;

others should be placed so as to be easily

visible when changing coils or making adjust-

ments requiring the operator to reach inside

the transmitter.

www.americanradiohistory.com

HANDBOOK Safety Precautions 389

113 VOLT SUPPLY FOR

ENTIRE TRANSMITTER

USES

SAFETY SWITCH

(SEE FIG.12)

AT OPERATING POSITION

STOP

L-t-~y-

TRANSMIT RECEIVE I

-CL ---121aCcH

THERMAL

TIME-DELAY

RELAY

,000, 1000, SW.,

ALL FILAMENT TRANSFORMERS

HIGH VOLT.

FILS. ON STANDBY (I13 V.)

INDICATOR LIGHTS

PROTECTIVE

INTERLOCKS

OVERLOAD'

CONTACTS

Lt ORECEIVER POWER

TRANSFORMER C.T.

(I TUNE-UP

SWITCH

IIII17 fllllll

EXCITER M.V.

TRANSFORMER HIGH VOLTAGE

TRANSFORMER

13 V. ANTENNA

CHANGEOVER

RELAY

Figure 7

PUSH -BUTTON TRANSMITTER -CONTROL CIRCUIT

Pushing the START button either at the transmitter or at the operating position will light all

filaments and start the time -delay r e I a y in its cycle. When the c y c l e has been completed, a

touch of the TRANSMIT button will put the transmitter on the air and disable the receiver. Push-

ing the RECEIVE button will disable the transmitter and restore the receiver. Pushing the STOP

button will instantly drop the entire transmitter from the a -c line. If desired, a switch may be

placed in series with the lead from the RECEIVE button to the protective interlocks; opening

the switch will make it impossible for any person accidentally to put the transmitter on the air.

Various other safety provisions, such as the protective- interlock arrangement described in the

text have been incorporated.

With the circuit arrangement shown for the overload -relay contacts, it is only necessary to use

a simple normally - closed d -c relay with a variable shunt across the coil of the relay. When the

current through the coil becomes great enough to open the normally-closed contacts the hold -

circuit on the plate -voltage relay will be broken and the plate voltage will be removed. If the

overload is only momentary, such as a modulation peak or a tank flashover, merely pushing the

TRANSMIT button will again put the transmitter on the air. This simple circuit provision elimi-

nates the requirement for expensive overload relays of the mechanically -latching type, but still

gives excellent overload protection.

button momentarily to light the transmitter fila-

ments and start the time -delay relay in its cy-

cle. When the standby light comes on it is only

necessary to touch the TRANSMIT button to

put the transmitter on the air and disable the

receiver. Touching the RECEIVE button will

turn off the transmitter and restore the receiver.

After a period of operation it is only necessary

to touch the STOP button at either the trans-

mitter or the operating position to shut down

the transmitter. This type of control arrange-

ment is called an electrically- locking push -to-

transmit control system. Such systems are fre-

quently used in industrial electronic control.

20 -3 Safety Precautions

The best way for an operator to avoid ser-

ious accidents from the high voltage supplies

of a transmitter is for him to use his head, act

only with deliberation, and not take unneces-

www.americanradiohistory.com

388 Transmitter Keying and Control THE RADIO

115 VOLT SUPPLY FOR

ENTIRE TRANSMITTER

FUSES

SAFETY SWITCH

(SEE FIGS )

\St HUSKY TOGGLE SWITCH

ON TRANSMITTER

THERMAL

TIME -DELAY

RELAY

,Qoo, ,000, -.000,

?-1

ALL r1 -AMENT TRANSFORMERS 3V.

HIGH VOLT.

FI LS. STANDS (115V.)

INDICATOR LIGHTS

PROTECTIVE

INTERLOCKS

O O

°RECEIVER POWER

TRANSFORMER C.T.

,Qoo)

TUNE -UP

SWITCH

EXCITER M.V.

TRANSFORMER

.000,

HIGH VOLTAGE

TRANSFORMER

TRANSMIT-

RECEIVE SWITCH

O

O11S V. ANTENNA

CHANGEOVER

O RELAY

Figure 6

TRANSMITTER CONTROL CIRCUIT

Closing S1 lights all filaments in the transmitter and starts the time -delay relay in its cycle.

When the time -delay relay has operated, closing the transmit -receive switch at the operating po-

sition will apply plate power to the transmitter and disable the receiver. A tune -up switch hos

been provided so that the exciter stages may be tuned without plate voltage on the final

amplifier.

mister on the air, has had the experience of

having to throw several switches and pull or

insert a few plugs when changing from receive

to transmit. This is one extreme in the direc-

tion of how not to control a transmitter. At the

other extreme we find systems where it is only

necessary to speak into the microphone or

touch the key to change both transmitter and

receiver over to the transmit condition. Most

amateur stations are intermediate between the

two extremes in the control provisions and use

some relatively simple system for transmitter

control.

In figure 5 is shown an arrangement which

protects mercury -vapor rectifiers against pre-

mature application of plate voltage without

resorting to a time -delay relay. No matter which

switch is thrown first, the filaments will be

turned on first and off last. However, double -

pole switches are required in place of the usual

single -pole switches.

When assured time delay of the proper inter-

val and greater operating convenience are de-

sired, a group of inexpensive a -c relays may

be incorporated into the circuit to give a con-

trol circuit such as is shown in figure 6. This

arrangement uses a 115 -volt thermal (or motor -

operated) time -delay relay and a d -p -d -t 115 -

volt control relay. Note that the protective

interlocks are connected in series with the

coil of the relay which applies high voltage to

the transmitter. A tune -up switch has been in-

cluded so that the transmitter may be tuned up

as far as the grid circuit of the final stage is

concerned before application of high voltage

to the final amplifier. Provisions for operat-

ing an antenna- changeover relay and for cut-

ting the plate voltage to the receiver when the

transmitter is operating have been included.

A circuit similar to that of figure 6 but in-

corporating push- button control of the trans-

mitter is shown in figure 7. The circuit features

a set of START -STOP and TRANSMIT -RE-

CEIVE buttons at the transmitter and a sepa-

rate set at the operating position. The control

push buttons operate independently so that

either set may be used to control the trans-

mitter. It is only necessary to push the START

www.americanradiohistory.com

HANDBOOK Transmitter Control 387

C

115V.A.0

LINE

TO EXCITER POWER SUPPLIES

TO N.V.

POWER SUPPLY

POWER CONTROL

RELAY

PLUG FOR

ABLE TO

VARIAC OR

ROWERSTAT

DUMMY PLUG FOR

STRAIGHT OPERATION

TO FILAMENT

TRANSFORMERS

EXTERNAL VARIAC

OR POWERSTAT

Figure 4

CIRCUIT WITH VARIABLE -RATIO

AUTO -TRANSFORMER

When the dummy plug is inserted into the re-

ceptacle on the equipment, closing of the

power control relay will apply full voltage

to the primaries. With the cable from the

Variac or Powerstat plugged into the socket

the voltage output of the high -voltage power

supply may be varied from zero to about IS

per cent above normal.

LINE

PLUGI

GREEN

PILOT

L 52

RED

PI LOT

J

',INTERLOCKS

Jt IN TRANSMITTER

TRANSMITTER

FILAMENT

TRANSFORMERS

115 V. TO EXCITER AND

HIGH.VOLTAGE RELAYS.

AND TO RECEIVER CON-

TROL AND ANTENNA

CHANGEOVER RELAYS

Figure 5

PROTECTIVE CONTROL CIRCUIT

With this circuit arrangement either switch

may be closed first to light the heaters of

all tubes and the filament pilot light. Then

when the second switch is closed the high

voltage will be applied to the transmitter and

the red pilot will light. With a 30- second de-

lay between the closing of the first switch

and the closing of the second, the rectifier

tubes will be adequately protected. Similarly,

the opening of either switch will remove

plate voltage from the rectifiers while the

heaters remain lighted.

One convenient arrangement for using a

Variac or Powerstat in conjunction with the

high -voltage transformer of a transmitter is

illustrated in figure 4. In this circuit a heavy

three -wire cable is run from a plug on the trans-

mitter to the Variac or Powerstat. The Variac

or Powerstat then is installed so that it is ac-

cessible from the operating desk so that the

input power to the transmitter may be con-

trolled during operation. If desired, the cable

to the Variac or Powerstat may be unplugged

from the transmitter and a dummy plug inserted

in its place. With the dummy plug in place the

transmitter will operate at normal plate voltage.

This arrangement allows the transmitter to be

wired in such a manner that an external Variac

or Powerstat may be used if desired, even

though the unit is not available at the time

that the transmitter is constructed.

Notes on the Use Plate voltage to the modula -

of the Variac tors may be controlled at the

or Powerstat same time as the plate volt-

age to the final amplifier is

varied if the modulator stage uses beam tetrode

tubes; variation in the plate voltage on such

tubes used as modulators causes only a mod-

erate change in the standing plate current.

Since the final amplifier plate voltage is being

controlled simultaneously with the modulator

plate voltage, the conditions of impedance

match will not be seriously upset. In several

high power transmitters using this system, and

using beam -tetrode modulator tubes, it is pos-

sible to vary the plate input from about 50

watts to one kilowatt without a change other

than a slight increase in audio distortion at

the adjustment which gives the lowest power

output from the transmitter.

With triode tubes as modulators it usually

will be found necessary to vary the grid bias

at the same time that the plate voltage is

changed. This will allow the tubes to be op-

erated at approximately the same relative point

on their operating characteristic when the plate

voltage is varied. When the modulator tubes are

operated with zero bias at full plate voltage, it

will usually be possible to reduce the modu-

lator voltage along with the voltage on the

modulated stage, with no apparent change in

the voice quality. However, it will be necessary

to reduce the audio gain at the same time that

the plate voltage is reduced.

20 -2 Transmitter

Control Methods

Almost everyone, when getting a new trans-

www.americanradiohistory.com

386 Transmitter Keying and Control THE RADIO

o

o 115 V.A.0

LINE

o

o 230 v. A C

SINGLE PHASE

WITH GROUNDED

NEUTRAL

s

TO EXCITER POWER SUPPLIES

T

HI -LO

POWER RELAY

POWER CONTROL RELAY

TO FILAMENT TRANSFORMERS

'Tv

K1 HI -LO

K2 POWER

RELAY

POWER CONTROL RELAY

TRANSFORMERS

Figure 3

FULL -VOLTAGE /HALF -VOLTAGE

POWER CONTROL SYSTEMS

The circuit at (A) is for use with o 115 -volt

a -c line. Transformer T is of the standard

type having two 11S -volt primaries; these

primaries are connected in series for half -

voltage output when the power control relay

Kt is energized but the hi -lo relay K2 is not

operated. When both relays are energized the

full output voltage is obtained. At (B) is a

circuit for use with a standard 230 -volt resi-

dence line with grounded neutral. The two

relays control the output of the power sup-

plies the some as at (A).

primaries in parallel will deliver full output

from the plate supply. Then when the two pri-

maries are connected in series and still oper-

ated from the 115 -volt line the output voltage

from the supply will be reduced approximately

to one half. In the case of the normal class C

amplifier, a reduction in plate voltage to one

half will reduce the power input to the stage

to one quarter.

If the transmitter is to be operated from a

230 -volt line, the usual procedure is to operate

the filaments from one side of the line, the

low- voltage power supplies from the other side,

and the primaries of the high -voltage trans-

former across the whole line for full power

output. Then when reduced power output is

required, the primary of the high -voltage plate

transformer is operated from one side to center

tap rather than across the whole line. This

procedure places 115 volts across the 230 -volt

winding the same as in the case discussed in

the previous paragraph. Figure 3 illustrates

the two standard methods of power reduction

with a plate transformer having a double pri-

mary; (A) shows the connections for use with

a 115 -volt line and (B) shows the arrangement

for a 230 -volt a -c power line to the transmitter.

The full- voltage /half- voltage methods for

controlling the power input to the transmitter,

as just discussed, are subject to the limitation

that only two levels of power input (full power

and quarter power) are obtainable. In many

cases this will be found to be a limitation to

flexibility. When tuning the transmitter, the

antenna coupling network, or the antenna sys-

tem itself it is desirable to be able to reduce

the power input to the final stage to a rela-

tively low value. And it is further convenient

to be able to vary the power input continuous-

ly from this relatively low input up to the full

power capabilities of the transmitter. The use

of a variable -ratio auto -transformer in the cir-

cuit from the line to the primary of the plate

transformer will allow a continuous variation

in power input from zero to the full capability

of the transmitter.

Variable -Ratio There are several types

Auto- Transformers of variable -ratio auto- trans-

formers available on the

market. Of these, the most common are the

Variac manufactured by the General Radio

Company, and the Pouerstat manufactured by

the Superior Electric Company. Both these

types of variable -ratio transformers are excel-

lently constructed and are available in a wide

range of power capabilities. Each is capable

of controlling the line voltage from zero to

about 15 per cent above the nominal line volt-

age. Each manufacturer makes a single -phase

unit capable of handling an output power of

about 175 watts, one capable of about 750 to

800 watts, and a unit capable of about 1500 to

1800 watts. The maximum power- output capa-

bility of these units is available only at ap-

proximately the nominal line voltage, and must

be reduced to a maximum current limitation

when the output voltage is somewhat above or

below the input line voltage. This, however, is

not an important limitation for this type of

application since the output voltage seldom

will be raised above the line voltage, and when

the output voltage is reduced below the line

voltage the input to the transmitter is reduced

accordingly.

www.americanradiohistory.com

HANDBOOK Transmitter Control 385

not drop more than 5 volts (assuming a 117 -

volt line) under load and the wiring does not

overheat, the wiring is adequate to supply the

transmitter. About 600 watts total drain is the

maximum that should be drawn from a 117 -volt

lighting outlet or circuit. For greater power,

a separate pair of heavy conductors should be

run right from the meter box. For a 1 -kw. phone

transmitter the total drain is so great that a

230 -volt "split" system ordinarily will be re-

quired. Most of the newer homes are wired with

this system, as are homes utilizing electricity

for cooking and heating.

With a three -wire system, be sure there is

no fuse in the neutral wire at the fuse box. A

neutral fuse is not required if both "hot" legs

are fused, and, should a neutral fuse blow,

there is a chance that damage to the radio

transmitter will result.

If you have a high power transmitter and do

a lot of operating, it is a good idea to check

on your local power rates if you are on a

straight lighting rate. In some cities a lower

rate can be obtained (but with a higher "mini-

mum") if electrical equipment such as an

electric heater drawing a specified amount

of current is permanently wired in. It is not

required that you use this equipment, merely

that it be permanently wired into the electrical

system. Naturally, however, there would be no

saving unless you expect to occupy the same

dwelling for a considerable length of time.

Outlet Strips The outlet strips which have

been suggested for installation

in the baseboard or for use on the rear of a desk

are obtainable from the large electrical supply

houses. If such a house is not in the vicinity

it is probable that a local electrical contractor

can order a suitable type of strip from one of

the supply house catalogs. These strips are

quite convenient in that they are available in

varying lengths with provision for inserting

a -c line plugs throughout their length. The

a -c plugs from the various items of equipment

on the operating desk then may be inserted

in the outlet strip throughout its length. In

many cases it will be desirable to reduce the

equipment cord lengths so that they will plug

neatly into the outlet strip without an excess

to dangle behind the desk.

Contactors and The use of power -control con -

Relays tactors and relays often will

add considerably to the oper-

ating convenience of the station installation.

The most practicable arrangement usually is

to have a main a -c line switch on the front of

the transmitter to apply power to the filament

transformers and to the power control circuits.

It also will be found quite convenient to have

a single a -c line switch on the operating desk

to energize or cut the power from the outlet

strip on the rear of the operating desk. Through

the use of such a switch it is not necessary to

remember to switch off a large number of sepa-

rate switches on each of the items of equip-

ment on the operating desk. The alternative

arrangement, and that which is approved by the

Underwriters, is to remove the plugs from the

wall both for the transmitter and for the oper-

ating -desk outlet strip when a period of oper-

ation has been completed.

While the insertion of plugs or operation of

switches usually will be found best for ap-

plying the a -c line power to the equipment, the

changing over between transmit and receive

can best be accomplished through the use of

relays. Such a system usually involves three

relays, or three groups of relays. The relays

and their functions are: (1) power control relay

for the transmitter -applies 115 -volt line to the

primary of the high- voltage transformer and

turns on the exciter; (2) control relay for the

receiver -makes the receiver inoperative by

any one of a number of methods when closed,

also may apply power to the v.f.o. and to a

keying or a phone monitor; and (3) the antenna

changeover relay- connects the antenna to the

transmitter when the transmitter is energized

and to the receiver when the transmitter is not

operating. Several circuits illustrating the ap-

plication of relays to such control arrangements

are discussed in the paragraphs to follow in

this chapter.

Controlling Transmitter It is necessary, in

Power Output order to comply with

FCC regulations, that

transmitter power output be limited to the mini-

mum amount necessary to sustain communica-

tion. This requirement may be met in several

ways. Many amateurs have two transmitters;

one is capable of relatively high power output

for use when calling, or when interference is

severe, and the other is capable of consider-

ably less power output. In many cases the

lower powered transmitter acts as the exciter

for the higher powered stage when full power

output is required. But the majority of the ama-

teurs using a high powered equipment have

some provision for reducing the plate voltage

on the high -level stages when reduced power

output is desired.

One of the most common arrangements for

obtaining two levels of power output involves

the use of a plate transformer having a double

primary for the high -voltage power supply. The

majority of the high -power plate transformers

of standard manufacture have just such a dual -

primary arrangement. The two primaries are

designed for use with either a 115 -volt or 230 -

volt line. When such a transformer is to be

operated from a 115 -volt line, operation of both

www.americanradiohistory.com

384 Transmitter Keying and Control THE RADIO

FROM LINE

__ TO OTHER

- HOUSE CIRCUITS

PLAN OA

Figure 1

THE PLAN (A) POWER SYSTEM

A -c line power from the main fuse box in the

house Is run separotely to the receiving

equipment and to the transmitting equipment.

Separate switches and fuse blocks then are

available for the transmitters and for the

auxiliary equipment. Since the fuses in the

boxes at the operating room will be in series

with those at the main fuse box, those in the

operating room should have a lower rating

than those at the main fuse box. Then It will

always be possible to replace blown fuses

without leaving the operating room. The fuse

boxes can conveniently be located alongside

one another on the walla the operating room.

SHORT CORDS FROM

RECEIVER VF.O..CLOCR

FRED METER, TO

OUTLET STRIP.

PLAN pB

Figure 2

THE PLAN (B) POWER SYSTEM

This system is less convenient than the (A)

system, but does not require extensive re-

wiring of the electrical system within the

house to accommodate the arrangement. Thus

it is better for a temporary or semi -permanent

installation. In most cases it will be neces-

sary to run an extra conduit from the main

fuse box to the outlet from which the trans-

mitter is powered, since the standard arrange-

ment in most houses is to run all the outlets

In one room (and sometimes all in the house)

from a single pair of fuses and leads.

type. It is possible also that the BX cable will

have to be permanently affixed to the trans-

mitter with the connector at the fuse -box end.

These details may be worked out in advance

with the electrical inspector for your area.

The general aspects of Plan ( B) are shown

in figure 2. The basic difference between the

two plans is that (A) represents a permanent

installation even though a degree of mobility

is allowed through the use of BX for power

leads, while plan (B) is definitely a temporary

type of installation as far as the electrical in-

spector is concerned. While it will be permis-

sible in most areas to leave the transmitter

cord plugged into the outlet even though it is

turned off, the Fire Insurance Underwriters

codes will make it necessary that the cord

which runs to the group of outlets at the back

of the operating desk be removed whenever the

equipment is not actually in use.

Whether the general aspects of plans (A) or

(B) are used it will be necessary to run a num-

ber of control wires, keying and audio leads,

and an excitation cable from the operating desk

to the transmitter. Control and keying wires

can best be grouped into a multiple -wire rubber -

covered cable between the desk and the trans-

mitter. Such an arrangement gives a good ap-

pearance, and is particularly practical if cable

connectors are used at each end. High -level

audio at a moderate impedance level (600 ohms

or below) may be run in the same control cable

as the other leads. However, low -level audio

can best be run in a small coaxial cable. Small

coaxial cable such as RG -58 /U or RG -59/U

also is quite satisfactory and quite convenient

for the signal from the v.f.o. to the r -f stages

in the transmitter. Coaxial -cable connectors of

the UG series are quite satisfactory for the

terminations both for the v -f -o lead and for any

low -level audio cables.

Checking an To make sure that an outlet will

Outlet with a stand the full load of the entire

Heavy Load transmitter, plug in an electric

heater rated at about 50 per cent

greater wattage than the power you expect to

draw from the line. If the line voltage does

www.americanradiohistory.com

CHAPTER TWENTY

20 -1

Transmitter Keying and Control

Power Systems

It is probable that the average amateur sta-

tion that has been in operation for a number

of years will have at least two transmitters

available for operation on different frequency

bands, at least two receivers or one receiver

and a converter, at least one item of monitor-

ing or frequency measuring equipment and

probably two, a v.f.o., a speech amplifier, a

desk light, and a clock. In addition to the

above 8 or 10 items, there must be an outlet

available for a soldering iron and there should

be one or two additional outlets available for

plugging in one or two pieces of equipment

which are being worked upon.

It thus becomes obvious that 10 or 12 out-

lets connected to the 115-volt a -c line should

be available at the operating desk. It may be

practicable to have this number of outlets in-

stalled as an outlet strip along the baseboard

at the time a new home is being planned and

constructed. Or it might be well to install

the outlet strip on the operating desk so as

to have the flexibility of moving the operating

desk from one position to another. Alterna-

tively, the outlet strip might be wall mounted

just below the desk top.

Power Drain When the power drain of all the

Per Outlet items of equipment, other than

transmitters, used at the oper-

ating position is totalled, you probably will

find that 350 to 600 watts will be required.

383

Since the usual home outlet is designed to

handle only about 600 watts maximum, the

transmitter, unless it is of relatively low power,

should be powered from another source. This

procedure is desirable in any event so that the

voltage supplied to the receiver, frequency con-

trol, and frequency monitor will be substan-

tially constant with the transmitter on or off

the air.

So we come to two general alternative plans

with their variations. Plan (A) is the more de-

sirable and also the most expensive since it

involves the installation of two separate lines

from the meter box to the operating position

either when the house is constructed or as an

alteration. One line, with its switch, is for the

transmitters and the other line and switch is

for receivers and auxiliary equipment. Plan

( B) is the more practicable for the average ama-

teur, but its use requires that all cords be re-

moved from the outlets whenever the station

is not in use in order to comply with the elec-

trical codes.

Figure 1 shows a suggested arrangement for

carrying out Plan (A). In most cases an instal-

lation such as this will require approval of the

plans by the city or county electrical inspector.

Then the installation itself will also require

inspection after it has been completed. It will

be necessary to use approved outlet boxes at

the rear of the transmitter where the cable is

connected, and also at the operating bench

where the other BX cable connccts to the out-

let strip. Also, the connectors at the rear of

the transmitter will have to be of an approved

www.americanradiohistory.com

HANDBOOK Deluxe Mobile Transceiver 563

A complete chassis -assembly mock -up should

be made up of cardboard sheets, and the

various parts laid out in order to ascertain

their final position. The tuning capacitor gang

is made up of two dual units and two single

units, with their shafts cut to length so that

the over -all depth of the gang allows room

for the p.a. plate coil and associated padding

capacitors.

Transceiver The under -chassis wiring may

Wiring be observed in figure 38. All

power wiring is laced to form

a harness that runs about the chassis in a

square loop centered about the coil assembly.

Small components are mounted directly to tube

socket pins, to lug terminal strips, or to small

phenolic terminal boards. Ground connections

are made to lugs placed beneath socket re-

taining bolts.

The r.f. components of the receiver occupy

the center portion of the chassis. Small inter -

stage shields made of durai separate the r.f.,

mixer, and oscillator stages, and an additional

shield plate covers the bottom of the 6AH6

v.f.o. compartment. To the rear of this com-

partment are the driver stages of the trans-

mitter section.

A wiring harness of the type used in this

transceiver may best be made up external to

the unit. A layout of the harness and the

terminations of the various wires is sketched

full -size on a large board and the wires are

then laid out on the board in their proper posi-

tions, cut to length, and laced. The completed

harness is then dropped into the equipment

and the terminations made. An amateur experi-

enced in equipment construction, or who has

done this type of assembly and wiring as a

vocation will find this style of construction

interesting and a challenge to his ingenuity.

The beauty of the final equipment is well

worth the time and study it takes to design

and lay out a unit of this order of complexity.

Transceiver When the transceiver is com-

Alignment pleted, all wiring should be

and Test checked and "rung out" to pre-

clude the possibility of wiring

errors or accidental grounds. The tubes are

now placed in the unit, and the various tuned

circuits adjusted to their approximate operat-

ing range by means of a grid -dip oscillator.

The transmitter and modulator tubes are re-

moved, and the receiver section is aligned in

the following manner: The first step is to

align the low frequency i.f. strip. A low level

modulated 260 kc. signal is injected into the

plate circuit of the 6BE6 second mixer and

transformers T.., T3, and T4 are adjusted for

maximum receiver output. Next, oscillator coil

L of the 6BE6 stage is adjusted for maxi-

mum #1 grid current and a 4.26 Mc. signal

is fed to the input circuit of the 6BA7 first

mixer. Transformer Tt is adjusted for

maximum signal strength.

A 29 Mc. signal is now applied to the

antenna circuit of the receiver, and the main

tuning dial is adjusted to this approximate

setting. Coil L3 and capacitor Ca of the master

oscillator are adjusted until the test signal is

heard. The tuned circuit of the oscillator is

aligned to cover the span of 23,740- 25,440

kc., with equal leeway on each end of the

range. The test signal is now placed on 29.5

Mc. and the padding capacitors of the r.f. and

Figure 39

REAR VIEW OF

TRANSCEIVER

Amplifier tuning and

loading controls are

mounted on rear of the

cabinet. Below (left to

right) are: antenna re-

ceptacle, power recep-

tacle, speaker receptacle,

and 5 -meter zero -set po-

tentiometer. Additional

ventilation is provided by

rows of holes across rear

of cabinet.

11 t f t t _ t! t!! tt

www.americanradiohistory.com

564 Receivers and Transceivers THE RADIO

mixer stage are adjusted for maximum signal.

The signal is next shifted to 28.5 Mc. and

the variable slugs of the r.f. and mixer coils

are adjusted in turn. This process is repeated

until the tuning range of the receiver is correct,

and the r.f. stages track properly across the dial.

The transmitter section may now be aligned.

The tubes are inserted in their sockets and

relay RY1 is activated. The screen power lead

to the 6146 is temporarily opened to disable

that stage. Once again, the transceiver is tuned

to 29.5 Mc. and the two padding capacitors of

the 6CL6 buffer stages are adjusted for maxi-

mum grid drive to the 6146 stage. (Note:

Grid current to the 6146 should be held to

less than 4 ma. at all times) . The dial is now

returned to 28.5 Mc. and the variable slugs

of the buffer circuits are adjusted for maxi-

mum grid drive. The adjustments are repeated

until reasonably constant grid drive occurs

across the tuning range. The buffer stage and

power amplifier are neutralized and screen

voltage is applied to the 6146 tube. The

transmitter frequency is set to 29.0 Mc. and

the amplifier is tuned and loaded by means

of the controls on the rear of the cabinet

(figure 39) . The frequency of the transceiver

is shifted to 29.5 Mc. and (without adjusting

the loading capacitor) resonance is again re-

established with the rear tuning capacitor, C.

Now, the frequency is shifted to 28.5 Mc.

and auxiliary capacitor C14 is adjusted for

resonance. This sequence of adjustment is re-

peated until proper resonance and loading

occurs across the dial. Resonant plate current

should be approximately 110 milliamperes

and grid current should be 2 to 3 milliamperes.

Modulator resting plate current is 25 milli-

amperes, rising to about 80 milliamperes

under full modulation.

The transmitter may be bench -tested with

an a.c. power supply and a dummy load before

it is placed in the automobile. Car mounting

is accomplished by means of two heavy alu-

minum rails bolted to the top of the trans-

ceiver case which slide into suitable clamps

affixed under the dash of the automobile as

shown in figure 31. A transistor -type power

supply or a dynamotor may be used. 250 volts

at about 150 milliamperes, and 500 -600 volts

at 200 milliamperes are required for operation

of the transceiver.

27 -7 A Deluxe

Receiver for the

DX Operator

The need exists for a high performance

receiver, suitable for s.s.b., a.m., and c.w.

operation that can be built in the home work-

shop at a modest price. The receiver should

have a high order of stability and sensitivity

and must have sufficient dynamic range to

protect it against excessive cross- modulation

caused by strong nearby signals. In addition,

it should be possible to build the receiver

without the use of special metal- handling tools.

The receiver described in this section was

designed to fill this need. It is a double con-

version superheterodyne, employing crystal con-

trol in the first conversion stage and a tunable

low frequency i.f. and mixer. This configura-

tion provides maximum stability and permits

the use of a dial calibrated directly in

frequency.

Collins mechanical filters and a Q- multi-

plier are used in the 455 kc. second inter-

mediate frequency amplifier to provide the

ultimate in selectivity and rejection and a

product detector is employed for c.w. and

Figure 40

FRONT VIEW OF DELUXE AMATEUR

COMMUNICATION RECEIVER

Six band receiver covers 80 -10 meters, with

extra bond for 15 Mc. reception of WWV

standard frequency signals. Collins mechanical

filters provide ultimate in selectivity for s.s.b.

a.m. phone, and c.w. The receiver employs a

crystal controlled first conversion oscillator

for high -order stability and "hang- a.g.c." for

improved sideband reception. A simplified

product detector is used for s.s.b. and c.w.

operation. The precision dial can be read to

one or two kilocycles on all bands. Room is

provided above main dial for inclusion of

v.h.f. converters for 2 and 6 meter operation,

if desired.

www.americanradiohistory.com

HANDBOOK DX Communication Receiver 565

V1

R

V2

IST MIX.

V4 VS

1ST I.F. 2ND MIX. -OSC.

(2.4 -2 9 MC. )

Ve

2ND I.F.

(33 KC.)

V7

3RD IF.

VIAL OSC. VS

X

(SEE PIG. 42)

RECEIVER

TUNING

1--15.0 XC. FILTER

(oSC1LLAFOA 1.945-2.445 MC. )

VIS V14A

VOLT. REGULATOR OSC.

100 RC 12AU7

VII

2ND AUDIO Ve

AM-CW-55B

V,4e

5 -METER ACC

12AU7t,

Figure 41

BLOCK DIAGRAM OF DELUXE AMATEUR COMMUNICATION RECEIVER

s.s.b. reception. An automatic gain control

circuit (a.g.c.) is provided for sideband, and

auxiliary equipment includes an S -meter and

100 kc. crystal calibrator. Reception of the 15

Mc. Standard Frequency (WWV) signal is

incorporated for receiver calibration purposes.

Construction is simplified by making the

receiver in modules that may be built and

tested one at a time.

The Receiver A block diagram of the re-

Circuit ceiver circuit is shown in figure

41. Fourteen tubes are used,

plus a voltage regulator. The power supply

utilizes semiconductors to reduce heating

effects.

The R.F. Section. The receiver covers the

amateur bands between 10 and 80 meters,

with an extra bandswitch position for "spot"

reception of WWV at 15 Mc. The r.f. stage

employs a 6DC6 semi -remote cutoff pentode

to provide maximum freedom from crosstalk

and front -end overload. A triode -connected

6AH6 serves as a low noise mixer stage, with

local oscillator injection on the control grid.

The first conversion oscillator is crystal con-

trolled using a 6BJ6 in a "hot cathode" circuit

operating on the low frequency side of the

received signal.

Receiver tuning is accomplished at the first

intermediate frequency range of 2.4 -2.9 mega-

cycles. Each tuning range thus covers 500

kilocycles. Any 500 kc. segment of the 10

meter band may be utilized by the proper

choice of the conversion crystal. The tunable

portion of the receiver consists of a 6BJ6 i.f.

amplifier, and a 6BE6 second mixer stage. The

oscillator portion of the 6BE6 tube tunes the

region 1.945 -2.455 Mcs. to provide a 455 kc.

intermediate frequency. Both oscillators are

voltage regulated for maximum frequency

stability.

The I.F. Section. Two i.f. stages are employed

to provide sufficient receiver gain. The first

stage uses a 6AH6 which directly follows the

mechanical filters and the Q- multiplier circuit.

The filters allow a choice of 0.5 kc. passband

for c.w., or a 3.0 kc. passband for sideband.

A.m. reception may be done by listening to

one of the two sidebands, or a 6.0 kc. band-

width filter may be substituted for the 3.0 kc.

unit. The Q- multiplier places a rejection

"notch" at any point in the filter passband to

eliminate heterodyne interference. The depth

of notch can be adjusted by an auxiliary

control.

The over -all gain of the receiver is set by

adjusting the "r.f. gain" control which fixes

the operating bias on the low frequency i.f.

www.americanradiohistory.com

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www.americanradiohistory.com

DX Communication Receiver 567

Figure 42

(See opposite page)

C1-C4 -50 Aµfd. National UM -50 or equivalent

C5-100 µofd. National UM -100 or equivalent

C} -E. F. Johnson SMB11, with 240 µµtd. silver mica

shunted across each section

Ci -22 gµl& silver mica capacitor with 7 -3S µµI&

ceramic trimmer connected in parallel

C, -120 µµtd. silver mica capacitor with 7 -35 µµfd.

ceramic trimmer connected in parallel

RFC -1 mh. J. W. Miller Co. 1:J300 -1000

S,4,n,e,n- Centralab PA -305 assembly with 6 -inch

shalt and six Centralab PA -17 ceramic sections

(60 degree index)

SSA,n,e --- Centralab PA -301 assembly with 4 -inch

shaft and two Centralab PA -0 ceramic sections

S3- Corner Plates of C, bent to short out filter

T1, T2 -J. W. Miller Co. #B -727RF coil with S -27

shield

Lr -J. W. Miller Co. re-727C coil with S -27 shield

X, Crystals -International Crystal Mfg. Co.

Dial- Eddystone. Available from British Radio Elec-

tronics, Ltd., 1833 Jefferson Place, N.W., Washing-

ton 6, D. C.

All bypass capacitors are .01 ;lid., disc ceramic, 600

volt. High frequency oscillator capacitors are silver

mica

Mechanical filters- Collins Radio Co., 455 kc., style K

stages and also on the tunable i.f, stage. The

front end of the receiver operates at maximum

sensitivity and gain at all times in order to

override the inherent tube noise level of the

various mixer stages.

The Detector and Audio Section. A 6BE6

mixer tube is employed as a hybrid detector.

For sideband and c.w. operation, it functions

as a product detector, with injection on the

#1 grid from the beat oscillator and signal

injection on the #3 grid. For a.m. service,

the beat oscillator is disabled, and the signal

is switched to the #1 grid. Thus one tube

serves two functions, and does both of them

well. The beat oscillator is a 6BJ6, with vari-

able injection taken from the plate circuit.

The oscillator frequency may be moved across

the passband of the i.f. system to provide a

choice of upper or lower sideband reception,

as desired.

The automatic gain control system employs

a separate 6BJ6 i.f. amplifier stage driving a

simple "hang- a.g.c." system of the type de-

scribed by W1DX in the January, 1957 issue

of QST magazine. The 6BJ6 stage isolates the

b.f.o. from the a.g.c. system and prevents

oscillator voltage from leaking into the a.g.c.

circuit. The latter circuit is especially designed

for s.s.b. and c.w. reception. It has a very

rapid response that prevents receiver overload

on a syllabic burst of s.s.b., instantly reducing

receiver gain to prevent overloading. The gain

reduction remains in effect as long as the

signal is in evidence, then "hangs" on for

about 0.5 second after the removal of the

signal. This sequence of action reduces to a

minimum the usual "thump" that occurs at

the start of a syllable and removes the "rush"

of background noise at the end of a syllable

that occurs with a conventional a.v.c. system.

A triple diode 6BC7 and one -half a 12AU7

double triode comprise the complete "hang-

a.g.c." system. The double diode system fol-

lowing transformer T., and the 470K/0.01

µfd. R -C network determine the "on" time

of the "attack" system, permitting the 0.1 µfd.

a.g.c. capacitor to charge up in a relatively

quick time. The capacitor remains charged, as

the 12AU7 triode is cut off by this action,

and there remains no discharge path to ground

in the a.g.c. circuit, even when the voltage

across the "attack" R -C network is removed.

The time constant of the "release" network

is considerably longer, and after a predeter-

mined period, the a.g.c. voltage across this

network decays sufficiently to permit the triode

section to conduct and discharge the a.g.c. line

capacitor. The proper ratio of voltages in the

two R -C circuits can easily be established by

proper adjustment of transformer T. A slight

degree of delayed a.g.c. action is provided by

applying fixed bias to the "attack" diode to

prevent the circuit from being tripped by

back,.grc und noise or weak signals.

The S- Meter, Audio System and Power Supply.

The S -meter circuit is a simple vacuum tube

voltmeter that compares the a.g.c. voltage

against a fixed reference voltage. The circuit

is balanced for a meter null with no signal

input to the receiver, and a.g.c. voltage un-

balances the circuit causing a reading on the

meter placed in the bridge of the circuit. The

meter may be used for all modes of reception,

providing usable readings on c.w. signals as

well as sideband or a.m.

A single 6AK6 provides sufficient audio

for earphone reception, or to drive a speaker

to good room volume. Ignition and other

pulse -type noise is effectively reduced by

means of a peak noise clipper made up of

two inexpensive semiconductor diodes.

The power supply is a voltage doubler type

utilizing inexpensive silicon rectifiers. High

www.americanradiohistory.com

¡00000 1

_Z .i

-14 ' .. ; -_

www.americanradiohistory.com

DX Communication Receiver 569

Figure 44

TUNABLE I.F. SECTION OF RECEIVER

The tunable i.f. section cf the receiver is

built upon a 3" x 5" x 7" aluminum chassis.

Input and output connections are made via

"phono- type" coaxial fittings and lengths of

RG -58 U coaxial line. Tube in foreground is

68E6 mixer (Vs), and tube in the rear is

6816 tunable i.f. (V4). Ceramic padding capa-

citors C, (two) and Cv are mounted at right

of chassis, with the three Li. coils atop the

chassis.

Figure 45

UNDER -CHASSIS

VIEW OF TUNABLE

I.F. SECTION OF

RECEIVER

Tunable i.f. stage is iso-

lated from second mixer

by shield partition across

middle of chassis box.

Mixer and oscillator sec-

tions of Vs are separated

by a small partition.

Tuning capacitors are

mounted to the shield

partitions and are driven

through metal shaft coup-

lings. Power receptacle is

at rear of chassis. Com-

plete assembly is fas-

tened to main chassis by

six sheet metal screws.

voltage is regulated by an 0A2 for the entire

receiver, and standby is accomplished by break-

ing the B -plus line from the supply. Three

separate filament windings on the power

transformer provide sufficient capacity to

power all the tubes. The use of 150 milli-

ampere filament tubes wherever possible re-

duces the filament drain considerably. The

whole receiver runs reasonably cool because

of the low plate voltage and choice of low

filament power tubes, achieving a high order

of thermal stability in a short period of time.

Receiver

Construction

A receiver such as this is a

complex device and its con-

struction should only be under-

taken by a person familiar with receiving

equipment and who has built equipment of

this category before.

The receiver is built upon an aluminum

chassis measuring 1531" x 11" x 3" in size,

and is contained within a ventilated cabinet

measuring 16" x 111'4" x 91 ') ". The tunable

i.f. system is built as a separate unit on an

aluminum chassis -box measuring 3" x 5" x

7" (figures 44 and 45). The mechanical

filter assembly is also built as a separate unit

in an aluminum box measuring 2" x 3" x

www.americanradiohistory.com

570 Receivers and Transceivers THE RADIO

51/4" ( figure 47) . The b.f.o. assembly is built

within an aluminum box measuring 11/2" " x

2" x 23/ " (figure 48). The remainder of the

receiver is built upon the main chassis.

No receiver is better than its tuning dial,

so the very excellent Eddystone geared slide

rule dial is used. The dial is centered hori-

zontally on the panel and vertical placement

is adjusted so that the drive engages the shaft

of the variable tuning capacitors of the tunable

i.f. system.

Figure 46

UNDER -CHASSIS VIEW OF

COMMUNICATIONS RECEIVER

The receiver is built in sections which may be

checked out one at a time for sake of sim-

plicity. Crystal controlled r.f. section is at

left, with coil slugs projecting from front and

back of assembly. Conversion crystals are

mounted in holders on front partition. Near

center of chassis is box containing mechanical

filters and switch (figure 47). At right is par-

tition holding Q- multiplier coil and potentiom-

eter, with auxiliary notch control located on

the panel. The product detector switch is

driven off- center by two flexible couplings.

Power supply, diode rectifiers and filter sec-

tion are at lower right, with audio stages

across bottom of chassis.

The chassis, panel, and tunable i.f. chassis

should be assembled and studied before any

chassis holes are drilled. The dial cut -out

should next be made, making sure of align-

ment of the dial with the variable capacitors.

Placement of the remainder of the components

is not at all critical.

The Tunable I.P. System. It is best to con-

struct this item first, as it determines dial

position and placement of other major parts.

A close -up of this assembly is shown in figures

44 and 45. The three variable capacitors are

ganged by means of brass shaft bushings. The

first capacitor is mounted to the front wall of

the chassis -box, and the other two are placed

on aluminum interior partitions. The 6BE6

mixer tube is mounted to the front with the

6BJ6 at the rear. Power connections are made

to a miniature connector on the rear of the

chassis, and input and output terminations are

made through "phono- type" coaxial connectors

and short lengths of RG -58/U coaxial line.

The Mechanical Filter Assembly. A partition

separates the input and output circuits of the

filter assembly, as shown in figure 47. Bulk-

www.americanradiohistory.com

HANDBOOK DX Communication Receiver 571

head mounting filters are used to achieve a

maximum degree of isolation across the filter.

The individual segments of the selectivity

switch (S_) are mounted in each compart-

ment, with the switch mechanism passing

through the bulkhead. A spring wiping contact

is made for the rotor arm of the switch,

grounding it at the center bulkhead to pre-

vent a leakage path around the filter from

being formed. Input and output terminations

are made via "phono- type" coaxial fittings

and RG -58/U coaxial line.

The input and output circuits of the filters

must be tuned to frequency. This is accom-

plished by a 50 µµtd. variable padding capa-

Figure 47

INTERIOR VIEW OF MECHANICAL

FILTER ASSEMBLY BOX

Bulkhead mounting mechanical filters are

mounted to interior partition which isolates

input and output sections. Drive shaft of selec-

tivity switch is grounded at point it passes

through partition by a wiping spring to achieve

maximum circuit isolation. Input and output

tuning capacitors of filters are made up of

SO ppld. variable ceramic trimmers connected

in parallel with 75 µpfd. silver mica capacitors.

Trimmers are adjusted for maximum signal

response, in same manner as i.f. transformer

capacitors.

citor placed across each circuit and adjustable

from the bottom of the receiver.

The R.T. Assembly. The r.f. assembly is con-

structed within the main chassis as shown in

figure 50. The sockets for the 6DC6 r.f. stage,

the 6AH6 mixer, and the 6BJ6 oscillator are

mounted on the main chassis and the asso-

ciated coils, tuning capacitors, and bandswitch

are mounted to four vertical partitions fixed

beneath the chassis. Slug -tuned coils are used

for all circuits and are mounted in a hori-

zontal position about the bandswitch. The r.f.

and mixer coils can be aligned by means of a

"TV- type" screwdriver thrust through holes in

the rear of the chassis, while the r.f. coils are

adjusted from the front of the assembly by

means of a short screwdriver. The partitions

are mounted so that a space of 2" exists be-

tween them, and the associated tube socket

falls in the center of each space. The switch

assembly passes through the partitions and, in

fact, holds them in position by virtue of the

switch arms and spacers. The individual switch

segments are placed so that they are near the

end of each coil. This results in a very com-

pact assembly having extremely short leads to

all coils. The coils are staggered about the cir-

cumference of a circle so that both the r.f.

and mixer slugs can be reached from the rear

11 www.americanradiohistory.com

572 Receivers and Transceivers THE RADIO

without interference between the coils.

The four partition plates are cut from

1/32 -inch aluminum stock, and follow the

layout of figure 51. They are not notched at

first. Rather, a cardboard template is cut out

and marked for drilling as shown. Then all

four partitions are clamped together and

drilled along with the template. Corner notches

are now cut and all edges filed so that all four

partitions are as identical in size and shape as

possible. Only the holes shown in figure 51

are common to all pieces. The front and rear

partitions have other holes - i.e., crystal

sockets, antenna input, power lead holes, etc.

These may be drilled during layout and assem-

bly of the unit as required. The 1/2-inch

flanges are then bent over, taking care to bend

the front and rear pieces in the proper

direction.

The coils should be wound to the data of

figure 49, before the unit is assembled. Only

three coils are used in the oscillator section

as an r.f. choke is employed on the 80 and

40 meter bands. The 14 Mc. coils are jum-

pered across the switch and used for the

WWV position on 15 Mcs. All coils should

be wired to the bandswitch before the tuning

Figure 48

REAR VIEW OF RECEIVER

Placement of major parts may be seen in

this view. B.f.o. components and tube are

mounted in small aluminum box next to the

front panel (left), with S -meter above main

tuning dial. At right on panel is standby con-

trol switch, with noise limiter switch beneath

it. Power transformer is at left rear of chassis.

On rear apron of chassis are placed !1. to r.):

115 volt power receptacle, utility socket,

break -in gain control, S -meter adjust poten-

tiometer, speaker terminals, and coaxial an-

tenna receptacle. At extreme right are pass -

through holes to permit alignment of high

frequency r.f. coils.

www.americanradiohistory.com

HANDBOOK DX Communication Receiver 573

Band L1

80 9t #24e

3/16" 1. closewound

Coil Table

Figure 49

Ls, L4

55t #30e

closewound

40 6t #24e 33t #24e

3/16" 1. closewound

L3 L5

20t #30e RFC

closewound

12t #24e RFC

closewound

20 4t #34e 16t #24e 8t #24e 40t #30e

3/16" 1. spaced length closewound closewound

of form (11600 kc.)

15 3-1/2t #24e 14t #24e 7t #24e 23t #24e

3/16" 1. spaced as above closewound spaced to cover

form

(18,600 kc.)

10 3t #24e 12t #24e 6t #24e 16t #24e

3/16" 1. spaced as above closewound spaced to cover

form

(25,600 kc.)

All coils wound on XR -SO forms.

Ls, L. wound first -then a loyer of 1/2" Scotch No. 33 tape wound on cold ends of L2, L4 coils

and LI, L3 primaries wound over tape. Small strip of tope plus coil cement secures the free

ends of LI, L3.

80 M coils L2, L4 have SOµ,.fd. padders soldered across terminals.

capacitors are finally mounted in place. The

last step is to use the unit as a template to

mark the clearance holes on the rear of the

chassis, which are drilled before the unit is

finally installed in the chassis.

Receiver Wiring. The remainder of the re-

ceiver wiring is simple and straightforward.

The sideband -a.m. switch (S4) is offset from

the panel hole to clear the Q- multiplier coil

(Ls) mounted on an L- shaped bracket be-

neath the chassis (figure 46) . The audio low -

pass filter coil (L9) is placed between the

6BE6 detector and 12AU7 audio socket. Long

runs of a.c. leads are done in shielded wire, as

are audio leads.

Receiver The receiver may be aligned in

Alignment sections. The first step is to

align the i.f. system and beat

oscillator. Next, the tunable i.f. stages should

be aligned and tracked. Finally, the r.f. sec-

tions are properly tuned.

The i.f. system should be aligned to the

center of the passband of the narrowest -band-

width mechanical filter. In the case of the

500 cycle filter, the center frequency must be

455.0 kilocycles with very little tolerance. The

system may be roughly aligned with the aid

of an external signal generator coupled into

the #3 grid of the 6BE6 second mixer. A

455 kc. signal of low amplitude is injected

into the input circuit and the tuning capa-

citors across the filter terminals, plus trans-

formers T3 and T4 are adjusted for maximum

response. The Q- multiplier should be out of

the circuit for this test (switch S3 closed).

Care should be taken not to overload the i.f.

system during alignment, so a relatively weak

signal should be used for this portion of the

adjustment. A.g.c. transformer T5 should then

be adjusted to provide the proper "attack" and

"release" time for the gain control circuit.

Finally, the slug of the b.f.o. coil (L10) is set

to place the beat oscillator signal at the center

of the i.f. passband with the b.f.o. panel

control set at mid- scale.

The signal generator is now shifted to the

input circuit of the 6BJ6 tunable i.f. stage.

The main tuning dial is set at 500 (minimum

circuit capacitance). The generator is ad-

justed to 2.90 Mc., and padding capacitor Cy

of the oscillator section is adjusted for signal

response. 1.f. and mixer padders C. are then

tuned for maximum signal. At a dial reading

www.americanradiohistory.com

574 Receivers and Transceivers THE RADIO

e

Figure 50

UNDER -CHASSIS

VIEW OF R.F.

COIL ASSEMBLY

The high frequency coils

are placed in a circle

about the bandswitch

(figure 51). Coils and

capacitors are mounted

on four shield partitions

which are located be-

tween the tube sockets.

R.f. stage socket (V1) is

at rear of chassis, with

mixer socket (V2) in cen-

ter, and crystal oscillator

socket (V3) nearest the

panel. Oscillator crystals

are mounted on front

partition. Entire assem-

bly is shielded by alu-

minum cover plate.

of zero (maximum circuit capacitance), the

tunable stages should resonate at 2.40 Mc.

Attention should now be given to the front -

end stages. It is a good idea and a time saver

to peak circuits L1 -L2 and L3 -L4 to the proper

frequency with the aid of a grid -dip oscillator.

Coil L; is adjusted for proper crystal oscillator

operation, which may be monitored in a

nearby receiver. The signal generator is now

set to the center frequency of the 500 kilo-

cycle band in use and a moderate signal is

injected into the antenna circuit of the receiver.

The main tuning dial is adjusted to receive

the signal, and the r.f. and mixer coils are

peaked for maximum response with the r.f.

tuning capacitors set at mid -scale.

Once alignment has been completed, the

operator should familiarize himself with re-

ceiver operation. The last step is to adjust

the "break -in" gain control so that the receiver

may be used to monitor c.w. transmissions. The

Figure 51

R.F. ASSEMBLY PLATES

Four assembly plates are required, as shown.

Each plate is drilled as necessary for mounting

of small components, etc.

PELATIvE POSiT,ON OF TUBES

WHEN A5SEMBLED

UM-50

OuTLINE

E --

OR -50 FORM OUTLINE - 6 (MAO ) USED PER STAGE.

12 HOLES 1 Di SPACED EVERT 50. ON

1 R NOTE OWL V SHOWN NERE

ROTOR

TB

TO

STATOR

TAB

L

z

i - -05WoTCNMOUNT

X SCREW BOA 5 121

B_ \ 2

ITCH e -p

TAB

www.americanradiohistory.com

HANDBOOK DX Communication Receiver 575

short across the control circuit is removed sensitivity level. The control may be shorted

from the utility socket on the rear apron and out by an external switch or relay during

the control adjusted for the desired standby periods of reception.

Figure 52

SCHEMATIC, POWER SUPPLY FOR

RECEIVER

L11 -4.5 H at 200 ma. Stancor C -1411

S, , A -2 pole, 3 position rotary switch

SRI, 2 -200 ma. rectifier. Sarkes- Tarzian M-

500 with dual mounting kit

T7-117 volts at 200 ma. Three 6.3 volt

windings at 2.0, 4.0, and 4.23 amperes,

respectively. Stancor P -8158

Ti 6. 3 V., 4. 2 5 A ., T O I.F. I AUDIO U D I O

(CREENLEADS)

63V,4.0 A.,TO R. F.

(BROWN LEADS)

6.3 V., 2 OA TO TUNABLE I. r

(YELLOW LEADS) L.

SR, OFF

5W

+ 40

200

STa

sreY.

oN-0 a+

350

www.americanradiohistory.com

CONVERSION

E & E TECHNI

TABLE - UNITS

-MILLIONTH

-THOUSANDTH

-SHEET

OF MEASUREMENT

MICRO = (µ) ONE

MILLI = (m) ONE

KILO (K) ONE THOUSAND

MEGA (M) ONE MILLION

TO CHANGE

FROM TO OPERATOR

UNITS MICRO -UNITS

MILLI -UNITS

KILO -UNITS

MEGA -UNITS

X 1,000,000 or X 108

X 1,000 or X 103

± 1,000 or X 10 -3

± 1,000,000 or X 10 -8

MICRO -UNITS MILLI -UNITS

UNITS - 1,000 or X 10 -3

± 1,000,000 or X 10 -6

MILLI -UNITS MICRO -UNITS

UNITS

X 1,000 or X 103

= 1,000 or X 10 -3

KILO -UNITS MEGA -UNITS

UNITS - 1,000 or X 10 -3

X 1,000 or X 103

MEGA -UNITS KILO -UNITS

UNITS

X 1,000 or X 103

X 1,000,000 or X 108

www.americanradiohistory.com

CHAPTER TWENTY -EIGHT

Low Power

Transmitters and Exciters

The transmitter is the "heart" of the ama-

teur station. Various forms of amplifiers and

power supplies may be used in conjunction

with basic exciters to form transmitters which

will fit almost any requirement. Several dif-

ferent types of transmitting equipment de-

signed to meet a wide range of needs are out-

lined in this chapter. A simple transistorized

transmitter for 50 Mc. is described. This unit

is a good introductory project for the amateur

to "cut his teeth on" relative to the field of

transistors. Also shown is a complete, TVI-

proof, medium- powered all -band phone and

c.w. transmitter. A "W9TO" electronic keyer

is illustrated, together with newly -developed

"Strip Line" circuits which are applicable to

the v.h.f. spectrum. For the amateur who is

interested in the construction phase of his

hobby, these units should offer interesting

ideas which might well fit in with the design

of his basic transmitting equipment.

Figure 1.

A POCKET -SIZE

50 MC. TRAN-

SISTORIZED PHONE

TRANSMITTER.

Capable of 100 milliwatts

input, this "collector

modulated" six meter

phone transmitter will

provide amazing results

when used with a good

antenna system. The com-

plete unit may be held

in the palm of the hand.

Panel controls are 11. to

r.): crystal oscillator tun-

ing (top) and audio gain

control (bottom), multi -

meter, amplifier tuning

(top) and loading (bot-

tom). Switch on left is

the multi -meter switch,

with power switch at

right. Microphone plug

is centered between

switches.

www.americanradiohistory.com

578 Low Power Transmitters THE RADIO

MIC

RCA

2N384

5O AK X1 (PNP) LI

RCA

2N3B4

(P)(P) L2

SO R

ADJUST

!/AS

t

OSC w--AMP

2 N44

(PNP)

ALL RESISTORS 1/2 WATT.

SCHEMATIC, 50 MC.

-ISV 15V.

AUX.

Figure 2.

TRANSISTORIZED TRANSMITTER.

L1-6 turns =18 wire, 58 inch diameter, S/8

inch long. (B8W miniductor 3007.) Top

three turns from transistor end

L2, L3 -Make both coils from a single piece

of 88W miniductor =3007. Use nine turns.

Cut coil between sixth and seventh turn,

making two coils having six and two turns,

respectively, separated by a distance of

one turn

M -0 -10 ma., d.c., 11/4" square meter

T1- Transistor transformer, SK to 80K. Thor -

darson TR -13

T2, Ta-Transistor transformer, 10K to 2K.

Triad TY -56X

28 -1 A Transistorized

50 Mc. Transmitter

and Power Supply

The simple 50 Mc. transistorized trans-

mitter shown in this section makes an inter-

esting project for the amateur who wishes to

familiarize himself with high frequency tran-

sistors. Capable of 100 milliwatts input, this

little phone transmitter will give a good

account of itself when it is used in conjunction

with a beam antenna. It may be run from

batteries or from a regulated a.c. power supply.

Circuit The transmitter circuit utiliz-

Description ing inexpensive PNP -type

transistors is shown in figure

2. The oscillator is crystal controlled, employ-

ing a 2N384 in conjunction with a 50 Mc.

third -overtone crystal connected between

collector and base of the drift transistor. Oper-

ating bias level is adjusted by a variable

potentiometer. The low impedance base of the

2N384 amplifier is tapped on the oscillator

coil to achieve a match to the higher imped-

ance collector circuit of the oscillator. The

amplifier collector output circuit is inductively

coupled to the antenna. It may be seen that

this configuration bears a close similarity to a

vacuum tube circuit in that the emitter of the

transistor resembles the cathode of the tube.

The base may be compared to the grid, and the

collector to the plate.

A two stage modulator section provides

sufficient gain to operate a dynamic micro-

www.americanradiohistory.com

HANDBOOK 50 Mc. Transmitter 579

phone. The audio stages are tranformer

coupled and base driven. A 1N34 diode is

used as a high level positive peak loading

device to prevent peak clipping at high modu-

lation levels. Positive peak clipping is em-

ployed since the collector supply voltage is

negative with respect to ground. A simple

metering system permits the operator to moni-

Figure 3.

REAR VIEW OF

TRANSISTORIZED TRANSMITTER.

The two r.f. transistors are mounted in sockets

on L- shaped bracket at the center of the

chassis. Directly below them is the oscillator

bias -potentiometer. Across the rear edge of

the chassis are the audio stages, with the

power terminals on the rear apron of the

chassis. Relative size of transmitter and com-

ponents may be judged from comparison with

standard coaxial receptacle at left of chassis.

Oscillator stage is at right, with amplifier

at left.

tor the collector current of the r.f. stages.

The positive terminal of the power supply

is at "ground," or chassis potential. If NPN-

type transistors are substituted for the speci-

fied units, battery polarity must be reversed.

Transmitter

Construction The complete transmitter is

built upon a small aluminum

chassis measuring x 31/2"

x 1" in size. The front panel measures 6" x

4 ". The two r.f, transistor sockets and ri.

components are mounted on an L- shaped alu-

minum bracket centered on the chassis, measur-

ing 2" high by 21/4" long. The right -angle

portion of the bracket holding the crystal

socket is 11/2" high by 1" wide. Miniature

transistor sockets are mounted in the top

corners of the bracket, with the oscillator bias

control centered beneath them (figure 3) .

www.americanradiohistory.com

580 Low Power Transmitters THE RADIO

The transistorized audio section is placed

across the rear of the chassis. Transformer

leads pass through small rubber grommets to

the under -chassis area. At one end of the

chassis is an aluminum bracket holding the

coaxial antenna receptacle. Small components

are mounted under the chassis on phenolic

terminal strips. Transmitter wiring is straight-

forward, and is done with #22 insulated wire.

Coil data is given in figure 2.

Shown in figures 5 and 6 is a simple volt-

age regulated power supply that provides 18

volts at 100 milliamperes. A 2N561 power

transistor is used as a series regulator, with a

2N44 serving as a regulator driver stage. The

control element is a Zener diode delivering a

constant source of 14.7 volts, which is used to

set the output voltage. As the transmitter is

operating near maximum transistor voltage

values, it is important that the power supply

Figure 4.

UNDER -CHASSIS VIEW OF TRANSMITTER.

Miniature components are mounted on phe-

nolic terminal strips beneath the chassis.

"Clipping" diode is at right, behind slide

switch. Audio leads are run in shielded wire.

voltage remain constant under varying loads.

A voltage surge could possibly damage the

transistors in the transmitter at this relatively

high operating potential.

The power supply is built upon an alu-

minum chassis measuring 51/2" x 31/2" x 1 ".

The 2N561 power transistor must be insu-

lated from the chassis by means of mica shims

or an anodized plate, as the collector element

is bonded to the case of the unit. The power

supply may be tested by placing a 350 ohm,

10 watt resistor across the output. 18 volts

should be developed across the resistor.

Transmitter

Adjustment

and Tune -up

When the transmitter wiring is

completed, it should be care-

fully checked, especially in the

area of the transistor sockets.

Insert the r.f. transistors and crystal in their

sockets and turn the oscillator bias potenti-

ometer to maximum resistance. Place the meter

switch in the oscillator position. Use a 52 ohm,

1 -watt composition resistor across the antenna

receptacle as a dummy load for these tests.

Turn the transmitter on and adjust the oscil-

lator tuning capacitor for oscillation ( jump in

collector current) as noted on the meter. Ad-

www.americanradiohistory.com

HANDBOOK 200 -Watt Transmitter 581

TI SRI -

Figure S.

SCHEMATIC,

VOLTAGE REGULATED POWER SUPPLY.

B -115 volt neon lamp in holder

5R, -4- Silicon rectifier, 400 v. p.i.v., S00 ma.

Sarkes -Tarzian CM -500

T,- Filament transformer. 26.8 volts at 1 a.

Triad F -40X

Z, -Zener diode, 15 volts, 1/2 watts, Moto-

rola 1.5M15Z (10% tolerance)

just the bias potentiometer for about 5 milli-

amperes oscillator current. Now, place the

meter switch in the amplifier position and

adjust the oscillator tuning capacitor for maxi-

mum meter reading. Adjust the amplifier tun-

ing capacitor for a meter dip. Finally, adjust

the antenna loading until the meter indicates

about 6 milliamperes, re- resonating the circuit

with the collector tuning capacitor. A field

strength meter is helpful for the initial

tune -up.

The signal may now be monitored in a

nearby 50 Mc. receiver. Connect a dynamic

microphone and modulate the transmitter, ad-

justing the audio gain control for good modu-

lation. The transmitter is now ready to be

connected to your station antenna.

28 -2 A Deluxe

200 -Watt Tabletop

Transmitter

This self contained, TVI- proof, tabletop

transmitter is designed for the amateur who

desires a compact station capable of running

sufficient power to provide consistent results

in today's busy amateur bands. Modern in

Figure 6.

VOLTAGE REGULATED POWER SUPPLY.

The silicon rectifiers are mounted above the

chassis for proper ventilation, with the two

transistors directly in front. 2N561 transistor

is insulated from the chassis by a mica shim.

styling, this deluxe unit is designed around the

7270 beam power tube and is capable of a

conservative input of 200 watts on phone,

and 250 watts on c.w. The transmitter covers

all amateur bands between 10 and 80 meters,

is v.f.o. controlled, and incorporates speech

clipping for maximum audio "punch." A semi-

conductor high voltage rectifier is used to

reduce heat and to provide improved voltage

regulation. "Break -in" c.w. keying is incorpor-

ated employing a time differential system that

results in chirp -free, clickless keying. Band

changing is simplified by ganging the exciter

switching circuits with the final amplifier pi-

network so that single control adjustment is

achieved. In short, the transmitter incorporates

all modern techniques to make it an up -to-

date, valuable item of station equipment that

will not become obsolete.

Circuit A block diagram of the table -

Description top transmitter is shown in

figure 8. Thirteen tubes are

employed, five in the r.f. section, five

in the audio section, and the remain-

der in the control and power supply sec-

tion. A complete schematic is shown in

figures 9 and 10. The RCA 7270 beam power

tube is employed in the final amplifier stage.

This compact tube has high -perveance and

good power gain. It can be operated at full

input above 50 Mc., and has a maximum plate

dissipation of 90 watts. At a plate potential

of 1000 volts, this miniature "bottle" is cap-

able of 250 watts input on c.w., and 200 watts

input on a.m. phone. In addition, the tube has

triple base -pin connections for the screen grid

to permit good r.f. grounding and has large

plate radiating fins for effective cooling. The

www.americanradiohistory.com

582 Low Power Transmitters T H E R A D I O

compact size makes it especially effective in

the high frequency portions of the communi-

cation spectrum. Driving requirements are

modest and permit the use of a simple band -

switching exciter.

The Exciter Section. The high stability, all -

band v.f.o. consists of a 6AH6 (V1) in a

"hot cathode" circuit, followed by a 6CL6

(V2) crystal oscillator- buffer stage. Very

high -C is used in the oscillator stage to swamp

out variations and changes in stray circuit and

tube capacitance. The frequency determining

circuit operates on 80 meters (L1 and asso-

ciated components) for 80, 40 and 10 meter

transmitter operation, and on 40 meters (L_

and associated components) for 20 and 15

meter transmitter operation. The circuit is a

modified version of the Clapp oscillator. The

tuning rate for each amateur band is changed

automatically so that each band is spread over

the entire portion of the tuning dial. Use of

the exceptionally smooth Eddystone dial with

a turn indicator makes it possible to read the

transmitter frequency within a kilocycle or

two. The oscillator is keyed by a section of the

12ÁU7 keyer tube (V6) for c.w. operation.

The crystal oscillator- buffer stage (6CL6,

V_) employs a broadly tuned 7 Mc. plate

circuit for operation on 40 meters and all

higher bands. For 80 meter operation, switch

section SIB inserts an r.f. choke in series with

the tuned circuit, dropping the r.f. output on

this band to the correct value, and eliminating

the necessity of tracking the stage across the

relatively wide band. Switch S2 disables the

v.f.o. and converts tube Vo into a 3.5 Mc.

crystal oscillator, with the choice of two

crystal frequencies.

Figure 7.

MODERN 200 WATT

ALL BAND TABLE -TOP TRANSMITTER.

Complete TVI -proof phone and c.w. transmit-

ter is housed in modern -style tabletop cabinet.

V.f.o. controlled, the transmitter covers all

amateur bands between 80 to 10 meters. High

level plate modulation with speech clipping is

used fcr ptione, and a time -sequence break -in

keyer is featured for c.w. operation. A stan-

dard 10 /s" x 19" panel is used in cose rock

mounting is desired. Multi -meter on the left

reads grid and screen current of amplifier

stage, or modulator plate current. Selector

switch is at left, directly below main tuning

dial.

Controls across bottom of panel are (I to r.):

audio -gain, microphone receptacle, filament -

on switch, amplifier plate tuning (top) and

amplifier plate loading (bottom), bandswitch,

v.f.o.- crystal switch (top) and amplifier arid

tuning (bottom), power switch (Ss) and pilot

light, c.w.- tune -a.m. switch, and key jack.

Below the tuning dial to the right is the grid

drive control, and at the for right is the

plate meter, M2.

www.americanradiohistory.com

HANDBOOK 200 -Watt Transmitter 583

VI

"FO - 7

MIC

Sz

V2

OSC.-SI/F.

V3

MUL r.

V DR/VER

® ® ®

Mf YER

VR

Ve

SPEECH

VP

CLIPPER

V7

GA IN CL/P.

3 RC

FILTER

-13PV.

1

RIAS

SUPPLY

V I 1

MOD.

V10

OR/ VER

VS

RF AMP.

PI NET.

+400 V.

i

V12

V13

5 V

L V. SUPPLY

+1000 V.

N.V. SUPPLY

Figure 8.

BLOCK DIAGRAM OF 200 WATT TABLE -TOP TRANSMITTER.

41.1 '

115V 1

The plate circuit of the 6CL6 multiplier

stage (V3) is untuned for 80 and 40 meter

operation, and is resonated to 14 Mc. for 20

and 10 meter operation by coil L4, and to 21

Mc. for 15 meter operation by coil L5. This

stage is block -grid keyed for c.w. operation.

A 2E26 (V4) is used as a driver for the

7270 amplifier. This stage is neutralized and

operates "straight through" on all bands except

10 meters, where it acts as a doubler from

14 Mc. A potentiometer control (grid drive)

in the screen circuit of the 2E26 determines

the excitation level to the final amplifier

stage. The 7270 (V3) serves as a neutralized

amplifier on all bands. Grid, screen and plate

current are monitored for proper operation. A

pi- network output circuit permits operation

into unbalanced loads having impedances in

the range of 50 to 75 ohms, and an s.w.r. value

of 2.5 to 1, or less. The screen circuit is pro-

tected by relay RY3 which is energized by

application of primary power to the high volt-

age plate supply. Thus, screen voltage cannot

be applied to the tube unless plate voltage is

also applied.

The Mode Switch, S3. For tune -up purposes,

amplifier screen voltage is dropped to a low

value by the c.w.- tune -a.m. mode switch sec-

tion S3,,. In the c.w. position, protective cut -off

bias is applied to the 7270 by switch section

S{i,. For phone operation, the amplifier screen

circuit is "self- modulated" by choke CHI

placed in the circuit by switch section Sac.

The keyer tube (V11, 12AU7) keys the

oscillator in addition to the 6CL6 multiplier

stage, and optimum break -in characteristics

may be set by the variable potentiometer

www.americanradiohistory.com

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www.americanradiohistory.com

586 Low Power Transmitters THE RADIO

labelled adjust keying. Switch section S3A

shorts out the keyer in the tune and a.m.

modes. Switch sections StD and S3F disable

the modulator and speech amplifier in the

c.w. and tune positions. Switch section S3E

activates the 400 volt power supply in the

Figure 11.

TOP VIEW

OF 200 WATT TRANSMITTER.

Layout of above- chassis ports is shown in this

view. Amplifier compartment is center rear,

with buffer compartment between it and v.f.o.

box near panel. Main power transformer is to

the left of amplifier compartment, with modu-

lation transformer and 811 -A tubes to the

right.

Adjacent to the horizontal oscillator tube (on

left of v.f.o. box) are the two 6CL6 amplifier

stages. The low- voltage power supply com-

ponents are on the separate chassis at the

left. On the side of the main chassis is the

large cutout for air intake to the ventilating

fan. To the right of the v.f.o. box are located

the 12AU7 keyer tube and the OB2 voltage

regulator. The adjust "keying" and "clipping"

controls are between these tubes and the

shielded speech amplifier tubes on the far

edge of the chassis. The 6DE7 is next to the

modulator tubes.

c.w. and tune positions, and the supply is

activated by push -to -talk relay RY1 when

switch S3 is set in the phone position. In

addition, the transmitter may also be activated

by the panel mounted power switch, S5 which

completes the relay control circuit.

The Modulator Section. A pair of zero bias

811A tubes (V11, V12) are used as class

B modulators, eliminating the need of bias

and screen power supplies which are costly

and expensive. A dual -purpose 6DE7 (V10)

serves as a speech amplifier and driver stage.

A 6AL5 double diode (V9) is a low level

audio peak clipper which serves to increase

the average level of modulation. This stage

is followed by a home -made low -pass filter

that restricts all audio frequencies above 3000

cycles. A 12AX7 dual triode (V8) provides

sufficient gain for proper transmitter opera-

tion from a low level crystal microphone. In

addition, a microphone push -to -talk circuit

may be used to energize d.c. relay RY1 by

means of a microphone control switch.

www.americanradiohistory.com

HANDBOOK 200 -Watt Transmitter 587

Power Supplies. A careful selection of power

supply components makes it possible to build

a transmitter of this capability in such a

small space. A cooling fan has been incor-

porated to insure that proper movement of air

is maintained, and components have been

selected for adequate safety margins and cool

operation. The chassis has several cut -out

openings on the sides and top for air circula-

tion, and the chassis bottom plate of the audio

section is made of perforated aluminum.

The low voltage and bias supplies are con-

ventional; however, the high voltage supply

makes use of a bridge circuit employing twelve

miniature silicon diode rectifiers. The center

tap of the transformer high voltage winding

is not used, and the 5 -volt winding is em-

ployed only to light a panel indicator lamp

when the high voltage is switched on. The

high voltage rectifier "stack" is protected from

accidental overloads by a 1/2- ampere fuse

placed in the B -plus lead to the filter system.

Three 40 pfd., 450 -volt electrolytic capaci-

tors are placed in series to provide approxi-

mately 12 izfd. at a working voltage of 1350.

Transmitter The entire transmitter, includ-

Construction ing power supplies, is built

upon a heavy aluminum chassis

measuring 13" x 17" x 3" in size. Shielded,

chassis -type construction is used, and no re-

liance is placed upon the cabinet for TVI-

reduction (figures 7 and 11) . The v.f.o. is

built as a separate unit in a 3" x 4" x 5"

aluminum box which is bolted to the main

chassis behind the geared dial.

The low voltage and bias supplies are built

on an aluminum chassis measuring 4" x 7" x

11/2", and may be seen in figure 11. The

5V4 -GA rectifier tube (V13) is mounted

"outboard" on a small L- shaped bracket beside

the power transformer (T2), fitting in nicely

between the supply chassis and the buffer

stage. The supply leads are brought through

grommeted holes in the main chassis to ter-

minal strips placed on the side apron of the

chassis.

The 7270 amplifier stage is entirely en-

closed in an aluminum box measuring 61/2

inches square and 614 inches high (figure

12) . The top and back of this enclosure are

fabricated from a single piece of perforated

aluminum. The other three sides of the box

are formed from an aluminum sheet, while

the main chassis serves as the bottom of the

enclosure.

The 2E26 buffer stage is mounted between

the v.f.o. enclosure and the final amplifier

compartment. The buffer tube is placed in a

horizontal position to best isolate the input

and output circuits, and to obtain short leads

in the plate circuit. The enclosure has screened

sides and measures 3" x 2" x 1/2 " in size.

The 2E26 tube projects into the box, with

the base connections remaining outside the

box in close proximity to the neutralizing

Figure 12.

CLOSEUP OF FINAL

AMPLIFIER

ASSEMBLY.

The top and front of the

final amplifier enclosure

have been removed to

show placement of major

components. The tank

coil is mounted in a ver-

tical position, bolted to

the side wall of the box.

The output loading capa-

citor is just below the

10 -meter coil section.

The neutralizing capa-

citor is mounted on in-

sulated pillars between

the 7270 tube and the

tank tuning capacitor.

Plate leads are made of

silver -plated copper

strap. The perforated

shield at front of photo

covers the horizontally

mounted 2E26 buffer

tube.

www.americanradiohistory.com

200 -Watt Transmitter 589

capacitor. The plate coils of the 2E26 are be-

neath the chassis, grouped about bandswitch

section S11). The buffer tuning capacitor

( labelled grid tuning) is adjacent to the band-

switch ( figure 13) .

Placement of the major components may be

seen in figure 11. The audio section is on

the right of the chassis (viewed from the

rear) and is separated from the r.f. section by

a partition running the entire depth of the

chassis on the underside. The 0A2 voltage

regulator tube (V7) and the 12AU7 differ-

ential keyer tube (V6) are also in the audio

section.

The remainder of the smaller components

are mounted beneath the chassis ( figure 13 ).

The modulator section is to the left, while

along the opposite side of the chassis are

located the small blower fan, the high voltage

silicon rectifiers, and the large filter choke.

The center portion of the chassis is reserved

for the r.f. section of the transmitter. The 7270

socket is centered towards the rear, directly

behind a horizontal partition that separates the

final amplifier components from the exciter

stages. Vent holes are cut in the side aprons

of the chassis (figure 11) and are covered

with screening.

In order to mount the 6.3 volt filament trans-

former (T4) on the side apron, a hole is drilled

in the side of the case and the transformer

leads are brought out through this hole, in-

stead of via the bottom hole. This same tech-

nique is used to mount the high voltage filter

choke, CH_. To facilitate mounting these

components, 6 -32 nuts are soldered to the

mounting flanges to accept the mounting bolts.

Panel Layout and The panel layout is dic-

Bandswitch tared by placement of the

Placement major components. The

v.f.o. tuning dial is cen-

tered on the panel near the top to allow proper

clearance for the drive mechanism. The dial

drive shaft, therefore, determines the position

of the dual v.f.o. tuning capacitor which is

mounted inside the enclosed oscillator assem-

bly. A flexible coupling is used to join the

dial to the capacitor to provide proper shaft

alignment and smooth tuning. The v.f.o. itself

is built as a separate unit after the position

of the oscillator tuning capacitor has been

determined.

The power amplifier output loading capa-

citor, plate tuning capacitor, and pi- network

coil switch (S1E) are controlled from the

front panel by means of right -angle drive sys-

tems placed beneath the chassis. The band-

switch S1 (centered on the panel) drives the

v.f.o. bandswitch through a right angle coupler,

in addition to driving the pi- network switch

of the amplifier stage. Two small bevel gears

are used for the oscillator drive, one mounted

on the main bandswitch assembly between

segments SIB and SID, and the other placed

on the shaft of switch SIA which is located in

the v.f.o. compartment (figure 14) . The oscil-

lator bandswitch is placed directly below the

v.f.o. tuning capacitor, with its shaft on the same

vertical center line as that of the capacitor. The

switch projects down through a 3A-inch match-

ing hole in the chassis, placing the shaft at

right angles to the center line of the main

bandswitch where it is driven by the bevel

gears. The main bandswitch assembly passes along

the center line of the chassis to the final am-

plifier area, extending through a shield parti-

tion which isolates the multiplier and driver

coils. An added section of shaft coupling drives

a set of Boston gears mounted on a small sup-

port bracket at the back of the chassis. The

gears have a 1:2 step -down ratio, as the final

amplifier bandswitch has 60- degree indexing,

whereas the main bandswitch has 30- degree

indexing.

It is a good idea to assemble the chassis,

panel, and v.f.o. box, and lay out the various

gear drive systems before other holes are

drilled or components mounted in place. The

final amplifier tuning and loading capacitors

are mounted alongside the pi- network coil

and their shafts project into the under -chassis

area where they are joined to right -angle

drives which bring the controls to the front

panel. The amplifier tuning capacitor is driven

with a set of Boston gears having a 2:1 step -

up ratio so that the dial turns 360- degrees

while the capacitor rotates 180- degrees. This

makes for easier adjustment of the circuit.

Placement of the remaining panel controls

and meters is not critical and is dictated by

good symmetry and eye -appeal. Panel and

chassis should be drilled together so that all

shaft holes are in alignment. Panel and chassis

are held together by two 13 -inch aluminum

angle brackets placed at the ends of the chassis.

www.americanradiohistory.com

590 Low Power Transmitters THE RADIO

Figure 14.

INSIDE THE V.F.O. ENCLOSURE.

The oscillator tube socket is mounted to the left wall of the box, with the tuning capacitor

adjacent to the terminals. The two one -inch diameter ceramic coil forms are mounted to the

opposite wall with the padding capacitors between them. At ftc bottom of the box is the

oscillator bandswitch, driven from the main bondswitch below deck by right -angle gears. Extra

bolts are used to fasten the sides of the box securely in place, and all paint is scraped off the

mating areas to ensure good contact.

www.americanradiohistory.com

HANDBOOK 200 -Watt Transmitter 591

1.0 JJF

600 V.

1 .0J.1 F

600V.

J

EA.

7.001

47 L

K [CH 5

l 1

47K

.01,600V.

K7',4K

i100 '

r--

270

K

1 .OJJF

600 V.

L_

Figure 15.

SPEECH AMPLIFIER TERMINAL BOARD.

Make of phenolic, or other insulating material.

Transmitter The transmitter is most easily

Assembly worked upon if the heavy

and Wiring transformers are left off the

chassis until the very last. The

v.f.o. and low voltage supplies can be wired

and tested as separate units before they are

affixed to the chassis.

The socket for the 7270 tube is recessed so

that the vent holes in the base are on the

underside of the chassis for passage of air

from the cooling fan. The variable neutralizing

capacitor for the amplifier stage is mounted

vertically between the socket and the plate

tuning capacitor (figure 12) and is adjust-

able from beneath the chassis. Space is limited

so a modified APC -type unit is used. It is

a 50 µpfd. size, with plates removed two at

a time to obtain a spacing of about 3/16 -

inch, leaving nine plates in all (4 rotor, 5

stator). The capacitor is connected to the low

potential (pi- network) side of the plate block-

ing capacitor so that d.c. plate voltage does not

appear across it.

Oscillator Construction. The whole v.f.o. unit

may be wired separate from the transmitter.

The tuning capacitor is a dual 25 µpfd. unit,

with two rotor plates removed from the front

section which tunes the 40 meter coil (L0).

Two ceramic coil forms are mounted on the

wall of the v.f.o. box opposite the tuning

capacitor and two MAPC -type adjustable pad-

ding capacitors are in a line between the coils.

The oscillator tube socket is on the side wall

below the tuning capacitor, and all associated

resistors and capacitors are mounted at the

socket, with the exception of the silver mica

capacitors which make up the various tuned

circuits. These are mounted on the band -

switch, or to the wires running between switch

www.americanradiohistory.com

592 Low Power Transmitters THE RADIO

CUT OUT (WARD

2-}"A 2"

RECTIFIER BOARD

FRAMEWORK

5+

TO N.V. TO N.V.

3

ALUMINUM NEAT BARRIER

I. I

204 204 204

s

401r

450 V. 4010

450V. 40Ur

450 V.

6-12

FILTER BOARD

Figure 17.

LAYOUT OF RECTIFIER AND

FILTER BOARDS.

and coils. All tuned circuit wiring is done

with #14 solid tinned copper wire. The v.f.o.

output lead to the next stage passes via a feed -

through insulator in the bottom of the box to

the under -chassis area. Filament and power leads

are brought out through a grommet to a

terminal strip beneath the main chassis.

The Exciter and Audio Circuits. The exciter

wiring is straightforward. The slug -tuned ex-

citer coils are grouped about the main band -

switch, and all r.f. leads are short and direct.

All r.f. bypass capacitors are mounted directly

on the socket terminals. Part of the speech

amplifier components (including the audio

filter) are pre- assembled on a phenolic board

which is mounted on the side apron of the

chassis ( figure 15) . Small components are

soldered directly to socket pins. Miniature

transistor -type cathode bypass capacitors are

used to conserve space. The clipping and

keyer controls are mounted on the chassis deck,

between the low level stages. Filter inductor

(CH5) is made from a Stancor TA -27 audio

transformer, using the entire secondary wind-

ing as the coil. The voltage dividers of the bias

supply and the dropping resistor for the regu-

lator tube are placed in this section.

The Power Supply and Control Circuits. The

twelve silicon diode rectifiers and the filter

network are placed below the high voltage

transformer. The diodes are mounted on a

perforated frame made from a sheet of fiber-

glass or phenolic material (figure 17) . The

diodes are supported by their leads from small,

hollow rivets employed as connecting points.

The diode leads should be left untrimmed, and

the leads are grasped with a pliers during the

soldering process to prevent the heat of the

iron from injuring the diode. The diode

mounting plate is attached to the side apron

of the chassis in front of the large air vent,

directly behind the ventilating fan.

The main filter capacitor consists of three

series connected 450 volt capacitors in parallel

with three wirewound resistors. These com-

ponents are wired as a unit and mounted on

a phenolic board on the rear apron of the

chassis, alongside the blower motor. The I/2-

ampere high voltage fuse holder is also

mounted on this board. A small aluminum

shield is placed between the resistors and

capacitors to act as a heat barrier ( figure 17)

Figure 16.

COIL DATA.

TABLE TOP TRANSMITTER.

L1-40 turns #22 enameled wire on I" ceramic

(National XR -62) Range: 3.5 -4.0 Mc.

L2 -17 turns #20 enameled wire on I" ceramic

(Notional XR -62) Range: 7.0 -7.175 Mc.

L3-40 turns »28 enameled wire on /2" form

(National XR -50) Ronge: 7.0 Mc.

L4 -20 turns #20 enameled wire on i /2" form

(National XR -50) Range: 14 Mc.

form

L5 -12 turns »20 enameled wire on /2" form

(National XR -50) Range: 21 Mc.

L6 -16 turns 3/4" diameter tapped at 9 and 12

turns from junction with LT

( #3011 B&W miniductor)

L7-38 turns #24 tapped at center 1" diameter

( »3016 B&W miniductor)

LB -» 851 B&W tank coil assembly

www.americanradiohistory.com

HANDBOOK 200 -Watt Transmitter 593

The a.c, line fuse holders, antenna connector,

relay connector J3, and Hy-pass feedthrough

capacitors for the power line are also mounted

on the rear apron. The coaxial antenna relay

(RY2) is placed on the outside of the apron

with a right -angle fitting added so that the

antenna connection is accessible when the

transmitter is placed in its cabinet.

Most of the power and control wiring fol-

lows along the front inside edge of the chassis.

Shielded wire is used for the 7270 filament

and screen leads, and filament circuits are

wired with #14 wire. The screen capacitor of

the 7270 stage consists of three separate .001

pfd., 3 kv. ceramic disc capacitors, one placed

from each screen socket pin to ground.

The multi -meter (M1) has a 5 milliampere

movement, and is converted into a low range

voltmeter by the addition of a 750 -ohm series

resistor. The voltage drops across shunts placed

in the grid and screen circuits of the ampli-

fier, and the plate circuit of the class B modu-

lator are measured in this fashion. The meter

scale is 0 -10 milliamperes when switch S4 is

in the grid position, 0 -40 milliamperes in the

screen position, and 0 -400 milliamperes in the

modulator position.

Tuning and When the transmitter is corn -

Adjusting the pleted, the wiring should be

Transmitter visually inspected, and circuits

"rung out" with an ohm-

meter. The next step is to test and adjust the

v.f.o. The fuse should be left out of the primary

circuit of the high voltage supply to disable

this section and to ensure that relay RY3 re-

mains open. The 2E26 screen control should

be set to remove screen voltage. Starting with

the 80 meter band and the v.f.o. dial set at

the low end (maximum v.f.o. tuning capaci-

tance), trimming capacitor C4 is adjusted for

3500 kc., as noted on a frequency meter. With

the specified coils, the 80 meter band extends

over the entire dial scale, with the slug almost

out of the coil form. Once the coverage is set,

the slug should be secured with an extra nut

to prevent movement. Capacitor C4 should not

be moved now, as it will be in the circuit for

the 40 and 10 meter adjustments. Next, the

bandswitch is placed in the 10 meter position,

the dial set at the low frequency end, and

trimmer capacitor C7 adjusted for 28.0 Mc.,

with the v.f.o. dial pointer in approximately

the same position as for 3.5 Mc. The 10 meter

band will now extend over almost the entire

scale. Next, the bandswitch is placed in the

40 meter position, and it will be noted that

the 7.0 Mc. position will fall very near the

28 Mc. mark, with the 40 meter band spread

over most of the scale.

The bandswitch should now be placed on

the 20 meter position, and trimmer capacitor

Cs adjusted so that 14.0 Mc. falls near the

3.5 Mc. dial point. The 15 meter calibration

is automatically set by this adjustment.

Once the v.f.o. has been calibrated, the 80

meter exciter circuits are tuned by simply ad-

vancing the 2E26 screen voltage potentiometer

and tuning the driver stage to resonance, as

indicated by grid current of the 7270 tube.

Grid current should be held to a maximum of

4 milliamperes. The bandswitch may now be

set to 40 meters and buffer coil L3 adjusted

for maximum amplifier grid current with the

v.f.o. set at 7.15 Mc. The 14 Mc. adjustments

are now made with the bandswitch in the 20

meter position, and coil L4 peaked for maxi-

mum amplifier grid current at 14.15 Mc.

Finally, the bandswitch is set to 15 meters and

the slug of coil L, is peaked at 21.2 Mc. The

driver stage is, of course, resonated for each

band. The ten meter band is tuned by merely

peaking the driver stage. Check both the low

and high ends of the 10 meter band and

equalize the grid drive by slight adjustments

to coils L3 and L4,

The 2E26 stage is neutralized in the 20

meter position by placing a temporary grid

meter in series with the "cold" end of the 22K

grid resistor and adjusting the neutralizing

capacitor for minimum meter kick when the

plate circuit is tuned through resonance.

Screen voltage should be removed for this test.

This setting will hold for all bands. Grid

current to the 2E26 should not run over 3

milliamperes.

The amplifier stage is now neutralized in a

similar manner, using meter M1 to observe

action of the grid current. This adjustment

should be done on the 10 meter band.

The final amplifier should not be operated

without a dummy antenna load of some kind.

Two 100 -watt lamp bulbs in parallel at the

end of a short length of coaxial line will make

a satisfactory load for preliminary adjustment

purposes. Place the high voltage primary fuse

in its receptacle and set the function switch S3

to the tune position. Grid current will now be

www.americanradiohistory.com

594 Low Power Transmitters THE RADIO

NOTE: DIMENS IONS OF FLANGE TO FIT

TUN /NG CAPACITOR TERMINALS.

C

-

A

"PLATE" LINE Lx

DRILL BOTH PLATES FOR INSULATED

BOLTS AND BUSHING.

D

ROUND

CORNERS

F

',CHOKE" LINE LI

CAPACITORS BU /LT WITHIN

E /MAC SOCKET.

s I

L I

+ SCR.

I 5001

IBIAS ,00

E %C.

L SUB -CHA SS /S

AREA.

LINE WITH

FINGER STOCK

DIMENSIONS

144 MC 220 MC.

A Cyy Dyy E F A B Cy/ D E F

10 , Byy

Et/i11L'tB 7} 21" MCI 2i 4 -

STRIP LINE CAVITY

RFC,

EQUIVALENT CIRCUIT

ANT

Figure 18.

SCHEMATIC AND EQUIVALENT CIRCUIT OF STRIP LINE AMPLIFIER.

The strip line amplifier is built within 3" x 5" x 13" aluminum chassis box (144 Mc.), or

2" x S" x 91/2" (220 Mc.). The plate tuning capacitor of the 144 Mc. assembly is a cut -down

Johnson 154 -11 having three plates, spaced 0.25" apart. The antenna "hairpin" loop is 1 turn,

4" long and 11/2" wide (144 Mc.), or 2" long 34" wide (220 Mc.) placed parallel to strip line.

Antenna resonating capacitor C2 is 35 µold. for either amplifier. Plate choke RFC, is Ohmite

Z -144 or Z -220. B -plus lead posses through insulated hole in chassis, or may pass through

feed- through type capacitor for low voltage operation (500 volts or less). The screen bypass

capacitors are built within the Eimac air system sockets. Input circuits and blower are placed in

sub- chassis enclosure.

B+

ANT.

observed on the final stage. The power switch

S5 is turned on energizing relay RY1, and the

final amplifier resonated and loaded to a plate

current of about 150 milliamperes. The series

screen resistor used in the tune mode limits

off - resonance amplifier plate current to less

than 200 milliamperes. The screen voltage tap

on the 2500 ohm, 25 watt resistor is now ad-

justed (with the transmitter off!) to place

about 320 volts on the 7270 screen circuit

with the function switch in the a.m. position,

and the amplifier loaded to 200 milliamperes

plate current. In the c.u'. position the screen

voltage will be slightly higher.

Maximum voltage (400 volts) is always

applied to the plate of the 2E26, and the drop-

ping resistors reduce this to about 260 volts

for the v.f.o., 6CL6 stages, and speech am-

plifier. The final plate voltage runs 1000

volts at a load current of 200 ma., and rises

to about 1200 volts in the c.w., key -up posi-

tion. Oscillator screen voltage is regulated at

105 volts. The bias supply delivers -135

volts, and the push -to -talk relay circuit is

tapped down on the bleed resistor to supply

about 100 volts to the d.c. relay RY1.

The c.w. keying characteristic is determined

by the adjustment of the keyer potentiometer,

and by the choice of the 0.1 pfd. capacitors in

the grid returns of the keyed tubes. For break -

in keying the "key -up" signal is monitored in

the receiver and the keyer potentiometer is

backed off until the oscillator signal just

disappears.

For phone operation, the modulator resting

plate current is about 20 ma., kicking up to

approximately 175 ma. on voice peaks. Maxi-

mum current excursions and modulation level

are set by the adjust clip control, and the

degree of modulation by the audio control.

www.americanradiohistory.com

HANDBOOK Strip Line Circuit 595

28 -3 Strip -Line

Amplifiers for VHF

Circuits

A major stumbling block in the design and

construction of high power v.h.f. transmitting

equipment is the assembly of a suitable am-

plifier plate tank circuit. Simple L -C tuned

circuits tend to assume microscopic propor-

tions in this region of the spectrum and are

incapable of handling large amounts of r.f.

energy. Coaxial circuits, on the other hand,

work well but are expensive, difficult to

build and bulky to handle.

A welcome compromise design is the simple

strip line tank circuit, illustrated in figure

18A. The circuit is a modified cavity, making

use of an inexpensive aluminum chassis as

the outer enclosure, and employing strips of

aluminum as the plate inductance. The line

assumes r.f. ground potential at the end oppo-

site the tube and is an approximate electrical

eighth -wavelength long. It becomes an elec-

trical quarter -wavelength when loaded by the

tube and tuning capacitor placed at the high

impedance end of the line. The line is made

of two aluminum plates, separated by insulat-

ing material. This "sandwich" may be visual-

ized as the equivalent circuit of figure 18B,

which permits plate voltage to be applied to

the amplifier tube via "plate line" L1 yet iso-

lates the tuning capacitor and plate inductance

from the d.c. voltage by means of a "distri-

buted" r.f. choke. The cavity is completed by

placing an aluminum cover plate over the

open side of the chassis.

A proper ratio of strip length and width

compared to the cavity dimensions must be

observed to determine the optimum line im-

pedance, but the parameters may be varied

sufficiently to permit the use of an inexpen-

sive ready -made chassis for the line cavity

without appreciable circuit degradation. Effi-

ciency of the strip line is high, comparing

favorably with conventional tank circuits

operating at intermediate frequencies.

The approximate characteristic impedance

of the strip line may be determined from the

following formula:

(1) Z "=" 377 S

W

where S is the spacing between the strip line

and the chassis, and W is the width of the

strip; the width being much greater than the

spacing. A practical strip line will be shorter

than a quarter wavelength by virtue of the

interelectrode capacitance loading of the asso-

ciated tube and the auxiliary tuning capacitor

placed across the line (figure 18A) . In this

case, the characteristic impedance of a loaded

strip line is approximately:

(2) Z = Z0 tan 131

0

where p = 3 , ) is the wavelength in

centimeters and 1 is the length of the line in

centimeters.

If the total capacitive reactance is set equal

to Z0, then tan /3, = 1, when the line length

is 1/2- wavelength. For example: Assume a

1/2- wavelength line having a width (W) of

3 inches and a spacing (S) of 1 inch. The

impedance, Zo is therefore (by formula 1)

about 127 ohms. The output capacitance of

a 4X250B is approximately 5 µµfd., represent-

ing an impedance value of about 220 ohms at

144 Mc. A parallel tuning capacitance of 5

µµfd. has the same impedance value, and the

combined parallel impedance is approximately

110 ohms. Therefore a 1/2- wavelength line of

the aforementioned dimensions could be used

to tune the 2 meter band with a 4X250B

tube. This line would be about 10 inches long,

so a standard chassis box measuring 3" x 5"

x 13" could be used for the plate cavity

assembly. The construction of such a unit is

described in this section.

Building the Shown in figure 19 are two

Strip Line strip line units for 144 Mc.

Circuit and 220 Mc. The amplifiers

are designed around the cera-

mic 4CX250B tube and may be operated at

power inputs up to 500 watts for c.w. service,

or 300 watts for a.m. phone. The limiting

factors for power input are the maximum

voltage rating of the plate bypass capacitor

(if used), tuning capacitor spacing, and the

voltage breakdown of the material employed

as the dielectric of the strip line circuit.

The units illustrated employ 10 -mil teflon

coated fiberglass as the strip line dielectric,

with fiber or teflon bushings and 4 -40 machine

screws holding the assembly together. It is

also possible to purchase teflon screws which

could be used to advantage in this assembly.

A sheet of 10 -mil mylar may be substituted for

the fiberglass.

Layout of the strip line units is shown in

www.americanradiohistory.com

596 Low Power Transmitters THE RADIO

Figure 19.

STRIP LINE AMPLIFIERS FOR 144 MC. AND 220 MC.

The simple mechanical assembly of the strip line tank circuit is especially suitable for home con-

struction. Using o standard aluminum chassis as the foundation, the strip line consists of two

aluminum plates separated by a dielectric. The line is supported from one end of the chassis

box, and the tube socket is mounted in the bottom, with the tuning capacitor at the opposite

end. At the near end of the assembly are the antenna resonating capacitor, the B -plus terminal

and the antenna coaxial receptacle. The tube plate "finger stock" connector is made by Eitel-

McCullough, Inc., San Carlos, California, part =008294, Anode Collet.

The dielectric material for the "sandwich" may be either 10 -mil (0.01 ") Mylar sheet, or 10 -mll

teflon coated fiberglass. The mylar may be obtained from: Milam Co., 1100 Elmwood St., Provi-

dence, R. I. The teflon coated fiberglass may be obtained from Dodge Fibers, Inc., Hoosick Falls,

N. Y. For maximum values of plate voltages, two layers of material should be used. Open side of

chassis is closed by cover plate.

figure 18. The "plate" section of the line (L2)

is bolted to one end of the chassis box, at the

proper height to encircle the anode of the tube

without actually touching it. The "hot" end

of this line is affixed to the stator of the plate

tuning capacitor. The capacitor of the 144 Mc.

amplifier has 0.25" spacing, as the unit is de-

signed for high power operation. The "choke"

plate of the "sandwich" line (L1) is shorter in

length and spaced away from the grounded

plate by means of the sheet fiberglass or mylar

insulator. One end of this plate is connected

to the B- supply through an auxiliary r.f. choke,

and the opposite end makes contact to the

anode of the tube by means of flexible metal

finger stock soldered to the plate (see parts

list) . Both plates are sanded smooth to ensure

that no metallic splinters or grains can punc-

ture the thin dielectric sheet.

The 220 Mc. unit is designed for low power

doubler service at 500 volts and therefore

makes use of a receiving -type capacitor in the

plate circuit. A capacitor having greater spac-

ing would be required for high voltage

operation.

The strip line amplifiers employ standard

Eimac v.h.f. air sockets to ensure stability of

operation. A standard grid circuit is employed

and if neutralization is desired, it is possible

to insert a probe into the strip line cavity and

www.americanradiohistory.com

HANDBOOK "9T0" Electronic Key 597

feed back a small amount of energy in the

proper phase to the grid circuit. A "hairpin"

loop (L3) provides coupling to the antenna

circuit, and the reactance of the loop is tuned

out by means of a series capacitor. The grid

circuit components are built within a small

chassis box placed beneath the strip line as-

sembly, with a cooling blower mounted on

the side of the box.

The dimensions given are correct for the

4X150A- 4CX250B type tube, but may be

varied for other tubes having slightly differ-

ent interelectrode capacitances. Length of the

strip line and the value of the tuning capacitor

determine the resonant frequency, with the

width of the center line and chassis spacing

determining line impedance and exhibiting a

second order frequency effect. It is therefore

possible to effect small changes in the fre-

quency of the circuit by varying the value of

the tuning capacitor or the width and chassis

spacing of the line if it is mechanically

awkward to adjust the length of the strip.

28 -4 A "9T0"

Electronic Key

The good c.w. operator is always trying to

improve his skill and increase his keying

speed. The modern way to do this is to use

an electronic key. The dots, dashes, and spaces

are all created electronically with a minimum

of effort on the part of the operator. A good

keyer has a "mechanical mind of its own" and

almost teaches the operator to send good code!

Shown in this section is a version of the

famous "9T0" keyer which provides the ulti-

mate in reliable, precise electronic code. The

keyer uses four tubes and two voltage regu-

lators, and is packaged in a cabinet only

slightly larger than a mechanical "bug" key.

Best of all, it is inexpensive to build and

fool -proof in operation.

Operation

of the Keyer One of the most reliable and

stable methods of generating

automatic and self- completing

dots and dashes is the multivibrator system

used in this keyer (figure 21). The keyer is

driven by a "sideswiper" key which completes

a control circuit to ground in either the "dot"

or "dash" position. Closing the key on the

"dot" side energizes the dot keyer tube

(V OA) which turns on the dot multivibrator

tube (VIA_B) to form a string of evenly

spaced dots. Once the action has started, this

generator will continue to form dots as long

as the key contact is closed and will complete

a full dot even if the key is released in the

middle of a dot or a space. The output of the

dot multivibrator is fed to the grid of the

relay tube (VIA), and the contacts of the

quick- acting relay in the plate circuit are used

Figure 20.

THE "9T0"

ELECTRONIC KEY.

This simple, inexpensive

electronic key generates

dots, dashes, and spaces

with a minimum effort

on the part of the oper-

ator. Four tubes and two

voltage regulators are

used in a simple and re-

liable circuit. The "side -

swiper" key is mounted

to an extension of the

bottom plate of the

keyer, making a unit

only slightly larger than

a mechanical "bug" key.

Panel controls are (I. to

r.): Weight control (with

on -off switch), monitor

speaker and speed con-

trol. Below these are the

zero -beat, or tune -up

button, and the neon

character indicator.

www.americanradiohistory.com

598 Low Power Transmitters THE RADIO

22 +150 V

HI B+

0. 1,200 V. T2 SPAR

VIA V1B

-12AU7

V3A V3B

- - 12AU7-

3 e

V2A

S, (ON WEIGHT

CONTROL)

11SV.1, 0

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ìF = MATCHED PAIR RESISTORS

ALL RESISTORS WATT UNLESS

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0A2

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5,1

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5,1

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Figure 21.

SCHEMATIC, ELECTRONIC KEY.

RYI -DPST, 5000 ohm relay. Potter -Brumfield SM -SLS. Other satisfactory (but larger)

relays are: Claire HG -1002 or W. E. 2766. The 15K series resistor may have to be

adjusted for different relay models. Weight of dots may be varied by changing

value of this resistor.

SRI, 3- Silicon rectifier. p.i.v. 400 volts @ S00 ma. Sackes-Tarzian «M -500.

TI -150 v. @ 50 ma., 6.3 v. @ 2a. Stancor PA -8421

T2- Push -pull replacement output transformer. Stancor A -3856

Key- Autronic sideswipes. Electrophysics Corp., 2500 West Coast Highway, Newport

Beach, California

H I B+

750 v

to key the transmitter and to activate an audio

tone oscillator (V4B) used as a monitor.

When the key is closed in the "dash" posi-

tion, the dash keyer tube (V.,B) is energized,

placing the dash multivibrator tube (V ;i A.B )

in readiness for operation, and at the same

time sending a pulse through the 1N34 diode

to start the dot multivibrator circuit again.

This, in turn, triggers the dash multivibrator,

turning it on with the start of the first dot

pulse, and turning it off with the end of the

second dot pulse. The dash multivibrator,

therefore, is an electronic switch which is

turned on and off by two dot pulses. A dash

of proper length and timing is created in this

manner because the time length of the second

"dot" adds to the "on" time of the switch cir-

cuit in holding the relay closed for the dash.

www.americanradiohistory.com

HANDBOOK "9T0" Electronic Key 599

Figure 23.

TERMINAL BOARD LAYOUT.

The parts shown outside of board are mounted

underneath it. The lines indicate connections

made to the board from tube pins or other

components.

The complete keyer configuration makes use

of four 12ÁU7 double triode tubes. The power

supply uses two silicon diodes to furnish both

a positive and a negative voltage, regulated by

the 0A2 and OB2 regulator tubes. Unregulated

voltage is supplied to the relay tube and the

tone oscillator. If desired, the 5963 computer -

type tube may be substituted for Vt, V., and

V3 for improved long term stability of

operation.

Keyer Construction The electronic keyer is

and Wiring built upon an aluminum

chassis measuring 6" x

4" x 2 ", having two

auxiliary end plates 5 inches high. A wrap-

over perforated aluminum cover screens the

top and sides providing maximum ventilation.

In addition, four large holes are punched in

the sides of the chassis for additional cooling.

The "sideswiper" key is mounted on an exten-

sion of the bottom plate of the chassis. The

wiring of the keyer is simplified by mounting

most of the multivibrator components on a

terminal board placed in the underchassis area

( figure 22) The board is mounted on two

pillars in the front -center of the chassis after

all other wiring has been done (figure 23).

The balance control is mounted on the rear

apron of the chassis, as it requires adjustment

only at intervals as the tubes age.

When the unit is completed, all wiring

should be checked. The unit is turned on and

after a short warm -up period the key lever

is held in the dash position and the balance

control adjusted until self -completing dashes

are formed. The neon lamp will flash at the

character rate. The speed and weight controls

are adjusted to suit the individual taste of the

operator.

Figure 22.

UN DE RCHASSIS

VIEW OF KEYER.

The resistors and capaci-

tors making up the mul-

tivibrators are mounted

on a terminal board sup-

ported below the chassis

on short pillars. The sili-

con diode power recti-

fiers are on the side

apron of the chassis ad-

jacent to the filter ca-

pacitors. The balance

potentiometer is on the

rear apron between the

keying lead and the

power cord.

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www.americanradiohistory.com

600 Low Power Transmitters

Figure 24.

TOP VIEW OF KEYER.

The keyer is built upon a 4" x 6" aluminum chassis. Layout of parts is not crowded. The audio

oscillator transformer is near the front panel below the controls, and the sealed high -speed relay

is in the center of the chassis with the 12AU7 tubes on either side. The two regulator tubes are

between the power transformer and the rear panel.

www.americanradiohistory.com

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www.americanradiohistory.com

CHAPTER TWENTY -NINE

The trend in design of transmitters for oper-

ation on the high frequency bands is toward

the use of a single high -level stage. The most

common and most flexible arrangement in-

cludes a compact bandswitching exciter unit,

with 15 to 100 watts output on all the high -

frequency bands, followed by a single power

amplifier stage. In many cases the exciter unit

is placed upon the operating table, with a co-

axial cable feeding the drive to the power am-

plifier, although some operators prefer to have

the exciter unit included in the main trans-

mitter housing.

This trend is a natural outgrowth of the in-

creasing importance of v -f -o operation on the

amateur bands. It is not practical to make a

quick change in the operating frequency of a

transmitter when a whole succession of stages

must be returned to resonance following the

frequency change. Another significant factor in

implementing the trend has been the wide ac-

ceptance of commercially produced 75 and

150 -watt transmitters. These units provide r -f

excitation and audio driving power for high -

level amplifiers running up to the 1000 -watt

power limit. The amplifiers shown in this

chapter may be easily driven by such exciters.

29 -1 Power Amplifier

Design

Choice of Either tetrode or triode tubes may

Tubes be used in high -frequency power

amplifiers. The choice is usually

dependent upon the amount of driving power

that is available for the power amplifier. If

a transmitter -exciter of 100 -watt power capa-

bility is at hand (such as the Heath TX-1)

it would be wise to employ a power ampli-

fier whose grid driving requirements fall in

the same range as the output power of the

exciter. Triode tubes running 1- kilowatt in-

put (plate modulated) generally require some

50 to 80 watts of grid driving power. Such

a requirement is easily met by the output

level of the 100 -watt transmitter which should

602

www.americanradiohistory.com

Design 603

TO ANTENNA

CIRCUIT

i

UNBALANCED COAXIAL

FEED SYSTEM

TO ANTENNA

CIRCUIT

t

BALANCED TWIN- LINE

,FED SYSTEM

Figure 1

LINK COUPLED OUTPUT CIRCUITS

FOR PUSH -PULL AMPLIFIERS

be employed as the exciter. Tetrode tubes

(such as the 4 -250A) require only 10 to

15 watts of actual drive from the exciter for

proper operation of the amplifier stage at 1-

kilowatt input. This means that the output

from the 100 -watt transmitter has to be cut

down to the 15 watt driving level. This is a

nuisance, as it requires the addition of swamp-

ing resistors to the output circuit of the trans-

mitter- exciter. The triode tubes, therefore,

would lend themselves to a much more con-

venient driving arrangement than would the

tetrode tubes, simply because their grid drive

requirements fall within the power output

range of the exciter unit.

On the other hand, if the transmitter- exciter

output level is of the order of 15 - 40 watts

(the Johnson Ranger, for example) sufficient

drive for triode tubes running 1- kilowatt input

would be lacking. Tetrode tubes requiring low

grid driving power would have to be employed

in a high -level stage, or smaller triode tubes re-

quiring modest grid drive and running 250

watts or so would have to be used.

Power Amplifier Either push -pull or single

Design- C,oice ended circuits may be em-

of Circuits ployed in the power ampli-

fier. Using modern tubes

and properly designed circuits, either type is

capable of high efficiency operation and low

harmonic output. Push -pull circuits, whether

using triode or tetrode tubes usually employ

link coupling between the amplifier stage and

the feed line running to the antenna or the an-

tenna tuner.

It is possible to use the link circuit in either

an unbalanced or balanced configuration, as

shown in figure 1, using unbalanced coaxial

line, or balanced twin -line.

Figure 2

CONVENTIONAL PUSH -PULL

AMPLIFIER CIRCUIT

The mechanical layout should be symmetrical

and the output coupling provision must be

evenly balanced with respect to the plate coil

C:- Approx. 1.5 1I.Id. per meter of wave-

length per section

Cr -Refer to plate tank capacitor design in

Chapter 11

Cr-May be S00 111.ld., 10,000 -volt type cer-

amic capacitor

NC -Max. usable capacitance should be great-

er, and min. capacitance less than rated

grid -plate capacity of tubes in amplifier.

50,-, greater air gap than C,.

R, -100 ohms, 20 watts. This resistor serves

as low Q r -f choke.

RFC, -All -band r -f choke suitable for plate

current of tubes

M, -M- Suitable meters for d -c grid and plate

currents

All low voltage .001 pfd. and .01 pfd. by -pass

capacitors are ceramic disc units (Centro -

lab DD or equiv.)

L, -50 -watt plug -in coil, center link

L-.- Plug -in coil, center link, of suitable power

rating.

Common technique is to employ plug -in

plate coils with the push -pull amplifier stage.

This necessitates some kind of opening for coil

changing purposes in the "electrically tight"

enclosure surrounding the amplifier stage. Care

must be used in the design and construction

of the door for this opening or leakage of har-

monics through the opening will result, with

the attendant TVI problems.

Single ended amplifiers may also employ

link -coupled output devices, although the trend

is to use pi- network circuits in conjunction

with single ended tetrode stages. A tapped or

otherwise variable tank coil may be used which

is adjustable from the front panel, eliminating

the necessity of plug -in coils and openings

into the shielded enclosure of the amplifier.

Pi- network circuits are becoming increasingly

popular as coaxial feed systems are coming

into use to couple the output circuits of trans-

mitters directly to the antenna.

www.americanradiohistory.com

604 H.F. Power Amplifiers T H E R A D I O

29 -2 Push -Pull Triode

Amplifiers

Figure 2 shows a basic push -pull triode

amplifier circuit. While variations in the meth-

od of applying plate and filament voltages and

bias are sometimes found, the basic circuit

remains the same in all amplifiers.

Filament Supply The amplifier filament trans-

former should be placed

right on the amplifier chassis in close proximity

to the tubes. Short filament leads are necessary

to prevent excessive voltage drop in the con-

necting leads, and also to prevent r -f pickup

in the filament circuit. Long filament leads

can often induce instability in an otherwise

stable amplifier circuit, especially if the leads

are exposed to the radiated field of the plate

circuit of the amplifier stage. The filament

voltage should be the correct value specified

by the tube manufacturer when measured at the

tube sockets. A filament transformer having a

tapped primary often will be found useful in

adjusting the filament voltage. When there is

a choice of having the filament voltage slight-

ly higher or slightly lower than normal, the

higher voltage is preferable. If the amplifier is

to be overloaded, a filament voltage slight-

ly higher than the rated value will give greater

tube life.

Filament bypass capacitors should be low in-

ternal inductance units of approximately .01

p,fd. A separate capacitor should be used for

each socket terminal. Lower values of capaci-

tance should be avoided to prevent spurious

resonances in the internal filament structure of

the tube. Use heavy, shielded filament leads for

low voltage drop and maximum circuit isola-

tion.

Plate Feed The series plate voltage feed

shown in figure 2 is the most

satisfactory method for push -pull stages. This

method of feed puts high voltage on the plate

tank coil, but since the r -f voltage on the coil

is in itself sufficient reason for protecting the

coil from accidental bodily contact, no addi-

tional protective arrangements are made neces-

sary by the use of series feed.

The insulation in the plate supply circuit

should be adequate for the voltages encoun-

tered. In general, the insulation should be

rated to withstand at least four times the max-

imum d-c plate voltage. For safety, the plate

meter should be placed in the cathode return

lead, since there is danger of voltage break-

down between a metal panel and the meter

movement at plate voltages much higher than

one thousand.

Grid Bias The recommended method of ob-

taining bias for c -w or plate mod-

ulated telephony is to use just sufficient fixed

bias to protect the tubes in the event of ex-

citation failure, and to obtain the rest by the

voltage drop caused by flow of rectified grid

current through a grid resistor. If desired, the

bias supply may be omitted for telephony if an

overload relay is incorporated in the plate cir-

cuit of the amplifier, the relay being adjusted

to trip immediately when excitation is re-

moved from the stage.

The grid resistor R. serves effectively as

an r -f choke in the grid circuit because the im-

pressed r -f voltage is low, and the Q of the

resistor is poor. No r -f choke need be used in

the grid bias return lead of the amplifier, other

than those necessary for harmonic suppression.

The bias supply may be built upon the am-

plifier chassis if care is taken to prevent r -f

from finding its way into the supply. Ample

shielding and lead filtering must be employed

for sufficient isolation.

The Grid Circuit As the power in the grid

circuit is much lower than

in the plate circuit, it is customary to use a

close- spaced split-stator grid capacitor with

sufficient capacitance for operation on the

lowest frequency band. A physically small ca-

pacitor has a greater ratio of maximum to

minimum capacitance, and it is possible to ob-

tain a unit that will be satisfactory on all bands

from 10 to 80 meters without the need for aux-

iliary padding capacitors. The rotor of the grid

capacitor is grounded, simplifying mounting of

the capacitor and providing circuit balance and

electrical symmetry. Grounding the rotor also

helps to retard v -h -f parasitics by by- passing

them to ground in the grid circuit. The L/C

ratio in the grid circuit should be fairly low,

and care should be taken that circuit reso-

nance is not reached with the grid capacitor at

minimum capacitance. That is a direct invita-

tion for instability and parasitic oscillations

in the stage. The grid coil may be wound of

no. 14 wire for driving powers of up to 100

watts. To restrict the field and thus aid in

neutralizing, the grid coil should be physical-

ly no larger than absolutely necessary.

Circuit Layout The most important consid-

eration in constructing a

push -pull amplifier is to maintain electrical

symmetry on both sides of the balanced cir-

www.americanradiohistory.com

HANDBOOK Design 605

cuit. Of utmost importance in maintaining elec-

trical balance is the control of stray capaci-

tance between each side of the circuit and

ground.

Large masses of metal placed near one side

of the grid or plate circuits can cause serious

unbalance, especially at the higher frequen-

cies, where the tank capacitance between one

side of the tuned circuit and ground is often

quite small in itself. Capacitive unbalance

most often occurs when a plate or grid coil is

located with one of its ends close to a metal

panel. The solution to this difficulty is to

mount the coil parallel to the panel to make

the capacitance to ground equal from each end

of the coil, or to place a grounded piece of

metal opposite the "free" end of the coil to

accomplish a capacity balance.

Whenever possible, the grid and plate coils

should be mounted at right angles to each

other, and should be separated far enough

apart to reduce coupling between them to a

minimum. Coupling between the grid and plate

coils will tend to make neutralization frequency

sensitive, and it will be necessary to readjust

the neutralizing capacitors of the stage when

changing bands.

All r -f leads should be made as short and

direct as possible. The leads from the tube

grids or plates should be connected directly

to their respective tank capacitors, and the

leads between the tank capacitors and coils

should be as heavy as the wire that is used

in the coils themselves. Plate and grid leads

to the tubes may be made of flexible tinned

braid or flat copper strip. Neutralizing leads

should run directly to the tube grids and plates

and should be separate from the grid and plate

leads to the tank circuits. Having a portion of

the plate or grid connections to their tank cir-

cuits serve as part of a neutralizing lead can

often result in amplifier instability at certain

operating frequencies.

Excitation In general it may be stated

Requirements that the overall power require-

ment for grid circuit excitation

to a push -pull triode amplifier is approximately

10 per cent of the amount of the power output

of the stage. Tetrodes require about 1 per cent

to 3 percent excitation, referred to the power

output of the stage. Excessive excitation to

pentodes or tetrodes will often result in, re-

duced power output and efficiency.

Push -Pull Symmetry is the secret of suc -

Amplifier cessful amplifier design. Shown

Construction in figure 3 is the top view of

a 350 watt push -pull all band

Figure 3

LAYOUT OF 350 -WATT PUSH -PULL

TRIODE AMPLIFIER

Two 811 -A tubes are employed in this circuit.

Plate tuning capacitor is at left of chassis,

with swinging -link type plug -in coil assembly

mounted above it. Rotor of split -stator capaci-

tor may be insulated from ground to increase

voltage breakdown rating of capacitor. Note

that pickup link is series -tuned to reduce cir-

cuit reactance. One corner of rotor plate of

series capacitor is bent so that capacitor

shorts itself out at maximum capacitance.

Grid circuit coil and capacitor are at right.

Center- linked plug -in coil is employed. Para-

sitic chokes are placed in grid leads adjacent

to the tube sockets, and tube filaments are

bypassed to ground with .01 ,.fd. ceramic

capacitors. Complete area above the chassis

is enclosed with perforated screen to reduce

radiation of r.f, energy.

amplifier employing 811 -A tubes. The circuit

corresponds to that shown in figure 2 except

that the 811 -A's are zero bias tubes. The bias

terminals of the circuit are therefore jumpered

together and no external bias supply is re-

quired at plate potentials less than 1300 volts.

All r -f components are mounted above

deck. The plate circuit tuning capacitor and

swinging link tank coil are to the left, with

the two disc -type neutralizing capacitors be-

tween the tank circuit and the tubes. At the

right of the chassis is the grid tank circuit.

Small parasitic chokes may be seen between

the tube sockets and the grid circuit. Plate

and grid meters are placed in the under -chassis

area where they are shielded from the r -f field

of the amplifier.

Larger triode tubes such as the 810 and

8000 make excellent r -f amplifiers at the kilo-

watt level, but care must be taken in ampli-

fier layout as the inter -electrode capacitance of

these tubes is quite high. One rube and one

neutralizing capacitor is placed on each side

of the tank circuit (figures 4 and 5) to permit

very short interconnecting leads. The relative

position of the tubes and capacitors is trans-

www.americanradiohistory.com

606 H.F. Power Amplifiers THE RADIO

Figure 4

UNIQUE CHASSIS LAYOUT PERMITS

SHORT LEADS IN KILOWATT

AMPLIFIER

Large size components required for high level

amplifier often complicate amplifier layout.

In this design, the plate tank capacitor sits

astride small chassis running lengthwise on

main chassis. Inductor is mounted to phenolic

plate atop capacitor. Variable link is panel

driven through right -angle gear drive. Plate

circuit is grounded by safety arm when panel

door is opened. Note that plate capacitor is

mounted on four TV -type capacitors which

serve to bypass unit, and also act as supports.

A small parasitic choke is visible next to the

grid terminal of the 810 tube.

posed on each side of the chassis, as shown in

the illustrations. The plate tank coil is mount-

ed parallel to the front panel of the amplifier

on a phenolic plate supported by the tuning

capacitor which sits atop a small chassis -type

box. The grid circuit tuning capacitor is located

within this box, as seen in figure 6. An ex-

ternal bias supply is required for proper ampli-

fier operation. Operating voltages may be de-

termined from the instruction sheets for the

particular tube to be employed.

Whenever the amplifier enclosure requires

a panel door for coil changing access it is wise

to place a power interlock on the door that

will turn off the high voltage supply whenever

the door is open!

Figure 5

LEFT -HAND VIEW OF KILOWATT

AMPLIFIER OF FIGURE 4

Above shielded meter box is the protective

"micro- switch" which opens the primary power

circuit when the panel door is not closed. Tube

sockets are recessed in the chassis so that

top of tube socket shells are about 1,2-inch

above chassis level. On right side of amplifier

(facing it from the rear) the tube socket is

nearest the panel, with the neutralizing ca-

pacitor behind it. On the opposite side, the

capacitor is nearest the panel with the tube

directly behind it. This layout transposition

produces very short neutralizing leads, since

connections may be made through the stator

of plate tuning capacitor.

29 -3 Push -Pull

Tetrode Amplifiers

Tetrode tubes may be employed in push -pull

amplifiers, although the modern trend is to

parallel operation of these tubes. A typical

circuit for push -pull operation is shown in

figure 7. The remarks concerning the filament

supply, plate feed, and grid bias in Section

29 -2 apply equally to tetrode stages. Because

of the high circuit gain of the tetrode ampli-

fier, extreme care must be taken to limit

interstage feedback to an absolute minimum.

Many amateurs have had bad luck with tet-

rode tubes and have been plagued with para-

sitics and spurious oscillations. It must be re-

membered with high gain tubes of this type

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HANDBOOT( P -P Tetrode Amplifier 607

Figure 6

UNDER CHASSIS VIEW OF

1- KILOWATT TRIODE AMPLIFIER

The grid circuit tuning capacitor and plate

circuit r -f choke are contained in the below

chassis enclosure formed by a small chassis

mounted at right angles to the front panel.

The bondswitch coil assembly for the grid

circuit is mounted on two brackets above this

cutout. A metal screen attached to the bottom

of the amplifier completes the TVI -proof

enclosure.

that almost full output can be obtained with

practically zero grid excitation. Any minute

amount of energy fed back from the plate cir-

cuit to the grid circuit can cause instability or

oscillation. Unless suitable precautions are in-

corporated in the electrical and mechanical de-

sign of the amplifier, this energy feedback will

inevitably occur.

Fortunately these precautions are simple.

The grid and filament circuits must be isolated

from the plate circuit. This is done by placing

these circuits in an "electrically tight" box.

All leads departing from this box are by -pass-

ed and filtered so that no r -f energy can pass

along the leads into the box. This restricts the

energy leakage path between the plate and grid

circuits to the residual plate -to -grid capacity

of the tetrode tubes. This capacity is of the

order of 0.25 µpfd. per tube, and under normal

conditions is sufficient to produce a highly re-

generative condition in the amplifier. Whether

or not the amplifier will actually break into os-

cillation is dependent upon circuit losses and

residual lead inductance of the stage. Suffice

to say that unless the tubes are actually neu-

tralized a condition exists that will lead to

circuit instability and oscillation under cer-

tain operating conditions. With luck, and a

Figure 7

CONVENTIONAL PUSH -PULL

TETRODE AMPLIFIER CIRCUIT

Push -pull amplifier uses many of the same

components required by triode tubes (see

figure 2). Screen supply is also required.

B- Blower for filament seals of tubes.

C. -Low internal inductance capacitor, .001

pfd., SKV. Centralab type 858S -1000.

NC -See text and figure 8.

PC- Parasitic choke. 50 ohm, 2 -watt compo-

sition resistor wound with 3 turns =12 e.

wire.

Note: Strap multiple screen terminals together

at socket with Je" copper ribbon. Attach

PC to center of strap.

heavily loaded plate circuit, one might be able

to use an un- neutralized push -pull tetrode am-

plifier stage and suffer no ill effects from the

residual grid -plate feedback of the tubes. In

fact, a minute amount of external feedback in

the power leads to the amplifier may just (by

chance) cancel out the inherent feedback of

the amplifier circuit. Such a condition, how-

ever, results in an amplifier that is not "re-

produceable." There is no guarantee that a du-

plicate amplifier will perform in the same, sta-

ble manner. This is the one, great reason that

many amateurs having built a tetrode amplifier

that "looks just like the one in the book" find

out to their sorrow that it does not "work like

the one in the book."

This borderline situation can easily be over-

come by the simple process of neutralizing the

high -gain tetrode tubes. Once this is done, and

the amplifier is tested for parasitic oscilla-

tions (and the oscillations eliminated if they

occur) the tetrode amplifier will perform in an

excellent manner on all bands. In a word, it

will be "reproduceable."

www.americanradiohistory.com

608 H.F. Power Amplifiers THE RADIO

1

Figure 8

REAR VIEW OF PUSH PULL

4 -250A AMPLIFIER

The neutralizing rods are mounted on ceramic

feedthrough insulators adjacent to each tube

socket. Low voltage power leads leave the

grid circuit compartment via Hypass capaci-

tors located on the lower left corner of the

chassis. A screen plate covers the rear of the

amplifier during operation. This plate was

removed for the photograph.

As a summation, three requirements must

be met for proper operation of tetrode tubes -

whether in a push -pull or parallel mode:

1. Complete isolation must be achieved be-

tween the grid and plate circuits.

2. The tubes must be neutralized.

3. The circuit must be parasitic -free.

Amplifier The push -pull tetrode ampli -

Construction fier should be built around

two "r -f tight" boxes for the

grid and plate circuits. A typical layout that has

proven very satisfactory is shown in figures 8

and 9. The amplifier is designed around a Bar-

ker & Williamson "butterfly" tuning capacitor.

The 4 -250A tetrode tubes are mounted at the

rear of the chassis on each side of the capaci-

tor. The base shells of the tubes are grounded

by spring clips, and short adjustable rods pro-

ject up beside each tube to act as neutralizing

capacitors. The leads to these rods are cross -

connected beneath the chassis and the rods

provide a small value of capacitance to the

plates of the tubes. This neutralization is nec-

essary when the tube is operated with high

1 ,,,

- .ita.... .:

Figure 9

UNDER CHASSIS VIEW OF

4 -250A AMPLIFIER

The bias supply for the amplifier is mounted

at the front of the chassis between the two

control shafts. A blower motor is mounted

beneath each tube socket. A screened plate

is placed on the bottom of the chassis to com-

plete the under -chassis shielding.

power gain and high screen voltage. As the

operating frequency of the tube is increased,

the inductance of the internal screen support

lead of the tube becomes an important part of

the screen ground return circuit. At some criti-

cal frequency (about 45 Mc. for the 4 -250A

tube) the screen lead inductance causes a

series resonant condition and the tube is said

to be "self- neutralized" at this frequency.

Above this frequency the screen of the tetrode

tube cannot be held at ground potential by the

usual screen by -pass capacitors. With normal

circuitry, the tetrode tube will have a tendency

to self -oscillate somewhere in the 120 Mc. to

160 Mc. region. Low capacity tetrodes that can

operate efficiently at such a high frequency are

capable of generating robust parasitic oscilla-

tions in this region while the operator is vain-

ly trying to get them operating at some lower

frequency. The solution is to introduce enough

loss in the circuit at the frequency of the

parasitic so as to render oscillation impos-

sible. This procedure has been followed in this

amplifier.

During a long series of experiments de-

signed to stabilize large tetrode tubes, it was

found that suppression circuits were most ef-

fective when inserted in the screen lead of the

www.americanradiohistory.com

HANDBOOK Pi- Network Amplifiers 609

tetrode. The screen, it seemed, would have r -f

potentials measuring into the thousands of volts

upon it during a period of parasitic oscillation.

By- passing the screen to ground with copper

strap connections and multiple by -pass capaci-

tors did little to decrease the amplitude of the

oscillation. Excellent parasitic suppression was

brought about by strapping the screen leads

of the 4 -250A socket together (figure 7) and

inserting a parasitic choke between the screen

terminal of the socket and the screen by -pass

capacitor.

After this was done, a very minor tendency

towards self -oscillation was noted at extreme-

ly high plate voltages. A small parasitic choke

in each grid lead of the 4 -250A tubes elimi-

nated this completely.

The neutralizing rods are mounted upon two

feedthrough insulators and cross -connected to

the 4 -250A control grids beneath the chassis.

These rods are threaded so that they may be

run up and down the insulator bolt for neu-

tralizing adjustment.

Because of the compact size of many tetrodes

it is necessary to cool the filament seals of the

tube with a blast of air. A small blower can be

mounted beneath the chassis to project cooling

air directly at the socket of the tube as shown

in figure 9.

Inductive Tuning of The plate tank circuit

Push -Pull Amplifiers of the push -pull ampli-

fier must have a low

impedance to ground at harmonic frequencies

to provide adequate harmonic suppression. The

usual split- stator tank capacitor, however, has

an uncommonly high impedance in the VHF

region wherein the interference -causing har-

monics lie. A push -pull vacuum -type capacitor

may be used as these units have very low in-

ternal inductance, but the cost of such a capac-

itor is quite high.

A novel solution to this problem is to em-

ploy a split stator capacitor made up of two

inexpensive fixed vacuum capacitors. Ampli-

fier adjustment can then best be accomplished

by inductive tuning of the plate tank coil as

Figure 10

INDUCTIVE TUNING MAY BE

EMPLOYED IN HIGH POWER

AMPLIFIER

Two fixed vacuum capacitors form split -stator

capacitance, providing very low inductance

ground path for plate circuit harmonics. Tun-

ing is accomplished by means of shorted,

single -turn link placed in center of tank

coil. Shorted link is made from 3k -inch section

cut from copper water pipe. Larger link out-

side of tank coil is antenna pick -up coil.

seen in figure 10. Two fixed vacuum capaci-

tors are mounted vertically upon the chassis

and the upper terminals are attached to the

plates of the amplifier tubes by means of low

impedance straps. Resonance is established by

rotation of a shorted copper loop located with-

in the amplifier tank coil. This loop is made

of a %8" long section of copper water pipe,

two inches in diameter. Approximate resonance

is established by varying the spacing between

the turns of the copper tubing tank coil. In-

ductive coupling is used between the tank coil

and the antenna circuit in the usual manner.

Sufficient range to enable the operator to cover

a complete high frequency band may be had

with this interesting tuning method.

29 -4 Tetrode Pi-

Network Amplifiers

The most popular amplifier today for both

commercial and amateur use is the pi- network

configuration shown in figure 11. This circuit

is especially suited to tetrode tubes, although

triode tubes may be used under certain circum-

stances.

A common form of pi- network amplifier is

shown in figure 11A. The pi circuit forms the

matching system between the plate of the am-

plifier tube and the low impedance, unbalanced

antenna circuit. The coil and input capacitor

www.americanradiohistory.com

610 H.F. Power Amplifiers THE RADIO

EXCITATION

- BIA' 15V.1,

LOW Z

OUTPUT

Figure 11

TYPICAL PI- NETWORK CONFIGURATIONS

A -Split grid circuit provides out -of -phase voltage for grid neutralization of tetrode tube. Rotary coil

is employed in plate circuit, with small, fixed auxiliary coil for 28 Mc. Multiple tuning grid tank TI

covers 3.5 - 30 Mc. without switching.

B- Tapped grid and plate inductors are used with "bridge type" neutralizing circuit for tetrode ampli-

fier stage. Vacuum tuning capacitor is used in input section of pi- network.

C- Untuned input circuit (resistance loaded) and plate inductor ganged with tuning capacitor comprise

simple amplifier configuration.

PCr, PC. -57 ohm, 2 watt composition resistor, wound with 3 turns o 18 c. wire.

www.americanradiohistory.com

HANDBOOK Pi- Network Amplifiers 611

of the pi may be varied to tune the circuit over

a 10 to 1 frequency range (usually 3.0 - 30

Mc.) . Operation over the 20 - 30 Mc. range

takes place when the variable slider on coil L2

is adjusted to short this coil out of the circuit.

Coil L. therefore comprises the tank inductance

for the highest portion of the operating range.

This coil has no taps or sliders and is con-

structed for the highest possible Q at the high

frequency end of the range. The adjustable

coil ( because of the variable tap and physical

construction) usually has a lower Q than that

of the fixed coil.

The degree of loading is controlled by ca-

pacitors C. and G. The amount of circuit ca-

pacity required at this point is inversely pro-

portional to the operating frequency and to

the impedance of the antenna circuit. A load-

ing capacitor range of 100 µµfd. to 2500

µµfd. is normally ample to cover the 3.5 - 30

Mc. range.

The pi circuit is usually shunt -fed to remove

the d.c. plate voltage from the coils and capa-

citors. The components are held at ground po-

tential by completing the circuit ground

through the choke RFC,. Great stress is placed

upon the plate circuit choke RFC:. This com-

ponent must be specially designed for this

mode of operation, having low inter -turn ca-

pacity and no spurious internal resonances

throughout the operating range of the ampli-

fier.

Parasitic suppression is accomplished by

means of chokes PC -I and PC -2 in the screen

and grid leads of the tetrode. Suitable values

for these chokes are given in the parts list of

figure 1L Effective parasitic suppression is de-

pendent to a large degree upon the choice of

screen bypass capacitor C2. This component

must have extremely low inductance through-

out the operating range of the amplifier and

well up into the VHF parasitic range. The ca-

pacitor must have a voltage rating equal to at

least twice the screen potential (four times the

screen potential for plate modulation). There

are practically no capacitors available that will

perform this difficult task. One satisfactory so-

lution is to allow the amplifier chassis to form

one plate of the screen capacitor. A "sandwich"

is built upon the chassis with a sheet of insu-

lating material of high dielectric constant and

a matching metal sheet which forms the screen

side of the capacitance. A capacitor of this

type has very low internal inductance but is

very bulky and takes up valuable space be-

neath the chassis. One suitable capacitor for

this position is the Centralab type 858S-1000,

B = BIAS SUPPLY

S SCREEN SUPPLY

P= PLATE SUPPLY

OPERATE

TUNE

"COMMON

MINUS"

LEAD

BY-

Figure 12

GROUNDED SCREEN GRID

CONFIGURATION PROVIDES HIGH

ORDER OF ISOLATION IN TETRODE

AMPLIFIER STAGE

A- Typical amplifier circuit has cathode re-

turn at ground potential. All circuits re-

turn to cathode.

B -All circuits return to cathode, but ground

point has been shifted to screen terminal

of tube. Operation of the circuit remains

the same, as potential differences between

elements of the tube are the same as in

circuit A.

C- Practical grounded screen circuit. "Com-

mon minus" lead returns to negative of

plate supply, which cannot be grounded.

Switch S: removes screen voltage for tune -

up purposes.

rated at 1000 µµfd. at 5000 volts. This com-

pact ceramic capacitor has relatively low inter-

nal inductance and may be mounted to the

chassis by a 6 -32 bolt. It is shown in various

amplifiers described in this chapter. Further

screen isolation may be provided by a shielded

power lead, isolated from the screen by a .001

pfd. ceramic capacitor and a 100 ohm car-

bon resistor.

Various forms of the basic pi- network am-

plifier are shown in figure 11. The A config-

uration employs the so- called "all- band" grid

tank circuit and a rotary pi- network coil in the

www.americanradiohistory.com

612 H.F. Power Amplifiers THE RADIO

plate circuit. The B circuit uses coil switching

in the grid circuit, bridge neutralization, and

a tapped pi- network coil with a vacuum tun-

ing capacitor. Figure 1 1C shows an interesting

circuit that is becoming more popular for

class ABI linear operation. A tetrode tube op-

erating under class ABI conditions draws no

grid current and requires no grid driving

power. Only r -f voltage is required for proper

operation. It is possible therefore to dispense

with the usual tuned grid circuit and neutraliz-

ing capacitor and in their place employ a sim-

ple load resistor in the grid circuit across

which the required excitation voltage may be

developed. This resistor can be of the order of

50 - 300 ohms, depending upon circuit re-

quirements. Considerable power must be dis-

sipated in the resistor to develop sufficicnt

grid swing, but driving power is often cheaper

to obtain than the cost of the usual grid cir-

cuit components. In addition, the low im-

pedance grid return removes the tendency to-

wards instability that is so common to the

circuits of figure 1IA and 11B. Neutralization

is not required of the circuit of figure 11C,

and in many cases parasitic suppression may

be omitted. The price that must be paid is the

additional excitation that is required to de-

velop operating voltage across grid resistor R..

The pi- network circuit of figure 11C is

interesting in that the rotary coil L. and the

plate tuning capacitor Ca are ganged together

by a gear train, enabling the circuit to be tuned

to resonance with one panel control instead of

the two required by the circuit of figure 11A.

Careful design of the rotary inductor will per-

mit the elimination of the auxiliary high fre-

quency coil L,, reducing the cost and complexi-

ty of the circuit.

The Grounded Screen For maximum shielding,

Configuration it is necessary to operate

the tetrode tube with the

screen at r.f. ground potential. As the screen

has a d.c. potential applied to it (in grid -

driven circuits), it must be bypassed to ground

to provide the necessary r.f. return. The by-

pass capacitor employed must perform effi-

ciently over a vast frequency spectrum that

includes the operating range, plus the region

of possible v.h.f. parasitic oscillations. This is

a large order, and the usual bypass capacitors

possess sufficient inductance to introduce re-

generation into the screen circuit, degrading

the grid -plate shielding to a marked degree.

Nonlinearity and self -oscillation can be the

result of this loss of circuit isolation. A solu-

tion to this problem is to eliminate the screen

bypass capacitor, grounding the screen ter-

minals of the tube by means of a low induc-

tance strap. Screen voltage is then applied to

the tube by grounding the positive terminal

of the screen supply, and "floating" the nega-

tive of the screen and bias supplies below

ground potential as shown in figure 12.

Meters are placed in the separate circuit

cathode return leads, and each meter reads

only the current flowing in that particular

circuit. Operation of this grounded screen cir-

cuit is normal in all respects, and it may be

applied to any form of grid- driven tetrode

amplifier with good results.

29 -5 Grounded -Grid

Amplifier Design

The grounded grid (g -g) amplifier has

achieved astounding popularity in recent years

as a high power linear stage for sideband ap-

plication. Various versions of this circuit are

illustrated in figure 13. In the basic circuit,

the control grid of the tube is at r.f. ground

potential and the exciting signal is applied to

the cathode by means of a tuned circuit. Since

the grid of the tube is grounded, it serves as

a shield between the input and output cir-

cuits, making neutralization unnecessary in

many instances. The very small plate to cathode

capacitance of most tubes permits a minimum

of intrastage coupling below 30 Mc. In addi-

tion, when zero bias triodes or tetrodes are

used, screen or bias supplies are not usually

required.

Feedthrough Power A portion of the exciting

power appears in the plate

circuit of the grounded grid (cathode driven)

amplifier and is termed feedthrough power.

In any amplifier of this type, whether it be

triode or tetrode, it is desirable to have a

large ratio of feedthrough power to peak grid

driving power. The feedthrough power acts

as a swamping resistor across the driving cir-

cuit to stabilize the effects of grid loading.

The ratio of feedthrough power to driving

power should be about 10 to 1 for best stage

linearity. The feedthrough power provides the

user with added output power he would not

obtain from a more conventional circuit. The

www.americanradiohistory.com

HANDBOOK Grounded -Grid Amplifier 613

E xC.

R.r. OUT

EXC.

EXC

©

R.r. OUT

Figure 13

THE GROUNDED -GRID AMPLIFIER

Widely used as a linear amplifier for sideband service, the grounded -grid circuit provides economy

and simplicity, in addition to a worthwhile reduction in intermodulation distortion. A -The basic

g -g amplifier employs tuned input circuit. B -A simplified circuit employs untuned r.f. choke in

cathode in place of the tuned circuit. Linearity and power output are inferior compared to circuit

of figure A. C- Simple high -C pi- network may be used to match output impedance of sideband

exciter to input impedance of grounded -grid stage. D- Parallel- tuned, High -C circuit may be

employed for bandswitching amplifier. Excitation tap is adjusted to provide low value of sm.,. on

exciter coaxial line.

driver stage for the grounded grid amplifier

must, of cous-se, supply the normal excitation

power plus the feedthrough power. Many com-

mercial sideband exciters have power output

capabilities of the order of 70 to 100 watts

and are thus well suited to drive high power

grounded grid linear amplifier stages whose

total excitation requirements fall within this

range.

Distortion Laboratory measurements made on

Products various tubes in the circuit of

figure 13A show that a distortion

reduction of the order of 5 to 10 decibels in

odd -order products can be obtained by operat-

ing the tube in grounded grid service as op-

posed to grid- driven service. The improvement

in distortion varies from tube type to tube

type, but some order of improvement is noted

for all tube types tested. Most amateur -type

transmitting tubes provide signal -to- distortion

ratios of -20 to -30 decibels at full output

in class ABI grid- driven operation. The ratio

increases to approximately -25 to -40 deci-

bels for class B grounded grid operation. Dis-

tortion improvement is substantial, but not as

great as might otherwise be assumed from

the large amount of feedback inherent in the

grounded grid circuit.

A simplified version of the grounded grid

amplifier is shown in figure 13B. This con-

figuration utilizes an untuned input circuit,

and is very popular as an inexpensive and

simplified form of the more sophisticated cir-

cuit of figure 13A. It has inherent limitations,

however, that should be recognized. In gen-

eral, slightly less power output and efficiency

is observed with the untuned cathode circuit,

odd -order distortion products run 4 to 6

decibels higher, and the circuit is harder to

drive and match to the exciter than is the

tuned cathode circuit of figure 13A. For maxi-

mum linearity and optimum operation, a cer-

tain amount of "flywheel" effect is required in

the cathode input that can only be supplied

by a high -C tuned circuit of some form.

Since the single ended class B grounded

grid linear amplifier draws grid current on

only one -half (or less) of the operating cycle,

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614 H.F. Power Amplifiers THE RADIO

-BIAS

EXC.

R.F.OUT.

Figure 14

TETRODE TUBES MAY BE USED IN GROUNDED -GRID AMPLIFIERS

A- Tetrodo tube may be used in cathode driven configuration, with bias and screen

voltages applied to elements which are at r.f. ground potential. B-Grid current of

grounded -grid tube is easily monitored by R -C network which lifts grid above ground

sufficiently to permit a millivoltmeter to indicate voltage drop across 1 -ohm resistor.

Meter is a 0 -1 d.c. milliameter in series with appropriate multiplier resistor.

the sideband exciter "sees" a low impedance

load during this time, and a very high im-

pedance load over the balance of the cycle.

Linearity of the exciter is thereby affected and

the distortion products of the exciter are en-

hanced. Thus, the driving signal is degraded

in the cathode circuit of the grounded grid

stage unless the unbalanced input impedance

can be modified in some fashion. A high -C

tuned circuit stores enough energy over the

operating r.f. cycle so that the exciter "sees"

a relatively constant load at all times. In addi-

tion, the tuned circuit may be tapped or other-

wise adjusted so that the standing wave ratio

on the coaxial line coupling the exciter to the

amplifier is relatively low. This is a great

advantage, particularly in the case of those

exciters having fixed -ratio pi- network output

circuits designed expressly for a 50 -ohm

termination.

Finally, it must be noted that removal of

the tuned cathode circuit breaks the amplifier

plate circuit return to the cathode, and r.f.

plate current pulses must return to the cathode

via the outer shield of the driver coaxial line

and back via the center conductor! Extreme

fluctuations in exciter loading, intermodula-

tion distortion, and TVI can be noticed by

changing the length of the cable between the

exciter and the grounded grid amplifier when

an untuned cathode input circuit is employed.

Grounded Grid Design features of the single -

Amplifier ended and push -pull ampli-

Construction fiers discussed previously

apply equally well to the

grounded grid stage. The g -g linear amplifier

may have either configuration, although the

majority of g -g stages are single- ended, as

push -pull offers no distinct advantages and

adds greatly to circuit complexity.

The cathode circuit of the amplifier is

resonated to the operating frequency by means

of a high -C tank (figure 13A). Resonance is

indicated by maximum grid current of the

stage. A low value of s.w.r. on the driver co-

axial line may be achieved by adjusting the

tap on the tuned circuit, or by varying the

capacitors of the pi- network (figure 13C).

Correct adjustments will produce minimum

s.w.r. and maximum amplifier grid current at

the same settings. The cathode tank should

have a Q of 2 or more.

The cathode circuit should be completely

shielded from the plate circuit. It is common

practice to mount the cathode components in

an "r.f. tight" box below the chassis of the

amplifier, and to place the plate circuit com-

ponents in a screened box above the chassis.

The grid (or screen) circuit of the tube is

operated at r.f. ground potential, or may have

d.c. voltage applied to it to determine the

operating parameters of the stage (figure

14A ). In either case, the r.f. path to ground

must be short, and have extremely low induc-

tance, otherwise the screening action of the

element will be impaired. The grid (and

screen) therefore, must be bypassed to ground

over a frequency range that includes the oper-

ating spectrum as well as the region of possible

v.h.f. parasitic oscillations. This is quite a

large order. The inherent inductance of the

usual bypass capacitor plus the length of

www.americanradiohistory.com

HANDBOOK Grounded -Grid Amplifier 615

element lead within the tube is often suffi-

cient to introduce enough regeneration into

the circuit to degrade the linearity of the am-

plifier at high signal levels even though the

instability is not great enough to cause para-

sitic oscillation. In addition, it is often desired

to "unground" the grounded screen or grid

sufficiently to permit a metering circuit to

be inserted.

One practical solution to these problems is

to shunt the tube element to ground by means

of a 1 -ohm composition resistor, bypassed with

a .01 µfd. ceramic disc capacitor. The voltage

drop caused by the flow of grid (or screen)

current through the resistor can easily be

measured by a milli -voltmeter whose scale

is calibrated in terms of element current

(figure 14B).

The plate circuit of the grounded grid am-

plifier is conventional, and either pi- network

or inductive coupling to the load may be used.

There is some evidence to support the belief

that intermodulation distortion products are

reduced by employing plate circuit Q's some-

what higher than normally used in class -C

amplifier design. A circuit Q of 15 or greater

is thus recommended for grounded grid

amplifier plate circuits.

Tuning the Since the input and output

Grounded Grid circuits of the grounded grid

Amplifier amplifier are in series, a cer-

tain proportion of driving

power appears in the output circuit. If full

excitation is applied to the stage and the out-

put circuit is opened, or the plate voltage re-

moved from the tube, practically all of the

driving power will be dissipated by the grid

of the tube. Overheating of this element will

quickly occur under these circumstances, fol-

lowed by damage to the tube. Full excitation

should therefore never be applied to a

grounded grid stage unless plate voltage is

applied beforehand, and the stage is loaded

to the antenna.

For best linearity, the output circuit of the

grounded grid stage should be overcoupled

so that power output drops about 2- percent

from maximum value. A simple output r.f.

voltmeter is indispensable for proper circuit

adjustment. Excessive grid current is a sign of

antenna undercoupling, and overcoupling is

indicated by a rapid drop in output power.

Proper grounded grid stage operation can be

determined by finding the optimum ratio be-

tween grid and plate current and by adjusting

the drive level and loading to maintain this

ratio. Many manufacturers now provide

grounded grid operation data for their tubes,

and the ratio of grid to plate current can be

determined from the data for each particular

tube.

Choice of Tubes

for G -G Service

distortion ratio

Not all tubes are suitable

for grounded grid service. In

addition, the signal- to -dis-

of the suitable tubes varies

over a wide range. Some of the best g -g per-

formers are the 811A, 813, 7094, 4 -125A,

4-250A, 4 -400A and 4- 1000A. In addition,

the 3 -400Z and 3 -1000Z triodes are specifi-

cally designed for low distortion, grounded

grid amplifier service. The older types 837

and 803 are used extensively for g -g operation

but are not recommended because of poor

signal -to- distortion ratios.

Certain types of tetrodes, exemplified by the

4 -65A, 4X150A, 4CX300A and 4CX1000A

should not be used as grounded grid ampli-

fiers unless grid bias and screen voltage are

applied to the elements of the tube (figure

14A). The internal structure of these tubes

permits unusually high values of grid current

to flow when true grounded grid circuitry is

used, and the tube may be easily damaged by

this mode of operation.

The efficiency of a typical grounded grid

amplifier runs between 55- and 65- percent,

indicating that the tube employed should have

plenty of plate dissipation. In general, the

p.e.p. input in watts to a tube operating in

grounded grid configuration can safely be

about 2.5 to 3 times the rated plate dissipa-

tion. Because of the relatively low average -to-

peak power of the human voice it is tempting

to push this ratio to a higher figure in order

to obtain more output from a given tube.

This action is unwise in that the odd -order

distortion products rise rapidly when the tube

is overloaded, and because no safety margin is

left for tuning errors or circuit adjustments.

Neutralization

of the G -G Stage

At some high frequency the

shielding action of the grid

of the g -g amplifier de-

teriorates. Neutralization may be necessary at

higher frequencies either because of the pre-

sence of inductance between the active grid

www.americanradiohistory.com

616 H.F. Power Amplifiers THE RADIO

EXC.

O

Figure 15

NEUTRALIZING CIRCUITS FOR GROUNDED -GRID STAGES

Neutralization of the g -g stage may be necessary at the higher frequencies. Energy fed

back in proper phase from plate to cathode is used to neutralize the unwanted energy

fed through the tube (A). Reactance placed in series with the grid return lead (B) will

accomplish the same result. The inductance L usually consists of the internal grid lead

of the tube, and capacitor C may be the grid bypass capacitor. A series resonant circuit

at the operating frequency is thus formed.

n. r. ouT.

element and the common returns of the input

and output circuit, or because of excessive

plate- cathode capacitance.

Neutralization, where required, may be ac-

complished by feeding out -of -phase energy

from the plate circuit to the filament circuit

(figure 15A) or by inserting a reactance in

series with the grid (figure 15B). For values

of plate- cathode capacitance normally encoun-

tered in tubes usable in g -g service, the resi-

dual inductance in the grid -ground path pro-

vides sufficient reactance, and in some cases

even series capacitance will be required. Typical

tube electrode capacitances are shown in figure

16A. These can be represented by an equiva-

lent star connection of three capacitors (figure

16B). If an inductance L is placed in series

with C, so that a resonant circuit is formed

(figure 16C), point O will be at ground

potential (figure 16D) . This will prevent the

transfer of energy from point P to point K,

since there now exists no common coupling

impedance. The determination of the value of

C, and L are shown in the drawing.

It is apparent that when the plate- cathode

capacitance of the tube is small as compared

to the plate -grid and the grid- cathode capaci-

tances, C, is a large value and the required

value of inductance L is small. In practical

cases the value of L is supplied by the tube

Figure 16

Tube electrode capacitances can be represented by an equivalent star connection of three capaci-

tors. If inductance is placed in series with C, so that a resonant circuit is formed (drawing C),

point 0 will be at ground potential.

CP-K

O

CC ° CG-P t CP-K X CG-K CG-K X C-G

C-K

t

L (2Rf)2 X Cc

www.americanradiohistory.com

HANDBOOK 350 Watt P.E.P. Amplifier 617

and lead inductance, and the grid to ground

impedance can be closely adjusted by proper

choice of the bias bypass capacitor (figure

15B). Below a certain frequency determined

by the physical geometry of the tube, neutral-

ization may be accomplished by adding induc-

tance to the grid return lead; above this

frequency it may be necessary to series tune

the circuit for minimum energy feedthrough

from cathode to plate. Most tubes are suffi-

ciently well screened so that series inductive

neutralization at the lower frequencies is un-

necessary, but series capacitance tuning of the

grid return lead may be required to prevent

oscillation at some parasitic frequency in the

v.h.f. range.

29 -6 A 350 Watt P.E.P.

Grounded -Grid

Amplifier

This section features an extremely stable,

five band, grounded grid linear amplifier for

sideband service. Employing the 7094 beam

power tube, the amplifier provides band -

switched operation on all bands between 80

and 10 meters. Power output is in excess of

200 watts, and third order distortion products

are better than -30 decibels below maximum

two -tone signal level.

High power gain, high efficiency, and low

distortion can be provided economically by a

high -s triode tube operating in grounded grid

configuration. Beam power tubes or tetrodes

(such as the 7094, 813 or 4 -250A) which

can be operated as high -µ triodes make excel-

lent grounded grid amplifiers. As a class B

linear amplifier in sideband service, a triode -

connected 7094 with forced air cooling of the

envelope can handle a conservative peak -

envelope -power input (p.e.p.) of 350 watts

with only I750 volts on the plate and zero

bias on the grids. For full input, a sideband

exciter capable of an output of only 15 watts

p.e.p. is required.

The amplifier, complete with power supply,

fits on a standard 101/2 -inch relay rack panel

which may be placed within a cabinet for use

directly on the operating table.

Amplifier The circuit of the amplifier and

Circuit power supply is shown in figure

17. The plate output circuit is a

bandswitching pi- network using two tapped

coils and a shorting switch. The position of the

taps are chosen to provide an operating Q of

15 or better on all bands with a 50 -ohm

antenna load. An auxiliary loading capacitor

is switched into the circuit in the 80 meter

position of the bandswitch. For low impedance

antennas (below 50 ohms) this capacitor

should be increased in value to 1000 ppfd.

The grid and screen of the 7094 tube are

at r.f. ground potential. The d.c. screen return

is to the cathode of the tube, and the panel

meter (M1) is switched so that it is pos-

sible to read either grid current or plate cur-

rent. The meter is a single- scale, 0 -300 d.c.

milliammeter. A lower range meter and ex-

ternal shunt were not considered necessary

because the normal peak grid current (80 ma.)

and peak plate current (200 ma.) can easily

be read on the same scale. A 1000 -ohm re-

sistor is connected between the positive ter-

minal of the meter and ground to prevent

high voltage from appearing at the cathode of

the tube in the event of switch failure.

An untuned input circuit is used in the

cathode for simplicity. An alternative tuned

input circuit is shown. Use of the tuned cir-

cuit will result in better linearity and lower

driving power requirements. If the tuned cir-

cuit is omitted, it may be necessary to "prune"

the coaxial line between the exciter and the

amplifier to achieve maximum driving voltage

in the cathode circuit. A circuit Q of two or

more is required in this tank.

The power supply is a conventional full

wave circuit with a choke input filter. Type

3B28 gas rectifier tubes are used in place of

866A's to eliminate the "hash" produced by

the mercury vapor tubes and to permit the

amplifier to be operated on its side during

tests and measurements. 866A's may be used

in place of the 3B28's without any circuit

changes provided the amplifier is always posi-

tioned so that the tubes are vertical.

The plate switch is connected in series with

the filament switch so that plate power cannot

be applied to the rectifier tubes until the

filament circuit is energized. Filaments should

be allowed to warm up for 30 seconds before

plate voltage is turned on.

www.americanradiohistory.com

618 H.F. Power Amplifiers THE RADIO

Amplifier Because of the simplicity of the

Construction circuit it is possible to con-

struct the amplifier and power

supply on a single 12" x 17" x 3" aluminum

chassis. The chassis is attached to a 101/2"

relay rack panel by means of two chassis

mounting brackets. The 7094 and plate tank

circuit components are enclosed in a 7" x 12"

x 91A " box made of 18 -gauge sheet aluminum.

The front of the box is mounted flush against

the rack panel, and both are drilled simul-

taneously for the shafts of the plate tuning

and loading capacitors and the bandswitch.

Half -inch wide flanges on the top and bottom

of the enclosure provide good r.f. contact to

the chassis and to the perforated aluminum

cover plate.

The small fan mounted on the rear wall of

the box provides forced -air cooling for the

7094. The air intake hole is 3- inches in dia-

meter and covered with perforated aluminum

stock.

Figure 17

SCHEMATIC, GROUNDED -GRID AMPLIFIER

C1 -100 ;LAM.,

(#155-10) 3 kv. Johnson 100E30

C2 -500 µpfd., 2 kv. Johnson 500E20

(#154-3)

C3 -See text. 500 ppfd. mica, 1250 volts

C4 -3 section b.c. capacitor, 1100 ppld.

Miller 2113

CH1 -8H, 250 ma. Thordarson 20056

F1 -5 amp. fuse, size 3AG

F2 -1 amp. fuse, size 3AG "slo -blo"

L1 -10 and 15 meter coil: 9 turns of 3/16 -

inch copper tubing, 2" inside diameter, 1/2-

inch spacing between turns. 10 meter tap is

4/2 turns from plate end al coil. 15 meter

tap is at junction between L1 and L2

L2-23 turns, B&W #3095 -1 inductor. 20 and

40 meter taps are 19 and 10 turns, respec-

tively, from the output end of coil. Number

12 wire, 2/2" diameter

L3 -18 turns # 16 wire, 1" diam., 3" long, 6

turns per inch (Air -Dux #806-T), 2.3 ph.

40 meter tap (1 ph) at 9 turns, 20 meter

top (0.5 ph) at 41/2 turns, I5 meter tap

(0.3 ph) at 3 turns, 10 meter tap (0.15 ph)

at 11/2 turns. All taps measured from

ground end of coil

P1,P2 -115 volt pilot lamp assembly

PC -1 turn of 1/2 -inch plate strap, 1/2-inch

diameter wound about three 100 ohms, 2

watt composition resistors in parallel

RFC1 -2.5 mh, 300 ma. National R -300,

placed between pins 4 and 7 of tube socket

RFC2 -0.225 mh, 800 ma. National R -175A

RFC3 -2.5 mh, 100 ma. National R -100

SI- Single pole, 5 position ceramic switch.

Ohmite #111 or equivalent

T,-6.3 volt @ 4 amp. Stancor P -4019

T2 -2.5 volt @ 10 a. Thordarson 21F02

T3- 2065 -0 -2065 volts @ 200 ma. (1750 v.,

d.c.) Stancor PT -8315

Blowers- Cooling motor and fan. Shaded -pole

induction motor, 2400 r.p.m. with 4- bladed

fan, 21/2" diem. Allied Radio Co., Chicago,

Ill. Part number 72P -715

www.americanradiohistory.com

HANDBOOK 350 Watt P.E.P. Amplifier 619

Figure 18B

REAR VIEW OF

AMPLIFIER

The power supply com-

ponents are grouped

about one end of the

chassis. R.f. input recep-

tacle and 115 -volt power

receptacle are placed on

rear apron of chassis.

Antenna receptacle is

mounted on rear wall of

shielded enclosure. Cera-

mic disc capacitor is

placed across meter leads

directly at terminals, and

leads are run in shielded

braid to under- chassis

area.

Figure 18A

350 WATT P.E.P.

AMPLIFIER AND

POWER SUPPLY

Using the 7094 beam

power tube, this com-

pact, grounded -grid am-

plifier may be driven to

full input with a 15 -watt

sideband exciter. The

complete amplifier and

power supply mount be-

hind a 101/2" relay rack

panel. Panel controls are

(I. to r.): meter switch,

plate tuning (above) and

filament switch (below),

bandswitch,ontennaload-

ing (above) and plate

switch (below).

www.americanradiohistory.com

620 H.F. Power Amplifiers T H E R A D I O

To hold r.f. loss to a minimum, connections

between the plate tank circuit components are

made of 1/4 -inch wide silver plated copper

strap. Your local jeweler can probably handle

the silver plating job for you.

A short length of RG -8 /U coaxial line is

used to make the connection between the load-

ing capacitor and the coaxial antenna recep-

tacle located on the rear of the enclosure. The

Figure 19

TOP VIEW OF LINEAR AMPLIFIER

The plate circuit is enclosed in r.f. -tight com-

partment bolted to the chassis deck. Power

supply choke is outside rear of compartment,

with plate transformer and 3B28 rectifier

tubes to the right. Enclosure is covered with

a piece of perforated aluminum plate for

maximum ventilation. The 7094 tube is at

center of enclosure, with 10 -15 meter coil be-

tween it and bandswitch. Loading capacitor is

to the left, with the 20 -80 meter coil directly

above it. Plate capacitor and r.f. choke are

at the right.

outer braid of the line is grounded at one end

to the frame of the capacitor and at the other

end to the shell of the coaxial receptacle.

A single 8 pfd. filter capacitor is too large

to fit beneath the chassis, so four 2 pfd. units

are wired in parallel to provide sufficient

capacity for good dynamic regulation. These

capacitors, together with the filament trans-

formers and bleeder resistors are placed in a

free corner of the under -chassis area.

Amplifier Tuning All wiring should be

checked before power is

applied to the amplifier.

The d.c. resistance to ground of the B -plus

line should be about 100,000 ohms. The am-

plifier is connected to the exciter and to the

antenna or to a 200 watt, 50 -ohm dummy

load. Mesh the plates of the loading capacitor

and place the meter switch in the plate current

and Adjustment

www.americanradiohistory.com

HANDBOOK 350 Watt P.E.P. Amplifier 621

position. Turn on the plate voltage and note

the resting plate current. It should be about

35 to 40 milliamperes. Apply a low level single

tone signal (carrier) to the amplifier and tune

the plate tank circuit to resonance. Switch the

meter to indicate grid current, and advance

the excitation level until the grid current read-

ing is about 50 milliamperes. Reduce the load-

ing capacitance, keeping the plate tank tuned,

until the plate current is approximately 100

milliamperes. Increase the excitation level to

obtain 80 mil:iamperes of grid current. Finally,

adjust loading and tuning to obtain a resonant

plate current of 200 milliamperes, keeping the

grid current at 80 milliamperes. Varying the

excitation level and the plate loading will per-

mit a 2.5 -to -1 ratio between plate and grid

current to be held. An exciter delivering less

than 15 watts may be used provided the load-

ing is sufficiently reduced to maintain the

same ratio between plate and grid current.

Under voice operation, meter readings will be

one -half (or slightly less) than the steady -

state readings indicated above. If a tuned

cathode circuit is used, it is resonated for

maximum grid current on each band.

Figure 20

UNDER -CHASSIS VIEW OF 7094

GROUNDED -GRID AMPLIFIER

Power supply components are grouped at left

side of chassis. The 0.01 pfd. ceramic bypass

capacitors are grouped about the socket to

keep all r.f. leads short. RFC, is mounted

directly on the socket between two pins. Fila-

ment transformer T, is at the right, with the

rectifier filament transformer mounted to the

rear wall of the chassis. Millen ceramic sockets

are used for the high -voltage rectifier tubes.

www.americanradiohistory.com

622 H.F. Power Amplifiers T H E R A D I O

29 -7 The "Tri- Bander"

Linear Amplifier

for 20 -15 -10

With the advent of the trap -tuned "tri -

bander" beam, many amateurs are concentrat-

ing their efforts on the 20, 15 and 10 meter

bands. In addition, low frequency operation

is often impractical for amateurs located on

small city lots and their activities must be con-

fined to the higher frequencies, This linear

amplifier is designed for the amateur whose

principal interest lies in the 14 -30 Mc. spec-

trum. An amplifier built specially for this

range can be made smaller and more inex-

pensively than one that covers the complete

3.5 -30 Mc. range.

The unit described in this section is a one

kilowatt p.e.p. class ABt cathode driven

linear amplifier using two compact, ceramic

4CX300A tubes. A novel and easily built

chassis- cabinet enclosure is employed, together

with the inexpensive model of the Eimac air -

system socket. The amplifier is small enough

so that it may be placed on the operating table

next to the sideband exciter and receiver. Pro-

visions are made for voice operation, or for

operating the s.s.b. exciter without the am-

plifier. At 2000 volts plate potential, third

order distortion products are better than -30

decibels below maximum signal input.

Amplifier Circuit A high perveance tube

such as the 4CX300A

cannot be used in a conventional class B

grounded grid circuit, as the element geometry

leads to high grid current and to destructive

values of grid dissipation. The distortion re-

duction characteristics of grounded grid cir-

cuitry, however, may be retained in an accept-

able cathode driven circuit, wherein grid and

screen operating potentials are applied to the

tube. The schematic of this amplifier which

makes use of such a circuit is illustrated in

figure 22. Two 4CX300A tubes are employed,

with the driving signal applied to the cathode

circuit as is done in the common grounded

grid configuration. Grid and screen elements

are at r.f. ground, while normal Class AB'

grid bias and screen potentials are applied to

the tubes. Under these conditions, the power

gain of the 4CX300.A is quite high; approxi-

r

Figure 21

TRI -BANDER LINEAR

AMPLIFIER FOR

10 -15 -20 METER

SIDEBAND

This one kilowatt p.e.p.

linear amplifier is de-

signed for those ama-

teurs interested in the

higher frequency DX

bands. Using two 4CX-

300A tubes, this compact

bandswitching unit is

ideally suited for exciters

having a p.c. p. output of

about 30 watts. Panel

controls are (I. to r.):

Screen meter, plate meter,

plate tuning, plate load-

ing. On the left is the

mode switch, Si; and on

the right is the band

switch, S2. Amplifier is

mounted on four rubber

"feet" so that cooling

air may be withdrawn

from under the cabinet.

Geared tuning dials,

switch knobs, and plate

bandswitch are salvaged

from surplus "TU" tun-

ing drawers from BC-

191;375 transmitter.

www.americanradiohistory.com

HANDBOOK Tri -Bander Amplifier 623

B+2 NV. 0

SIA

, RFC.

sew*1 0= u =

2 RFC1

ANT RELAYQ ITT

B+ SCR. 3 RFC,

T T

4 RFCI

-BIAS _ _ ,OOK

2K

RFC,

500 000 500 o

Hy -r T,0 KV °RFC

-_ll 1_

E AC.

INPUT RY R.P.

OUTPUT

RFC3

50 250

C, 52 C2

RFCI

115 VA, i25t) F0rT

Sie

MA>

RFC3

X

o, AT EACH

SOCKET

C

1

E

AU*.^ B RFC,

CONTROL o 1rs

TUNE T RFC,

CONTROL 6001

O B C

CONTROL T

CND. 9 -11I

4

MS-1

SWITCH 1

1. Off

2 AUX.

3. PI LS.

4. TUNE

5. OPERATE

NOTE : 1. CAPACITORS -C-ARE .0011/F., 400 V,

DISC CERAMIC.

2. ONE CAPACITOR "C" ON EACH GRID

AND SCREEN SOCKET TERMINAL (1.1-)

3.- INDICATES PEEDTHRU CAPACITOR

Figure 22

SCHEMATIC, TRI -BANDER LINEAR AMPLIFIER

C1 -50 III,fd., 3 kv. Johnson

155 -8 (50F30), 0.075" spacing

C2 -250 /IIId., 2 kv. Johnson

155 -6 (250F20), 0.045" spac-

ing

L, -10 mete,. section: 31/2 turns,

3/16" copper tubing, wound

11/4" i.d. Adjust length to

resonate with C1 25% meshed.

15 -20 meter section: S turns,,

Vs" copper tubing, wound

21/4" i.d. 15 meter tap 3

turns from "cold" (output)

end

M, -0 -5C d.c. milliammeter. Re-

calibrated to -20 to +30 ma.

M2 -0 -S00 d.c. milliammeter

MS1 -SPST lever -type "Micro -

switch"

PC- Parasitic choke. Two turns

«12, 1/2 -inch diem., wound

about 47 ohm, 2 watt com-

position resistor

RFC1 -VHF choke. Ohmite Z -144

RFC2 -44 ph., S00 ma., Ohmite

Z -14

RFC3 -2.5 mh, 300 ma. Na-

tional R -300

RY1 -DPDT, 115 volt coil, an-

tenna relay. Advance AM-

2C-115VA

Si-Two pole, 5 position pro-

gressively shorting switch. Two

Centralab « P -1 decks, with

P -121 Index Assembly

S2 -- Single -pole, 5 position cera-

mic switch from surplus "TU"

tuning unit, or Centralab

«2550

T,-6.3 volt at 6 amp. Stancor

P -6456. Adjust primary resis-

tor to deliver 6.0 volts at

tube sockets under load

Blower -35 cubic feet per min-

ute. 6000 r.p.m., 115 volts

a.c. Ripley «8445 -E

Feedthrough capacitors -Each of

the eight control leads, plus

the two leads to the relay

coil pass through 0.001 Aid.

ceramic feedthrough capaci-

tors. Centralab type FT -1000

Sockets: Eimac 5K -760 air socket.

Place one 0.001 Aid., 600

volt ceramic capacitor from

each screen terminal to

ground

mately 30 watts p.e.p. drive being required

for full output.

The amplifier plate circuit is a simple three

band pi- network, designed for a circuit Q of

15. As the low frequency bands are not in-

cluded, only two small self- supporting air

wound coils are required. In addition, the size

of the pi- network loading capacitance is con-

siderably smaller than a capacitor necessary for

all band operation.

www.americanradiohistory.com

624 H.F. Power Amplifiers T H E R A D I O

The amplifier is controlled by a two deck

progressively- shorting switch (S1) that re-

motely controls the auxiliary equipment and

provides the operator with a choice of "tune"

or "operate" modes. All control and low volt-

age power leads are suitably filtered by L -C

networks to suppress radiation of TVI-

producing harmonics.

The "Tri- bander" linear amplifier construc-

tion is novel in that no regular chassis deck is

employed. The amplifier is built within an

enclosure made up of two aluminum chassis,

each measuring 10" x 14" x 3 ". One chassis

is inverted and serves as a pan within which

the components are mounted. The second

chassis is placed atop the first and serves as a

top shield cover. This chassis assembly is

hinged along the rear edge, and opens up much

in the manner of a suitcase. A single -piece

front panel made of aluminum is fixed to the

lower chassis. The front apron of the top

section is cut away to provide clearance for

the meters, switches and capacitors. When the

top section is closed, the cabinet is sealed by

a strip of finger stock that runs around the

inside edges of the lower chassis box. A length

of "piano -type" hinge fastens the rear edges

of the two chassis together, and the enclosure

halves are held in place by five panel bolts

which screw into nut plates riveted to the lip

of the lid, or top section.

An aluminum partition divides the interior

of the enclosure into two compartments

(figure 23). The smaller compartment con-

tains the blower motor, filament transformer,

panel meters, auxiliary control relay, function

switch, and power lead filters. The larger com-

partment contains the two 4CX300A tubes,

the plate circuit pi- network components and

the antenna relay. The partition is shaped to

fit around the housing holding the tetrode

tube sockets. As the standard air system socket

with built -in screen bypass capacitor is both

expensive and bulky, the smaller phenolic

socket having no screen capacitor was used as

an inexpensive substitute. Two of these sockets

will mount atop an oscillator shield can taken

from a defunct surplus "Command" trans-

mitter. The can makes an inexpensive and r.f.-

tight shield for the grid and cathode compon-

ents, and is mounted directly to the bottom

chassis "pan." The pi- network capacitors and

bandswitch are panel mounted, and the re-

maining compartment area is taken up by the

plate coils, r.f. choke, and the plate blocking

capacitors. Antenna relay RY, is mounted

within a small aluminum shield box placed at

the back of the compartment.

Transmitter wiring is simple and straight-

forward. All connections in the meter com-

partment are made with unshielded wire. The

relay leads pass through the internal shield

partition via high frequency feedthrough

capacitors, and the exciter switching leads to

the contacts of the relay pass through short

lengths of RG -58/U coaxial line. The outer

braided conductor of the line is soldered to a

u.h.f. -type "hood" (Amphenol type 83 -1H) to

ensure r.f. -tight connections where the cables

enter and leave the amplifier compartment.

The three ceramic capacitors that make up

the plate blocking unit are mounted atop the

plate r.f. choke, and are fastened to the main

tuning capacitor by means of an aluminum

strap visible in figure 23.

Connection is made to the anode of each

tube by means of a 1/2 -inch wide copper strap

encircling the air cooler structure. Air is

drawn through 1/4 -inch holes in the bottom

pan by the blower, forced into the grid com-

partment, circulated upward through the tube

socket and cooling anode, and exhausted via

1/4-inch vent holes drilled in the top lid of

the enclosure. The blower motor goes on when-

ever the filaments of the tubes are lit.

Transmitter Control Switch S1 controls the

Circuits and

Power Supply transmitter and auxiliary

equipment. All circuits

are off in the first posi-

tion. In the second position, an auxiliary cir-

cuit is completed which can turn on the station

receiver or sideband exciter. The third position

turns on the amplifier tube filaments and

energizes the blower motor to cool the tubes.

Cut -off bias is applied to the tubes to eliminate

diode noise often noticed in standby operation.

The fourth position applies full plate voltage

and reduced screen voltage to the amplifier for

tuning operations, and the fifth switch position

applies full screen voltage. Cut -off bias is re-

moved by the voice -actuated relay in the power

supply. Screen and plate currents are con-

tinually monitored by the two panel meters.

The screen meter is recalibrated to have an

www.americanradiohistory.com

HANDBOOK Tri -Bander Amplifier 625

Figure 23

INTERIOR VIEW OF LINEAR AMPLIFIER

The r.l. components are contained in the compartment to the right of the shield par-

tition. Antenna relay RYA is placed in small aluminum box mounted to rear wall of

cabinet directly behind antenna loading capacitor. The two 4CX300A tube sockets are

mounted on top of aluminum shield can taken from oscillator coil section of surplus

"command" transmitter. Micro- switch on partition removes high voltage when cover is

opened. Midget relay adjacent to switch is added for auxiliary control circuits and is

not required. At extreme left rear cre feedthrough capacitors mounted on aluminum

plcte, with r.l. chokes beneath them. Filament transformer is in corner of compartment,

in back of mode selector switch. Pi- network components are at right, with three plate

blocking capacitors mounted to aluminum strip supported by plate tank capacitor.

elevated zero point and reads -20 to +30

milliamperes. Under certain conditions, nega-

tive screen current can flow and it is important

to monitor this sensitive indicator of amplifier

operation.

The power supply schematic is shown in

figure 24. The high voltage supply uses 3B28

"hash -free" gas rectifier tubes and provides

2000 volts d.c. at 500 ma. and regulated 360

volts at 30 milliamperes. "Jumpers" in the

base of the regulator tubes are wired in series

with the primary relay circuit so that the

supply cannot be energized unless the tubes

are in their sockets. A smaller half -wave semi-

conductor supply provides operating and cut-

off bias for the amplifier. The bias relay may

be actuated by the voice circuit of the exciter

to drop the bias to the correct amount during

the time the voice circuit is energized.

www.americanradiohistory.com

626 H.F. Power Amplifiers THE RADIO

n5V.'NJ

T,

CH1 B+ 211V.

RY 2A

6-1-SCR. (360 V.)

(s3)

0.15 12

411V 311V

T3

p 100

p 2W

20 I °

SR + 100

ADJ.

BIAS

1011

4W

. 7 4 11

2W

20

Roo

R105

GN D.

(a9)

VR105

GRID

METER

RESISTOR

RY4 RY2

VOX - MANUAL

SWITCH 6--ce- JUMPERS IN VR TUBES

BIAS

(a4)

01,5 V. WI

ANT. RELAY

(ell

RY2B (a21

-O 15 V.*2

TO

R

PR

ECE

1.

I

C11TVER (a6)

0 -A U X CONTROL

TUNE' CONTROL

(a7)

O OPERATE CONTROL

(.6)

3 7 3 7 3 7

TO VOX

CONTROL RELAY

CONTACTS

CH1-6 h. at S00 ma. Chicago R -65

P,, P3 -115 volt pilot lamp and receptacle

RY2 -DPST, 11S volt coil. Potter -Brumfield

MRSA, 115 volt a.c.

RY3 -SPST, 115 volt coil, 20 amp. contact.

Potter -Brumfield PR3AY, 115 volt, a.c.

RY4 - -DPDT, 115 volt coil. Potter -Brumfield

MR11A, 115 volt a.c.

SR- Selenium rectifier, S00 ma. Sarkes- Tarzian

M -500

T1 -2.5 volts at 10 a., 10 kv. insulation.

Chicago F -210H

T2- 2900 -2300 volts each side of c.t. at 500

ma. 115 -230 volt primary. Chicago P -2126

T3 -125 v., 50 ma. Stancor PA -8421

Extra contact set of RY4 is placed in series

with antenna relay control lead (17 2) and

RY2R contacts to actuate antenna relay R Y,

(figure 22) by VOX circuit.

Figure 24

SCHEMATIC, LINEAR AMPLIFIER POWER SUPPLY

Transmitter The only initial adjustment is

Adjustment to set the operating bias level

and Tuning by means of the potentiometer.

Initially, the arm should be set

at the high potential end of the potentiometer

to apply full bias to the tubes. The filaments

and blower are turned on, and the high volt-

age and bias supply energized. Using a volt-

meter, the potentiometer should be set to

provide about -60 volts on the arm. The

voice relay is energized dropping the cut -off

bias out and the potentiometer is carefully

reset to provide a static plate current of 200

ma. as read on the meter. Indicated screen cur-

rent (bleeder current) should be about 22

ma. When the voice relay drops out, the plate

current should fall to zero.

The amplifier is now fed a small exciting

signal (single tone) and tuned and loaded for

a maximum plate current of 500 milliamperes.

Screen current should now be approximately

30 ma. (This is a total of screen and bleeder

current.) The output coupling is now increased

slightly so that r.f. output (as read on an r.f.

ammeter, or output voltmeter) drops about

2 percent. Maximum linearity is obtained when

the amplifier is slightly overcoupled. Under

voice conditions, plate current peaks should

reach approximately 250 ma., as read on the

meter. No grid current should be read on a

0 -1 d.c. milliammeter placed across the grid

current terminals in the power supply. Any

flicker of grid current indicates the amplifier

is being overdriven, with a consequent severe

www.americanradiohistory.com

HANDBOOK 813 Linear Amplifier 627

increase in distortion. Under voice conditions,

indicated screen current will be relatively con-

stant, as actual current drawn by the screen

of the tubes will be less than + or - 10 ma.,

and this small value is swamped out by the

bleeder current, which is constant at 22 ma.

Low values of screen meter current (indicating

that the tubes are drawing negative current)

indicates excessive loading; high values of

screen current indicate insufficient plate circuit

loading.

Never apply excitation to this (or any

other) grounded grid amplifier without all

operating potentials applied to the tubes.

Figure 25

THE 813 GROUNDED -GRID LINEAR

AMPLIFIER

Two 813's are used in this simple and effec-

tive linear amplifier. Built on a 101'2 -inch

rock panel, the amplifier may be placed in a

metal cabinet for desktop operation. Capable

of operation on all amateur bands between

80 and 10 meters, this unit may be driven by

the popular 75 to 100 watt sideband exciters.

Panel controls are (I. to r.): bandswitch, plate

tuning (top) and antenna loading (bottom),

meter switch (top) and bias control (bottom).

Front bushing of linkage shalt for switch S2

passes through panel between tuning and

loading controls and is camouflaged with

small knob.

29 -8 An 813 Grounded -

Grid Linear Amplifier

The popular amateur s.s.b. transmitters in

the 75- to 100 -watt power class provide a

ready -made exciter when the time comes to

add a more powerful final amplifier to the

amateur station. Because tetrodes have low

power drive requirements, a power dissipating

device must be employed when these tubes are

driven from a 100 -watt class transmitter. A

suitable dissipation device is usually fragile,

expensive, and difficult to construct. In addi-

tion, the tetrode tube requires bias and screen

power supplies which are bulky and expensive.

A grounded grid amplifier circuit provides

a satisfactory solution to these problems as no

power dissipating device is required, and

screen and bias supplies may be eliminated.

Certain tetrodes and pentodes operate well as

zero -bias, grounded grid triodes, and the 813

is one of these. This tube operates efficiently

in class B grounded grid service at plate poten-

www.americanradiohistory.com

628 H.F. Power Amplifiers THE RADIO

325 CAI

ALTERNATIVE TUNED

CATHODE CIRCUIT

.ó1

= 1.2Kv

813 813 EA.

.001

57/V LI

500 7L15_2

10 KV C1

RFC4 50-760

Tool

CAPAC ITOR

FIL. CET.

ADJ. B /AS

R,

II RFC2

I

R r.

OUT.

RFC

1500

C2

L. t

50Ó 0

KK V

13+ 14.V.

.01 .01

Io O

2

ro VOX

RELAY

3 4

115 V.1,

TS2

ITS,

CND

NOTES

1. 52 CLOSED IN 80 t 40M POSITIONS. OPEN /1120,

154 10 M. POSITIONS OF BANOSW/rCN SI.

2. ALL 01 CAPACITORS ARE BOO V CERAMIC UNITS.

3. JUMPER TERMINALS / B 2 ON TERMINAL STRIP

ro REMOVE BLOCK /NG B /AS.

Figure 26

SCHEMATIC, 813 LINEAR AMPLIFIER

B- Tube -cooling motor and fan. Shaded pole

induction motor, 2400 r.p.m., with 4 -blade

fan, 21/2" diom. Allied Radio Co., Chicago.

Part 5 72P715

C1- Two- section variable capacitor. Front sec-

tion (added for 40 -80 meters): 28 -160 ppfd.

Rear section: 7 -50 ppfd. 0.125" spacing.

Barker & Williamson. A conventional split -

stator capacitor may be substituted. John-

son 1;l54 -3 (100E045) is recommended. In-

stall the switch between the stators, on the

studs supporting the stator plates at the

middle of the capacitor. Change length of

linkage to fit new layout.

C2 -1500 ppfd., 0.03" spacing. Barker & Wil-

liamson 51241. A four section, b.c. -type

variable capacitor (J. W. Miller ,V.2l04) with

sections in parallel may be substituted.

C3 -1260 µµId. Three section b.c. -type capa-

citor (J. W. Miller 52113) with sections in

parallel

C4 -325 µµId., 0.024" spacing. Hammarlund

MC -325M

L, -10.5 µh. transmitting inductance. Barker

& Williamson 850A. Air -Dux 5195 -2 coil

may be substituted. This coil should be

trimmed and topped to resonate as follows:

80 meters, 210 ppfd.; 40 meters, 105 ppfd.;

20 meters, S2 ppfd.; 15 meters, 30 µpfd.;

10 meters, 30 ppfd. Above capacities in-

clude output capacitance of tubes

L3 -10 meter section: 0.44 ph. S turns 512

e., 1" diam., 1" long, space -wound S turns

per inch. Tapped section: 4.2 ph. 17 turns

5 16 tinned, 11/2" diom., 21/4" long, space -

wound 8 turns per inch. Tapped 2 (21 Mc.),

4 (14 Mc.), and 10 (7 Mc.) turns from 10

meter end of coil. B&W 53018 miniductor

M, -0 -1 d.c. milliammeter

RFC, -0.5 mh., 300 ma. choke. National

R -300

RFC2 -15 ampere filament choke. B&W type

FC -15

RFC3 -200 ph. choke. B&W type 800, or

National R -175A

RFC4,3-1 mh., 300 ma. choke. National

R -300

S, -Part of L,. An Ohmite type 111 -5, S

position ceramic switch may be used with

Air -Dux coil. Switch should be mounted on

an insulated brocket and driven with an

insulated coupling

S2- Special switch. See text for details

S3- Single -pole, S position ceramic. Centralab

52500

SR -130 volt, 75 ma., replacement -type se-

lenium rectifier

T, -10 volt, 10 amperes. Thordarson 21F19

T2 -115 volt, 50 ma. Stancor PA -8421

711,2-Insulated terminal strips. Cinch -Jones

Knobs -B&W 5901 (11/2" diam., 3 req.) B&W

»903 (11/16" diam., 3 req.)

www.americanradiohistory.com

HANDBOOK 813 Linear Amplifier 629

Figure 27

LEFT REAR VIEW OF AMPLIFIER

A I á -inch thick sheet of aluminum 13 inches by 17 inches in size forms the main chassis and is

fastened to the panel with chassis support brackets. Connection between plate r.f. choke, blocking

capacitors, plate tuning capacitor and plate coil are made with copper strap. Plate leads from

tubes to strap are made with =10 flexible braided wire. Coaxial r.f. input receptacle is next to

11S -volt line cord, and antenna receptacle is mounted on angle bracket at end of sub -chassis.

Switch S, is at rear of bandswitching inductor.

bals up to 3000 volts. Two 813's in parallel

at 2500 volts will provide a p.e.p. input of

1500 watts (750 watts, single tone) provided

cooling air is circulated about the tubes. At

3000 volts, a p.e.p. input of 2000 watts (1000

watts, single tone) may be run but the plate

dissipation of the tubes exceeds the recom-

mended maximum figure. If plenty of cooling

air is used, this does not seem to shorten tube

life. Under these two operating conditions,

third order distortion products are better than

-30 decibels below maximum power level.

Amplifier Circuit The circuit of this linear

amplifier is shown in

figure 26. The basic amplifier employs an un-

tuned cathode input circuit for simplicity and

low cost, although an alternative tuned input

configuration is shown. Improved intermodu-

lation distortion suppression and less driving

power can be gained with the use of the tuned

circuit.

The screen and beam -forming plates of the

813's are grounded directly at the socket. The

www.americanradiohistory.com

630 H.F. Power Amplifiers THE RADIO

grids are bypassed to ground and receive a

small amount of negative bias from the built -

in bias supply. The exact bias level may be

set by the potentiometer. In addition, when

the connection between terminals 1 and 2 on

the terminal strip is broken, the tubes are

biased to cut -off to eliminate troublesome diode

standby noise. When these terminals are

shorted by the contacts of the voice relay, the

bias is reduced to the operating value deter-

mined by the setting of the potentiometer.

Separate metering of current in the grid and

plate circuits is accomplished by switching a

single meter (M) across shunt resistors. The

0 -1 d.c. milliammeter is converted into a low -

range voltmeter by the addition of the 1.2K

series multiplier resistor, and the voltage drop

across grid and plate shunt resistors is mea-

sured. In the grid position, the meter reads

0 -100 ma., and in the plate position it reads

0 -500 ma.

A pi- network plate tank circuit is employed.

Optimum plate load impedance for this cir-

cuit is about 5000 ohms, and the Q should

be held to a figure of 15 or better. These

requirements may be met with the specified

components, or with less expensive substitutes,

as outlined in the parts list.

High voltage is applied to the parallel -con-

nected 813's through the plate r.f. choke. Three

blocking capacitors in parallel keep high volt-

age from reaching the pi- network plate tank

circuit. A tapped coil and two section tuning

capacitor provide nearly optimum L/C ratio

on all amateur bands from 80 to 10 meters.

Only one section of the tuning capacitor is in

the circuit on the 10, 15 and 20 meter bands

when the automatic switch S2 is open. Both

capacitor sections are in parallel on 40 and 80

meters where greater maximum tuning capaci-

tance is required, S2 being closed by a me-

chanical linkage from the main bandswitch, S,.

A large variable pi- network output capacitor

(1500 µµfd.) eliminates the need for several

fixed capacitors and a tap switch to add them

to the circuit as needed. The output circuit

will match load impedances in the range of

50 to 75 ohms having an s.w.r. of 2/1 or less.

Figure 28

RIGHT REAR VIEW

OF AMPLIFIER

Main tuning capacitors

are mounted on vertical

end -brackets made of

!Vs -inch sheet aluminum.

The copper nngle brac-

kets on the plate capaci-

tor plus U- shaped bracket

on switch linkage form

Sy. In foreground, moun-

ted on sub -chassis are

the filament transformer,

bias supply filter capaci-

tor, high voltage ter-

minal, and plate r.f.

choke. Bottom chassis

plate is drilled beneath

fan to permit cooling air

to be drawn into sub -

chassis area.

www.americanradiohistory.com

HANDBOOK 813 Linear Amplifier 631

REAR SUPPORT

PLATE IOR CI

AND C2-AY 7Y}

ALUMINUM

FRONT PLATE

ON LI

TOP VIEW

SO

PANEL

} Y f !BASS STRIP LONG

E- LUCITE 2 +LONG

U -CLIP FORMED FROM

FROM it2Y=r SPRING

!BASS 1 LONG

FRONT SUPPORT

FOR CI C2

2 7 sRAS3 STRIP

If LONG

PANEL

POSITION OF LINRAGC

IN 14,21 ANC 211-MC

POSITIONS OF LI"

Alt

Vir m;p=A=1R>=::

POSITION O \

LINKAGE IM 3 ].]\

AND 1 MC POSITIONS

OF LIST

FRONT VIEW

Figure 29

DETAIL DRAWING OF

SWITCH S, LINKAGE

Three ! /e" x /2" brass strips, sol-

dered to brass shaft couplings

make up the linkage arms. Plastic

arm supports U -clip which closes

circuit between copper angle brac-

kets mounted on main tuning capa-

citor in 80 and 40 meter positions

of bandswitch.

Amplifier Amplifier construction is quite

Construction simple due to the utilization of

standard, readily available com-

ponents. The main chassis is a 14" x 17" x

1,á -inch thick sheet of aluminum fastened with

its bottom surface !48-inch above the lower

edge of a 101/2" x 19" aluminum relay rack

panel. Only the pi- network components, meter,

and meter switch are mounted to the main

chassis, the remaining components being assem-

bled on the 6" x 11" x 21/2" aluminum sub -

chassis. The photographs and drawings illus-

trate the placement of the major components.

The end plates of the tuning capacitors are

fastened to / -inch aluminum brackets seven

inches high and four inches wide (figure 30).

The shaft on which the linkage for switch S2

is supported also runs between these brackets.

The parts of this linkage, and assembly details

are shown in figure 29. A U- shaped clip, made

from spring brass or phosphor bronze, com-

pletes the connection between copper angle

brackets fastened to the two stator sections on

the main tuning capacitor when the bandswitch

is in the 80 and 40 meter positions. The short,

rotary arm on the bandswitch is adjusted so

that it engages the forked arm, as shown in

solid lines in the sketch when the bandswitch

is in the 40 meter position. Both arms should

then move up so that the forked arm is in the

position indicated by the dotted lines when

the bandswitch is in the 20 meter position.

The rest of the plate circuit wiring is done

with silver plated ', -inch copper strap. The

strap is ordinary flexible copper "flashing" cut

into strips and silver plated by a local utensil

replating company.

www.americanradiohistory.com

632 H.F. Power Amplifiers T H E R A D I O

PANEL LAYOUT

813 GROUNDED GRID

AMPLIFIER

SUPPORT

BRACKET FOR ,

CI 1.C2

PLATE

CITUNING

2 I DIA KNOBS

TUNT HOLE DIA

22 ¡LARGER

r,

A

THAN METER

--J-L-1

-1 ri GRID PLATE CASE

ALOA ING

I

MET SW I

LaW)

I( S3 2

t RI BIAS

2Ii-

S 4 4

L`-

3

Figure 30

PANEL LAYOUT FOR

AMPLIFIER

The linkage for capacitor

switch pivots on shaft lo-

cated between main tuning

capacitors. Drill 3/4-inch holes

for this shaft, and the shafts

of the capacitors, plus the

meter s w itch. Aluminum

chassis -deck is positioned

I9 -inch above bottom edge

of panel.

Sub -chassis assembly and wiring is shown

in figure 31. The ceramic sockets for the 813

tubes are sub -mounted on metal pillars to

bring the top of the socket shell level with the

under side of the top of the chassis. Under -

chassis wiring, with the exception of the #12

filament leads is run with #18 insulated wire.

The filament choke and bias transformer are

mounted on opposite walls of the chassis. A

small, 115 volt blower motor and fan draws

air up through 1/4-inch holes drilled in the

bottom chassis plate and exhausts the air

through the holes cut in the sub -chassis for

the 813 tubes.

Socket pins 3 and 5 are connected together

and grounded to each of the two adjacent

socket bolts. A jumper runs between the #4

pins, each of which are bypassed to ground by

a .001 µfd. ceramic disc capacitor. Each capa-

citor must be a 1.2 KV type in order to carry

the r.f. charging current existing in the grid

circuit. In addition, a small 50 gefd, ceramic

capacitor is connected between pins 1 and 4

of the tube socket nearest the filament choke.

This capacitor stabilizes the amplifier in the

28 Mc. region.

The 10 volt filament transformer for the

813's is placed above the chassis, as are the

plate r.f. chokes and bypass capacitors. The bias

filter capacitor is a can -type unit which mounts

adjacent to the filament transformer. Various

meter leads are brought out of the chassis via

a terminal strip mounted on the side opposite

the power cable and coaxial input receptacle.

In a TV fringe area, it may be necessary to

completely shield the amplifier with perforated

aluminum sheet. Amplifier harmonic content

is low, and complete shielding is not necessary

in an area of strong TV signals.

Testing and Once construction is finished,

Operating the check the filament and bias

Amplifier circuits before connecting the

high voltage supply to the

amplifier. A power supply with provision for

reducing the output to about one -half of maxi-

mum voltage is recommended, especially if

the operating voltage is 2500 or higher. Con-

nect a dummy load or antenna to the output

receptacle.

Caution: Never apply full excitation to this

or any other grounded grid amplifier without

the plate circuit tuned to resonance, and plate

voltage on the stage. Damage to the amplifier

tubes may result if this rule is violated.

Tune -up for sideband operation consists of

applying full plate voltage and (with terminals

1 and 2 on the power strip shorted) setting

the bias potentiometer for 55 milliamperes of

resting plate current with the meter switch

set in the "plate" position. Only a few volts

of bias are required, and the potentiometer arm

will fall very near one end of the swing. Set

the bandswitch to the frequency of the exciter

and apply a small amount of driving power by

injecting carrier in the s.s.b. exciter. Place the

loading capacitor at full capacitance, and ad-

just the plate tank capacitor for resonance

(minimum plate current). Apply more drive

www.americanradiohistory.com

HANDBOOK 813 Linear Amplifier 633

Figure 31

TOP AND BOTTOM V EWS OF SUB- CHASSIS

Filament transformer and filter capacitor are placed at left edge of chassis. 813 socket holes are

2 -9, 16- inches in diameter, placed 214- inches frcm opposite end of chassis. Small plate choke is

supported on bypass capacitor terminals. Bias transformer and filament choke are mounted to

underside of chassis, as is blower fan.

www.americanradiohistory.com

634 H.F. Power Amplifiers THE RADIO

to obtain about 75 ma. grid current and de-

crease the loading capacitor until resonant

plate current rises to about 200 ma. Finally,

increase the drive and increase the loading

until plate current reaches 400 ma. (300 ma.

at a plate potential of 3000 volts). Grid cur-

rent should be approximately 100 ma. Slightly

overcouple the antenna circuit until the output

(as measured on an r.f. ammeter) drops about

2 percent. This will be the condition of maxi-

mum linearity. Now, switch the exciter to

s.s.b. With speech, the plate current of the

linear amplifier should kick up to about 135

to 150 ma.; while with a steady whistle the

plate current should reach nearly 400 ma.

Tune -up for c.w. operation is similar, except

that the bias potentiometer is adjusted for zero

(cut -off) resting plate current. With full plate

voltage (2500) , the resonant plate current

should be about 375 ma., with 100 ma. of

grid current. At a plate potential of 3000, the

plate current should be reduced to 300 ma.

29 -9 The KW -2. An

Economy Grounded -

Grid Linear Amplifier

The KW -2 sideband amplifier is designed

for use with 4 -400A, 4 -250A or 4 -125A tubes,

and will operate on the 80, 40, 20, 15 and

10 meter amateur bands. A pi- network output

circuit is used, capable of matching 52 -ohm

or 75 -ohm coaxial antenna circuits. Maximum

power input is 2 kilowatts (p.e.p.) or 1

kilowatt, c.w. The amplifier may be driven by

any of the popular s.s.b. exciters having 70 to

100 watts output.

Full input may be achieved with the use

of 4 -400A tubes, but the unit may be run at

reduced power rating with 4 -250A or 4 -125A

tubes. No circuit alterations are necessary when

tube types are changed.

The amplifier employs a passive (untuned)

input circuit, and an adjustable pi- network

output circuit. Air tuning capacitors are used

in the network in the interest of economy and

Figure 32

REAR VIEW OF

AMPLIFIER PLATE

CIRCUIT

Sub- chassis has been re-

moved to show ventila-

tion holes in chassis -

deck. Plate bypass capa-

citors are supported by

t/ -inch copper strap

leads.

www.americanradiohistory.com

HANDBOOK KW -2 Amplifier 635

with no sacrifice in performance. The complete

amplifier is housed in a TVI- suppressed per-

forated metal cabinet measuring 171/4" x 12"

x 121/i" - small enough to be placed on the

operating table next to your receiver.

Amplifier Circuit. The schematic of the am-

plifier is shown in figure 34, Two tetrode

tubes are operated in parallel, cathode driven,

with grid and screen elements grounded. The

sideband exciting signal is applied to the fila-

ment circuit of the tubes, which is isolated

from ground by an r.f. choke. The resistance

of the windings of the choke must be limited

to .01 ohms or less, as filament current is 30

amperes for two 4 -250A or 4 -400A tubes.

Neutralization is not required because of the

excellent circuit isolation afforded by the

grounded elements of the tubes.

The Input Circuit. The input signal is fed in

a balanced manner to the filament circuit of

the two tubes. Ceramic capacitors are placed

between the filament pins of each tube socket,

and excitation is applied to each tube through

two 1250 volt, mica capacitors. The latter are

employed because of the relatively high value

s

of excitation current which may cause capa-

citor heating if ceramic units are employed

at this point.

The filament circuit is wired with #10

stranded insulated wire to hold voltage drop

to a minimum. The leads from the choke to

the filament transformer are run in shielded

loom which is grounded to the chassis at each

end of the wire. The use of shielded leads for

all low voltage d.c. and a.c. power wiring does

much to reduce TVI -producing harmonics.

Figure 33

THE KW -2 LINEAR AMPLIFIER

This two kilowatt p.e.p. amplifier uses two

4 -400A tubes in a grounded -grid circuit. Other

tetrodes, such as the 4 -125A and 4 -250A

may be used without modification to the unit.

At full output, distortion products are better

than -30 decibels below peak power level.

Panel components are (I. to r.): Plate current

meter (top) and output meter (bottom), meter

switch and pilot lamp, plate tuning, band -

switch, and plate loading. At lower right is a

tuning chart for the various bands.

Chassis is bolted directly to the front panel,

allowing about / -inch clearance along bottom

edge to permit edge of shield cage to pass

between chassis and panel lip.

v

www.americanradiohistory.com

636 H.F. Power Amplifiers THE RADIO

exc

N OTE:

RI +METER RES /STANCE

. 100.11

4 -400A 4 -400A ANT

C2 C

Figure 34

SCHEMATIC, KW -2 AMPLIFIER

C -0.001 pfd., 600 volt disc ceramic

C,,C2,C3 -0.1 pfd., 600 -volt coaxial capacitor.

Sprague "Hypass" z80P3

C4 -150 µpfd., 4500 volt. Johnson =1501345

(153 -8)

Cs -SO µµId., surplus vacuum capacitor (see

text)

Ce -1000 µµId., 1250 -volt mica capacitor (see

text)

C7-1500 f.µfd. Barker & Williamson »51241

or 4 -gong b.c. capacitor. Miller 2104

L,- Kilowatt pi- network coil. Air -Dux #195 -

2S (silver plated). Modify as follows: Strap

coil: 3 turns, 13/4" diameter. Wire coil: Re-

move turns from free end, leaving 111/2

turns, counting from junction with tubing

coil. Tap placements: 10 meters, 13/4 turns

from junction of tubing coil and strop coil.

IS meters, 31/4, as above. 20 meters, 11/2

turns of wire coil, counting from junction

with tubing coil. 40 meters, 53/4, as above.

80 meters, complete coil in use

RFC1 -30- ampere filament choke. B&W zFC-

30

RFC2- Kilowatt r.f. choke. Raypar, or B&W

2800

RFC3- v.h.f. choke. Ohmite z I -50

T1-5 volts at 30 amperes. Stancor P -6468

PC -31/2 turns z 12e, r /e" diam., 2" long.

Wound around three 220 -ohm, 2 -watt com-

position resistors connected in parallel

M1-0 -1000 ma. Triplett

M2-0 -1 ma. Triplett

X1- Diode, type 11434

The Grid Circuit. The grid circuit of this am-

plifier is simplicity itself. Screen terminals of

both sockets are grounded to the chassis of the

amplifier. The best and easiest way to accom-

plish this is to bend the terminal lead of the

socket down so that it touches the chassis.

Chassis and lead are then drilled simultane-

ously for a 4 -40 machine screw. Low induc-

tance ground paths are necessary for the high

order of stability required in grounded grid

service.

It is helpful to monitor the control grid

current for tuning purposes, and also to hold

the maximum current within the limits given

in the data chart. Maximum grid current for

the 4 -400A is 100 milliamperes. Under nor-

mal voice conditions this will approximate a

peak meter reading of 50 milliamperes.

Grid current can be observed by grounding

the control grid of each tube through a 1 -ohm

composition resistor, bypassed by a .01 pfd.

disc capacitor. The voltage drop across the

www.americanradiohistory.com

HANDBOOK KW -2 Amplifier 637

resistor is measured by a simple voltmeter

calibrated to read full scale when 100 milli-

amperes of grid current are flowing through

the resistor. A double throw switch will permit

monitoring grid current of either tube. With

incorrect antenna loading, it is possible to

exceed maximum grid current rating with

some of the larger size s.s.b. exciters. No cir-

cuit instability is introduced by this metering

technique.

The Plate Circuit. Power is applied to the

plate circuit via a heavy duty r.f. choke by-

passed at the "cold" end by a 500 µµfd., 10

kv. "TV -type" ceramic capacitor. In addition,

a v.h.f. choke and capacitor are used to sup-

press high frequency harmonics that might

pass down the plate lead and be radiated

through the power supply wiring. Two .001

cfd., 5 kv. ceramic capacitors in parallel are

used for the high voltage plate blocking

capacitor, and are mounted atop the plate

choke.

)

The pi- network coil is an Air -Dux

#195 -2S inductance, designed for service at a

kilowatt level, and silver plated for minimum

circuit loss. Use of the cheaper model having

tinned wire is not recommended for con-

tinuous service at maximum power. The band

switch is a Radio Switch Corp. #88 high

voltage, ceramic switch.

Figure 35

REAR VIEW OF AMPLIFIER

The tube sockets are placed at the right end

of the chassis, with plate r.f. choke centered

between the tubes. The two plate coupling

capacitors are mounted to top terminal of the

choke by means of a brass strap. A "TV-

type" 500 0, fd. capacitor is placed at the

foot of the choke. The two panel meters are

mounted orte above the other. An aluminum

shield plate is placed around the rear of the

meters to protect them from the strong r.f.

field of the tubes. Meter terminals are by-

passed, and the meter lecds are run in shielded

braid. Power, control terminals, fuse and

coaxial receptacles are mounted on rear apron

of chassis.

www.americanradiohistory.com

638 H.F. Power Amplifiers THE RADIO

SHAFT OF SWITCH S I

f0 -32

BRASS BOLT

AND NUT

TOP VIEW

TEFLON. PHENOLIC OR

/ OTHER INSULATING

MATERIAL

MODIFIED INSULATED

COUPLING.

SWITCH ARM

CERAMIC PILLAR Ir SHAFT OF SI

,TOCS

to

SWITCH

ARM

4 `MAMI TWO CONTACTS

OF SPRING BRASS.

,ALUMINUM BRACKET, BOLTED TO

CORNER OF C4 FRAME

SWITCH, SHOWN IN CLOSED POSITION

TOP VIEW

Figure 36

AUXILIARY PADDING SWITCH, PART OF BANDSWITCH

Construction of padding capacitor switch made from parts of on insu-

lated, flexible shaft coupler. Contacts are mode from 1/2 -inch wide

strip of spring brass mounted on small ceramic insulators attached to

main tuning capacitor. Contacts are shorted in 80 meter position of

bandswitch.

A circuit Q of 15 was chosen to permit a

reasonable value of capacitance to be used at

80 meters. In this case, a 150 µµfd. variable

air capacitor is employed for operation above

80 meters, and an additional 50 µµfd. parallel

capacitance is switched in the circuit for 80

meter operation. The 50 µµfd. padding capaci-

tor is the small vacuum capacitor found in the

"Command" set antenna relay boxes. These

capacitors seem to be plentiful and inexpen-

sive. A satisfactory substitute would be a

50 µµfd. 5 kv. mica capacitor, also available

on the surplus market.

The pi- network output capacitor is a 1500

µµfd. unit. It is sufficiently large to permit

operation at 80 meters into reasonable an-

tenna loads. For operation into very low im-

pedance antenna systems that are common on

this band, the loading capacitor should be

paralleled with a 1000 µµfd., 1250 volt mica

capacitor. This capacitor may be connected to

the unused 80 meter position of the band -

switch.

The Metering Circuits. It is always handy to

have an output meter on any linear amplifier.

A simple r.f. voltmeter can be made up of a

germanium diode and a 0 -1 d.c. milliammeter.

The scale range is arbitrary, and may be set

to any convenient value by adjusting the po-

tentiometer mounted on the rear apron of the

chassis. Once adjusted to provide a convenient

reading at maximum output level of the am-

plifier, the control is left alone. Under proper

operating conditions, maximum output meter

reading will concur with resonant plate

current dip.

It is dangerous practice to place the plate

current meter in the B -plus lead to the am-

plifier unless the meter is insulated from

ground, and is placed behind a protective

panel so that the operator cannot accidentally

touch it. If the meter is placed in the cathode

return the meter will read the cathode current

which is a combination of plate, screen and

grid current. This is poor practice, as the

reading is confusing and does not indicate the

true plate current of the stage. A better idea

is to place the meter in the B -minus lead be-

tween the amplifier chassis ground and the

power supply. The negative of the power

supply thus has to be "ungrounded," or the

meter will not read properly (figure 37) . A

protective resistor is placed across the meter

to ensure that the negative side of the power

supply remains close to ground potential.

Make sure that the negative lead between the

power supply and the amplifier is connected

at all times.

www.americanradiohistory.com

HANDBOOK KW -2 Amplifier 639

PLATE SWITCH OR

VOX RELAY

CONTROL

RELAY T,

866'5

OR

3824.5

4

CHI

843000 V.

EA.

3011

50W

15 0

ti

IIS

PRIMARY

CONTROL

SWITCH

100

20 W CHASSIS

GROUND

Figure 37

SCHEMATIC, POWER SUPPLY FOR LINEAR

AMPLIFIER

CH, -6 H, 500 ma. Chicago R -65

T,- 3450 -2850 volts each side of center tap, 500 ma. 115 -230

volt primary. Chicago P -3025

72 -2.5 volts, 10 a. 9 kv. insulation. Chicago FH -210H

The Cooling System. It is necessary to provide

a current of cool air about the base seals and

plate seal of the 4 -250A and 4 -400A tubes.

If small blowers are mounted beneath each

tube socket it is possible to dispense with the

special air sockets and chimneys, and use the

inexpensive "garden variety" of socket. A

Barber Coleman type DYAB motor and im-

peller is mounted in a vertical position cen-

tered on the socket, and about an inch below

it. Cooling air is forced up through the socket

and around the envelope of the tube. The

perforated metal enclosure provides maximum

ventilation, yet effectively "bottles up" the

r.f. field about the amplifier. In order to per-

mit air to be drawn into the bottom of the

amplifier chassis, small rubber "feet" are

placed at each corner of the amplifier cabinet,

raising it about 1/2 -inch above the surface

upon which it sits.

Amplifier

Construction The amplifier is built upon an

aluminum chassis measuring

13" x 17" x 3 ". Input circuit

components, power circuits, and the blower

motors are mounted below the chassis, and

the plate circuit components are mounted

above the deck. Placement of parts is not

critical, except that the leads beween the band-

switch and the plate coil must be short, heavy

and direct. One -half inch, silver plated copper

strap is used. The straps are bolted to the

bandswitch with 4 -40 nuts and bolts. Each

lead is tinned and wrapped around the proper

coil turn and soldered in place with a large

iron. The operation should be done quickly

to prevent softening of the insulating coil

material. Low resistance joints are imperative

at this point of the circuit. To play safe, you

can submerge the coil in a can of water, with

just the top of the turns showing above the

surface. This will prevent the body of the coil

from overheating during the soldering pro-

cess. It is also helpful to depress a turn on

each side of the tap in order to provide

sufficient clearance for the soldering iron. This

may be done by placing the blade of a screw

driver on the wire, and hitting it with a

smart tap.

The coil assembly is supported on four

ceramic pillars, and placed immediately be-

hind the band change switch, which is

mounted on a sturdy aluminum bracket. The

coil is positioned so that the taps come off

on the side nearest the switch.

A set of auxiliary contacts are required to

switch the padding capacitor into the circuit

when the bandswitch is thrown to the 80

www.americanradiohistory.com

640 H.F. Power Amplifiers THE RADIO

r

t urre

Figure 38

UNDER -CHASSIS VIEW OF AMPLIFIER

The filament transformer is mounted to the side apron, with the filament choke placed between

the transformer and the tube sockets. The two blower motors are attached to an aluminum strip

that holds them in position under the tube sockets, on a level with the bottom edge of the chassis.

This strip is bolted to the chassis flange with flat -head bolts. The bolts holding the blowers pass

through rubber grommets mounted on the strip to deaden blower noise. All low- voltage power

leads run through shield braid which is grounded to the chassis by means of aluminum clamps

mode from scrap material. B -plus lead is a section of RG -8 /U coaxial cable. Diode voltmeter com-

ponents are mounted to a phenolic board attached to the side apron at right.

meter position. A simple switch may be made

up from the metal portions of an insulated

coupling and a block of insulating material,

such as teflon, lucite, or micarta (figure 36).

The insulated disc of the coupling is removed,

and an oval of insulating material is substi-

tuted. This assembly is placed on the shaft

of the bandswitch. A set of spring contacts

are mounted on small stand -off insulators

attached to the side of the tuning capacitor

and positioned so that the oval rotates be-

tween the contacts as the switch is turned. A

hole is drilled in the oval, and a flat -head

8 -32 brass machine screw is passed through

it. A nut is run onto the screw, and screw

end and nut head are filed flat. When the

www.americanradiohistory.com

HANDBOOK KW -2 Amplifier 641

Figure 39

PLATE TANK CIRCUIT ASSEMBLY

The plate bandswitch is supported on a l's -inch thick aluminum bracket. The 80 meter padding

capacitor is mounted on the front of the bracket. Silver -plated copper strap is used to make

connections between the switch and the coil. Switch connections are made with 4 -40 hardware, and

then soldered securely. Auxiliary padding capacitor switch may be seen on shaft of bandswitch,

directly in front of bracket. Plate switch is made by Radio Switch Corp., Marlboro, N.I.

1

www.americanradiohistory.com

642 H.F. Power Amplifiers T H E R A D I O

Figure 40

OPERATING CHARACTERISTICS, GROUNDED -GRID CONFIGURATION

4 -125A

D.c. Plate Voltage 2000 2500 3000 volts

Zero -Signal Plate Current 10 15 20 ma.

Single -Tone Plate Current 105 110 115 ma.

Single -Tone Screen Current 30 30 30 ma.

Single -Tone Grid Current 55 55 55 ma.

Single -Tone Driving Power 16 16 16 watts

Driving Impedance 340 340 340 ohms

Load Impedance 10,500 1 3,500 15,700 ohms

Plate Input Power 210 275 345 watts

Plate Output Power 145 190 240 watts

4 -400A

(ratings apply to 4 -250A, within plate dissipation rating of 4 -250A)

D.c. Plate Voltage 2000 2500 3000 volts

Zero -Signal Plate Current 60 65 70 ma.

Single -Tone Plate Current 265 270 330 ma.

Single -Tone Screen Current 55 55 55 mo.

Single -Tone Grid Current 100 100 100 ma.

Single -Tone Driving Power 38 39 40 watts

Driving Impedance 160 150 140 ohms

Load Impedance 3950 4500 5000 ohms

Plate Input Power 530 675 990 watts

Plate Output Power 325 435 600 watts

4 -1000A

D.c. Plate Voltage 3000 4000 5000 volts

Zero -Signal Plate Current 100 120 150 ma.

Single -Tone Plate Current 700 675 540 ma.

Single -Tone Screen Current 105 80 55 ma.

Single -Tone Grid Current 170 150 115 ma.

Single -Tone Driving Power 130 105 70 watts

Driving Impedance 104 106 110 ohms

Load Impedance 2450 3450 5550 ohms

Plate Input Power 2100 2700 2700 watts

Plate Output Power 1475 1870 1900 watts

switch is rotated to the 80 meter position,

contact is made between the two spring arms

through the body of the screw, which com-

pletes the circuit between the switch contacts.

Amplifier Typical operating conditions

Adjustment for various tubes are tabulated

in figure 40. For initial ad-

justment, four or five hundred volts plate

potential is applied to the amplifier, and

sufficient grid drive is supplied (five watts

or so) to provide an indication on the plate

meter. The loading capacitor is set at maxi-

mum capacitance, and the tuning capacitor is

adjusted for resonance, which is indicated by

the customary dip in plate current. After reso-

nance is found full plate voltage should be

applied to the amplifier, and resting plate

current compared with the value shown in the

table. If all is well, a carrier is applied to the

amplifier for adjustment purposes. The signal

may be generated by carrier injection, or by

www.americanradiohistory.com

HANDBOOK 4 -400A Amplifier 643

tone modulation of a sideband exciter.

Caution! Do not apply full excitation to

any grounded grid amplifier without plate

voltage on the stage, or with the stage im-

properly loaded. Under improper conditions,

driving power normally fed to the output cir-

cuit becomes available to heat the control grid

of the tube to excessive temperature, and such

action can destroy the tube in short time.

Adjustable control of the excitation level is

mandatory.

The amplifier is now loaded to full, single

tone input. (In the case of two 4- 400A's this

will be 3000 volts at 333 ma., 2500 volts at

400 ma., or 2000 volts at 500 ma.) Driving

power will be approximately 30 watts per

tube. Under these conditions, power input will

be 1000 watts p.e.p. for sideband operation.

To properly load the amplifier for 2 kw.

p.e.p, operation it is necessary to have a

special test signal. Tuning of this (or any

other linear amplifier) is greatly facilitated

by the use of an oscilloscope and envelope

detectors. Even with two -tone or carrier input

signal, however, it is difficult to establish the

proper ratio of grid drive to output loading.

In general, antenna coupling should be quite

heavy: to the point where the power output

of the amplifier has dropped about two per-

cent. This point may be found by experiment

for power levels up to 1 kw. p.e.p. However,

since neither this amplifier, nor most power

supplies, are designed for continuous carrier

service at two kilowatts and since this average

power level is illegal, some means must be

devised to tune and adjust a "legal" two

kilowatt p.e.p. linear amplifier without ex-

ceeding the limitations of the amplifier, and

without breaking the law. A proper test signal

having high peak to average power ratio will

do the job, permitting the amplifier to run

at less than a kilowatt d.c. input while allow-

ing the 2 kw. peak power level to be reached.

This type of signal can be developed by an

audio pulser, such as was described in QST

magazine, August, 1947 ( figure 41) . The

duty cycle of this simple pulser is about 0.44.

This means that when the amplifier is tuned

up for a d.c. indicating meter reading 800

watts, using the pulser and single tone audio

injection, the peak envelope power will just

reach the 2 kw. level. An oscilloscope and

AUDIO

INPUT

0010

6J5

PULSED AUDIO

OUTPUT

MEA /r.P -3045

OR

FOUI VALENT

Figure 41

AUDIO PULSER FOR HIGH POWER

TUNE -UP OF AMPLIFIER

This simple audio pulser modifies the audio

signal to the sideband exciter so that it has a

high peak -to- average power ratio. Amplifier

may be thus tuned for two kilowatt p.e.p.

input without violating the one kilowatt

maximum steady state condition.

audio oscillator are necessary for this test,

but these are required items in any well

equipped sideband station. Loading and drive

adjustments for optimum linearity consistent

with maximum power output may be con -

ducted by this method.

29 -10 A Pi- Network

Amplifier for C -W,

A -M, or SSB

This all- purpose amplifier covers the 3.5-

29.7 Mc. range, and is designed for one kilo-

watt c.w. or s.s.b. operation, and 825 watts

input plate modulated a.m. service. Using a

single 4 -400A tetrode tube, this grid- driven

amplifier may be driven by an exciter having

a power output of approximately 15 watts.

Two mechanical designs are discussed, one

using variable vacuum tuning capacitors, and

the other employing the less expensive vari-

able air capacitors. The latter design is highly

recommended as an inexpensive and foolproof

amplifier for the amateur wishing to go high

power on a lean purse!

Amplifier Circuit The schematic of the am-

plifier is shown in figure

43. Bandswitching is employed in the grid

and plate circuits, and the tetrode tube is

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644 H.F. Power Amplifiers THE RAD ;O

Figure 42

4 -400A ALL -BAND AMPLIFIER

This compact amplifier is designed for operation in the 3.5 -29.7 Mc. ronge. Using bandswitching in

the grid and plate circuits, the unit is capable of a full kilowatt input on c.w. and s.s.b., and 825

watts a.m. phone. The amplifier employs variable vacuum tuning capacitors, but an alternative

design uses inexpensive air capacitors. Panel controls are (I to r.): plate current meter (top), grid

bandswitch (center), and grid tuning (bottom). Screen current meter (top), plate tuning (center),

and plate loading (bottom). Grid current meter (top), plate bandswitch (center), and filament

switch and pilot lamp (bottom).

www.americanradiohistory.com

HANDBOOK 4 -400A Amplifier 645

Ext. S1A

60

L1

4- 250A /4 -400A EA

300

,0 KV L 2

470

27.1V.

RFC,

12 ,011

zwll

SOO

SEE 110MV

NOTE Hn

B+HV

OC

--IHII

470

Uri

O

+SCR CND.

+BIAS

-SCR.

C

-BIAS B MV

Figure 43

SCHEMATIC, 4 -400A AMPLIFIER

C1 -140 ppfd. Hammarlund APC -1408

C2- Neutralizing capacitor. 10 µpfd. Millen

#15011, or Johnson N -250

C3 -250 µµid. Jennings UCS -250 variable

vacuum capacitor. Johnson 250070(153 -13)

C4 -1500 ppfd. Jennings UCSL -1200 variable

vacuum capacitor. J. W. Miller #2I04 air

capacitor may be substituted

L1 -50 turns, #24, 13/4" long, 3/4" diam. Tap

S, 8, 13, and 25 turns from grid end.

Wound on ceramic form. Link coil is 4 turns

#18 insulated wire, wound on "cold" end

of LI, tapped at center of winding

L2- Barker d Williamson #850 pi- network in-

ductor. 80 meters, 13.5 ph.; 40 meters, 6.S

ph.; 20 meters, 1.75 ph.; 15 meters, 1.0 ph.;

10 meters, 0.8 ph.

MI -0 -50 d.c. milHammeter

M2 -0 -100 d.c. milliammeter

M3 -0 -800 d.c. milliammeter

PC -4 turns, 1" diam. wound about four 220

ohm, 2 watt composition resistors in parallel

RFC,,s -2.5 mh. National R -100

RFC2 -BSW #800 plate choke, or National

R -175A

S1 -2 pole, S position ceramic switch. Cen-

tralab 2002

T1 -5 volts @ 15 amperes. Triad F -9U

Blower- Shaded pole induction motor, 2400

r.p.m. 4 blade fan, 21/2" diam. Allied Rodio

Co., Chicago, part number 72P -715

Counter dials: Grath Mfg. Co.

115V."1.

neutralized to achieve maximum stability of

operation. Link coupling from the external

exciter is used, and a tuned grid circuit offers

ISO

CS 2 1500

S4

RF

OUT.

RFC

NOTE.

C IS .01 Ur CORAM IC, 600 VOLT.

SCLr MODULATION CIRCUIT

FOR SCRCCN LOAD.

X

OMONE

maximum rejection to any spurious harmonics

or unwanted emissions of the exciter. Capaci-

tive bridge neutralization is employed, with

a 250 ¡yid. mica capacitor forming the

ground leg of the bridge in the grid circuit.

Each screen terminal of the tube socket is

bypassed to ground with a low inductance

high voltage ceramic capacitor, and the screen

power lead is harmonic filtered by a simple

R -C network. Grid and screen currents are

separately metered. To aid circuit stability in

the region of v.h.f. parasites, one leg of the

filament is grounded, and the opposite ter-

minal is bypassed to ground at the tube socket.

In addition, simple parasitic chokes are used

in the grid and plate circuits as a safety

measure. The plate circuit is the popular pi-

network configuration, and will match 50-

or 75 -ohm antenna loads having an s.w.r. of

less than 2 to 1.

Amplifier plate current is metered in the

B -minus lead to the power supply in order

to remove the meter from the high potential

B -plus circuit. By returning the bias and

screen supplies to the cathode circuit (ground)

the plate meter reads only the true plate cur-

rent and not the cathode current, which is

the sum of grid, screen, and plate currents.

The reader is referred to the discussion of

this subject in a previous section of this

chapter.

www.americanradiohistory.com

646 H.F. Power Amplifiers THE RADIO

Figure 44

TOP VIEW OF 4 -400A AMPLIFIER

R.f. circuits atop the chassis are enclosed in ventilated box made of perforated aluminum. Band -

switching inductor is at the right, with coaxial antenna receptacle directly to the rear, mounted

on aluminum plate. To left of variable vacuum capacitor is the disc -type neutralizing capacitor.

Plate r.f. choke is directly behind tube. Panel meters are isolated from r.f. field by aluminum

sub -panel.

Amplifier

Construction

The amplifier is constructed

upon an aluminum chassis

measuring 15" x 17" x 4 1/2".

Standard, TVI -proof construction is used, as

outlined in the Workshop Practice chapter

of this Handbook. The above -chassis cir-

cuitry is enclosed in a perforated aluminum

enclosure measuring 13 t/4" x 17" x 9 ". The

frame of the enclosure is made of t/2 -inch

aluminum angle stock, with corner gusset

plates. Perforated sheets form the sides and

top and are held in position with sheet metal

screws spaced about three inches apart along

the edges of the material. A sub -panel made

of I/8-inch aluminum is placed about 13/4

www.americanradiohistory.com

HANDBOOK 4 -400A Amplifier 647

inches behind the main panel. The area be-

tween the two panels is taken up by the three

meters, and the gear drive system for the grid

bandswitch. The panels are held in position

by metal spacers located at the extreme top

corners of the assembly.

Placement of the major components may

be seen in the photographs. The pi- network

tuning capacitors are centered on the panel,

with the bandswitch controls placed symmetri-

cally about the tuning capacitor. Below deck

the output loading capacitor is contained

within a small shielded compartment formed

from sheet aluminum. As the grid input cir-

cuit is adjacent to this capacitor, it is impor-

tant that :here be no leakage of r.f. energy

from input to output circuits. The bottom

plate of the chassis is a solid piece of alu-

minum, with a 4 -inch hole cut in it directly

below the blower for the tube socket. The hole

is covered with perforated aluminum stock,

and the bottom plate is firmly bolted to the

chassis lip, and also to the flanges of the box

screening the output loading capacitor. An

"r.f.- tight" box thus surrounds the capacitor.

Connection between the capacitor and the pi-

network circuit above the deck is made via a

ceramic feedthrough insulator mounted in

the deck.

The blower motor is mounted in a vertical

position below the ceramic tube socket (figure

44A ). A strip of aluminum supports the motor

between the lip of the chassis and a lip of

the capacitor compartment. The bracket is

mounted with flat -head bolts, and the motor

bolts are run through rubber grommets

mounted in the strip. The power leads to the

motor, as well as all other low voltage power

wiring beneath the chassis, are run in shielded

braid with the lead bypassed to the braid at

each end of the run.

The grid circuit components are mounted

to an aluminum plate spaced away from the

panel by four aluminum posts. The grid capa-

citor is driven by two flexible couplings from

the tuning dial, which is positioned on the

panel below the bandswitch and meter. The

grid bandswitch is driven from atop the

chassis by means of two right -angle gear

drives. One drive is below the chassis and

the second is placed in the meter compartment

behind the bandswitch dial.

Placement of the major plate circuit com-

ponents may be seen in figure 44. The

tuning capacitor is centered on the chassis with

the tube and neutralizing capacitor on one

side, and the plate tank inductor on the oppo-

site side. The ceramic plate circuit coupling

capacitors are mounted between two aluminum

plates, forming a "sandwich" supported on one

side by a 1/2-inch wide copper strap from the

plate r.f. choke, and on the other side by a

similar strap affixed to the plate tank

capacitor.

Bias and Screen The bias and screen supply

Supply described in the next sec-

tion of this chapter may

be used for all- purpose amplifier operation.

Screen protective relay RY1 should be adjusted

to cut out at a maximum screen current of

50 milliamperes. If sideband operation is not

contemplated, it is possible to eliminate the

voltage regulator tubes in the screen supply

and substitute a simpler unit that will pro-

vide 400 volts d.c. at 50 milliamperes. This

will be suitable for either phone or c.w. opera-

tion. For the former, it is necessary to allow

the screen to "self- modulate" itself to obtain

100 percent plate modulation. This is done

by inserting a 10 -henry filter 100 ma. choke

in the screen lead at the point marked "X"

(figure 43) . The choke is shorted out for

c.w. operation.

Use of Air In order to reduce the cost of

Capacitors the amplifier, it is possible to

substitute air capacitors for the

variable vacuum units. A Johnson #250D70

(153 -13) will serve as the plate capacitor,

and a four gang b.c. -type capacitor, such as

the J. W. Miller #2104 will replace the

vacuum output capacitor. In addition, the in-

expensive Air -Dux inductor and the ceramic

switch described in the "KW -2" amplifier

may be used as a substitute for the more

expensive bandswitch assembly shown here.

Amplifier Tuning The amplifier should be

and Adjustment neutralized in the manner

described in the next sec-

tion of this chapter. Proper neutralization is

indicated during operation of the amplifier

by detuning the plate tuning capacitor a small

amount each side of resonance. The point of

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648 H.F. Power Ampli=fiers THE RADIO

Figure 44A

LAYOUT OF UNDER -CHASSIS COMPONENTS

The pi- network loading capacitor is mounted on angle plates within the shielded compartment at

center. The grid circuit components a-e at the left, in fient of blower fa- and motor. The filament

transformer is mounted to the wall at right side of chassis. Shielded wire .s used for all low- voltage

power leads.

www.americanradiohistory.com

HANDBOOK Kilowatt Amplifier 649

minimum plate current should coincide with

the point of maximum grid current. If grid

current increases when the plate circuit is

tuned either side of resonance, the setting of

the neutralizing capacitor should be varied

slightly until the two readings coincide at one

capacitor setting.

The bias supply is adjusted to provide ap-

proximately -120 volts of cut -off bias. Full

screen voltage may be applied as long as cut-

off bias is on the stage. Full excitation, how-

ever, should never be applied in the presence

of screen voltage unless full plate voltage is

on, and the amplifier is properly loaded.

Screen current is a very sensitive indicator of

proper operation. High values of screen cur-

rent point to insufficient antenna loading, or

to excess drive. Low screen current indicates

excessive antenna loading or insufficient drive.

If the plate current seems normal, the drive

level should be adjusted to provide proper

screen current.

29 -11 Kilowatt Amplifier

for Linear or Class C

Operation

A pair of 4 -250A or 4 -400A tetrode tubes

may be employed in a pi- coupled amplifier

capable of running one kilowatt input, c -w or

plate modulated phone, or two kilowatts

p.e.p. for sideband operation. Correct choice of

bias, screen, and exciting voltages will permit

the amplifier to function in either class A, B,

or class C mode. The amplifier is designed

to operate at plate potentials up to 4000 volts,

and excitation requirements for class C oper-

ation are less than 25 watts.

A bandswitching type of pi- network is em-

ployed in the plate circuit of such an ampli-

fier, shown in figure 45 The pi- network is

an effective means of obtaining an impedance

match between a source of r.f. energy and a

low value of load impedance. A properly de-

signed pi- network is capable of transformation

ratios greater than 10 to 1, and will provide

approximately 30 decibels or more attenuation

to the second harmonic output of the amplifier

as compared to the desired signal outpiat. Since

the second harmonic level of the amplifier

tube may already be down some 20 db, the

actual second harmonic output of the network

will be down perhaps 50 db from the funda-

mental power level of the transmitter. Atten-

uation of the third and higher order harmonics

will be even greater.

Figure 45

GENERAL PURPOSE

AMPLIFIER

OPERATES IN CLASS

A, 8, OR C MODE

This kilowatt amplifier

employs a pair of

4- 250A's in a pi- network

circuit. Mode of opera-

tion may be set by selec-

tion of proper screen and

bias voltages. Grid, plate,

and screen current me-

ters are mounted on

plastic plate behind panel

cut -out, and tubes are

visible through shielded

panel opening. Across

bottom of panel (left to

right' ore bandswitch,

grid tuning, plate tun-

ing, loading, and primary

power control circuits.

Plate tuning knob is at-

tached to small counter

dial.

www.americanradiohistory.com

650 H.F. Power Amplifiers

L L2 L3 L4 Ls

4 -250A

4 -400A 4 -250A

4 -4004

INPUT

JI

01

EACH

001

5NV S

H(

RFC 2

500

20NV

Le

OUTPUT

S 1 J2

RFCI

5ó1 1

NOTE: SCREEN BYPASS CAPACITORS

ARE CENTRALAB TYPE 838

/M3 y 01

+T

Hl1

TS1 L -BIAS CON- +SCREEN GN0

rROL 115V 115v.

ti ti

B+2500-

3500

Figure 46

SCHEMATIC, GENERAL PURPOSE KILOWATT AMPLIFIER

CI -100 µp /d. Hammar-

lund HF -100

Cr-200 µµId., 10KV var-

iable vacuum capaci-

tor. Jennings UCS -200

Cs -1500 µtd., variable

capacitor. Cardwell

8013

C.-Neutralizing capaci-

tor, disc. Millen 15011

C1 -300 µµtd., mica,

1250 volt

L1 -L,, -See coil table

PC -47 ohm, 2 watt S -Two pole, 6 position T -S volt, 20 ampere.

composition resistor switch. Two Centralab Stancor P -6492

wound with 6 turns PA -17 decks, with PA-

= 18e. 301 index assembly

RFC: -2.5 mh. choke.

National R -100

RFC. -Heavy duty, wide -

band r.i. choke. Bar-

ker 8 Williamson type

800

S: -Two pole, 6 position

high voltage switch.

Communication Prod-

ucts Co. type 88 two

gang switch

51 -Four pole, three po-

RFC -VHF choke. Ohm- sition switch. Centra -

ite Z -144 lab

M: -0 - 50 ma. d.c.

Triplett

- 150 ma. d.c.

Triplett

M -0 - 750 ma. d.c.

Triplett

Gears -2 required. Bos-

ton Gear CG -465 and

=G-466

The peak voltages encountered across the

input capacitor of the pi- network are the same

as would be encountered across the plate tun-

ing capacitor of a single -ended tank used in

the same circuit configuration. The peak volt-

age to be expected across the output capacitor

of the network will be less than the voltage

across the input capacitor by the square

root of the ratio of impedance transformation

of the network. Thus if the network is trans-

forming from 5000 ohms to 50 ohms, the ra-

tio of impedance transformation is 100 and

the square root of the ratio is 10, so that the

voltage across the output .capacitor is 1 /10

that across the input capacitor.

A considerably greater value of maximum

capacitance is required of the output capacitor

than of the input capacitor of a pi- network

when transformation to a low impedance load

is desired. For 3.5 Mc. operation, maximum

values of output capacitance may run from

500 µµfd. to 1500 µµfd., depending upon the

ratio of transformation. Design information

covering pi- network circuits is given in an earl-

ier chapter of this Handbook.

Illustrated in this section is an up -to-

date version of an all -band pi- network ampli-

fier, suited for sideband or class -C operation.

The unit is designed for TVI -free operation

over this range.

Circuit The schematic of the general

Description purpose amplifier is shown in

figure 46. The symmetrical

panel arrangement of the amplifier is shown

in the front view (figure 45) and the rear

view (figure 47) . A 200 µµfd. variable va-

cuum capacitor is employed in the input side

of the pi- network, and a 1500 µµfd. variable

air capacitor is used in the low impedance out-

put side. The coils of the network are switched

in and out of the circuit by a two pole, five

www.americanradiohistory.com

Figure 47

REAR VIEW OF GENERAL PURPOSE

AMPLIFIER WITH SHIELD

REMOVED

The pi- network circuit is built from an inex-

pensive high voltage rotary switch, and five

inductors. The switch is panel driven by a

gear and shaft system shown in figure 38.

Variable vacuum capacitor is mounted ver-

tically between the tubes, directly in back of

the plate r.f. choke. Neutralizing capacitor is

at right, connected to plates of tubes with a

wide, silver plated copper strap. Meters are

enclosed by aluminum shield partition running

the width of the enclosure, with conduit car-

rying meter loads to under- chassis area at

left, front of chassis. Metal shells of tube

bases are grounded by spring contacts.

position high voltage ceramic rotary switch.

Each coil is adjusted for optimum circuit Q,

resultine in no tank circuit compromise in ef-

ficiency at the higher frequencies. A close -up

of the tank circuit is shown in figure 47. The

plate blocking capacitor is made of two .001

µfd., 5 kv. ceramic capacitors connected in series.

Special precautions are taken to insure op-

erating stability over the complete range of

amplifier operation. The screen terminals of

each tube socket are jumpered together with

Ye" copper strap and a parasitic choke (PC)

is inserted between the center of the strap and

the screen bypass capacitor. In addition, sup-

O

u

pressor resistors are placed in the screen leads

after the bypass capacitor to isolate the sensi-

tive screen circuit from the external power

leads. A third parasitic choke is placed between

the grid terminals of the tubes and the tuned

grid circuit.

The five coils of the grid circuit are en-

closed in a small aluminum shield placed ad-

jacent to the tube sockets (figure 48 and figure

50). The amplifier is neutralized by a capaci-

tive bridge system consisting of neutralizing

Figure 48

PLACEMENT OF

PARTS IN UNDER -

CHASSIS AREA

Grid tuned circuit is en-

closed in separate enclo-

sure at left. Bandswitch

projects out the rear of

case, and is gear driven

by same shaft that act-

uates the plate band -

switch. Switches are driv-

en through right -angle

gear drives and gears.

Output capacitor of pi-

network is shielded from

rest of under -chassis

components.

The screen terminals of

each tube socket are

strapped together with

3 /e" copper ribbon, and

low inductance screen

bypass capacitor is

grounded t o socket

mounting bolt. Screen

parasitic choke mounts

between strap and ca-

pacitor terminal. All

power leads beneath the

chassis are run in shield-

ed braid, grounded to

chassis at convenient

points. B -plus lead is

made of section of RG-

8,'U coaxial cable, with

outer sheath and braid

removed.

www.americanradiohistory.com

652 H.F. Power Amplifiers T H E R A D I O

FIGURE 49

COIL TABLE FOR KILOWATT AMPLIFIER

GRID COILS

Li- (SO METERS) : 40 TURNS N24 E, 3/4' O /A., 1" LONG ON

AMPNENOL POLYSTYRENE FORM.

L2 -(40 METERS I: 3O TURNS 24 E, .7/4 "DIA., 3/4- LONG

ON AMPNENOL POLYSTYRENE FORM.

L3-(20 METERS): 72 TURNS, 82W 3071 MINIDUCTOR,

S/4" OM., 3/4' LONG.

L4 -(1S METERS): 7TURNS, 88 3O1O MINIDUCTOR,

3/4' DIA., 7 /e' LONG.

LE -(10 METERS): 3 TURNS. AS ABOVE.

ALL COILS HAVE 3 TURN LINKS MADE OF HOOKUP WIRE.

PLATE COILS

LS- (SO METERS) : 77 TURNS !O, 3 -O O., 4 TURNS PER INCH

-AIR -DUX -

L7- (40 METERS): IO TURNS IO, 3" OD., 5 TURNS PER /NCH

L! -lao METERS): P TURNS, 3/7" COPPER TURING,

2 1/2' O.O., 3" LONG.

LP -(1S METERS): 7 TURNS, 1/4" COPPER TURING.

2 1/4" 0.0., .1" LONG.

L10 -00 METERS): S TURNS, I /4- COPPER TURING.

2 I/4" 0.0., 3 "LONG.

capacitor G and Cr., the grid circuit bypass ca-

pacitor.

Screen voltage may be removed for tune -up

purposes by control switch 5., section B. The

screen circuit is grounded in the "off" and

"fil" positions by means of switch section C.

Amplifier

Construction The complete amplifier is

built upon an aluminum chas-

sis measuring 13" x 17" x 3"

and has a 14" standard relay rack panel. The

Figure 50

GRID TANK CIRCUIT ASSEMBLY

Coils are mounted to the ceramic switch decks

by their leads. A small aluminum plate at-

tached to rear of the switch assembly rods

supports grid tuning capacitor which projects

out rear of shielded enclosure. Entire assembly

may be pre -wired before placing in enclosure.

grid circuit components are mounted within

an aluminum box measuring 3" x 4" x 4 ".

Plate loading capacitor G, r.f. choke RFC -1,

and output connector J2 are placed within an

enclosure measuring 6" x 6" x 3", made up

of aluminum angle sections and sheet material.

The plate circuit shielding is made of Reynold's

"Do- it- yourself" aluminum stock, available at

most hardware stores.

Layout of the major components can be seen

in figure 47. The two tube sockets are placed

directly behind the panel opening, with the

plate r.f. choke between them, and the variable

vacuum capacitor is mounted vertically to the

chassis directly behind the sockets, on the cen-

ter line of the chassis. To the right of the sock-

ets is neutralizing capacitor C.. The high volt-

age ceramic coil switch SA-B is placed directly

behind the vacuum capacitor, mounted in a

vertical position.

The variable vacuum capacitor is panel driv-

en by a counter -type dial, through a miniature

right angle gear drive, as seen in the under -

chassis view (figure 48). The plate and grid

band switches are ganged and switched in uni-

son by means of a shaft acting through two

right angle gear drives and two bevel gears.

Both circuits are thus switched by the "Band -

switch" control located in the lower left corner

of the front panel.

It is necessary to apply forced air to the

sockets of the amplifier tubes. A large 115

volt a.c. operated blower is therefore mounted

in the center of the bottom shield plate. The

under -chassis area is thus pressurized and the

majority of the air escapes through the socket

ventilation holes located near the pins of the

tubes. All wiring beneath the chassis (with the

exception of the filament leads) is done with

5KV insulated wire, encased in metallic braid

which is grounded to the chassis every inch or

so. The B -plus wiring from the high voltage

terminal to the plate current meter is done

with a section of RG -8 /U coaxial line from

which the outer braid has been removed. A

similar piece of line is run from the ,peter to

the plate r.f. choke, RFC -2.

The three meters are mounted upon a lu-

cite sheet placed behind a second lucite sheet

mounted behind a cut -out in the front panel.

The meters are shielded from the plate circuit

of the amplifier by an aluminum enclosure

that covers the wiring and meters, running the

full length of the chassis. The meter leads pass

through the plate circuit area via a short length

of 1,1,-inch aluminum conduit that is threaded

www.americanradiohistory.com

HANDBOOK Kilowatt Amplifier 653

4-250A /4-400A OPERATING CHARACTERISTICS

(2 TUBES )

ITEM MODE

55861 33682 PHONE C.W.

PLATE VOLTAGE 3000 3000 2500 3000

PLATE CURRENT (MA.) 110 -420 260 -440 400 330

SCREEN VOLTAGE 600 S00 400 500

SCREEN CURRENT (MA.) 24 1.0 60 70

GRID BIAS -110 -60 -200 -200

PROTECTIVE BIAS -110 -60 -120 -120

GRID CURRENT (MA.) 0 0 20 20

POWER OUTPUT(WATTS) 600 700 770 600

Ti 5R4 -GY CH, R1 5K RY1

sow

2

111

o loF 5

1 nv T

VR -1150 5

I

U

5GR

1oó 1s

/W

9

115V.

VR -150

1.3 v.

VR OR

V R- 150 ID

T2 VR -90 5

OR

VR -150

106.3V.

` \ \\

GND

20LIF

(

5Y3-GT ((

* 5K/25w

R3 +

R4

St

6

450 V. R2

LW I

e -BIAS

Figure 51

OPERATING DATA AND SCHEMATIC,

SCREEN AND BIAS SUPPLY

T:- 870 -410 -0- 410 -870 volts of 150 ma. and

60 ma. 5 volts, 2 o., 6.3 v. 3.5 a.

Stancor P -8307

Te- 235 -0 -235 volts at 40 ma. Stanco, PC-

8401

CH -7 henry at 150 ma. Stancor C -1710

CH -7 henry at 50 ma. Stancor C -1707

RY,- Overload relay, adjustable 100 -250 ma.

Note: J, is insulated from chassis.

at each end and bolted to the chassis and the

meter shield. Plate circuit wiring above the

chassis is done with 1/2-inch silver plated cop-

per strap.

Amplifier

Neutralization After the amplifier is wired

and checked, it should be

neutralized. This operation

can be accomplished with no power leads at-

tached to the unit. The tubes are placed in

their sockets, and about 10 watts of 30 Mc.

r.f. energy is fed into the plate circuit of the

amplifier, via the coaxial output plug J,. The

plate and grid circuits are resonated to the

frequency of the exciting voltage with the aid

of a grid -dip meter. Next, a sensitive r.f. volt-

meter, such as a 0 - 1 d -c milliammeter in

series with a 1N34 crystal diode is connected

to the grid input receptacle (J1) of the am-

plifier. The reading of this meter will indicate

the degree of unbalance of the neutralizing

circuit. Start with a minimum of applied r.f.

excitation to avoid damaging the meter or the

diode. Resonate the plate and grid circuits for

maximum meter reading, then vary the setting

of neutralizing capacitor G until the reading

of the meter is a minimum. Each change in G

should be accompanied by re- resonating the

grid and plate tank circuits. When a point of

minimum indication is found, the capacitor

should be locked by means of the auxiliary set

screw.

Complete neutralization is a function of the

efficiency of the screen bypass system, and

substitution of other capacitors for those noted

in the parts list is not recommended. Mica,

disc -type, or other form of bypass capacitor

should not be substituted for the units speci-

fied, as the latter units have the lowest value of

internal inductance of the many types tested

in this circuit.

Bias and The amplifier requires -60 to

Screen Supply -110 volts of grid bias, and

plus 300 to 600 volts of

screen potential for optimum characteristics

when working as a class AB1 linear ampli-

fier. Screen voltage for class C operation

(phone) is 400 volts. The voltage may be

raised to 500 volts for c.w. operation, if de-

sired, although the higher screen voltage does

little to enhance operation. Approximately -120

volts cut -off bias is required for either phone

or c -w operation. A suitable bias and screen

power supply for all modes of operation is

shown in figure 51, together with an operating

chart for all operating voltages. The supply

furnishes slightly higher than normal screen

voltage which is dropped to the correct value

by an adjustable series resistor, R1. This series

resistor is adjusted for 30 milliamperes of cur-

rent as measured in meter jack J1 when the

supply is disconnected from the amplifier.

Series bias resistor R2 is adjusted for the same

current in jack J. under the same conditions.

The value of protective bias may now be set

by adjusting potentiometer R3.

Additional bias is required for class C oper-

ation which is developed across series resistor

R.. Switch S1 is open for class C operation and

closed for sideband operation.

It is imperative that the screens of the tet-

www.americanradiohistory.com

654 H.F. Power Amplifiers T H E R A D I O

rode amplifier tubes be protected from exces-

sive current that could occur during tuning

adjustments, or during improper operation of

the amplifier. The safest way to accomplish

this is to include an overload relay that will

open the screen circuit whenever the maximum

screen dissipation point is reached. Two 4-

250A tubes or 4 -400A tubes have a total screen

dissipation rating of 70 watts, therefore relay

RY -1 should be adjusted to open the screen

circuit whenever the screen current reaches

approximately 100 milliamperes.

29 -12 A 2- Kilowatt

P.E.P. All -Band

Amplifier

Described in this section is a deluxe all -

band linear amplifier suited for s.s.b. or c.w.

operation up to the maximum legal power

limit. A 4CX1000A ceramic power tetrode is

employed in a basic passive grid circuit shown

earlier in this chapter in figure 11C.

The 4CX1000A is a ceramic and metal,

forced air -cooled, radial beam tetrode with a

rated maximum plate dissipation of 1000

watts. It is a medium voltage, high current

tube specifically designed for Class AB1 r.f.

linear amplifier service where its high gain

and low distortion characteristics may be used

to advantage. At the maximum rated plate

voltage of 3000, the tube is capable of 1680

watts p.e.p. output in sideband service. Maxi-

mum grid dissipation of the 4CX1000A is

zero watts. The design features which make

the tube capable of maximum power opera-

tion without driving the grid into the positive

region also makes it necessary to avoid positive

grid operation.

This efficient amplifier covers the 3.5 -29.7

Mc. amateur range and may be driven by any

modern sideband exciter having a power out-

put of 75 watts, p.e.p. In addition to sideband

operation, the amplifier may be used as an

a.m. linear, providing a carrier power of about

350 watts.

Circuit Description The circuit of this all -

band amplifier is shown

in figure 53. A resistance loaded, passive grid

configuration is employed in conjunction with

a pi- network output circuit. Grid drive re-

quirements are about 60 volts peak, developed

across resistor R1 which has a value of 50

ohms. This corresponds to approximately 72

watts p.e.p., all of which is dissipated in the

grid resistor. Average power dissipated in this

resistor is about 30 watts under voice wave-

form conditions. It is possible to tune up and

adjust the exciter with the plate and screen

voltages removed from the 4CX1000A, using

this resistor as a dummy load.

The amplifier plate circuit is the popular

pi- network configuration employing a tapped

Figure 52

DELUXE 4CX1000A

SIDEBAND

AMPLIFIER

Constructed in a desk-

top cabinet, this 5 -bond

sideband amplifier is

rated at 2 kilowatt p.e.p.

level. Panel controls are

(I. to r.): meter switch,

plate tuning (top), fila-

ment and plate switches

and pilots (center), plate

loading control (bottom)

and bandswitch.

www.americanradiohistory.com

HANDBOOK 4CX 1000A Amplifier 655

RF

IN .001

1 zKV

4CX1000A EA 500

R FC 2

= 0.1

500.

SEE TEXT

tKx42

W

E'

E

ADJ.

B /AS 15+ HV

rDW

,0K

,ow

R

5000

2w

E'

S,*

l

SIB

0 -1

2

-BIAS A.C. PIL

HEUT SW B+

ON 5V. B+

N SCR.

Figure 53

SCHEMATIC, 4CX1000A AMPLIFIER

C1 -500 µµtd., 5 kv. Jennings Radio Co., type

UCSL

C2 -2000 NF,td., 2 kv. Jennings Radio Co.,

type UCSL

L,- Barker 8 Williamson x.852 turret

RFC,-2.5 mh. National R -100

RFC2- Transmitting type r.f. choke. National

R -175A

T, -6.0 volts @ 11 a. Stancor P -6463

Blower -S0 cu. ft. min. Ripley »81 or equi-

valent

coil and variable vacuum capacitors. A simple

diode voltmeter is used to monitor the r.f.

output voltage of the amplifier. The network

is capable of matching antenna loads of 50 -75

ohms, which exhibits an s.w.r. of less than 2/1.

Metering and Control Circuits. The amplifier

unit contains two panel meters (figure 54) .

A 0 -1 d.c. ammeter placed in the B -minus leg

of the power supply serves as a plate meter.

The second meter is a 0 -1 d.c, milliameter

connected as a multi -purpose indicator. Panel

switch S1 places the meter across shunt and

multiplier resistors in various circuits.

The basic control circuits are shown in

figure 55. A "fail- safe" design utilizes control

B

B- GND.

NOTE

R F OUTPUT

RF OUT

MULTIMETER SWITCH S1

A - 0 -, MA., GRID

B- 0 -1 MA., PF OUT.

C - 0 -40 MA , SCREEN

D- 0- 500V., 111, SCREEN

e-0 -5KV., DA PLATE

0.1 CAPACITORS ARE SPRAGUE

NYPASS 80P -3

relays energized from low voltage d.c. supplies.

If one of the supplies fails, or a relay becomes

inoperative for any reason, the 4CX1000A

tube is protected from abuse. Inexpensive 115

volt a.c. relays are used, which operate satis-

factorily from a d.c. source of about 30 volts.

Series resistors may be used with the relay

coils to provide the correct pull -in voltage.

Relay RY, is the main power relay. When the

"Filament On" switch on the amplifier is

thrown, the bias supply ( -150 volts) is ener-

gized. Power is applied to relay RY1 through

the overload relay contacts (RY_B) and the

time delay relay, TD. For usual voice opera-

tion, the plate supply is left on at all times.

Relay RY1 may be released by the overload

relay RY., whose actuator coil is placed in

series with the B -minus lead of the high volt-

age supply. The overload relay is adjusted to

trip at a plate current of approximately 850

ma. Once the relay is tripped, it is reset by

the auxiliary reset coil by momentarily throw-

ing off the filament switch.

Screen voltage of the 4CX1000A is ob-

tained from the high voltage supply through

www.americanradiohistory.com

656 H.F. Power Amplifiers T H E R A D I O

an adjustable dropping resistor and is con-

trolled by two OD3 regulator tubes. With this

regulator and divider combination, the screen

voltage is stabilized at 300 volts, yet a maxi-

mum of only 10 watts may be drawn from the

supply. This protects the 4CX 1000A under any

operating conditions, as the maximum screen

dissipation rating is 12 watts. In the event the

plate supply fails, the tube is protected from

screen overload, as the screen voltage is also

removed at the same time. In case of bias

failure, the plate circuit relay is de- energized,

as it receives power from the bias supply.

The screen current of the 4CX1000A varies

over a wide range, depending upon the tube

operating conditions, and may approach

Figure 54

HINGED FRONT PANEL REVEALS

BIAS AND VOLTMETER

ADJUSTMENTS

The main panel is hinged at the lower edge,

and is cut out to permit meter switch and

band switch knobs to project through. Special

dial plates are cut from lucite and engraved

for the two controls. At left, the two control

potentiometers are mounted below meter

switch.

moderate negative values if the tube is lightly

loaded. It is convenient, therefore, to be able

to monitor negative values of screen current.

A bleeder resistor is placed directly at the tube

socket after the screen meter shunt. With 300

volts applied to the screen, this resistor is

adjusted to provide a static reading of 20

milliamperes on the meter. Thus, 20 ma. must

be subtracted from the meter reading to ob-

tain the actual screen current. When the screen

current is negative, the meter reading will drop

below 20 milliamperes. A negative screen

current of 18 ma., for example, will result in

an indication of plus 2 ma. on the meter.

Negative screen currents as great as -20 ma.

can be monitored in this manner.

The amplifier is cut off during standby

periods by means of relay RY3, which boosts

the grid bias to -150 volts. This prevents the

amplifier from generating troublesome diode

noise during periods of reception. Full operat-

ing potentials are applied to the amplifier at

all times, and the amplifier is activated by re-

moving the blocking bias. Relay RY3 may be

controlled by an external voice circuit; it is

only necessary to ground pin #2 to pin #3 on

the control strip (figure 55) to activate the

www.americanradiohistory.com

HANDBOOK 4CX1000A Amplifier 657

20A. RYIA

C.C.

115V Ow--471---1 ' - 42 01

I I 3000 V.

0 O- I RY,eI SUPPLY

115 V 42301

I SEE

I TEXT

A RYI RY2e

TI r7, SRI

TO A.C.

NEUT.

1500

IÓ Ww

250

100 W EA 251c

200w

300 V.

RY2

-150V.

ADJUST

OVERLOAD

,0

25 w

20 w

OD3

RY3

TO M.V.

RECTIFIER

FIL, AROMA

O 0+3000 V.

RY2

RESET COIL

TO A.C. NEUT.

SEE TEXT I

RY2A

O O

BIAS 2

A.C.

NEUT.

J 4

FIL. e+

SW. ON 115V. e+

SCR. e-

SEE

TEXT

RY3

SRz 1ón -

5w

- e

250 V.

e

GND.

CONT

STRIPO

O

2 3

TO CONTROL SW.,

ANT. RELAY, ETC.

Figure 55

SCHEMATIC, AMPLIFIER CONTROL CIRCUIT

RY1- Primary control relay. 115 volt a.c. coil or d.c. coil chosen to work with bias supply voltage.

13 ampere contacts. Potter it Brumfield type PR7AY

RY3- Overload relay, l IS volt reset coil. Potter 8 Brumfield type GCIIA

RY3 -SPST, 115 volt a.c. relay. Potter B Brumfield type KLSA

SR1,2 -500 ma. rectifier. Sarkes- Tarzian M -500

T1 -150- 160 -170 volts @ 0.S amp. Triad R -93A Set at 170 volt tap

TO- Thermal delay unit. Amperite 115 -NO -180

amplifier. The coil of the antenna relay may

be placed in parallel with that of relay RY3

for completely automatic voice operation.

Amplifier

Construction This amplifier is an excellent

example of high -grade amateur

construction. The unit is housed

within an aluminum cabinet measuring 10"

high, 15" wide and 151/2" deep. A false front

panel, hinged at the lower edge (figure 54)

is employed for decorative purposes. An auxi-

liary panel is placed behind this, holding the

panel meters, control switches and the counter

dials (figure 56). This auxiliary panel is

spaced in front of the amplifier enclosure

(figure 57) . Electrical connections between

the amplifier and the auxiliary panel equip-

ment are made by means of two disconnect

plugs, permitting the auxiliary panel to be

wired and tested as a complete sub -assembly,

or to be removed for servicing.

Placement of the major components within

the amplifier box may be seen in the top view,

figure 58. No chassis is used, other than two

shield boxes which enclose the tube socket and

the power receptacles on the rear of the

cabinet.

The 4CX1000A tube is mounted on the top

of an aluminum box measuring 6" square and

4" high. Only four connections pass into this

compartment: Filament, screen, and bias leads

(via coaxial capacitors seen in figure 57);

and the excitation lead (via a coaxial plug and

receptacle placed beneath the blower motor).

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658 H.F. Power Amplifiers T H E R A D I O

Power and control leads from the shielded

receptacle at the rear of the cabinet pass

through a 3/4" aluminum conduit tube to the

various circuitry mounted on the auxiliary

front panel. The high voltage lead leaves the

shield box via a short length of copper tub-

ing to the bottom of the plate r.f. choke,

which is supported from the rear wall of the

cabinet.

The variable vacuum capacitors are mounted

to the tube socket assembly box by means of

a heavy aluminum bracket, and are driven by

the counter dials through flexible shaft

couplers.

The space between the front panel and the

enclosure is about 31/4" and the filament

transformer is mounted in the lower right por-

tion of this area (figure 57). The enclosure

is sealed by a hinged lid, which is TVI- proofed

by phosphor- bronze finger stock fastened

around the edge of the cabinet opening.

The passive grid resistor (R1) is made up

of nine 470 -ohm, 2 -watt composition resistors

Figure 56

REAR VIEW OF METER PANEL

Counter dial mechanisms, pilot lamps, meters

and switches are mounted on aluminum sub -

panel. Meter switch, potentiometers, and meter

resistors are mounted on phenolic panel at

lower right. Panels have disconnect plugs so

that they may be wired separately.

placed in parallel (figure 59) . The resistor

leads are clipped short, and the units are

mounted between two copper discs, 11/4" in

diameter. The assembly is enclosed in a copper

tube measuring 11/2" inside diameter, 21/2"

long, and having a 1/16" wall. After the re-

sistor assembly is completed, it is bolted to

a plate which fastens to one end of the tube.

The container is then filled with transformer

oil through a vent hole in the top. When it

is full, it is placed in a pan of water which

is heated to the boiling point. The oil expands

and drives the air out through the second vent

hole. Before the unit cools, the vent holes are

closed with solder. This compact assembly will

handle up to nearly 100 watts on an inter-

mittent basis.

If it is desired to make a less complex re-

sistor assembly, thirty 1500 -ohm, 2 -watt com-

position resistors may be connected in parallel

between two copper plates, in the manner

shown in the photograph. This arrangement is

cooled by the flow of air past the resistors.

Amplifier Adjustment Before power is applied

and Tuning to the amplifier, fila-

ment voltage should be

adjusted to provide 6.0 volts at the socket of

the 4CX 1000A. Voltage should be held within

plus -or -minus 5 percent for maximum tube

life, so an accurate voltmeter is required for

this check.

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HANDBOOK 4CX 1000A Amplifier 659

Figure 57

MAIN PANEL OF

AMPLIFIER

Meter multiplier and filament transformer (right) are mounted to the main panel. At left are

feedthrough capacitors for various supply leads. Disconnect plugs to auxiliary panel are at center.

Bandswitch shalt bracket is mounted to top of transformer.

The amplifier is attached to a dummy an-

tenna or other r.f. load. The sideband exciter

may now be tuned and loaded, using the pas-

sive input resistor of the amplifier as a dummy

load. The filament of the 4CX1000A is turned

off, and high voltage applied to the amplifier.

The reading of the screen current meter is

noted, and the high voltage turned off and the

screen bleeder adjusted until the meter indi-

cates 10 milliamperes of bleeder current.

The filament is now turned on, and the

plate voltage applied and checked. The "Ad-

just Bias" potentiometer at the rear of the am-

plifier is set to provide a static plate current

reading of 0.3 ampere. (Note that 60 ma. of

indicated meter reading is current drawn by

the screen regulator tubes and bleeder. This

current is constant, regardless of plate current,

and must be subtracted from the meter reading

to obtain true plate current.)

Next, apply a small carrier signal to the

amplifier to increase the plate current indica-

tion by about 50 ma. A large value of negative

screen current will be noted. Quickly set the

loading capacitor to full scale, and adjust the

plate tank capacitor for plate current dip,

which will be very small.

Monitor the screen current and advance the

grid drive until about plus 10 to 20 milli-

amperes of screen current are noted. Decrease

the capacitance of the loading capacitor (in-

crease loading) slowly, and observe that the

screen current decreases as the loading in-

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660 H.F. Power Amplifiers TH E RADIO

creases. Screen current will approach zero, and

perhaps go slightly negative. Re- resonate plate

tank, and increase excitation until plate cur-

rent reaches 0.75 ampere. Screen current can

be adjusted by alternately varying grid drive

Figure 58

TOP VIEW OF AMPLIFIER CABINET

Placement of the major components may be

seen in this view. At left is 4CX1000A tube

and socket, with blower immediately behind

it. Atop the blower are the plate circuit ri.

choke and blocking capacitors. Ten meter sec-

tion of tank coil is mounted in a vertical

position behind vacuum capacitor, which in

turn is mounted to tube enclosure. At right

is the tank inductor, with an auxiliary switch

deck (not used) mounted on rear of assembly.

This deck may be employed to switch antenna

relays. Lid of cabinet is perforated to provide

ventilation. Air intake is on left side of the

cabinet beside blower cage.

and antenna loading. The sequence of events

is to tune, load, and vary the drive until a

plate meter reading of 0.75 ampere is achieved,

with an indicated screen current of approxi-

mately 0 to plus 20 ma. When excitation is

removed, screen current will drop to about

18 ma. This indicates a true resting screen

current of -2 ma., plus a bleeder current of

20 ma. Grid current, of course, is zero.

The carrier signal may now be removed, and

voice excitation applied to the amplifier. Plate

current may be "talked" up to about 0.38 am-

pere on voice peaks. True screen current will

run -5 ma. to plus 20 ma. on voice peaks,

depending upon the degree of loading and the

exact ratio between loading and grid drive.

Under optimum conditions, screen current will

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HANDBOOK 3 -1000Z Amplifier 661

rest at 8 ma., and drop to about -2 ma. under

voice waveforms.

29 -13 A 3 -1000Z

Linear Amplifier

The 3 -1000Z is a compact power triode in-

tended to be used as a zero bias class B am-

plifier in audio or r.f. applications. Grounded

grid linear service is especially attractive, as

full legal input may be run at a plate potential

of only 2500 volts, yet the power gain of the

tube is high enough to allow sideband exciters

of the "100 watt" -type to drive it to full out-

put. Neutralization of the grounded grid stage

is not necessary, as the excellent internal shield-

ing of the tube reduces intra -stage feedback

to a minimum. Distortion products of this

amplifier are better than 35 decibels below

maximum p.e.p. level. A tuned cathode tank

circuit is employed in order to obtain greatest

linearity and power output.

Amplifier Circuit The 3 -1000Z amplifier

covers all amateur bands

between 3.5 Mc. and 29.7 Mc. with generous

overlaps. Bandswitching circuits are used, and

the unit is designed to operate with unbalanced

coaxial antenna systems having an s.w.r. of less

than 2/1. The complete schematic is given

in figure 61. A high -C, bandswitching cathode

circuit is used for best linearity (figure 62) .

The driving impedance of the 3 -1000Z is ap-

proximately 55 ohms, providing a close match

to the great majority of sideband exciters. The

"flywheel effect" of the cathode tank prevents

input waveform distortion caused by the half -

cycle loading of the class B amplifier. Filament

voltage is fed to the tube via a shunt choke

(L2) placed in parallel with the tuned circuit.

The cathode coil is tapped for the various

amateur bands, and extra shunt capacity is

placed in the circuit to maintain the proper

C/L ratio at 3.5 Mc.

Plate current metering is accomplished in

the B -minus lead to the power supply to

remove dangerous voltages from the meter

movement. The meter is shunted with a wire -

wound resistor as a safety measure. For stand-

by operation, the cathode to ground return of

the stage is opened by means of the voice

relay, and a small amount of idling current

flowing through a 50K cathode resistor pro-

vides sufficient bias to prevent annoying diode

noise from being generated during listening

periods. The voice relay shorts out the resistor

to allow normal operation of the stage.

It is necessary to "unground" the grounded

grid sufficiently to permit measurements of

grid current to be made. The 3 -1000Z has

Figure 59

NONINDUCTIVE

50 OHM GRID

RESISTOR

Nine 2 -watt composition

resistors are immersed in

oil bath to provide high -

peak level of dissipation.

Exciter may be tuned up

using this resistor as a

dummy load, if desired.

ii 1(

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662 H.F. Power Amplifiers

three grid pins, and each corresponding socket

terminal is bypassed to ground with a low

impedance r.f. shunt made of a 4.7 -ohm com-

position resistor and a 0.01 pfd., 1.2 kv.

ceramic disc capacitor connected in parallel.

Figure 60

2 KILOWATT P.E.P. GROUNDED -

GRID LINEAR AMPLIFIER

The Eimac 3 -10002 zero bias triode tube is

designed for grounded -grid linear amplifier

service, and is capable of full input at a

plate potential as low as 2500 volts. This

3 -1000Z amplifier covers all amateur bands

between 10 and 80 meters. Panel meters and

controls are (I. to r.): plate, Arid and output

meter; plate tuning (center); bandswitch;

cathode bandswitch and tuning (lower left);

antenna loading (center) and output volt-

meter adjustment. Complete amplifier is en-

closed in screen made of perforated Reynolds

aluminum sheet.

The voltage drop across the resulting resistance

(1.6 ohms) is measured by a simple d.c. volt-

meter made up of a 0 -1 d.c. milliammeter

with a series multiplier chosen to provide a

full scale reading when 300 ma. of grid cur-

rent develops 0.64 volts across the shunt. The

internal resistance of the meter is subtracted

from the value of the required series multiplier.

A pi- network tank circuit is used, with an

additional loading capacitor that can be

switched in the circuit to match low impedance

antenna loads commonly encountered on the

80 meter band. In addition, a diode voltmeter

is included to monitor the output voltage of

the amplifier.

The 3 -1000Z requires forced -air cooling to

maintain the base seals at a temperature below

200 °C, and the plate seal at a temperature

www.americanradiohistory.com

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www.americanradiohistory.com

664 H.F. Power Amplifiers THE RADIO

below 225 °C. When using the Eimac SK -510

Air Socket and SK -516 Chimney, a minimum

air flow of 25 cubic feet per minute is re-

quired to provide adequate cooling. In order

to overcome back pressure, a blower of twice

this capacity is recommended as most small

blowers lose efficiency rapidly when run with

any degree of back pressure. Cooling air must

be supplied to the tube even when the fila-

ment alone is on during standby periods.

When a socket other than the SK -510 is used,

provisions must be made for equivalent cool-

ing of the base, envelope, and the plate lead.

The 3 -1000Z amplifier is designed for

operation over the plate potential range of

2500 to 3000 volts. Operation above 3500

volts is not recommended with this circuit, as

the high value of static plate current boosts

the resting plate current and plate dissipation

to abnormally high values.

Amplifier The linear amplifier is con -

Construction strutted upon a 14" x 17" x

4" aluminum chassis. The front

panel measures 141/2" high and

171/2" wide and is cut from a standard relay

rack panel. If the amplifier is to be rack

mounted, the panel need not be cut down in

width. A solid aluminum bottom plate pres-

surizes the under -chassis area, and the above -

chassis components are enclosed in a per-

forated aluminum shield cover, which is sup-

ported by a framework made of 1/2-inch alu-

minum angle stock. The counter dial and panel

meters are isolated from the r.f. circuits by a

sub -panel which completes the r.f. enclosure.

The pi- network output loading capacitor

(C3) is mounted beneath the chassis in a

small aluminum box. When the bottom plate

is in place, the box isolates the capacitor from

the nearby input circuitry. Connection from

the capacitor to the plate inductor is made

via a feed- through insulator placed in the

chassis deck.

Shown in figure 62 is the tuned cathode

circuit, which is assembled upon a small alu-

minum plate and mounted beneath the chassis

in an upside -down position. Because of the

extremely high C/L ratio, placement of taps

on coil L1 is fairly critical. It is therefore re=

commended that the circuit be grid- dipped to

the center of each amateur band with the tun-

ing capacitor pre -set to the value of capaci-

tance given in figure 61. Placement of coil

Figure 62

TUNED CATHODE

CIRCUIT FOR

G -G AMPLIFIER

Flywheel effect of high -

C cathode tank circuit

prevents waveform dis-

tortion caused by half -

cycle input loading of

class B linear stage. Com-

ponents are mounted on

small L- shaped bracket

which is placed near in-

put circuit of the tube.

Coil is mounted on two

1/2-inch ceramic insula-

tors.

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HANDBOOK 3 -1000Z Amplifier 665

Figure 63

TOP VIEW OF 3 -1000Z LINEAR

AMPLIFIER

Sub -panel construction is employed to isolate

panel meters from strong r.f. field of ampli-

fier plate circuit. The tube (chimney removed

for photograph) is at left rear, with blower

motor directly in front. Blower is mounted on

cork pad to reduce noise. Vacuum tuning

capacitor is panel -driven by counter dial and

flexible coupling, and is mounted on heavy

aluminum bracket. Plate coupling capacitors

are affixed to rear stator of capacitor by alu-

minum angle plate. Plate r.f. choke is directly

behind capacitor. Heavy -duty tank coil is at

right, with the auxiliary 10 -meter coil sup-

ported by the rear flange of the capacitor.

taps is critical to within 1/8 turn in order to

achieve optimum C/L ratio. A second seg-

ment of the bandswitch is used to parallel the

tuning capacitor with a high voltage mica

unit to provide sufficient capacitance for 80

meter operation. A transmitting -type capaci-

tor is used to handle the high value of r.f.

current flowing in this circuit. A common

ground connection is made at one end of the

tuning capacitor and the shield of the RG-

58/U coaxial line from the input receptacle

is terminated at this point.

The tuned circuit is coupled to the filament

of the 3 -1000Z through a 0.02 pfd., 1.2 kv.

mica capacitor which is capable of handling

the r.f. current peaks. All leads in this circuit

are short, heavy and direct, as it is essential

www.americanradiohistory.com

666 H.F. Pcwer Amplifiers THE RADIO

.ole J111111.

to keep the circuit inductance in the coil where

it belongs, rather than in the leads and switch

connections.

Filament connections to the tube socket re-

ceptacles are made by means of flexible braid,

and the grid shunts are attached to the recep-

tacles so that the latter are capable of move-

ment. No lateral pressure should be applied

to the tube pins, so it is essential that the

socket receptacles be free to move about to

conform with the pin alignment.

Figure 64

UNDER -CHASSIS VIEW OF

AMPLIFIER

The pi- network output capacitor is placed in

a shield box at front center of the chassis.

To the left is the cathode tank assembly,

mounted in place on metal spacers. The

3 -10002 air socket is in the corner, with grid

R -C bypass networks placed directly between

socket pins and ground. Use of the new inex-

pensive Eimac SK -510 Air Socket should be

made in place of the SK -500 (shown here).

Filament choke is at center of chassis, with

the filament transformer and primary voltage

control transformer at the right.

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HANDBOOK 3 -1000Z Amplifier 667

c

Figure 65

REAR VIEW OF LINEAR AMPLIFIER

Power leads pass into chassis enclosure via Sprague "Hypass" capacitors. Terminals (I. to r.) ore:

filament center tap, 115 volt primary terminals, plate voltage negative return, B -plus terminal,

and coaxial input receptacle.

A small auto -transformer may be placed in

the primary filament circuit to adjust the volt-

age to the correct value. All low voltage leads

in the filament and metering circuits pass

through flexible braid, and each lead is by-

passed at both ends by means of 0.001 pfd.

ceramic capacitors affixed between the lead

and the braid.

Components of the diode voltmeter are

mounted on a small ceramic board fastened to

the inner side wall of the chassis near the

coaxial output receptacle.

Amplifier Adjustment After wiring is com-

pleted and checked, the

cathode and plate tuned circuits should be

grid- dipped to the operating frequency. A low

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668 H.F. Power Amplifiers

value of plate voltage is applied, and suffi-

cient carrier (or steady tone) inserted into

the exciter to raise the amplifier plate current

by approximately 100 ma. The cathode circuit

is resonated for maximum grid current, and

plate circuit resonance is determined. Plate

voltage is now raised to the operating value

and excitation and loading are adjusted until

the desired input level is reached. For example,

at a plate potential of 2500 volts and a single

tone (carrier) source of drive, maximum plate

current should be 800 ma., and grid current

will be about 250 ma. This ratio of about

3 ma. of plate current for 1 ma. of grid cur-

rent should be held regardless of the loading

level. This will insure that the proper propor-

tion between grid drive and antenna loading

is maintained. Finally, the amplifier should be

overcoupled to the antenna until r.f. output

drops about 2 percent to attain a condition of

maximum linearity. Under voice conditions,

peak plate current (as read on the meter)

will approximate 350 to 400 ma., and grid

current will be about one -third this value.

Intermodulation products will be better than

35 decibels below p.e.p. level.

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CHAPTER THIRTY

Speech and

Amplitude Modulation Equipment

Amplitude modulation of the output of a

transmitter for radiotelephony may be accom-

plished either at the plate circuit of the final

amplifier, commonly called high -level AM or

simply plate modulation of the final stage, or

it may be accomplished at a lower level. Low -

level modulation is accompanied by a plate -

circuit efficiency in the final stage of 30 to 45

per cent, while the efficiency obtainable with

high -level AM is about twice as great, running

from 60 to 80 per cent. Intermediate values of

efficiency may be obtained by a combination

of low -level and high -level modulation; cath-

ode modulation of the final stage is a common

way of obtaining combined low -level and high -

level modulation.

High -level AM is characterized by a require-

ment for an amount of audio power approxi-

mately equal to one -half the d -c input to the

plate circuit of the final stage. Low -level mod-

ulation, as for example grid -bias modulation of

the final stage, requires only a few watts of

audio power for a medium power transmitter

and 10 to 15 watts for modulation of a stage

with one kilowatt input. Cathode modulation

of a stage normally is accomplished with an

audio power capability of about 20 per cent

of the d -c input to the final stage. A detailed

discussion of the relative advantages of the

different methods for accomplishing amplitude

modulation of the output of a transmitter is

given in an earlier chapter.

Two trends may be noted in the design of

systems for obtaining high -level AM of the

final stage of amateur transmitters. The first

is toward the use of tetrodes in the output

stage of the high -power audio amplifier which

is used as the modulator for a transmitter. The

second trend is toward the use of a high -level

splatter suppressor in the high- voltage circuit

between the secondary of the modulation trans-

former and the plate circuit of the modulated

stage.

30 -1 Modulation

Tetrode In regard to the use of tetrodes,

Modulators the advantages of these tubes

have long been noted for use in

modulators having from 10 to 100 watts out-

put. The 6V6, 6L6, and 807 tubes have served

well in providing audio power outputs in this

range. Recently the higher power tetrodes such

as the 4 -65A, 813, 4 -125A, and 4 -250A have

come into more general use as high -level au-

dio amplifiers. The beam tetrodes offer the

advantages of low driving power (even down

to zero driving power for many applications)

as compared to the moderate driving power re-

quirements of the usual triode tubes having

equivalent power- output capabilities.

On the other hand, beam tetrode tubes re-

quire both a screen -voltage power supply and

a grid -bias source. So it still is expedient in

many cases to use zero -bias triodes or even

669

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670 Speech and A. M. Equipment T H E R A D I O

low -mu triodes such as the 304TL in many

modulators for the medium -power and high -

power range. A list of suggested modulator

combinations for a range of power output capa-

bilities is given in conjunction with several

of the modulators to be described.

Increasing the It has long been known

Effective Modu- that the effective modu-

lation Percentage lation percentage of a

transmitter carrying un-

altered speech waves was necessarily limited

to a rather low value by the frequent high -

amplitude peaks which occur in a speech

waveform. Many methods for increasing the

effective modulation percentage in terms of

the peak modulation percentage have been

suggested in various publications and subse-

quently tried in the field by the amateur

fraternity. Two of the first methods suggested

were Automatic Modulation Control and Vol-

ume Compression. Both these methods were

given extensive trials by operating amateurs;

the systems do give a degree of improvement

as evidenced by the fact that such arrangements

still are used in many amateur stations. But

these systems fall far short of the optimum be-

cause there is no essential modification of the

speech waveform. Some method of actually

modifying the speech waveform to improve the

ratio of peak amplitude to average amplitude

must be used before significant improvement

is obtained.

It has been proven that the most serious ef-

fect on the radiated signal accompanying over -

modulation is the strong spurious -sideband ra-

diation which accompanies negative -peak clip-

ping. Modulation in excess of 100 per cent in

the positive direction is accompanied by no

undesirable effects as far as the radiated sig-

nal is concerned, at least so long as the linear

modulation capability of the final amplifier is

not exceeded. So the problem becomes mainly

one of constructing a modulator -final amplifier

combination such that negative -peak clipping

(modulation in excess of 100 per cent in a

negative direction) cannot normally take place

regardless of any reasonable speech input

level.

Assymetrical The speech waveform of the

Speech normal male voice is charac-

terized, as was stated before,

by high -amplitude peaks of short duration. But

it is also a significant characteristic of this

wave that these high -amplitude peaks are poled

in one direction with respect to the average am-

plitude of the wave. This is the "lopsided" or

assymetrical speech which has been discussed

and illustrated in an earlier chapter.

The simplest method of attaining a high

average level of modulation without negative

peak clipping may be had merely by insuring

that these high -amplitude peaks always are

poled in a positive direction at the secondary

of the modulation transformer. This adjust-

ment may be achieved in the following man-

ner: Couple a cathode -ray oscilloscope to the

output of the transmitter in such a manner that

the carrier and its modulation envelope may

be viewed on the scope. Speak into the micro-

phone and note whether the sharp peaks of

modulation are poled upward or whether these

peaks tend to cut the baseline with the "bright

spot" in the center of the trace which denotes

negative -peak clipping. If it is not obvious

whether or not the existing polarity is correct,

reverse the polarity of the modulating signal

and again look at the envelope. Since a push -

pull modulator almost invariably is used, the

easiest way of reversing signal polarity is to

reverse either the leads which go to the grids

or the leads to the plates of the modulator

tubes.

When the correct adjustment of signal po-

larity is obtained through the above procedure,

it is necessarily correct only for the specific

microphone which was used while making the

tests. The substitution of another microphone

may make it necessary that the polarity be re-

versed, since the new microphone may be con-

nected internally in the opposite polarity to

that of the original one.

Low -Level The low -level speech clip-

per is, in the ideal case, a

very neat method for ob-

taining an improved ratio of average -to -peak

amplitude. Such systems, used in conjunction

with a voice -frequency filter, can give a very

worthwhile improvement in the effective mod-

ulation percentage. But in the normal amateur

transmitter their operation is often less than

ideal. The excessive phase shift between the

low -level clipper and the plate circuit of the

final amplifier in the normal transmitter re-

sults in a severe alteration in the square -wave

output of the clipper -filter which results from

a high degree of clipping. The square -wave

output of the clipper ends up essentially as a

double saw -tooth wave by the time this wave

reaches the plate of the modulated amplifier.

The net result of the rather complex action of

the clipper, filter, and the phase shift in the

succeeding stages is that the low -level speech

clipper system does provide an improvement

Speech Clipping

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HANDBOOK Design 671

MODULATOR R F FINAL

5R4 -GY

OR 836

2

*B B I A C.

MOD. R F 115V

FINAL

Figure 1

HIGH -LEVEL SPLATTER SUPPRESSOR

The high- vacuum diode acts as a series lim-

iter to suppress negative -peak clipping in

the modulated r -f amplifier as a result of

large amplitude negative -peak modulating

signals. In addition, the low -pass filter fol-

lowing the diode suppresses the transients

which result from the peak- clipping action of

the diode. Further, the filter attenuates all

harmonics generated within the modulator

system whose frequency lies above the cut-

off frequency of the filter. The use of an

appropriate value of capacitor, determined

experimentally as discussed in Chapter

Fifteen, across the primary of the modulation

transformer (C ) introduces further attenua-

tion to high -frequency modulator harmonics.

Chokes suitable for use at L are manufactured

by Chicago Transformer Corp.. The correct

values of capacitance for Cr, C,, Cr, and C

are specified on the installation sheet for the

splatter suppressor chokes for a wide variety

of operating conditions.

in the effective modulation percentage, but it

does not insure against overmodulation. An

extensive discussion of these factors, along

with representative waveforms, is given in

Chapter Fifteen. Circuits for some recommend-

ed clipper- filter systems will also be found

in the same chapter.

High -Level One practicable method

Splatter Suppressor for the substantial elim-

ination of negative -peak

clipping in a high -level AM transmitter is the

so- called high -level splatter suppressor. As

figure 1 shows it is only necessary to add a

high- vacuum rectifier tube socket, a filament

transformer and a simple low -pass filter to an

existing modulator -final amplifier combination

to provide high -level suppression.

The tube, V,, serves to act as a switch to

cut off the circuit from the high- voltage power

supply to the plate circuit of the final ampli -

fer as soon as the peak a -c voltage across

the secondary of the modulation transformer

has be -orne equal and opposite to the d -c volt-

age be'ng applied to the plate of the final am-

plifier stage. A single- section low -pass filter

serves to filter out the high- frequency compo-

nents result ng from the clipping action.

Tube V, may be a receiver rectifier with a

5 -volt filament for any but the highest power

transmitters. The 5Y3 -GT is good for 125 ma.

plate current to the final stage, the 5R4 -GY

and the 5U4 -G are satisfactory for up to 250

ma. For high -power high -voltage transmitters

the best tube is the high- vacuum transmitting

tube type 836. This tube is equivalent in

shape, filament requirements, and average -cur-

rent capabilities to the 866A. However, it is

a vacuum rectifier and utilizes a large -size

heater -type dual cathode requiring a warm -up

time of at least 40 seconds before current

s hou 1 d be passed.. The tube is rated at an

average current of 250 ma. For greater current

drain by the final amplifier, two or more 836

tubes may be placed in parallel.

The filament transformer for the cathode of

the splatter- suppressor tube must be insulated

Figure 2

TOP VIEW OF THE

6L6 MODULATOR

www.americanradiohistory.com

Figure 3

UN DERCHASSIS

OF THE

6L6 MODULATOR

A 4- connector plug is

used for filaments and

p la t e voltage to the

speech amplifier, while a

6 -wire terminal strip is

used for the high -voltage

connections and the trans-

mitter-control switch.

for somewhat more than twice the operating

d -c voltage on the plate modulated stage, to

allow for a factor of safety on modulation

peaks. A filament transformer of the type nor-

mally used with high -voltage rectifier tubes

will be suitable for such an application.

30 -2 Design of Speech

Amplifiers and Modulators

A number of representative designs for

speech amplifiers and modulators is given in

this chapter. Still other designs are included

in the descriptions of other items of equipment

in other chapters. However, those persons who

wish to design a speech amplifier or modulator

to meet their particular needs are referred to

Chapter Six, Vacuum Tube Amplifiers, for a

detailed discussion of the factors involved in

the design of such amplifiers, and for tabular

material on recommended operating conditions

for voltage and power amplifiers.

10 to 120 Watt It is difficult to surpass

Modulator with the capabilities of the

Beam Power Tubes reliable beam power

tube when an audio

power output of 10 to 120 watts is required

of a modulator. A pair of 6L6 tubes operating

in such a modulator will deliver good plate

circuit efficiency, require only a very small

amount of driving power, and they impose no

serious grid -bias problems.

Circuit Included on the chassis of the

Description modulator shown in figures 2

and 3 are the speech amplifier,

the driver and modulation transformers for the

6SJ- 615

POWER CONNECTIONS

A- GROUND

B-6 3v

C -OF250-300v

D- BIAS

E-Bf150-7S0 v

1 SEE F/GU/+E 5,

D PAIR OF RESISTORS, 1%

Figure 4

SCHEMATIC OF BEAM POWER TUBE MODULATOR

M-0 - 250 d.c. milliammeter

T,- Driver transformer. Stancor A -4701, or UTC S -10.

T:- "Poly -pedance" Modulation transformer

60 -watt level - Stancor A -3893, or UTC 5 -20

125 -watt level Stancor A -3894

www.americanradiohistory.com

General Purpose Modulator 673

output tubes, and a plate current milliammeter.

The power supply has not been included. The

6SJ7 pentode first stage is coupled through

the volume control to the grid of a 6J5 phase

inverter. The output of the phase inverter is

capacitively coupled to the grids of a 6SN7 -GT

which acts as a push -pull driver for the output

tubes. Transformer coupling is used between

the driver stage and the grids of the output

tubes so that the output stage may be operated

either as a class AB, or class AB, amplifier.

The Output

Stage Either 6L6, 6L6 -G or 807

tubes may be used in the out-

put stage of the modulator. As

a matter of fact, either 6V6 -GT or 6F6 -G

tubes could be used in the output stage if

somewhat less power output is required. The

807 tube is the transmitting -tube counterpart

of the 6L6 and carries the same ratings and

recommended operating conditions as the 6L6

within the ratings of the 6L6. But the 807

does have somewhat greater maximum ratings

when the tube is to be used for ICAS (Inter-

mittent Commercial and Amateur Service) op-

eration. The 6L6 and 807 retail to the amateur

for essentially the same price, although the 807

is available only from transmitting tube dis-

tributors. The 6L6 -G tube retails for a some-

what lower price; hence it is expedient to pur-

chase 6L6 -G tubes if 360 to 400 volts is the

maximum to be used on the output stage, or

807 tubes if up to 750 volts will be applied.

Tabulated in figure 5 are a group of recom-

mended operating conditions for different tube

types in the output stage of the modulator. In

certain sets of operating conditions the tubes

will be operated class AB1, that is with in-

FIGURE 5

RECOMMENDED OPERATING CONDITIONS FOR

MODULATOR OF FIG.4 FOR DIFFERENT TUBE TYPES

TUBES CLASS PLATE SCREEN GRID PLATE-TO PUTE POWER 1

V,.V2 VOLTS VOLTS BIAS PLATE IDAD CURRENT TPUT

(EI ICI IDI (ONM31 IMA I (WATTS)

6V6GT ABI 250 250 -15 10,000 70-80 IO

6V6GT ARI 265 245 - 19 4000 70 -95 IS

6L6 A131 360 270 -23, 6.600 85-135 27

6L6 AB2 360 270 -23 3,600 65205

1

47

807 AB 1 600 300 -34 r 10,000 35-140 56

807 A131 750 300 -35 12,000 30 -140 75

807 A132 750 300 11 -35 7,300 30-240 120

creased plate current with signal but with no

grid current. Other operating conditions speci-

fy class AB_ operation, in which the plate cur-

rent increases with signal and grid current

flows on signal peaks. Either type of operation

is satisfactory for communication work.

30 -3 General Purpose

Triode Class B Modulator

High level class B modulators with power

output in the 125 to 500 watt level usually

make use of triodes such as the 809, 811,

8005, 805, or 810 tubes with operating plate

voltages between 750 and 2000. Figures 6,

7, and 8 illustrate a general purpose modula-

tor unit designed for operation in this power

range. Figure 8 gives a group of suggested

operating conditions for various tubes. The

size of the modulation transformer will of

course be dependent upon the amount of audio

power developed by the modulator. In the case

Figure 6

REAR VIEW OF THE

GENERAL -PURPOSE MODULATOR

www.americanradiohistory.com

674 Speech and A. M. Equipment

CRYSTAL

MIC.

J1 47N

6SJ7 li7

°r GAIN

604 -G Ti V1 Tz

01 701(

3

604 -G

10 LF m 4.711

6027

6N 7

R N

1

x

1 0 U F 5 N

450 T

5Y3 -G7

2

CH,

VVO,

+350v

-, 13+1000-1250 V.

~e/AS SUPPLY IF USED,

OTHERWISE SNORT

TERMINALS.

V2

2011F 20UF

450 450

Y Y

M = MATCHED PAIR

RESISTORS, ?SG

50LF

150

115V 1.

Figure

SCHEMATIC OF GENERAL

M -0 - 500 ma.

T -Driver transformer. Stancor A -4761

T1- "Poly -pedance" Modulation transformer.

300 watt rating, Stancor A -3898.

500 watt rating, Stancor A -3899.

T, -360 - 0 - 360 volts, 150 mo. Stancor PC -8410

000'

7

PURPOSE MODULATOR

T.- Suitable for tubes used.

For 811 -A's 6.3 volt, 8 amp. Stancor P -6308

For 810's 10 volt, 10 amp. Stancor P -6461

CH -14 henry, 100 ma. UTC S -19. Stancor C -1001

R -I K, 10 watts, adjustable. Set for plate current

of 80 ma. (no signal) to 684 -G tubes

(approximately 875 ohms).

V ,V; -See figure 8.

of the 500 watt modulator ( figures 9 and 10 )

the size and weight of the components require

that the speech amplifier be mounted on a

separate chassis. For power levels of 300 watts

or less it is possible to mount the complete

speech system on one chassis.

FIGURE 8

SUGGESTED OPERATING CONDITIONS

run

TUBES PLATE C^

V1, V 2 I VOLTAGE

.

_

GRID v

BIAS

(VOLTS)

^ PLATE

CURRENT

(MA 1

v

Y PLATE TO

PLATE LOAD

(OHMS)

SINE WAVE

POWEROUTPUT'

(WATTS)

809 700 0 70 -250 6,200 120

750 0 30 -350 5,100 175

811 -A

' 811 -A 1000 0 45 -350 7,400 245

811 -A 1250 0 50 -350 9,200 310

811-A 1500 -4.5 32 -315 12,400 340

805 1250 0 14e -400 6,700 300

805 1500 -16 64 -400 13,200 370

810 2000 -50 60 -420 Iap0o 450

510 2500 -75 50 -420 17,500 500

8005 1500 -67 40-330 9,eoo 330

Circuit Description The modulator unit

of General Purpose shown in figure 6 is

Modulator complete except for the

high voltage supply re-

quired by the modulator tubes. A speech am-

plifier suitable for operation with a crystal

microphone is included on the chassis along

with its own power supply. A 6SJ7 is used

as a high gain preamplifier stage resistance

coupled to a 6N7 phase inverter. The audio

level is controlled by a potentiometer in the

input grid circuit of the 6N7 stage. Push -pull

6B4 -G low la triodes serve as the class B driver

stage. The 6B4's are coupled to the grids of

the modulator tubes through a conventional

multi -purpose driver transformer. Cathode bias

is employed on the driver stage which is capa-

ble of providing 12 watts of audio power for

the grid circuit of the modulator.

The modulator illustrated in figures 9 and

10 is designed for use with class B 810 triode

tubes operating at a plate potential of 2500

volts. Maximum audio power available is 500

watts, sufficient to 100% modulate a transmit-

ter running one kw. input. The modulator may

be driven by the simple speech amplifier of fig-

ure 12, or by the clipper- amplifier of figure 15.

www.americanradiohistory.com

Figure 9

REAR VIEW OF 810

500 -WATT CLASS B

MODULATOR

Bias control is directly be-

low the 810 tubes with bias

supply to right.

otno N

The modulator chassis includes a small low

impedance bias supply capable of delivering

approximately -75 volts under the changing

load conditions imposed by the varying grid

circuit impedance of the class B modulator.

A low pass audio filter network is placed

after the modulator stage to reduce harmonics

generated in the audio system which would

normally show up as side -band "splatter" on

the phone signal. Frequencies above 3500

cycles are attenuated 15 decibels or more by

the simple pi- network filter shown in figure

10.

SEE FIG 12 .

DRIVER

TRANSFORMER

TI

-65 TO 75 V -

5Y3.T

810

Irz

LP=1SMi p65M

c b C

R1

ADJUST

BIAS

RI-M w

C H 2 400

50W

--' 3brv,`-

" J !'oJ "2ó T -

350

tiv t"l' 'I vT"V j2

40UF

50

-)H 0

CzJ¡ 05 5 nV

r0 R-F AMP

CHI CW

- 006 C3 006 SIA

5 v TSNv

B.- 2500

Figure 10

SCHEMATIC OF 500 WATT TRIODE MODULATOR WITH BIAS SUPPLY

T,- Driver transformer, 15 watts. Stanco, A -4761

T:-Modulation transformer, 500 watt. Chicago Transformer Co. CMS -3

T,-5 volts, 3 amp. Stanco, P -6467

T.- Tapped bias transformer, 15 -100 volts. UTC S -51

T -10 volt, 10 amp. Stanco, P -6461

CH -500 ma. "splatter choke" Chicago Transformer Co. SR -500

CH: -7 henry, 150 ma. Stanco, C -1710

M -0 - 500 d.c. milliammeter

S;A -B -High voltage switch (see text)

www.americanradiohistory.com

676 Speech and A. M. Equipment T H E R A D I O

rAuD1O

osc. I1I

1SPEE

pM.I

A

500

10KV

B+

. 500V TO

OSCILLOSCOPE

Figure 11

TEST SETUP FOR 500 -WATT

MODULATOR

For c -w operation the secondary of the class

B modulation transformer is shorted out and

the filament and bias circuits of the modulator

are disabled. Switch S,A must have 10,000

volt insulation rating. A suitable switch may

be found in the war surplus BC -306A antenna

loading unit.

All low voltage connections to the modula-

tor are brought out to a six terminal phenolic

strip on the back of the chassis. A 0 -500 ma.

d -c meter is placed in the filament circuit of

the 810 tubes. The meter is placed across a

50 ohm, 1 watt resistor so that the filament

return circuit of the modulator is not broken

if the meter is removed.

The modulation transformer, T,, is designed

for plate -to -plate loads of either 12,000 ohms

or 18.000 ohms when a 6250 ohm load is

placed across the secondary terminals. The 810

tubes are correctly matched when the 18,000

ohm taps are used at a plate potential of

2500 volts, or when the 12,000 ohm taps are

used at a plate potential of 2000 volts or 2250

volts.

Modulator

Construction Because of the great weight of

the modulator components it is

best to use a heavy -duty steel

chassis. A 13" x 17" x 4" chassis (Bud CB-

643) , a 14" steel panel (Bud PS- 1257G) and

a pair of Bud MB -449 mounting brackets make

up the assembly for this particular modulator.

As seen in figure 9, the CMS -3 modulation

transformer is mounted in the left -front corner

of the unit. The secondary terminals of T: are

to the front of the chassis, clearing the front

panel by about r To the right of T: is

placed the high level audio filter choke, CH,.

The two 810 tubes are mounted in back of T:.

To the right of the 810 tubes is the 10 volt

filament transformer, T5. To the right of T5 is

the 5Y3 -GT bias rectifier tube. Between T5

and CH, are mounted the mica bypass capaci-

tors which make up the high level filter net-

work. Two .003 µfd., 5000 volt mica capacitors

are paralleled for C, and also for G. C_ is made

from a .001 µfd. capacitor and a .0015 pfd. ca-

pacitor which are connected in parallel. All

of these capacitors are mounted upon a ply-

wood bracket which insulates them from the

metal chassis. This prevents insulation break-

down within the capacitors which might occur

if they were fastened directly to the metal

chassis.

The bias supply components are mounted

beneath the four -inch deep chassis. Placement

of these parts is not critical. The bias adjust-

ment control, R,, is mounted on the back lip

of the chassis as are the two high voltage ter-

minals. Millen 37001 high voltage connectors

are used for the two high voltage leads. High -

voltage TV wire should be employed for all

leads in the 810 plate circuit.

Modulator

Adjustment When the modulator has been

wired and checked, it should be

tested before being used with

an r -f unit. A satisfactory test set -up is shown

in figure 11. A common ground lead should

be run between the speech amplifier and the

modulator. Six 1000 ohm 100 watt resistors

should be connected in series and placed across

the high voltage terminals of the modulator

unit to act as an audio load. The first step is to

place the 810 tubes in their sockets and turn

switch S, to the "phone" position. The 810

filaments should light, and switch section S.

should remove the short across the secondary

of T:. R, should be adjusted to show -75

volts from each 810 grid terminal to ground

as measured with a high resistance voltmeter.

If an oscilloscope is available, it should be cou-

pled to point "A" on the load resistor ( figure

11) through a 500 gpfd. ceramic TV capacitor

of 10,000 volts rating. The case of the oscil-

loscope should be grounded to the common

ground point of the modulator.

A plate potential of 2500 volts is now ap-

plied to the modulator, and R, adjusted for

a resting plate current of 50 milliamperes as

read on the 500 milliampere meter in the cath-

ode circuit of the modulator. Be extremely

careful during these adjustments, since the

plate supply of the modulator is a lethal weap-

on. Never touch the modulator when the plate

voltage supply is on! Be sure you employ the

TV blocking capacitor between the oscillo-

scope and the plate load resistors, as these

load resistors are at high voltage potential!

If a high resistance a -c voltmeter is available

that has a 2000 volt scale, it should be clipped

www.americanradiohistory.com

HANDBOOK 10 -Watt Amplifier- Driver 677

MAC

JACK

6B4G

5v3Gr

NOTE ADJUST FOR PLATE

CURRENT OF80 MA.

TO 084 -G TUOES

(APPROR. 673 OHMS)

65J,

6J5

6646

6046

110 V ti 11)

10,250 SW ON VOL. fen] T,

/ CONTROL

DRIVER

TRANSFORMER

(l1II K)

c_JII

cJll r ftllr7

_

MEASURE 6816 PLATE

CURRENT NE RE

I+ 1-+SSO V.

1*

T---T CA. 20,450V

ALL RESISTORS AS WATT UNLESS

OTHERWISE NOTED

Ti. 360-0 -380 VOLTS AT 150 MA

STANCOR PC -8410

T2=a:1ANTERSTAGE TRANSFORMER

STANCOR A-4210

CHI= 7-HENRY AT 140 MA.

STANCOR C-1421

Figure 12

10-WATT SPEECH AMPLIFIER DRIVER

between the high voltage Itrm,nals of the m,,,1-

ulator, directly across the dummy load. Do not

touch the meter when the high voltage supply

is in operation! An audio oscillator should be

connected to the audio input circuit of the ex-

citer- transmitter and the audio excitation to

the high level modulator should be increased

until the a -c voltmeter across the dummy load

resistor indicates an R -M -S reading that is

equal to 0.7 (70 %) of the plate voltage applied

to the modulator. If the modulator plate volt-

age is 2500 volts, the a -c meter should indi-

cate 1750 volts developed across the 6000

ohm dummy load resistor. This is equivalent

to an audio output of 500 watts. With sine

wave modulation at 1000 c.p.s. and no speech

clipping ahead of the modulator, this voltage

should be developed at a cathode meter cur-

rent of about 350 ma. when the plate -to -plate

modulator impedance of the modulator is 18,-

000 ohms. Under these conditions, the oscillo-

scope may be used to observe the audio wave-

form of the modulator when coupled to point

"A" through the 10,000 volt coupling capac-

itor. When the frequency of the audio oscillator

is advanced above 3500 cycles the output level

of the modulator as measured on the a -c volt-

meter should drop sharply indicating that the

low pass audio network is functioning properly.

With speech waveforms and no clipping the

modulator meter will swing to approximately

150 - 200 milliamperes under 100% modula-

tion at a plate potential of 2500 volts. With

speech waveforms and moderate clipping the

modulator meter will swing to about 300 ma.

under 100% modulation.

30 -4 A 10 -Watt

Amplifier - Driver

A simple speeCI) amplifier- driver for a medi-

um powered class B modulator is shown in fig-

ure 12. The amplifier is designed to work with

a crystal microphone. The first stage utilizes

a 6SJ7. The gain control is between the 6SJ7

plate circuit and the grid of the 6J5 second

stage amplifier. The output tubes are a pair

of 6B4 -G low -mu tubes operating with a self -

bias resistor in their common filament return

circuit. Operating in this manner the 6B4's

have an undistorted output of approximately

10 watts. This is sufficient power to drive

most class -B modulators whose output is 500

watts or less. The driver transformer for cou-

pling the plates of the 6B4 -G tubes to the

grids of the Class B stage is not shown, as

it had been found more convenient to locate

this transformer at the grids of the modulator

tubes rather than in the speech amplifier. The

correct transformer step -down ratio for driving

most class B tubes has been set down in tabular

form by the various transformer manufacturers.

When the driver transformer is purchased one

should be obtained which has the proper turns

ratio for the class B tubes to be used.

www.americanradiohistory.com

678 Speech and A. M. Equipment THE RADIO

CRYSTAL

M C.

JACKS. 2

1 2AX7 6A..S 12AU7

0.1

6B4 -G

3

Ta

2

330

K

EA. -T IK

0,430 V. MAX.

ADJUST

CLIPPING

T1 - 450 -0 -450 VOLTS AT 105 MA.

CHICAGO TRANSFORMER PSR -I05

T2 -CLASS S DRIVER TRANSFORMER

CHICAGO TRANSFORMER COS-1

T3-125 V. AT 15 MA.

STANCOR PS-0415

LPF-2- LOW PASS FILTER UNIT

CHICAGO TRANS. LPF -2

NO MA. A

CHIC, AISGO TRNS. RC -12130

SR -50 MA. REPLACEMENT TYPE

SELENIUM RECTIFIER 1113V ,N

FILS.

UNLESS OTHERWISE INDICATED;

ALL RESISTORS 0.3 WATT

ALL CAPACITORS IN LF

RESISTORS MARKED WITH ASTERISK K

ARE BALANCED PA /RS.

221(.2

22K,2w

000.2w

EA. AO

WS _ 250 V.

s 50

»> 1 W.

Z SR

2A. Ti

DAG 1 2 TD

ONO. 0- Figure 13

SCHEMATIC, 15 WATT CLIPPER -AMPLIFIER

TO

CLASS IS

MODULATOR

GRID CIRCUIT

A three wire shielded cable should be used

to connect the 6B4 -G tubes to the driver trans-

former located at the grids of the class B

tubes. This cable may be any reasonable

length up to 25 or 30 feet. Any of the modula-

tor configurations shown in figure 8 may be

driven with this simple speech amplifier.

30 -5 A 15 -Watt

Clipper - Amplifier

The near -ultimate in 'talk power" can be

obtained with low level clipping and filtering

combined with high level filtering. Such a

modulation system will have real "punch,"

yet will sound well rounded and normal. The

speech amplifier described in this section

makes use of low level clipping and filtering

and is specifically designed to drive a pair of

push -pull 810 modulators such as shown in

Section Three.

Circuit

Description The schematic of the speech

amplifier -clipper is shown in

figure 13. A total of six tubes,

including a rectifier are employed and the unit

delivers 15 watts of heavily clipped audio.

A 12AX7 tube is used as a two stage mi-

crophone pre -amplifier and delivers approxi-

mately 20 volts (r.m.s.) audio signal to the

6AL5 series clipper tube. The clipping level

is adjustable between 0 db and 15 db by clip-

ping control, R2. Amplifier gain is controlled

by R,, in the grid circuit of the second section

of the 12AX7. A low pass filter having a 3500

cycle cut -off follows the 6AL5 clipper stage,

with an output of 5 volts peak audio signal

under maximum clipping conditions. A double -

triode 12AU7 cathode follower phase- inverter

follows the clipper stage and delivers a 125

volt r.m.s. signal to the push -pull grids of the

6B4 -G audio driver tubes. The 6B4 -G tubes

operate at a plate potential of 330 volts and

have a -68 volt bias voltage developed by a

small selenium rectifier supply applied to their

grid circuit. An audio output of 15 watts is de-

veloped across the secondary terminals of the

class B driver transformer with less than 5 per

cent distortion under conditions of no clipping.

A 5U4 -G and a choke input filter network pro-

vide unusually good voltage regulation of the

high voltage plate supply.

Amplifier

Construction The clipper -amplifier may be

built upon the same chassis as

the power supply, provided

the low level stages of the amplifier are spaced

away from the power transformers and filter

chokes of the supply. All capacitors and resis-

tors of the audio section should be mounted as

close to the respective sockets as is practical. For

minimum hum pickup, the filament leads to the

low level stages should be run in shielded braid.

www.americanradiohistory.com

HANDBOOK 15 -Watt Clipper -Amplifier 679

Those resistors in the 12AU7 phase inverter

plate circuit and the grid circuit of the 6B4 -G

tubes should be matched to achieve best phase

inverter balance. The exact value of the paired

resistors is not important, but care should be

taken that the values are equal. Random re-

sistors may be matched on an ohmmeter to find

two units that are alike in value. When these

matched resistors are soldered in the circuit,

care should be taken that the heat of the sol-

dering iron does not cause the resistors to

shift value. The resistors should be held firm-

ly by the lead to be soldered with a long nose

pliers, which will act as a heat -sink between

the soldered joint and the body of the resistor.

If this precaution is taken the two phase in-

verter outputs will be in close balance.

Adjustment of When the wiring of the

the Speech speech amplifier has been

Amplifier completed and checked, the

unit is ready to be tested.

Before the tubes are plugged in the amplifier,

the bias supply should be energized and the

voltage across the 600 ohm bleeder resistor

should be measured. It should be -68 volts.

If it is not, slight changes in the value of the

series resistor, Rr, should be made until the

correct voltage appears across the bleeder re-

sistor. The tubes may now be inserted in the

amplifier and the positive and cathode voltages

checked in accordance with the measurements

given in figure 13. After the unit has been

tested and is connected with the modulator,

R2 should be set so that it is impossible to over -

modulate the transmitter regardless of the set-

ting of R1. The gain control (R.) may then be

adjusted to provide the desired level of clip-

ping consistent with the setting of R2.

30 -6 A 200 -Watt

811 -A De -luxe Modulator

One of the most popular medium power r -f

amplifier stages consists of a single tetrode

tube, such as the 4 -125A, 813, or 7094 oper-

ating at a plate potential of 1200 - 1700 volts

and a plate current of 150 - 275 milliamperes.

Such an amplifier requires a minimum of r -f

driving power, allows an input of 300 to 400

watts, and yet employs power supply com-

ponents that are relatively modest in cost. The

5 -db signal increase between a 300 watt trans-

mitter and a 1000 watt transmitter is very

expensive when one considers the additional

cost of modulator and power supply equipment.

Additional economy may be achieved if

the modulator and final amplifier are operated

from the same power supply. The new series of

Chicago- Standard plate transformers provide

voltage ranges in the 1000 to 1500 volt re-

gion and are well suited for the combination

of this modulator and the aforementioned r -f

tubes. Within this voltage range, the 811 -A

triode is an excellent choice for the modulator

tubes. Zero bias operation may be had up to

Figure 14

REAR VIEW OF

811 -A MODULATOR

Modulator tubes and voltage

regulator are at right with

high level filter at center of

chassis. Plug -in speech am-

plifier is to left of clipper.

Gain and clipping controls

are atop the small chassis.

6L6'5881 is used as cathode

follower driver stage for

modulator.

www.americanradiohistory.com

680 Speech and A. M. Equipment THE RADIO

CRYSTAL 12AX7

MIC

503 S01!PLI

-- SN 2

PLUG -IN SPEECH AMPLIFIER

6AL5 v L1 12AU7 Ti

SO4

12

6

2

3

7

e

10

1M

L NOTES

I -ALL RESISTORS 1/Z WATT UNLESS

OTHERWISE SPECIFIED.

2 -SW ITCH Si (PHONE-CW) SHOWN

IN PHONE POSITION.

3- RY1 SHOWN ENERGIZED.

10UF

- 25

SK

1W

+ 1.6 N

611F- 61JF- 2v

450 450 T

- E6.36 Dc1_

L v J

6L6GB

AND

2D21 FILS

Y _

0ÚF

PL2,, _J

SOS

1 j63 V. ti OUT

2-

6L6GB /5881 3- GROUND. 6'-

_ / /3V ti /N

6- 8#330 V TO EXCITER

7 -1 I CW -PHONE SWITCH

10 OF 6-

3 0

4W

811 -A

ISO 9- GROUND, B-

10-

11- RY1 CONTROL

12-MIC. CONTROL PUSH -TO-TALK CIRCUIT

004

3NV

A

811

TS 2D21

Figure 15

SCHEMATIC, 200 WATT 811 -A MODULATOR

1, -1:3 Interstage Transformer. Stanco, A -53

Tr- "Poly -pedante" class B driver transformer. 2:1 ratio. Stanco, A -4761

T,-200 watt modulation transformer. 9 K primary. 5K secondary. Stanco, A -3829

T.-400 - 0 - 400 volts, 250 ma., 6.3 volts, 5 amperes. Stanco, PC -8413

T,-6.3 volts, 10 amperes. Stanco, P -6308

CH,-4 henry, 250 ma. Stanco, C -1412

L,-Low pass filter, 3000 cycle cut off. Chicago LPF -2.

L,- "Splatter" filter, 300 ma. Stanco, C -2317

RY, -SPDT relay, high voltage insulation, 115 volt coil. Leach =1723 with 374 coils, or equivalent

+HV OUr TO

AMPLIFIER

+HV IN

RY1

1250 volts, and only -4.5 volts is required

for 1500 volt operation. Bias voltage may be

obtained from flashlight batteries or other low

impedance source.

Modulator The 200 watt de -luxe modulator

Circuit is illustrated in figures 14 and

16 and the schematic is given

in figure 15. The low level audio stages are

similar to those of the speech amplifier shown

in Section Six. A 12AX7 is employed as two

stages of R -C amplification driving a 6AL5

speech clipper tube. A 3500 cycle low pass

filter follows the clipper, removing all high

order products of clipping action. A parallel -

connected 12AU7 follows the filter and is

transformer -coupled to a 5881 (6L6 -GB) ca-

thode follower driver stage. The impedance of

the cathode circuit of the driver stage is ex-

tremely low; it provides an excellent driving

source for the class B modulator grid circuit.

www.americanradiohistory.com

HANDBOOK 81 1 -A Modulator 681

Two 811 -A tubes are employed in the class

B stage. When operated at 1000 volts, no bias

supply is needed. At voltages of 1200 or above,

approximately 9 volts of bias is required. This

is supplied by a voltage divider composed of

a 20K, 10 watt resistor and a 2D21 thyratron

tube. When the miniature 2D21 is connected

as a triode, it acts as a voltage regulator tube

with a constant voltage drop of almost 9 volts

from plate to cathode. The tube will regulate

over 300 milliamperes of current while main-

taining a reasonably constant voltage drop

across its terminals. The center tap of the

811 -A filament transformer (Ta) is thus held

at a positive potential with respect to ground.

Since the center tap of the 811 -A driver trans-

former (T2) is grounded, the modulator tubes

are biased at a constant negative voltage equal

to the voltage drop across the 2D21 regulator

tube in the cathode circuit of the class B stage.

The plate to plate load impedance of the

811 -A tubes when operating at 1500 volts is

approximately 12,000 ohms. A multi -match

type modulation transformer may be employed

if desired, but in this case a Stancor A -3829

unit was used. This transformer is designed to

match the plate -to -plate load impedance of

9,000 ohms to a secondary load of 5000 ohms.

With the 12,000 ohm load of the 811 -A tubes,

a secondary load of 7,500 ohms must be used

to maintain the same primary to secondary im-

pedance ratio. This secondary load can be ob-

tained with a single 7094 tube operating at

1500 volts and 200 milliamperes of plate cur-

rent (300 watts input) . Other tubes and load

impedances can also be used, providing the

r -f input to the modulated stage does not ex-

ceed 400 watts. For example, a 4 -125A tube

operating at 2000 volts and 165 ma. (330

watts) may be modulated by this audio unit.

The secondary winding of the modulation

transformer can pass a maximum of 300 mil-

liamperes with safety.

The audio output from the 811 -A stage is

passed through a high level low -pass "splatter

suppressor" which attenuates all audio fre-

quencies above 3500 cycles. The use of both

low level and high level audio filters does

Figure 16

UNDER -CHASSIS VIEW OF

811 -A MODULATOR

High voltage relay is between 811 -A tube

sockets, and low voltage components ore at

opposite end of chassis.

much to reduce the broad sidebands and co-

channel interference that seems to be so com-

mon on the amateur phone bands.

A high voltage relay RN', is employed to

short the secondary of the modulation trans-

former and remove plate potential from the

modulator tubes for c -w operation. The relay

is actuated by the "phone -c.w." switch on the

front panel of the modulator. Other segments

of this switch turn off the modulator fila-

ments and provide extra contacts to control

auxiliary equipment.

A 350 volt supply is incorporated in the

modulator unit to power the speech amplifier

and driver stage and to provide power for

the r -f exciter stages of the transmitter. The

various power and control leads are brought

out to a multi -connector plug mounted on the

rear of the modulator deck.

www.americanradiohistory.com

682 Speech and A. M. Equipment T H E R A D I O

Modulator

Construction The modulator is constructed

upon a steel chassis measur-

ing 8" x 17 "x2 ".A 101/2"

aluminum panel is bolted to the chassis with

two mounting brackets to form a rugged as-

sembly. Placement of the major parts may be

seen in figures 14 and 16. The modulation

transformer T3 and the 811 -A tubes occupy

the right end of the chassis, balanced in weight

by the power transformer T. and modulator

filament transformer T. at the opposite end of

the chassis. The center space is taken by the

plug -in speech amplifier, the high level splat-

ter filter assembly and the 5881 driver stage.

The speech amplifier is constructed as a sep-

arate unit on a small aluminum utility box

measuring 5" x 3" x 2 ". The bottom of the

box holds two male plugs which match two

receptacles mounted on the amplifier chassis.

The speech amplifier, therefore, may be wired

and tested as a separate unit. Clipping and

audio level controls are mounted atop the

amplifier box as long usage of clipper circuits

has proven that these controls need not be re-

adjusted once they are properly set.

The phone -c.w. switch, relay RY-1 and var-

ious small components are mounted beneath

the chassis (figure 16) . The input receptacle

for the speech amplifier box is located adja-

cent to the microphone receptacle on the front

panel of the modulator making the intercon-

necting lead less than two inches long. Also

placed beneath the chassis are the filter choke

for the low voltage supply and the various by-

pass and filter capacitors.

Wiring and Testing The speech amplifier

the Modulator should be wired first.

The small resistors and

capacitors are mounted either between the tube

socket pins, or between terminals of small

phenolic tie -point strips. Transformer T. is

fastened within the amplifier box and is wired

in the circuit after all other wiring is com-

pleted. Plugs PL, and PL: are mounted on the

bottom portion of the box; the plug pins are

wired to the proper points of the speech am-

plifier with short lengths of wire that allow

the bottom plate to be removed for inspection

and testing without the necessity of unsolder-

ing any connections to the plugs.

The modulator chassis should be wired next.

All leads to T3, RY -3, and the low pass filter

should be carefully insulated from the chassis.

High voltage "5000 volt test" cable should be

employed for these connections. The capaci-

tors that make up the high level audio filter

are mounted directly to the terminals of the

filter choke which is mounted above the chas-

sis on 1/2-inch ceramic insulators. High voltage

connections to the modulator are made through

Millen 37001 safety terminals.

When the wiring has been completed and

checked, the 12AX7, 6AL5, 12ÁU7, 5881,

and 5V4 -GB tubes should be inserted in their

sockets and the speech amplifier is plugged

into the modulator receptacles. The vertical

amplifier of an oscilloscope should be connect-

ed to one grid terminal of the 811 -A stage.

Plate voltage of the 5881 should be approxi-

mately 370 volts. A low level 1000 cycle tone

(approximately 0.05 volts, r.m.s.) is applied

to the amplifier input. The output level of the

speech amplifier is controlled by the setting

of the clipping control Rz and the audio gain

is controlled by potentiometer R1 in the grid

circuit of the 12AX7. The clipping control

should be set so that not more than 60 volts

r.m.s. is developed from one 811 -A grid to

ground. The modulator tubes may now be

plugged in their sockets. A 7K, 200 watt re-

sistor should be placed between the "H.V.

Out" and H.V.

" In" terminals, serving as a

dummy load, and 1500 volts applied to the

latter terminal. With no audio signal the

resting plate current of the modulator stage

should be approximately 15 milliamperes,

kicking up to about 160 milliamperes under

full output conditions. Final adjustment of the

clipping control may be made when the mod-

ulator is placed in use with the r -f section of

the transmitter. Potentiometer Rz is then ad-

justed to limit the peak modulation level under

sine wave modulation to approximately 90%.

TO

DRi VER

STAGE

V+

Figure 17

ZERO BIAS TETRODE MODULATOR

ELIMINATES SCREEN AND BIAS

SUPPLIES

Low driving power and simplicity are key

features of this novel modulator. Tubes rang-

ing in size from 6AQ5's to 813's may be

employed in this circuit.

T -Class 8 driver transformer

Tr- Modulation transformer

V,, V. -6AQ5, 6L6, 807, 803, 813, etc.

R, R: -Not used with 803 and 813

www.americanradiohistory.com

HANDBOOK 683

6 -WATT

SPEECH

AMP.

(F /G /2)

803's

T 1 2

: -

II -

OPERATING CHARACTERISTICS

EGG (RMS) _ /70 VOLTS

DRIVING POWER= 7 -6 WATTS

RESTING PLATE CUR.= SOMA.

MAX. PLATE CUR =340 MA.

POWER OUTPUT = 5/0 WATTS

SUPPRESSOR VOLT = 260 -340v

ZS =6 25K

TO

MODULATOR

LOAD

8+2500 V.

IIS U 1

Figure 18

INEXPENSIVE 500 WATT

MODULATOR USING 803 TUBES

T,- "Poly -pedance" Class B driver transform-

er 2:1 ratio. Stancor A -4761

Tr-500 watt output transformer. 18K primary,

6.25K secondary. Chicago CMS -3

T, -10 volts, 10 amperes. Stancor C -6461

M-0 - 500 mo.

30 -7 Zero Bias

Tetrode Modulators

Class B zero bias operation of tetrode tubes

is made possible by the application of the

driving signal to the two grids of the tubes as

shown in figure 17. Tubes such as the 6AQ5,

6L6, 807, 803, and 813 work well in this

circuit and neither a screen supply nor a bias

supply is required. The drive requirements

are low and the tubes operate with excellent

plate circuit efficiency. The series grid resis-

tors for the small tubes are required to balance

the current drawn by the two grids, but are not

needed in the case of the 803 and 813 tubes.

Of great interest to the amateur is the cir-

cuit of figure 18, wherein 803 tubes are used

as high level modulators. These tubes will de-

liver 500 watts of audio in this configuration,

yet they require no screen or bias supply, and

can be driven by an 8 watt amplifier stage.

The use of 803 tubes (in contrast to 813's)

requires a higher level of driving power which

is offset by the fact that these tubes can often

be purchased "surplus" for less than four dol-

lars. A pair of 6B4 tubes operating with ca-

thode bias (figure 12) will suffice as a driv-

ing stage for the 803's. The power supply of

the speech amplifier provides high voltage for

the suppressors of the modulator stage.

Shown in figure 19 is a high level modu-

lator using 813 tubes. A full 500 watts of

audio may be obtained at 2500 volts plate po-

tential. Grid driving power is 5.5 watts. A

single 807 operating as a cathode follower

at 400 volts will provide sufficient drive for

the modulator stage. Plate to plate load im-

pedance for the 813's is not critical. The Chi-

cago CMS -3 500 watt modulation transformer

having a primary impedance of 18,000 ohms

has been used with success, although the opti-

mum plate load impedance of the modulator

is closer to 20,000 ohms.

6C4,6J5 Erc. T

807

813,5

+400

DRIVER STAGE,OPERATING NOTES

VOLTS,

I -EXACT VALUE OF 607 CATHODE RESISTOR

MEASURED TO GROUND. DEPENDS UPON RESISTANCE OF PRIMA RV

PIN 2 300 WINDING OF T2. ADJUST RESISTOR FOR

807 PIN .3 CATHODE 8 /AS OF 26.5 VOL rs, PLATE CUR-

RENT OF 53 -55 MA.

PIN 26,5 2 -RMS OPERATING VOLTAGES AT MAXIMUM

OUTPUT SHOWN ON SCHEMATIC.

X X

111

11SV.rt.

Figure 19

500 WATT MODULATOR USING 813 TUBES

T,-1:3 interstage transformer. Stancor A -53

Tr- "Poly -pedance" Class B driver transformer. 2:1 ratio. Stancor A -4761

T,-500 watt output transformer. 18K primary, 6.25K secondary. Chicago CMS -3

T.-10 volts, 10 amperes, Stancor C -6461

M-0 - 500 ma.

350 watts of audio are obtainable from this circuit at plate potential of 2000 volts.

TO

MODULATOR

LOAD

www.americanradiohistory.com

CHAPTER THIRTY -ON E

In view of the high cost of iron -core com-

ponents such as go to make up the bulk of a

power supply, it is well to consider carefully

the design of a new or rebuilt transmitter in

terms of the minimum power supply require-

ments which will permit the desired perform-

ance to be obtained from the transmitter. Care-

ful evaluation of the power supply require-

ments of alternative transmitter arrangements

will permit the selection of that transmitter

arrangement which requires the minimum of

power supply components, and which makes

most efficient use of such power supplies as

are required.

31 -1 Power Supply

Requirements

A power supply for a transmitter or for a

unit of station equipment should be designed

in such a manner that it is capable of deliver-

ing the required current at a specified voltage,

that it has a degree of regulation consistent

with the requirements of the application, that

its ripple level at full current is sufficiently

low for the load which will be fed, that its in-

ternal impedance is sufficiently low for the

job, and that none of the components shall be

overloaded with the type of operation contem-

plated.

The meeting of all the requirements of the

previous paragraph is not always a straight-

forward and simple problem. In many cases

compromises will be involved, particularly

when the power supply is for an amateur sta-

tion and a number of components already on

hand must be fitted into the plan. As much

thought and planning should be devoted to the

power- supply complement of an amateur sta-

tion as usually is allocated to the r -f and a -f

components of the station.

The arrival at the design for the power sup-

ply for use in a particular application may best

be accomplished through the use of a series

of steps, with reference to the data in this

chapter by determining the values of compo-

nents to be used. The first step is to estab-

lish the operating requirements of the power

supply. In general these are:

1. Output voltage required under full load.

2. Minimum, normal, and peak output cur-

rent.

3. Voltage regulation required over the cur-

rent range.

684

www.americanradiohistory.com

Requirements 685

Figure 1

POWER SUPPLY

CONTROL PANEL

A well designed supply con-

trol panel has separate primary

switches and indicator lamps

for the filament and plate cir-

cuits, overload circuit breaker,

plate voltage control switch

and primary circuit fuses.

FILAMENT

VOLTAGE

PLATE

VOLTAGE

0 3500 411

PLATE VOLTAGE

EMISSION

FINAL BIAS SCREEN MOD FONECA

2 AMP 2 AMP 2 AMP. 2AMMP. \

MN MN

4. Ripple voltage limit.

5. Rectifier circuit to be used.

The output voltage required of the power

supply is more or less established by the oper-

ating conditions of the tubes which it will sup-

ply. The current rating of the supply, however,

is not necessarily tied down by a particular

tube combination. It is always best to design

a power supply in such a manner that it will

have the greatest degree of flexibility; this

procedure will in many cases allow an exist-

ing power supply to be used without change

as a portion of a new transmitter or other item

of station equipment. So the current rating of

a new power supply should be established by

taking into consideration not only the require-

ments of the tubes which it immediately will

feed, but also with full consideration of the

best matching of power supply components in

the most economical current range which still

will meet the requirements. It is often long -

run economy, however, to allow for any likely

additional equipment to be added in the near

future.

Current- Rating

Considerations The minimum current drain

which will be taken from a

power supply will be, in most

cases, merely the bleeder current. There are

many cases where a particular power supply

will always be used with a moderate or heavy

load upon it, but when the supply is a portion

of a transmitter it is best to consider the mini-

mum drain as that of the bleeder. The mini-

mum current drain from a power supply is of

importance since it, in conjunction with the

nominal voltage of the supply, determines the

minimum value of inductance which the input

choke must have to keep the voltage from soar-

ing when the external load is removed.

The normal current rating of a power supply

usually is a round -number value chosen on the

basis of the transformers and chokes on hand

or available from the catalog of a reliable man-

ufacturer. The current rating of a supply to

feed a steady load such as a receiver, a speech

amplifier, or a continuously -operating r -f stage

should be at least equal to the steady drain of

the load. However, other considerations come

into play in choosing the current rating for a

keyed amplifier, an amplifier of SSB signals,

or a class B modulator. In the case of a sup-

ply which will feed an intermittent load such

as these, the current ratings of the transform-

ers and chokes may be lets than the maximum

current which will be taken; but the current

ratings of the rectifier tubes to be used should

be at least equal to the maximum current which

will be taken. That is to say that 300 -ma.

transformers and chokes may be used in the

supply for a modulator whose resting current

is 100 ma. but whose maximum current at

peak signal will rise to 500 ma. However, the

rectifier tubes should be capable of handling

the full 500 ma.

www.americanradiohistory.com

686 Power Supplies T H E R A D I O

The iron -core components of a power supply

which feeds an intermittent load may be cho-

sen on the basis of the current as averaged

over a period of several minutes, since it is

heating effect of the current which is of great-

est importance in establishing the ratings of

such components. Since iron -core components

have a relatively large amount of thermal in-

ertia, the effect of an intermittent heavy cur-

rent is offset to an extent by a key -up period

or a period of low modulation in the case of a

modulator. However, the current rating of a

rectifier tube is established by the magnitude

of the emission available from the filament of

the tube; the maximum emission must not be

exceeded even for a short period or the recti-

fier tube will be damaged. The above consider-

ations are predicated, however, on the assump-

tion that none of the iron -core components will

become saturated due to the high intermittent

current drain. If good -quality components of

generous weight are chosen, saturation will

not be encountered.

Voltage

Regulation The general subject of voltage

regulation can really be divided

into two sub -problems, which dif-

fer greatly in degree. The first, and more

common, problem is the case of the normal

power supply for a transmitter modulator,

where the current drain from the supply may

vary over a ratio of four or five to one. In

this case we desire to keep the voltage change

under this varying load to a matter of 10 or

15 per cent of the operating voltage under full

load. This is a quite different problem from

the design of a power supply to deliver some

voltage in the vicinity of 250 volts to an os-

cillator which requires two or three milliam-

peres of plate current; but in this latter case

the voltage delivered to the oscillator must be

constant within a few volts with small varia-

tions in oscillator current and with large varia-

tions in the a -c line voltage which feeds the

oscillator power supply. An additional voltage

regulation problem, intermediate in degree be-

tween the other two, is the case where a load

must be fed with 10 to 100 watts of power at

a voltage below 500 volts, and still the voltage

variation with changes in load and changes

in a -c line voltage must be held to a few

volts at the output terminals.

These three problems are solved in the nor-

mal type of installation in quite different man-

ners. The high -power case where output volt-

age must be held to within 10 to 15 per cent is

normally solved by using the proper value of

inductance for the input choke and proper

value of bleeder at the output of the power

supply. The calculations are simple: the in-

ductance of the power -supply input choke at

minimum current drain from the supply should

be equal in henries to the load resistance on

the supply (at minimum load current) divided

by 1000. This value of inductance is called

the critical inductance and it is the minimum

value of inductance which will keep the out-

put voltage from soaring in a choke -input

power supply with minimum load upon the

output. The minimum load current may be

that due to the bleeder resistor alone, or it

may be due to the bleeder plus the minimum

drain of the modulator or amplifier to which

the supply is connected.

The low- voltage low- current supply, such as

would be used for a v.f.o. or the high- frequen-

cy oscillator in a receiver, usually is regu-

lated with the aid of glow- discharge gaseous -

regulator tubes. These regulators are usually

called "VR tubes." Their use in various types

of power supplies is discussed in Section 31 -9.

The electronically -regulated power supply,

such as is used in the 10 to 100 watts power

output range, also is discussed later on in this

chapter,

Ripple The ripple -voltage limitation

Considerations imposed upon a power supply

is determined by the load

which will be fed by the supply. The tolerable

ripple voltage from a supply may vary from

perhaps 5 per cent for a class B or class C

amplifier which is to be used for a c -w stage

or amplifier of an FM signal down to a few

hundredths of one per cent for the plate -volt-

age supply to a low -level voltage amplifier in

a speech amplifier. The usual value of ripple

voltage which may be tolerated in the supply

for the majority of stages of a phone transmit-

ter is between 0.1 and 2.0 per cent.

In general it may be stated that, with 60-

cycle line voltage and a single -phase rectifier

circuit, a power supply for the usual stages in

the amateur transmitter will be of the choke -

input type with a single pi- section filter fol-

lowing the input choke. A c -w amplifier or

other stage which will tolerate up to 5 per cent

ripple may be fed from a power supply whose

filter consists merely of an adequate -size in-

put choke and a single filter capacitor.

A power supply with input choke and a

single capacitor also will serve in most cases

to feed a class B modulator, provided the out-

put capacitor in the supply is sufficiently large.

The output capacitor in this case must be

capable of storing enough energy to supply the

www.americanradiohistory.com

HANDBOOK Requirements 687

peak- current requirement of the class B tubes

on modulation peaks. The output capacitor for

such a supply normally should be between 4

pfd. and 20 pfd.

Capacitances larger than 20 µfd. involve a

high initial charging current when the supply

is first turned on, so that an unusually large

input choke should be used ahead of the ca-

pacitor to limit the peak- current surge through

the rectifier tubes. A capacitance of less than

4 pfd. may reduce the power output capability

of a class B modulator when it is passing the

lower audio frequencies, and in addition may

superimpose a low -frequency "growl" on the

output signal. This growl will be apparent only

when the supply is delivering a relatively high

power output; it will not be present when mod-

ulation is at a low level.

When a stage such as a low -level audio am-

plifier requires an extremely low value of rip-

ple voltage, but when regulation is not of im-

portance to the operation of the stage, the high

degree of filtering usually is obtained through

the use of a resistance- capacitance filter.

This filter usually is employed in addition to

the choke -capacitor filter in the power supply

for the higher -level stages, but in some cases

when the supply is to be used only to feed

low- current stages the entire filter of the pow-

er supply will be of the resistance- capacitance

type. Design data for resistance -capacitance

filters is given in a following paragraph.

When a low- current stage requires very low

ripple in addition to excellent voltage regula-

tion, the power supply filter often will end

with one or more gaseous -type voltage- regula-

tor tubes. These VR tubes give a high degree

of filtering in addition to their voltage -regu-

lating action, as is obvious from the fact that

the tubes tend to hold the voltage drop across

their elements to a very constant value regard-

less of the current passing through the tube.

The VR tube is quite satisfactory for improv-

ing both the regulation and ripple character-

istics of a supply when the current drain will

not exceed 25 to 35 ma. depending upon the

type of VR tube. Some types are rated at a

maximum current drain of 30 ma. while others

are capable of passing up to 40 ma. without

damage. In any event the minimum current

through the VR tube will occur when the as-

sociated circuit is taking maximum current.

This minimum current requirement is 5 ma.

for all types of gaseous -type voltage -regulator

tubes. Other types of voltage -regulation systems,

in addition to VR tubes, exhibit the added

TO FULL- WAVE

RECTIFIER

5 -25

NY

TO FULL-WAVE

RECTIFIER

RIPPLE IN TERMS OF C AT FULL LOAD

CAPACITANCE, C

2 UF

3 LF

25000 UF

6 L:

TO FULL -WAVE

RECTIFIER

25000

FIGURE 2

PERCENT RIPPLE

13.1

6.5

6.2

4.0

RIPPLE IN TERMS OF LOAD RESISTANCE

LOAD. ONMS PERCENT RIPPLE

25000 (BLEEDER ONLY) 0.02

15000 0.04

10000 0.06

5000 0.1

3000 O 17

2 000 0.25

FIGURE 3

RIPPLE IN TERMS OF CI AND C2 AT FULL LOAD

25000

C1

2

3 4

B

FIGURE 4

C2 PERCENT RIPPLE

2 1.2

0.7

a 0 25

6 0.06

characteristic of offering a low value of rip-

ple across their output terminals. The elec-

tronic -type of voltage -regulated power supply

is capable of delivering an extremely small

value of ripple across its output terminals,

even though the rectifier- filter system ahead

of the regulator delivers a relatively high

value of ripple, such as in the vicinity of 5 to

10 per cent. In fact, it is more or less self

evident that the better the regulation of such

a supply, the better will be its ripple charac-

teristic. It must be remembered that the ripple

output of a voltage -regulated power supply of

any type will rise rapidly when the load upon

the supply is so high that the regulator begins

to lose control. This will occur in a supply of

the electronic type when the voltage ahead of

the series regulator tube falls below a value

equal to the sum of the minimum drop across

the tube at that value of current, plus the out-

put voltage. In the case of a shunt regulator

of the VR -tube type, the regulating effect will

fail when the current through the VR tube

falls below the usual minimum value of about

5 ma.

Calculation Although figures 2, 3 and 4

of Ripple give the value of ripple volt-

age for several more or less

standard types of filter systems, it is often of

value to be able to calculate the value of rip-

ple voltage to be expected with a particular

set of filter components. Fortunately, the ap-

proximate ripple percentage for normal values

www.americanradiohistory.com

688 Power Supplies T H E R A D I O

TO FULL-WAVE

RECTIFIER

e Hr 12 HT

Figure 5

SAMPLE FILTER FOR

CALCULATION OF RIPPLE

of filter components may be calculated with

the aid of rather simple formulas. In the two

formulas to follow it is assumed that the line

frequency is 60 cycles and that a full wave or

a full -wave bridge rectifier is being used. For

the case of a single -section choke -input filter

as illustrated in figure 2, or for the ripple at

the output of the first section of a two -section

choke input filter the equation is as follows,

118

Per cent ripple = LC -1

where LC is the product of the input choke

inductance in henrys (at the operating current

to be used) and the capacitance which follows

this choke expressed in microfarads.

In the case of a two -section filter, the per

cent ripple at the output of the first section is

determined by the above formula. Then this

percentage is multiplied by the filter reduction

factor of the following section of filter. This

reduction factor is determined through the use

of the following formula: LC -1

Filter reduction factor - 1.76

Where LC again is the product of the in-

ductance and capacitance of the filter section.

The reduction factor will turn out to be a deci-

mal value, which is then multiplied by the per-

centage ripple obtained from the use of the

preceding formula.

As an example, take the case of the filter

diagrammed in figure 5. The LC product of the

first section is 16. So the ripple to be expected

at the output of the first section will be: 118/

(16 -1) or 118/15, which gives 7.87 per cent.

Then the second section, with an LC product

of 48, will give a reduction factor of: 1.76/

(48 -1) or 1.76/47 or 0.037. Then the ripple

percentage at the output of the total filter will

be: 7.87 times 0.037 or slightly greater than

0.29 per cent ripple.

Resistance- In many applications where the

Capacitance current drain is relatively small,

Filters so that the voltage drop across

the series resistors would not be

excessive, a filter system made up of resistors

and capacitors only may be used to advantage.

In the normal case, where the reactance of the

shunting capacitor is very much smaller than

the resistance of the load fed by the filter sys-

tem, the ripple reduction per section is equal

to 1/ (2TrRC). In terms of the 120 -cycle ripple

from a full -wave rectifier the ripple -reduction

factor becomes: 1.33 /RC where R is expressed

in thousands of ohms and C in microfarads.

For 60 -cycle ripple the expression is: 2.66/RC

with R and C in the same quantities as above.

Filter System

Resonance Many persons have noticed,

particularly when using an in-

put choke followed by a 2 -µfd.

first filter capacitor, that at some value of

load current the power supply will begin to

hum excessively and the rectifier tubes will

tend to flicker or one tube will seem to take

all the load while the other tube dims out. If

the power supply is shut off and then again

started, it may be the other tube which takes

the load; or first one tube and then the other

will take the load as the current drain is

varied. This condition, as well as other less

obvious phenomena such as a tendency for the

first filter capacitor to break down regardless

of its voltage rating or for rectifier tubes to

have short life, results from resonance in the

filter system following the high -voltage rec-

tifier.

The condition of resonance is seldom en-

countered in low -voltage power supplies since

the capacitors used are usually high enough

so that resonance does not occur. But in high -

voltage power supplies, where both choke in-

ductance and filter capacitance are more ex-

pensive, the condition of resonance happens

frequently. The product of inductance and ca-

pacitance which resonates at 120 cycles is

1.77. Thus a 1 -pfd. capacitor and a 1.77 henry

choke will resonate at 120 cycles. In almost

any normal case the LC product of any section

in the filter system will be somewhat greater

than 1.77, so that resonance at 120 cycles will

seldom take place. But the LC product for

resonance at 60 cycles is about 7.1. This is a

value frequently encountered in the input sec-

tion of a high- voltage power supply. It occurs

with a 2 -pfd. capacitor and a choke which has

3.55 henrys of inductance at some current

value. With a 2-pfd. filter capacitor following

this choke, resonance will occur at the current

value which causes the inductance of the choke

to be 3.55 henrys. When this resonance does

occur, one rectifier tube (assuming mercury-

www.americanradiohistory.com

HANDBOOK Rectification Circuits 689

vapor types) will dim and the other will be-

come much brighter.

Thus we see that we must avoid the LC

products of 1.77 and 7.1. With a swinging -type

input choke, whose inductance varies over a

5 -to -1 range, we see that it is possible for

resonance to occur at 60 cycles at a low value

of current drain, and then for resonance to oc-

cur at 120 cycles at approximately full load

on the power supply. Since the LC product

must certainly be greater than 1.77 for satis-

factory filtering along with peak- current limi-

tation on the rectifier tubes, we see that with

a swinging -type input choke the LC product

must still be greater than 7.1 at maximum cur-

rent drain from the power supply. To allow a

reasonable factor of safety, it will be well to

keep the LC product at maximum current drain

above the value 10.

It is possible to place the filter choke

in the B -minus lead of the power supply

which reduces the voltage potential appearing

from choke winding to ground. However, the

back e.m.f. of a good choke is quite high and

can develop a dangerous potential from the

center tap to ground on the secondary winding

of the plate transformer. If the transformer is

not designed to withstand this potential, it

is possible to break down the insulation at

this point.

31 -2 Rectification

Circuits

There are a large number of rectifier circuits

that may be used in the power supplies for sta-

tion equipment. But the simpler circuits are

more satisfactory for the power levels up to

the maximum permitted the radio amateur. Fig-

ure 6 shows the three most common circuits

used in power supplies for amateur equipment.

Half -Wave A half -wave rectifier, as shown in

Rectifiers figure 6A, passes one half of the

wave of each cycle of the alter-

nating current and blocks the other half. The

output current is of a pulsating nature, which

can be smoothed into pure, direct current by

means of filter circuits. Half -wave rectifiers

produce a pulsating current which has zero

output during one -half of each a -c cycle; this

makes it difficult to filter the output properly

into d.c. and also to secure good voltage regu-

lation for varying loads.

Full -Wove

Rectifiers A full -wave rectifier consists of

a pair of half -wave rectifiers

working on opposite halves of the

cycle, connected in such a manner that each

Figure 6

MOST COMMON RECTIFIER CIRCUITS

(A) shows a half -wove rectifier circuit, (8) is

the standard full -wove rectifier circuit used

with a dual rectifier or two rectifier tubes,

and (C) is the bridge rectifier circuit.

half of the rectified a -c wave is combined in

the output as shown in figure 7. This pulsat-

ing unidirectional current can be filtered to

any desired degree, depending upon the par-

ticular application for which the power supply

is designed.

A full -wave rectifier may consist of two

plates and a filament, either in a single glass

or metal envelope for low- voltage rectification

or in the form of two separate tubes, each hav-

ing a single plate and filament for high -voltage

rectification. The plates are connected across

the high -voltage a -c power transformer wind-

ing, as shown in figure 6B. The power trans-

former is for the purpose of transforming the

110 -volt a -c line supply to the desired second-

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690 Power Supplies T H E R A D I O

TRANSFORMER

SECONDARY

VOLTAGE

RECTIFIED VOLTAGE

PLATE N1

RECTIFIED VOLTAGE

PLATE N 2

COMBINED RECTIFIED

VOLTAGE

PLATES W1 l2

0

+ VOLTAGE

AFTER FIRST SECTION

OF FILTER

oL

D.C. VOLTAGE

AVAILABLE FOR

RADIO USE

Figure 7

FULL -WAVE RECTIFICATION

Showing transformer secondary voltage, the

rectified output of each tube, the combined

output of the rectifier, the smoothed voltage

after one section of filter, and the substan-

tially pure d.c. output of the rectifier- filter

after additional sections of filter.

ary a -c voltages for filament and plate sup-

plies. The transformer delivers alternating cur-

rent to the two plates of the rectifier tube; one

of these plates is positive at any instant dur-

ing which the other is negative. The center

point of the high- voltage transformer winding

is usually grounded and is, therefore, at zero

voltage, thereby constituting the negative B

connection.

While one plate of the rectifier tube is con-

ducting, the other is inoperative, and vice ver-

sa. The output voltages from the rectifier tubes

are connected together through the common

rectifier filament circuit. Thus the plates al-

ternately supply pulsating current to the out-

put (load) circuit. The rectifier tube filaments

or cathodes are always positive in polarity

with respect to the plate transformer in this

type of circuit.

The output current pulsates 120 times per

second for a full -wave rectifier connected to

a 60 -cycle a -c line supply; hence the output of

the rectifier must pass through a filter to

smooth the pulsations into direct current. Fil-

ters are designed to select or reject alternating

currents; those most commonly used in a -c

power supplies are of the low -past type.

Bridge The bridge rectifier (figure 6C)

Rectification is a type of full -wave circuit

in which four rectifier elements

or tubes are operated from a single high -volt-

age winding on the power transformer.

While twice as much output voltage can be

obtained from a bridge rectifier as from a cen-

ter- tapped circuit, the permissible output cur-

rent is only one -half as great for a given power

transformer. In the bridge circuit, four recti-

fiers and three filament heating transformer

windings are needed, as against two rectifiers

and one filament winding in the center- tapped

full -wave circuit. In a bridge rectifier circuit,

the inverse peak voltage impressed on any one

rectifier tube is halved, which means that tubes

of lower peak inverse voltage rating may be

used for a given voltage output.

Note that the center of the high voltage

winding of the bridge transformer (figure

6C) is not at ground potential. Many trans-

formers having a center -tapped winding are

not designed for bridge service as the insula-

tion between the center tap point and ground

is inadequate. Lack of insulation at this point

does no harm in a full -wave circuit, but may

cause breakdown when the transformer is used

in bridge configuration.

31 -3 Standard Power

Supply Circuits

Choke input is shown for all three of the

standard circuits of figure 6, since choke input

gives the best utilization of rectifier -tube and

power transformer capability, and in addition

gives much better regulation. Where greater

output voltage is a requirement, where the load

is relatively constant so that regulation is not

of great significance, and where the rectifier

tubes will be operated well within their peak -

current ratings, the capacitor -input type of fil-

ter may be used.

The capacitor -input filter gives a no -load

output voltage equal approximately to the peak

voltage being applied to the rectifier tubes.

At full -load, the d -c output voltage is usually

slightly above one -half the secondary a -c volt-

age of the transformer, with the normal values

of capacitance at the input to the filter. With

large values of input capacitance, the output

voltage will run somewhat higher than the

r -m -s secondary voltage applied to the tubes,

but the peak current flowing through the rec-

tifier tubes will be many times as great as the

d -c output current of the power supply. The

half -wave rectifier of figure 6A is commonly

used with capacitor input and resistance- capaci-

tance filter as a high -voltage supply for a

cathode -ray tube. In this case the current drain

www.americanradiohistory.com

HANDBOOK Standard Circuits 691

OA HALF AND FULL VOLTAGE BRIDGE POWER SUPPLY

Eoo., T osE_

+ Eoo,EL

+Eoo,

BO TWO VOLTAGE BRIDGE POWER SUPPLY

+EOo,-{3-

OD CENTER TAPPED METHOD FOR UNTAPPED TRANSFORMERS

FO SPECIAL FILTER CIRCUIT FOR BRIDGE RECTIFIER

Figure 8

SPECIAL SINGLE PHASE RECTIFICATION CIRCUITS

A description of the application and operation of each of these special circuits is given in

accompanying text. the

is very small so that the peak -current rating

of the rectifier tube seldom will be exceeded.

The circuit of figure 6B is most commonly

used in medium -voltage power supplies since

this circuit is the most economical of filament

transformers, rectifier tubes and sockets, and

space. But the circuit of figure 6C, commonly

called the bridge rectifier, gives better trans-

former utilization so that the circuit is most

commonly used in higher powered supplies.

The circuit has the advantage that the entire

secondary of the transformer is in use at all

times, instead of each side being used alter-

nately as in the case of the full -wave rectifier.

As a point of interest, the current flow through

the secondary of the plate transformer is a sub-

stantially pure a -c wave as a result of better

transformer utilization, instead of the pulsat-

ing d -c wave through each half of the power

transformer secondary in the case of the full -

wave rectifier.

The circuit of figure 6C will give the great-

est value of output power for a given trans-

former weight and cost in a single -phase power

supply as illustrated. But in attempting to

bridge -rectify the whole secondary of a trans-

former designed for a full -wave rectifier, in

order to obtain doubled output voltage, make

sure that the insulation rating of the trans-

former to be used is adequate. In the bridge

rectifier circuit the center of the high -voltage

winding is at a d -c potential of one -half the

total voltage output from the rectifier. In a

normal full -wave rectifier the center of the

high -voltage winding is grounded. So in the

bridge rectifier the entire high -voltage second-

ary of the transformer is subjected to twice

the peak- voltage stress that would exist if the

same transformer were used in a full -wave rec-

tifier. High -quality full -wave transformers will

withstand bridge operation quite satisfactorily

so long as the total output voltage from the

supply is less than perhaps 4500 volts. But

inexpensive transformers, whose insulation

is just sufficient for full -wave operation, will

break down when bridge rectification of the

entire secondary is attempted.

Special Single -

Phase Rectification

Circuits

Figure 8 shows six cir-

cuits which may prove

valuable when it is de-

sired to obtain more than

www.americanradiohistory.com

692 Power Supplies T H E R A D I O

PRIMARY

OA 3 -PHASE STAR

Eo = 1.17 Es

Is = 0.577 ID C.

RIPPLE FREQUENCY= 3F

RIPPLE PERCENT = 18.3

PEAK INVERSE 2.09 Eo

TUBE VOLTAGE = = 2.44 Es

Eo

Eo. 1.35 Es

Is = 0.4os ID . C.

RIPPLE FREQUENCY = 6F

RIPPLE PERCENT 4.2

PEAK INVERSE 2.09 Eo

TUBE VOLTAGE ' 2.53 Es

PRIMARY

+Eo

Eo = 2.34 Es

Is o.81e I D.0

RIPPLE FREQUENCY =6F

RIPPLE PERCENT* 4.2

PEAK INVERSE 1.03 Eo

TUBE VOLTAGE . 2.44 Es

Figure 9

COMMON

POLYPHASE -

RECTIFICATION

CIRCUITS

These circuits are used

when polyphase power is

available for the plate

supply of a high -power

transmitter. The circuit

at (B) is also called a

three -phase full -wave

rectification system. The

circuits are described in

the accompanying text.

one output voltage from one plate transformer

or where some special combination of voltages

is required. Figure 8A shows a more or less

common method for obtaining full voltage and

half voltage from a bridge rectification circuit.

With this type of circuit separate input chokes

and filter systems are used on both output

voltages. If a transformer designed for use

with a full -wave rectifier is used in this cir-

cuit, the current drain from the full -voltage

tap is doubled and added to the drain from the

half -voltage tap to determine whether the rat-

ing of the transformer is being exceeded. Thus

if the transformer is rated at 1250 volts at 500

ma. it will be permissible to pull 250 ma. at

2500 volts with no drain from the 1250 -volt

tap, or the drain from the 1250 -volt tap may

be 200 ma. if the drain from the 2500 -volt

tap is 150 ma., and so forth.

Figure 813 shows a system which may be

convenient for obtaining two voltages which

are not in a ratio of 2 to 1 from a bridge -type

rectifier; a transformer with taps along the

winding is required for the circuit however.

With the circuit arrangement shown the volt-

age from the tap will be greater than one -half

the voltage at the top. If the circuit is changed

so that the plates of the two rectifier tubes

are connected to the outside of the winding

instead of to the taps, and the cathodes of the

other pair are connected to the taps instead

of to the outside, the total voltage ouput of

the rectifier will be the same, but the voltage

at the tap position will be lest than half the

top voltage.

An interesting variable- voltage circuit is

shown in figure 8C. The arrangement may be

used to increase or decrease the output volt-

age of a conventional power supply, as repre-

sented by transformer T,, by adding another

filament transformer to isolate the filament

circuits of the two rectifier tubes and adding

another plate transformer between the filaments

of the two tubes. The voltage contribution of

the added transformer T may be subtracted

from or added to the voltage produced by T1

www.americanradiohistory.com

HANDBOOK Standard Circuits 693

simply by reversing the double -pole double -

throw switch S. A serious disadvantage of this

circuit is the fact that the entire secondary

winding of transformer T_ must be insulated

for the total output voltage of the power sup-

ply. An arrangement for operating a full -wave

rectifier from a plate transformer not equipped

with a center tap is shown in figure 8D. The

two chokes L, must have high inductance rat-

ings at the operating current of the plate sup-

ply to hold down the a -c current load on the

secondary of the transformer since the total

peak voltage output of the plate transformer

is impressed across the chokes alternately.

However, the chokes need only have half the

current rating of the filter choke L2 for a cer-

tain current drain from the power supply since

only half the current passes through each

choke. Also, the two chokes L, act as input

chokes so that an additional swinging choke

is not required for such a power supply.

A conventional two- voltage power supply

with grounded transformer center tap is shown

in figure 8E. The output voltages from this

circuit are separate and not additive as in

the circuit of figure 8B. Figure 8F is of ad-

vantage when it is desired to operate Class B

modulators from the half- voltage output of a

bridge power supply and the final amplifier

from the full voltage output. Both L, and L2

should be swinging chokes but the total drain

from the power supply passes through L, while

only the drain of the final amplifier passes

through 1.2. Capacitors C, and C2 need be rated

only half the maximum output voltage of the

power supply, plus the usual safety factor.

This arrangement is also of advantage in hold-

ing down the "key -up" voltage of a c -w trans-

mitter since both L, and L2 are in series, and

their inductances are additive, insofar as the

"critical inductance" of a choke -input filter

is concerned. If 4 µfd. capacitors are used at

both C, and C2 adequate filter will be obtained

on both plate supplies for hum -free radiophone

operation.

Polyphase It is usual practice in commer-

Rectification cial equipment installations

Circuits when the power drain from a

plate supply is to be greater

than about one kilowatt to use a polyphase rec-

tification system. Such power supplies offer

better transformer utilization, less ripple out-

put and better power factor in the load placed

upon the a -c line. However, such systems re-

quire a source of three -phase (or two -phase

with Scott connection) energy. Several of the

more common polyphase rectification circuits

with their significant characteristics are

shown in figure 9. The increase in ripple fre-

quency and decrease in percentage of ripple

is apparent from the figures given in figure 9.

The circuit of figure 9C gives the best trans-

former utilization as does the bridge circuit

in the single -phase connection. The circuit

has the further advantage that there is no aver-

age d -c flow in the transformer, so that three

single -phase transformers may be used. A tap

at half -voltage may be taken at the junction

of the star transformers, but there will be d -c

flow in the transformer secondaries with the

power supply center tap in use. The circuit of

figure 9A has the disadvantage that there is an

average d -c flow in each of the windings.

Rectifiers Rectifying elements in high -volt-

age plate supplies are almost in-

variably electron tubes of either the high -vacu-

um or mercury -vapor type, although selenium

or silicon rectifier stacks containing a large

number of elements are often used. Low -volt-

age high -current supplies may use argon gas

rectifiers (Tungar tubes), selenium rectifiers,

or other types of dry -disc rectification ele-

ments. The xenon rectifier tubes offer some

advantage over mercury -vapor rectifiers for

high- voltage applications where extreme tem-

perature ranges are likely to be encountered.

However, such rectifiers (3B25 for example)

are considerably more expensive than their

mercury -vapor counterparts.

Peak Inverse Plate In an a -c circuit, the maxi -

Voltage and Peak mum peak voltage or cur -

Plate Current rent is V 2 or 1.41 times

that indicated by the a -c

meters in the circuit. The meters read the root -

mean- square (r.m.s.) values, which are the

peak values divided by 1.41 for a sine wave.

If a potential of 1,000 r.m.s. volts is ob-

tained from a high -voltage secondary winding

of a transformer, there will be 1,410 -volts peak

potential from the rectifier plate to ground. In

a single -phase supply the rectifier tube has

this voltage impressed on it, either positively

when the current flows or "inverse" when the

current is blocked on the other half -cycle. The

inverse peak voltage which the tube will stand

safely is used as a rating for rectifier tubes.

At higher voltages the tube is liable to arc

back, thereby destroying or damaging it. The

relations between peak inverse voltage, total

transformer voltage and filter output voltage

depend upon the characteristics of the filter

and rectifier circuits (whether full- or half -

wave, bridge, single -phase or polyphase, etc.).

www.americanradiohistory.com

694 Power Supplies T H E R A D I O

=LINE VOLTS - HEATER VOLTS

HEATER AMPERES

C,.., CzT

1- 1- o

+ Q

LINE RECTIFIER

SELENIUM

LINE RECTIFIER

VOLTAGE DOUBLER

FULL -WAVE

e

VOLTAGE DOUBLER

HALF -WAVE

SELENIUM

RECTIFIER

VOLTAGE

QUADRUPLER

Figure 10

TRANSFORMERLESS POWER- SUPPLY

CIRCUITS

Circuits such as shown above are also fre-

quently called line -rectifier circuits. Seleni-

um rectifiers, vacuum diodes, or gas diodes

may be used as the rectifying elements in

these circuits.

Rectifier tubes are also rated in terms of

peak plate current. The actual direct load cur-

rent which can be drawn from a given rectifier

tube or tubes depends upon the type of filter

circuit. A full -wave rectifier with capacitor in-

put passes a peak current several times the

direct load current.

In a filter with choke input, the peak cur-

rent is not much greater than the load current

if the inductance of the choke is fairly high

(assuming full -wave rectification).

A full -wave rectifier with two rectifier ele-

ments requires a transformer which delivers

twice as much a -c voltage as would be the

case with a half -wave rectifier or bridge rec-

tifier.

Mercury -Vapor The inexpensive mercury -va-

Rectifier Tubes por type of rectifier tube is

almost universally used in

the high- voltage plate supplies of amateur and

commercial transmitters. Most amateurs are

quite familiar with the use of these tubes but

it should be pointed out that when new or

long -unused mercury -vapor tubes are first

placed in service, the filaments should be op-

erated at normal temperature for approxi-

mately twenty minutes before plate voltage

is applied, in order to remove all traces of

mercury from the cathode and to clear any

mercury deposits from the top of the envelope.

After this preliminary warm -up with a new

tube, plate voltage may be applied within 20

to 30 seconds after the time the filaments

are turned on, each time the power supply

is used. If plate voltage should be applied

before the filament is brought to full tem-

perature, active material may be knocked from

the oxide -coated filament and the life of the

tube will be greatly shortened.

Small r -f chokes must sometimes be con-

nected in series with the plate leads of mer-

cury -vapor rectifier tubes in order to prevent

the generation of radio- frequency hash. These

r -f chokes must be wound with sufficiently

heavy wire to carry the load current and must

have enough inductance to attenuate the r -f

parasitic noise current to prevent it from flow-

ing in the filter supply leads and then being

radiated into nearby receivers. Manufactured

mercury -vapor rectifier hash chokes are avail-

able in various current ratings from the James

Millen Company in Malden, Mass., and from

the J. W. Miller Company in Los Angeles.

When mercury -vapor rectifier tubes are op-

erated in parallel in a power supply, small

resistors or small iron -core choke coils should

be connected in series with the plate lead of

each tube. These resistors or inductors tend

to create an equal division of plate current be-

tween parallel tubes and prevent one tube from

carrying the major portion of the current.

When high vacuum rectifiers are operated in

parallel, these chokes or resistors are not re-

quired.

Transformerless Figure 10 shows a group of

Power Supplies five different types of trans-

formerless power supplies

which are operated directly from the a -c line.

Circuits of the general type are normally found

in a.c. -d.c. receivers but may be used in low -

powered exciters and in test instruments. When

circuits such as shown in (A) and (B) are

operated directly from the a -c line, the rec.

www.americanradiohistory.com

H A N D B O O K Standard Circuits 695

tifier element simply rectifies the line voltage

and delivers the alternate half cycles of energy

to the filter network. With the normal type

of rectifier tube, load currents up to approxi-

mately 75 ma. may be employed. The d -c

voltage output of the filter will be slightly

less than the r -m -s line voltage, depending

upon the particular type of rectifier tube em-

ployed. With the introduction of the miniature

selenium rectifier, the transformerless power

supply has become a very convenient source

of moderate voltage at currents up to perhaps

500 ma. A number of advantages are offered

by the selenium rectifier as compared to the

vacuum tube rectifier. Outstanding among

these are the factors that the selenium recti-

fier operates instantly, and that it requires no

heater power in order to obtain emission. The

amount of heat developed by the selenium rec-

tifier is very much less than that produced by

an equivalent vacuum -tube type of rectifier.

In the circuits of figure 10 (A), (B) and

(C) , capacitors G and G should be rated

at approximately 150 volts and for a normal

degree of filtering and capacitance, should be

between 15 to 60 ,dd. In the circuit of figure

10D, capacitor C, should be rated at 150

volts and capacitor G should be rated at 300

volts. In the circuit of figure 10E, capacitors

C, and G should be rated at 150 volts and

G and G should be rated at 300 volts.

The d -c output voltage of the line rectifier

may be stabilized by means of a VR tube.

However, due to the unusually low internal

resistance of the selenium rectifier, transform -

erless power supplies using this type of rec-

tifying element can normally be expected to

give very good regulation.

Voltage -Doubler Figures IOC and 1OD illus-

Circuits trate two simple voltage -

doubler circuits which will

deliver a d -c output voltage equal approximate-

ly to twice the r -m -s value of the power line

voltage. The no -load d -c output voltage is

equal to 2.82 times the r -m -s line voltage

value. At high current levels, the output volt-

age will be slightly under twice the line volt-

age. The circuit of figure IOC is of advantage

when the lowest level of ripple is required

from the power supply, since its ripple fre-

quency is equal to twice the line frequency.

The circuit of figure 10D is of advantage when

it is desired to use the grounded side of the

a -c line in a permanent installation as the re-

turn circuit for the power supply. However,

with the circuit of figure IOD the ripple fre-

quency is the same as the a -c line frequency.

OUTSIDE COLLECTOR

INSIDE COLLECTOR

SELENIUM COAT

100 - 90

60

Lß 70

60

> U 50

W 40

U 30

20

W

PHENOLIC WASNEF

BASE PLATE

SELENIUM RECTIFIER CELL

00 50 100 150 200 250

C, RELATIVE LOAD CURRENT,

PERCENT' OF FULL LOAD

300

Figure 11

THE SELENIUM RECTIFIER

A -The selenium rectifier is a semi -conductor

stack built up of nickel plated aluminum

discs coated on one side with selenium

alloy.

8- Rectifier efficiency is high, reaching 70";

for single phase service, dropping slightly

at high current densities.

Voltage

Quadrupler The circuit of figure 10E illus-

trates a voltage quadrupler cir-

cuit for miniature selenium rec-

tifiers. In effect this circuit is equivalent to

two voltage doublers of the type shown in fig-

ure 10D with their outputs connected in series.

The circuit delivers a d -c output voltage under

light load approximately equal to four times

the r -m -s value of the line voltage. The no-

load d -c output voltage delivered by the quad-

rupler is equal to 5.66 times the r -m -s line

voltage value and the output voltage decreases

rather rapidly as the load current is increased.

In each of the circuits in figure 10 where

selenium rectifiers have been shown, conven-

tional high -vacuum rectifiers may be substi-

tuted with their filaments connected in series

and an appropriate value of the line resistor

added in series with the filament string.

31 -4 Selenium and

Silicon Rectifiers

Selenium rectifiers are characterized by long

life, dependability, and maintenance -free op-

eration under severe operating conditions. The

www.americanradiohistory.com

696 Power Supplies THE RADIO

O 50 ¶00 ¶50 200 250 300

LOAD CURRENT, PEPCEN7 OF FULL LOAD

Figure 12

VOLTAGE REGULATION OF

SELENIUM CELL

This graph applies to single phase lull wove

bridge, and center -tap circuits which utilize

both halves of the input wave. In single phase

hall wave circuits the regulation will be poorer.

selenium rectifier consists of a nickel -plated

aluminum base plate coated with selenium

over which a low temperature alloy is sprayed.

The base plate serves as the negative electrode

and the alloy as the positive, with current

flowing readily from the base plate to the

alloy but encountering high resistance in the

opposite direction (figure 11A). This action

results in effective rectification of an alternat-

ing input voltage and current with the efficien-

cy of conversion dependent to some extent

upon the ratio of the resistance in the con-

ducting direction to that of the blocking di-

rection. In normal power applications a ratio

of 100 to 1 is satisfactory; however, special

applications such as magnetic amplifiers often

require ratios in the order of 1000 to 1.

The basic selenium rectifier cell is actually

a diode capable of half wave rectification.

Since many applications require full wave rec-

tification for maximum efficiency and mini-

mum ripple, a plurality of cells in series, paral-

lel, or series -parallel combinations are stacked

in an assembly.

Selenium rectifiers are operated over a wide

range of voltages and currents. Typical appli-

cations range from a few volts at milliamperes

of current to thousands of amperes at rela-

tively high voltages.

The efficiency of high quality selenium rec-

tifiers is high, usually in the order of 90%

in three phase bridge circuits and 70% in

single phase bridge circuits. Of particular in-

terest is the very slight decrease in efficiency

even at high current overloads (figure 11B).

Threshold Voltage A minimum voltage is re-

and Aging quired to permit a selen-

ium rectifier to conduct

in the forward direction. This voltage, com-

monly known as the threshold voltage, pre-

cludes the use of selenium rectifiers at ex-

POSITIVE TERMINAL CONTACT

q,yp\ww

My%/ Irrror

11711.

\\dA \J

NEGATIVE TERMINAL

SILICON CELL SPRING

Figure 13

THE SILICON CELL

The common silicon rectifier is a pressure

contact device capable of operation in am-

bient temperatures as high as 150 °C. Heavy

end ferrules that fit standard fuse clips are

large enough to provide "heat sink" action.

The positive ferrule is grooved to provide

polarity identification and prevent incorrect

mounting.

tremely low ( less than one volt) applications.

The threshold voltage will vary with tem-

perature and will increase with a decrease in

temperature.

Under operating conditions, and to a lesser

extent when idle, the selenium rectifier will

age. During the aging period the forward

resistance will gradually increase, stabilizing

at a new, higher value after about one year.

This aging will result in approximately a 7%

decrease in output voltage.

Voltage The selenium rectifier has ex-

Regulation tremely low internal impedance

which exhibits non -linear charac-

teristics with respect to applied voltage. This

results in good voltage regulation even at large

overload currents. Figure 12 shows that as the

load is varied from zero to 300% of normal,

the output voltage will change about 10 %.

It should be noted that because of non -linear

characteristics, the voltage drop increases rapid-

ly below 50% of normal load.

Silicon Of all recent developments in the

Rectifiers field of semi -conductors, silicon

rectifiers offer the most promising

range of applications; from extreme cold to

high temperature, and from a few watts of

output power to very high voltage and cur-

rents. Inherent characteristics of silicon allow

junction temperatures in the order of 200 °C

before the material exhibits intrinsic proper-

ties. This extends the operating range of sili-

con devices beyond that of any other efficient

semi -conductor and the excellent thermal range

coupled with very small size per watt of out-

put power make silicon rectifiers applicable

where other rectifiers were previously con-

sidered impractical.

Silicon The current density of a sili-

Current Density con rectifier is very high, and

on present designs ranges

www.americanradiohistory.com

HANDBOOK Mobile Power Supply 697

from 600 to 900 amperes per square inch of

effective barrier layer. The usable current den-

sity depends upon the general construction of

the unit and the ability of the heat sink to

conduct heat from the crystal. The small size

of the crystal is illustrated by the fact that a

rectifier rated at 15 amperes d.c., and 150

amperes peak surge current has a total cell

volume of only .00023 inches. Peak currents

are extremely critical because the small mass

of the cell will heat instantaneously and could

reach failure temperatures within a time lapse

of microseconds. The assembly of a typical

silicon cell is shown in figure 13.

Operating

Characteristics The reverse direction of a sili-

con rectifier is characterized

by extremely high resistance,

up to 10" ohms below a critical voltage point.

This point of avalanche voltage is the region

of a sharp break in the resistance curve, fol-

lowed by rapidly decreasing resistance (figure

15A). In practice, the peak inverse working

voltage is usually set at least 20% below the

avalanche point to provide a safety factor.

The forward direction, or direction of low

resistance determines the majority of power

loss within the semi -conductor device. Figure

15B shows the static forward current charac-

teristics versus applied voltage. The threshold

voltage is about 0.6 volts.

Since the forward resistance of a semi-con-

ductor is very low, any unbalance between

threshold voltages or internal voltage drop

would cause serious unbalance of load distri-

bution and ultimate failure of the overloaded

section. A small resistance should therefore

be placed in series with each half wave section

Figure 14

MINIATURE SEMI- CONDUCTOR

TYPE RECTIFIER

Raytheon CK -777 power rectifier bolts to

chassis to gain large "heat sink" area. Low

internal voltage drop and high efficiency

permit small size of unit.

operating in parallel to balance the load cur-

rents. Some interesting and practical semi- conduc-

tor power supplies are shown in figure 16.

Remember that the circuits of figure 16A and

B, and those of figure 10 are "hot" with re-

spect to one side of the power line.

31 -5 100 Watt Mobile

Power Supply

High efficiency and compact size are the

most important factors in the design of mobile

power supplies. The power package described

in this section meets these stringent require-

W

o: W t

1 W K

Vt

O4 WF

>z

FW

<CC

JC WD

1:0UQ 0 25

PERCENT

00

! . MIS Mg'

to

.o / / ZSc

l ®

SO 75

RATED PEAK

too /as ISO

INVERSE VOLTAGE FORWARD VOLTAGE DROP

vairs, D.C.

Figure 15

SILICON RECTIFIER CHARACTERISTICS

A- Reverse direction of silicon rectifier is characterized by extremely high resistance up to point of

avalanche voltage.

B- Threshold voltage of silicon cell is about 0.6 volt. Once device starts conducting the current in-

creases exponentially with small increments of voltage, then nearly linearly on o very steep slope.

www.americanradiohistory.com

698 Power Supplies

*

Di 60 UF

D 45óv.

A

OA DUAL VOLTAGE DOUBLER

WITH COMMON B -MINUS

l2 .1/7

+ D3 6H,450MA

J L_ .05

117V.

ti 150LF y

250V z

+265 V.

150 MA.

265 V.

150 MA

150 LW 4011F

450v.T SOV1-

4-325V.

450 MA

Figure 16

SEMI -CONDUCTOR POWER SUPPLIES

A- Voltage quadrupler circuit. It point "A" is taken as ground instead of point "B," supply will deliver

530 volts at ISO ma. from 115 volt a -c line. Supply is "hot" to line.

B- Voltage tripler delivering 325 volts at 450 ma. Supply is "hot" to line.

C -900 watt supply for sideband service may be mode from two voltage quadruplets working in series

from inexpensive "distribution- type" transformer. Supply features good dynamic voltage regulation.

PARTS LIST:

D,- Sarkes Tarnion Model 150 selenium cell or Model M -500 silicon cell.

Dr- Sarkes Tarzian Model 500 selenium cell or Model M -500 silicon cell.

T, -Power distribution transformer, used backwards. 230/460 primary, 115/230 secondary, 0.75 KVA.

Chicago PCB -24750.

ments, delivering 100 watts at various voltages

to run both the mobile transmitter and re-

ceiver. The input power required by the sup-

ply is almost directly proportional to the out-

put power drain, and only a small amount of

power is wasted to actuate the vibrator reeds.

The large demand for efficient and power-

ful mobile radio equipment has led to the

development of new heavy -duty, vibrator -type

power supply components; these are used to

advantage in this unique design.

The Split -Reed The new split -reed dual- inter-

Vibrator rupter vibrator overcomes the

power capacity limitations of

the older type vibrators. In addition, this vi-

brator permits the design of power supplies

requiring no component changes for operation

from either the 6- or 12 -volt d.c. power sys-

tems with which most automobiles are

equipped.

Until recently, the majority of vibrator sup-

plies have been designed around the synchon-

ous type vibrator. A simplified diagram of

such a vibrator is shown in figure 18A. One

set of vibrator contacts switches the battery

current alternately through two opposed pri-

mary windings of a transformer, thus inducing

a square -wave a.c. voltage in the secondary.

This a.c. secondary voltage is then rectified by a

second set of contacts on the vibrator arma-

ture. The limitations of this circuit are low

power handling capacity and the need to use

a different number of turns on the transformer

primary when the battery voltage is changed

from six to twelve volts.

In the split -reed vibrator (figure 18B), two

sets of double -throw contacts are electrically

isolated from each other. Each set has much

greater current carrying capacity than the

synchronous vibrator. However, a power trans-

former having two center - tapped primary

windings is required. One set of contacts

switches the d.c. power alternately between

halves of one primary winding, and the other

set simultaneously switches the other winding.

Therefore, the primaries can be connected in

parallel for 6 -volt operation, or in series for

12 -volt operation. No wiring changes are

needed in the supply if the battery is properly

connected to the power input terminals.

The Selenium A selenium rectifier system

Rectifier System is employed in this supply.

Since the vibrator contacts

open and close abruptly, the periodically in-

www.americanradiohistory.com

Figure 17

100 WATT MOBILE

POWER SUPPLY

This high efficiency mo-

bile supply features 6 or

12 volt input, and will

completely power both

the mobile transmitter

and receiver. 450, 300,

and 240 volts are avail-

able. At right is control

box for complete mobile

system. Overall size of

supply is only 6 "x6 "x6 ".

terrupted voltage impressed on the transform-

er primary has practically a square waveform.

The secondary voltage also has nearly a square

waveform as a result and thus the peak volt-

age on the rectifiers is only slightly higher

than the average voltage of the waveform.

This means that the rectifiers in a vibrator -

type power supply can be operated on a square

wave voltage close to their maximum peak

inverse voltage rating, instead of considerably

below this rating, as when a sine wave a.c.

voltage is applied to them.

Power Supply This power supply has three

Circuit high voltage sections as shown

in figure 19. The 250 volt

receiver supply is shown in figure 19A, the

300 volt low voltage transmitter supply in

figure 19B, and the 450 volt high voltage

transmitter supply is shown in figure 19C.

Note that each rectifier has two sections. The

part numbers correspond to the markings

shown in the complete schematic of figure

20. In the 250 volt circuit, the full trans-

former secondary voltage is applied to a full

wave rectifier consisting of half of rectifiers

SRI and SIL. These two rectifiers form a com-

mon portion of all three rectifier circuits. The

junction of the two rectifiers is grounded and

the positive 250 volt output is taken from

the center -tap transformer lead. This is the

opposite of the usual full wave rectifier cir-

cuit. The 300 volt d.c. output is obtained from

a bridge rectifier circuit (figure 19B) con-

sisting of one -half of SR, and SR_ in the

ground legs, plus SR3 in the two bridge legs

from which the positive voltage is obtained.

Another bridge rectifier circuit is employed

for the 450 volt d.c. output, again with the

SYNCHRONOUS

VIBRATOR VIBRATOR

TRANSFORMER

+H V

OUT

SPLIT REED

VIBRATOR DUAL PRIMARY

VIBRATOR TRANSFORMER

O

1_1010t1 6 VOLT CONNECTIONS

I 1 1 I

IIIIIIIIII + 12 VOLT CONNECTIONS

Figure 18

SUPPLY FEATURES THE NEW SPLIT -REED VIBRATOR CAPABLE OF HANDLING

100 WATTS OF POWER AT EITHER 6 OR 12 VOLTS

A- Simplified diagram of typical synchronous vibrator and rectifier circuit.

B- Simplified diagram of typical split -reed vibrator and 6 - 12 volt d -c. changeover system.

www.americanradiohistory.com

700 Power Supplies THE RADIO

Ti

3

250 VOLT SUPPLY

SRin

SR 2A

300 VOLT SUPPLY

SR.

SR2A

450 VOLT SUPPLY

Figure 19

SIMPLIFIED CIRCUITS OF RECTIFIER SYSTEMS USED IN HIGH EFFICIENCY

MOBILE SUPPLY

A -Full wave, 250 volt rectifier.

B -300 volt bridge circuit tapped across portion of the high voltage winding.

C-450 volt bridge rectifier circuit.

two grounded legs of the circuit passing

through SR. and SR:. The other two sections

of these rectifiers form the legs of the bridge

from which the positive d.c. voltage is taken.

Since there is little ripple voltage in the

d.c. output from the rectifiers, a single 40

µfd. filter capacitor on the 300 volt section

of the supply is adequate. Two 100 µfd. ca-

pacitors are placed in series across the 450

volt section and a capacitor input filter is

employed for the low voltage section.

The Control A "push -to- talk" power switch -

Circuit ing circuit is included in the

power supply so that the 250

volt output may be applied to a mobile re-

ceiver or converter. This is accomplished with

one pair of contacts on a d.p.d.t. relay, RY.,

which also turns on the 300 volt output to a

transmitter exciter when its relay coil is en-

ergized. A second d.p.d.t. relay, RY_, turns

on the 450 volt output to a transmitter final

amplifier and modulator when its coil is ener-

gized. It also provides a power- reducing fea-

ture when the coil is not energized by apply-

ing 300 volts to the 450 volt output terminal

and 250 volts to the 300 volt output terminal.

Since all rectifiers have high voltage on

them continuously when the supply is operat-

ing, the idling current flow through the un-

used rectifiers when receiving is reduced to

almost zero by disconnecting them from the

filter capacitors and bleeder resistors by the

action of the relays.

A special 20 volt secondary winding on the

transformer provides transmitter negative bias

through rectifier SR. and a 50 µfd. filter ca-

pacitor.

Circuit Details -- In this power supply,

Low Voltage Section changing from 6- to 12-

volt operation is accom-

plished with a 12 -pin Cinch -Jones type 300

power plug and socket, P. and J., respectively.

A separate low voltage input cable is used

for each voltage as shown in figure 20. Leads

from P. to the main power relay RY3 should

be made as short as possible to reduce the

voltage drop. This is particularly important

with a six volt power source, where a cable

resistance of only 0.04 ohm will cause a one

volt primary drop when the power supply

is operating at full load.

The six volt input plug connects the alter-

nate pairs of vibrator contacts and the halves

of the transformer primary windings in paral-

lel. These units are connected in series for

twelve volt operation as shown in figure 18B.

Six volts for the relay coils is supplied from

the power cable through pin 6 on J.. With the

twelve volt plug, a voltage dropping resistor,

R., is connected in series with the relay coils.

If the power supply is to be operated exclu-

sively from the higher primary voltage, re-

lays having twelve volt coils should be used,

thus eliminating R..

There usually is sparking at the vibrator

contacts so various capacitors and resistors

are incorporated in the primary circuit to sup-

press any radio noise generated by vibrator

action. These components are placed close

to the pins of the vibrator socket.

Component The heart of this power supply

Parts is the vibrator transformer de-

signed specially for two -way mo-

bile radio equipment. It is readily available

from many of the 3000 General Electric Co.

mobile radio service stations, or it may be

ordered from the address given in the parts

list of figure 20.

Power Supply All components except the

Construction power transformer are enclosed

inside a 6 "x6 "x6" aluminum

utility box for maximum protection against

dirt and dust. A sub -chassis made from 1/16-

www.americanradiohistory.com

HANDBOOK Mobile Power Supply 701

T1 10 TS,

t00LF

43T

2 3 5 6 7 6 9 10 11 12

1 2 3 4 S 6 7 6 9 10 11 12

JIPOWERINPUT 40,14T TROLIF

450V 450 V

V Y Y I RI 6 VOLT PLUG

ONO TO AUTO

6V. To Pz R Y3

1 2 3 4 5 6 7 6 9 10 1 1 12

GND TO AUTO

Ft

rRY2Al±

40Lgin

450 V.

TO 6 VOLTS

TO S3, R2.

(SEE F /G. 238)

PI la VOLT PLUG

12 V TO R2 RY3 F2 TO 12 VOLTS

TO S3, P2

(SEE FIG. 238)

RY1

e

O 2

RYI

NOTES

1- RELAYS SHOWN /N UNENERG /ZED POS /T /ON.

2 -ALL RES/SrORS I WATT UNLESS OTHERWISE NOTED

A.- MOUNT AT TRANSFORMER TERMINAL

25V BIAS

4-450/300V.

TRANSMI T.

GROUND

4-300V

TRANSMIT.

+250 V.

TRANSMIT.

. 250 V

RECEIVER

TO SI

TO S2

Figure 20

SCHEMATIC, 6 -12 VOLT MOBILE SUPPLY

E,- Vibrator, dual interrupter, split -reed; 6 volt coils, 116,c.p.s. reed frequency, 7 pin base. Mallory type

1701, Oak V -6853, Radiart 5722, or G.E. A- 7141584 -P3.

F, -30 ampere cartridge fuse.

Fr-15 ampere cartridge fuse.

J, -12 pin male chassis plug. Cinch -Jones P- 312 -AB

P, -72 pin female cable socket. Cinch -Jones 5 -312 -CCT

L, -7.5 henry, 100 ma. Stancor C -1421

L - -7 ph., 1 amp. Ohmite 2 -50 r -f choke

RY,, RY- D.P.D.T. relay, 6 volt coil. Advance MG '2c 6VD, Potter B Brumfield MR- 11D -6V, Guardian

200 -6D Coil and 200 -2 assembly, or Ohmite DOSX -158T, or equiv.

SR,, SR -Two section selenium rectifier, 150 ma., 380 volts peak inverse per section, connected for

doubler service. G.E. A- 7144141 -P2, or two Federal 1005 -A rectifiers in series for each leg (8 in all).

SR, -Two section selenium rectifier, 150 mo., 380 volts peak inverse per section, connected for center -tap

service. G.E. A- 7144141 -P1, or two Federal 1005 -A rectifiers with red terminals connected together.

SR.-150 ma., 64 volt peak inverse selenium rectifier. G. E. A- 7140806 -P1 or Federal 1015.

TI- Vibrator -type power transformer. Dual center- tapped 6 volt primaries. Secondaries: 420 volts c.t.

with 150 volt taps, 300 ma. and 20 volt, 150 ma. bias voltage winding. G.E. B- 7486449 -P1.

15, -8 terminal barrier strip. Cinch -Jones 8- 141 -Y.

Note: General Electric parts may be ordered from: D. S. Clark, G.E. Co., Product Service Renewal Parts

Section of Communication Products 509 Kent St., Utica, N.Y. Current prices plus shipping charges

are: T,, $14.70; E;, $5.75; SR SR,, SR,, $7.75 each; SR., $1.40.

inch thick aluminum is cut as shown in fig-

ures 21 and 22. A cut -out in one corner is

necessary to clear the lower portion of the

power transformer. All chassis drilling and

the cutouts for the power transformer, power

input plug, and output terminal strip should

be completed before the chassis is permanent-

ly fastened in place with small aluminum

angle brackets, two inches from the bottom of

the box.

One electrolytic capacitor, C,, is mounted

on an insulated plate furnished with the ca-

pacitor. When twisting the locking lugs, make

sure that they do not come near the metal

www.americanradiohistory.com

Figure 21

UNDER -CHASSIS

VIEW OF

POWER SUPPLY

The vibrator socket is

hidden by the resistors

and disc ceramic capaci-

tors of the hash filter.

R -F choke L: and the

25 pfd. capacitors are

above these parts on the

wall of the case. Resis-

tor R, is mounted to the

wall, with SR,, SR,, and

SR, in line below it.

chassis. The can of this capacitor is more than

200 volts "above ground" and should be in-

sulated with a fibre sleeve.

The large seven pin socket into which the

vibrator plugs should have two soldering lugs

placed under each mounting bolt for the by-

pass capacitors which are later wired to the

socket pins. If the power supply is to be

mounted with the chassis in a vertical plane,

pins 1 and 4 of the vibrator socket should be

in a vertical plane. The smaller components

below the chassis should not be installed until

parts above the chassis have been assembled

and wired.

The transformer primary leads are wired

to the input plug and vibrator socket with

shortest possible leads before other parts are

wired. Placement of the under -chassis parts

may be seen in figure 21. Layout of the parts

is not critical.

Testing the After the wiring has been

Supply checked, the power supply should

be checked at half input voltage,

but with full voltage applied to the vibrator

coil. This is done by temporarily removing

the jumper between pins 7 and 8 on the

twelve volt power plug, P,. Pin 8 is then

Figure 22

TOP VIEW OF

POWER SUPPLY

INTERIOR

Relays RY, and RY, are

at center above choke L,.

Power transformer T, is

mounted on end of cab-

inet. Across top of

chassis (left to right) are

vibrator and filter ca-

pacitors

www.americanradiohistory.com

Transistorized Power Supply 703

FROM

BATTERY

POWER

RELAY

AND

FUSE

3 WIRE CABLE

ev - se

12 V - 12

4 WIRE CABLE

TO RECEIVER

2 WIRE CABLE

*6 FOR 6 V.

*10 FOR 12V Pw J1 MOBILE

SUPPLY

TSt

PIJa J3 23_

CONTROL

BOX

PS _a__ J4

LOW VOLTAGE INPUT PLUG J2

1 2 3 4 5 6 7 6

S WIRE

HI- VOLTAGE

BIAS CABLE

4 WIRE HEATER

P4 6 RELAY CABLE

J3 FROM M.V. SUPPLY

1

g2 3 4

Seo-

HI -LO

11 TRANS

- MIT

1 2 3 4

JS TO RECEIVER

A VOLT POWER CABLE

1 2 3 4 5 6 7 6

TO RY3

CO L

(F/G 20)

TO HEATERS.

6 VOLTS

FROM PI

(FIG 20 )

Pa

rd

1 2 3 4

J4 TO TRANSMITTER

12 VOLT POWER CABLE

t0 RY3 TO HEATERS,

COIL t2 VOLTS

FROM P1

(F /G. 20) (F10.201

Figure 23

CONTROL WIRING

A -Block diagram of suggested power and

control cables and switching system for

mobile power supply.

B- Schematic diagram of suggested control

box including power plugs for changing

heaters for 6 or 12 volt operation. S,

"transmit- receive switch," Sr is "high -low"

switch, and S, is main power switch for

RY,.

C-6 and 12 volt power cables.

jumped to pin 9. The cable is attached to J1 on

the supply and to a six volt power source. Ap-

proximately half the rated voltage should be

measured at the output terminal strip if the

supply has been properly wired.

Replace the original connections on the

power plug and test the supply with full

input voltage. A 2500 ohm, 100 watt resistor,

or four 25 watt 115 volt lamp bulbs in series

make a good load resistance. The output volt-

ages should measure close to 450, 300, and

240 volts under load. Additional .02 µfd. by-

pass capacitors at RY3, the output terminal

strip, and the control box should eliminate

Figure 24

CABLE HARNESS, MAIN POWER

RELAY, RY;;, AND REAR OF

POWER CONTROL BOX

High voltage for the receiver, plus the control

switch circuits for RY and RY:, are brought

into the control box from the power supply

through

High voltage leads for the transmitter are

run directly from the supply, but the trans-

mitter heater power and transmit- receive"

control circuit is run to the transmitter

through control box circuit J, (figure 23A).

any hash still present during reception. Every

experienced "mobileer" will agree that noise

in each mobile installation usually must be

eliminated on a "search and filter" basis.

Installation This power supply may be op-

in the Car erated in conjunction with the

suggested configuration shown in

figure 23A. Note that a separate cable is

recommended for the heater power circuit to

reduce vibrator hash pickup, and to minimize

heater voltage variations when the supply is

switched from receive to transmit.

Provision has been made for changing the

mobile receiver and transmitter power circuits

for either six or twelve volt operation as shown

in the schematic of the control box in figure

23B. An eight contact plug and socket auto-

matically make the proper connections when

the six or twelve volt plug is attached.

31 -6 Transistorized

Power Supplies

The vibrator type of mobile supply achieves

an overall efficiency in the neighborhood of

70 %. The vibrator may be thought of as a

mechanical switch reversing the polarity of

the primary source at a repetition rate of 120

transfers per second. The switch is actuated by

a magnetic coil and breaker circuit requiring

appreciable power which must be supplied by

the primary source.

One of the principal applications of the

transistor is in switching circuits. The tran-

sistor may be switched from an "off" con-

dition to an "on" condition with but the ap-

www.americanradiohistory.com

704 Power Supplies THE RADIO

Figure 25

TRANSISTORS CAN REPLACE VIBRATOR IN MOBILE POWER SUPPLY SYSTEM

A- Typical vibrator circuit.

B- Vibrator can be represented by two single -pole single throw switches, or transistors.

C- Push -pull square wave "oscillator" is driven by special feedback windings on power transformer.

D- Addition of bias in base- emitter circuit results in oscillator capable of starting under full load.

plication of a minute exciting signal. When

the transistor is nonconductive it may be con-

sidered to be an open circuit. When it is in

a conductive state, the internal resistance is

very low. Two transistors properly connected,

therefore, can replace the single pole, double

throw mechanical switch representing the vi-

brator. The transistor switching action is many

times faster than that of the mechanical vi-

brator and the transistor can switch an ap-

preciable amount of power. Efficiencies in the

neighborhood of 95% can be obtained with 28

volt primary -type transistor power supplies,

permitting great savings in primary power

over conventional vibrators and dynamotors.

Transistor The transistor operation resembles

Operation a magnetically coupled multi -vi-

brator, or an audio frequency

push -pull square wave oscillator (figure 25C).

A special feedback winding on the power trans-

w

a F 40

35

Cr O 30

w 25

J 20

Ú 'S

O 10

1-

W 5

tr/ o

CO 0 5 VOLT

2.5 VOLTS `RISE TIME =l01/S

*200-2000+

!/s

TIME

.200-2000r

Us

Figure 26

BASE -COLLECTOR WAVEFORM OF

SWITCHING CIRCUIT,

FOR 12 VOLT CIRCUIT

Square waveshape produces almost ideal

switching action. Small "spike" on leading

edge of pulses may be reduced by proper

transformer design.

former provides 180 degree phase shift volt-

age necessary to maintain oscillation. In this

application the transistors are operated as on-

off switches; i.e., they are either completing

the circuit or opening it. The oscillator output

voltage is a square wave having a frequency

that is dependent upon the driving voltage,

the primary inductance of the power trans-

former, and upon the peak collector current

drawn by the conducting transistor. Changes

in transformer turns, core area, core material,

and feedback turns ratio have an effect on the

frequency of oscillation. Frequencies in corn -

mon use are in the range of 120 c.p.s. to

3,500 c.p.s.

The power consumed by the transistors is

relatively independent of load. Loading the

oscillator causes an increase in input current

that is sufficient to supply the required power

to the load and the additional losses in the

transformer windings. Thus, the overall effi-

ciency actually increases with load and is great-

est at the heaviest load the oscillator will sup-

ply. A result of this is that an increase in load

produces very little extra heating of the tran-

sistors. This feature means that it is impossi-

ble to burn out the transistors in the event of

a shorted load since the switching action mere-

ly stops.

Transistor The power capability of the

Power Rating transistor is limited by the

amount of heat created by the

current flow through the internal resistance

of the transistor. When the transistor is con-

ducting the internal resistance is extremely

low and little heat is generated by current

flow. Conversely, when the transistor is in a

cut -off condition the internal resistance is very

high and the current flow is extremely small.

Thus, in both the "on" and "off" conditions

www.americanradiohistory.com

HANDBOOK Transistorized Power Supplies 705

2N278

SO MA = 12 WATTS

+500 V. AT ISO MA = 75 WATTS

2N27B OR 2N301A

T2

TOTAL POWER = E7 WATTS

+275 V. AT 125 MA = 35 WATTS

IV/

Figure 27

PRACTICAL TRANSISTOR POWER SUPPLIES

Te- Chicago DCT -2 transistor transformer

Tr--Chicago OCT-1 transistor transformer

D,- D.- Sorkes- Tarzian M -S00 silicon rectifier or equivalent.

the transistor dissipates a minimum of power.

The important portion of the operating cycle

is that portion when the actual switching from

one transistor to the other occurs, as this is

the time during which the transistor may be

passing through the region of high dissipation.

The greater the rate of switching, in general,

the faster will be the rise time of the square

wave ( figure 26) and the lower will be the

internal losses of the transistor. The average

transistor can switch about eight times the

power rating of class A operation of the

unit. Two switching transistors having 5 watt

class A power output rating can therefore

switch 80 watts of power when working at

optimum switching frequency.

Self -Starting The transistor supply shown in

Oscillat. rs figure 25C is impractical be-

cause oscillations will not start

under load. Base current of the proper po-

larity has to be momentarily introduced into

the base -emitter circuit before oscillation will

start and sustain itself. The addition of a bias

resistor ( figure 25D) to the circuit results

in an oscillator that is capable of starting

under full load. R1 is usually of the order of

10 -50 ohms while R: is adjusted so that ap-

proximately 100 milliamperes flows through

the circuit.

The current drawn from the battery by this

network flows through R: and then divides

between RI and the input resistances of the

two transistors. The current flowing in the

emitter -base circuit depends upon the value

of input resistance. The induced voltage across

the feedback winding of the transformer is a

square wave of such polarity that it forward -

biases the emitter -base diode of the transistor

that is starting to conduct collector current,

and reverse -biases the other transistor. The

forward -biased transistor will have a very low

input impedance, while the input impedance

of the reversed- biased transistor will be quite

high. Thus, most of the starting current drained

from the primary power source will flow

in R1 and the base -emitter circuit of the for-

ward- biased transistor and very little in the

other transistor. It can be seen that R, must not

be too low in comparison to the input re-

www.americanradiohistory.com

706 Power Supplies T H E R A D I O

Figure 28

35 WATT TRANSISTOR

POWER SUPPLY

Two 2H301A power transistors are used in

this midget supply. Transistors are mounted

on sanded portion of chassis deck which acts

as "heat sink." See text for details.

sistance of the conducting transistor, or else

it will shunt too much current from the tran-

sistor. When switching takes place, the trans-

former polarities reverse and the additional

current now flows in the base -emitter circuit

of the other transistor.

The Power The power transformer in a

Transformer transistor -type supply is designed

to reach a state of maximum flux

density (saturation) at the point of maximum

transistor conductance. When this state is

reached the flux density drops to zero and

reduces the feedback voltage developed in the

base winding to zero. The flux then reverses

because there is no conducting transistor to

sustain the magnetizing current. This change

of flux induces a voltage of the opposite po-

larity in the transformer. This voltage turns

the first transistor off and holds the second

transistor on. The transistor instantly reaches

a state of maximum conduction, producing a

state of saturation in the transformer. This

action repeats itself at a very fast rate. Switch-

ing time is of the order of 5 to 10 micro-

seconds, and saturation time is perhaps 200

to 2,000 microseconds. The collector wave-

form of a typical transistor supply is shown

in figure 26. The rise time of the wave is

about 5 microseconds, and the saturation time

is 500 microseconds. The small "spike" at

the leading edge of the pulse has an ampli-

tude of about 21/2 volts and is a product of

switching transients caused by the primary

leakage reactance of the transformer. Proper

transformer design can reduce this "spike" to

a minimum value. An excessively large "spike"

can puncture the transistor junction and ruin

the unit.

31 -7 Two

Transistorized

Mobile Supplies

The new Chicago- Standard Transformer Co.

series of power transformers designed to work

in transistor -type power supplies permits the

amateur and experimenter to construct ef-

ficient mobile power supplies at a fraction

of their former price. Described in this section

are two power supplies designed around these

efficient transformers. The smaller supply de-

livers 35 watts (275 volts at 125 milliam-

peres) and the larger supply delivers 85 watts

(500 volts at 125 milliamperes and 250 volts

Figure 29

SCHEMATIC, TRANSISTOR

POWER SUPPLY FOR

12 VOLT AUTOMOTIVE

SYSTEM

T -Transistor power transformer.

12 volt primary, to provide 275

volts at 125 ma. Chicago Stand-

ard DCT -1.

D -D -Sarkes- Tarzian silicon recti-

fier, type M -500, or equivalent.

2N301 A

Ti

DI D2

275 V.

AT 125101A.

www.americanradiohistory.com

HANDBOOK Components 707

at 50 milliamperes, simultaneously). Both

power units operate from a 12 volt primary

source.

The 35 Watt The 35 watt power unit uses

Supply two inexpensive RCA 2N301A

P -N -P -type power transistors

for the switching elements and four silicon

diodes for the high voltage rectifiers. The

complete schematic is shown in figure 29.

Because of the relatively high switching fre-

quency only a single 20 µfd. filter capacitor

is required to provide pure d.c.

Regulation of the supply is remarkably good.

No load voltage is 310 volts, dropping to

275 volts at maximum current drain of 125

milliamperes.

The complete power package is built upon

an aluminum chassis -box measuring 54 "x

3 "x2 ". Paint is removed from the center por-

tion of the box to form a simple heat sink

for the transistors. The box therefore conducts

heat away from the collector elements of the

transistors. The collector of the transistor is

the metal case terminal and in this circuit is

returned to the negative terminal of the pri-

mary supply. If the negative of the automobile

battery is grounded to the frame of the car

the case of the transistor may be directly

grounded to the unpainted area of the chassis.

If the positive terminal of the car battery is

grounded it is necessary to electrically insulate

the transistor from the aluminum chassis, yet

at the same time permit a low thermal barrier

to exist between the transistor case and the

power supply chassis. A simple method of

accomplishing this is to insert a thin mica

sheet between the transistor and the chassis.

"Two Mil" (0.002 ") mica washers for tran-

sistors are available at many large radio sup-

ply houses. The mica is placed between the

transistor and the chassis deck, and fibre wash-

ers are placed under the retaining nuts hold-

ing the transistor in place. When the tran-

sistors are mounted in place, measure the

collector to ground resistance with an ohm-

meter. It should be 100 megohms or higher

in dry air. After the mounting is completed,

spray the transistor and the bare chassis sec-

tion with plastic Krylon to retard oxidation..

Several manufacturers produce anodized alumi-

num washers that serve as mounting insula-

tors. These may be used in place of the mica

washers, if desired. The under -chassis wiring

of the supply may be seen in figure 30.

The 85 Watt Figure 31 shows the schematic

Supply of a dual voltage transistorized

mobile power supply. A bridge

AIL '.s,,t Faczi.

Figure 30

UNDER -CHASSIS LAYOUT OF PARTS

Two 10 pfd. capacitors are connected in

parallel for 20 pfd. output filter capacitor.

Silicon rectifiers are mounted in dual fuse

clips at end of chassis. Transistors are insu-

lated from chassis with thin mica sheets and

fibre washers if supply is used with positive -

grounded primary system.

rectifier permits the choice of either 250 volts

or 500 volts, or a combination of both at a

total current drain that limits the secondary

power to 85 watts. Thus, 500 volts at 170

milliamperes may be drawn, with correspond-

ingly less current as additional power is drawn

from the 250 volt tap.

The supply is built upon an aluminum box -

chassis measuring 7"x5"x3", the layout close-

ly following that of the 35 watt supply. Delco

2N278 P -N -P -type transistors are used as the

switching elements and eight silicon diodes

form the high voltage bridge rectifier. The

transistors are affixed to the chassis in the

same manner as the 2N301A mounting de-

scribed previously.

31 -8 Power Supply

Components

The usual components which make up a

power supply, in addition to rectifiers which

have already been discussed, are filter ca-

pacitors, bleeder resistors, transformers, and

chokes. These components normally will be

purchased especially for the intended appli-

cation, taking into consideration the factors

discussed earlier in this chapter.

Filter There are two types of filter ca-

Capacitors pacitors: (1) paper dielectric

type, (2) electrolytic type.

Paper capacitors consist of two strips of

metal foil separated by several layers of special

www.americanradiohistory.com

708 Power Supplies T H E R A D I O

2N278

+

20LF

4S0 V.V.

Da

D4

+ +

Do

De

2500

500v

Figure 31

SCHEMATIC,

85 WATT

TRANSISTOR

POWER SUPPLY

FOR 12 VOLT

AUTOMOTIVE

SYSTEM

T -Transistor power

transformer. 12 volt

primary to provide 275

volts at 125 mo.

Chicago Standard

DCT -2.

D -D -Sarkes- Tarzian

silicon rectifier, type

M -500

paper. Some types of paper capacitors are

wax -impregnated, but the better ones, especial-

ly the high- voltage types, are oil- impregnated

and oil -filled. Some capacitors are rated both

for flash test and normal operating voltages;

the latter is the important rating and is the

maximum voltage which the capacitor should

be required to withstand in service.

The capacitor across the rectifier circuit in

a capacitor -input filter should have a wo -king

voltage rating equal at least to 1.41 times the

r -m -s voltage output of the rectifier. The re-

maining capacitors may be rated more nearly

in accordance with the d -c voltage.

The electrolytic capacitor consists of two

aluminum electrodes in contact with a conduct-

ing paste or liquid which acts as an electro-

lyte. A very thin film of oxide is formed on the

surface of one electrode, called the anode.

This film of oxide acts as the dielectric. The

electrolytic capacitor must be correctly con-

nected in the circuit so that the anode always

is at a positive potential with respect to the

electrolyte, the latter actually serving as the

other electrode (plate) of the capacitor. A re-

versal of the polarity for any length of time

will ruin the capacitor.

The dry type of electrolytic capacitor uses

an electrolyte in the form of paste. The die-

lectric in electrolytic capacitors is not perfect;

these capacitors have a much higher direct

current leakage than the paper type.

The high capacitance of electrolytic ca-

pacitors results from the thinness of the film

which is formed on the plates. The maximum

voltage that can be safely impressed across

the average electrolytic filter capacitor is be-

tween 450 and 600 volts; the working voltage

is usually rated at 450. When electrolytic ca-

pacitors are used in filter circuits of high -

voltage supplies, the capacitors should be con-

nected in series. The positive terminal of one

capacitor must connect to the negative ter-

minal of the other, in the same manner as

dry batteries are connected in series.

It is not necessary to connect shunt resis-

tors across each electrolytic capacitor section

as it is with paper capacitors connected in

series, because electrolytic capacitors have

fairly low internal d -c resistance as compared

to paper capacitors. Also, if there is any varia-

tion in resistance, it is that electrolytic unit

in the poorest condition which will have the

highest leakage current, and therefore the volt-

age across this capacitor will be lower than

that across one of the series connected units

in better condition and having higher internal

resistance. Thus we see that equalizing re-

sistors are not only unnecessary across series -

connected electrolytic capacitors but are ac-

tually undesirable. This assumes, of course,

similar capacitors by the same manufacturer

and of the same capacitance and voltage rat-

ing. It is not advisable to connect in series

electrolytic capacitors of different make or

ratings.

There is very little economy in using elec-

trolytic capacitors in series in circuits where

more than two of these capacitors would be

required to prevent voltage breakdown.

Electrolytic capacitors can be greatly re-

duced in size by the use of etched aluminum

foil for the anode. This greatly increases the

surface area, and the dielectric film covering

it, but raises the power factor slightly. For

this reason, ultra- midget electrolytic capacitors

ordinarily should not be used at full rated d -c

voltage when a high a -c component is present,

www.americanradiohistory.com

HANDBOOK Special Supplies 709

such as would be the case for the input ca-

pacitor in a capacitor -input filter.

Bleeder A heavy duty resistor should be

Ree stors connected across the output of a

filter in order to draw some load

current at all time. This resistor avoids soar-

ing of the voltage at no load when swinging

choke input is used, and also provides a means

for discharging the filter capacitors when no

external vacuum -tube circuit load is connected

to the filter. This bleeder resistor should nor-

mally draw approximately 10 per cent of the

full load current.

The power dissipated in the bleeder resistor

can be calculated by dividing the square of

the d -c voltage by the resistance. This power

is dissipated in the form of heat, and, if the

resistor is not in a well -ventilated position,

the wattage rating should be higher than the

actual wattage being dissipated. High -voltage,

high- capacitance filter capacitors can hold a

dangerous charge if not bled off, and wire -

wound resistors occasionally open up without

warning. Hence it is wise to place carbon re-

sistors in series across the regular wire -wound

bleeder.

When purchasing a bleeder resistor, be sure

that the resistor will stand not only the re-

quired wattage, but also the voltage. Some re-

sistors have a voltage limitation which makes

it impossible to force sufficient current through

them to result in rated wattage dissipation.

This type of resistor usually is provided with

slider taps, and is designed for voltage divider

service. An untapped, non -adjustable resistor

is preferable as a high voltage bleeder, and is

less expensive. Several small resistors may be

connected in series, if desired, to obtain the

required wattage and voltage rating.

Transformers Power transformers and fila-

ment transformers normally

will give no trouble over a period of many

years if purchased from a reputable manufac-

turer, and if given a reasonable amount of

care. Transformers must be kept dry; even a

small amount of moisture in a high -voltage

unit will cause quick failure. A transformer

which is operated continuously, within its

ratings, seldom will give trouble from mois-

ture, since an economically designed trans-

former operates at a moderate temperature

rise above the temperature of the surrounding

air. But an unsealed transformer which is

inactive for an appreciable period of time in

a highly humid location can absorb enough

moisture to cause early failure.

Filter Choke Filter inductors consist of a

Coils coil of wire wound on a lami-

nated iron core. The size of

wire is determined by the amount of direct

current which is to flow through the choke

coil. This direct current magnetizes the core

and reduces the inductance of the choke coil;

therefore, filter choke coils of the smoothing

type are built with an air gap of a small frac-

tion of an inch in the iron core, for the pur-

pose of preventing saturation when maximum

d.c. flows through the coil winding. The "air

gap" is usually in the form of a piece of fiber

inserted between the ends of the laminations.

The air gap reduces the initial inductance of

the choke coil, but keeps it at a higher value

under maximum load conditions. The coil must

have a great many more turns for the same

initial inductance when an air gap is used.

The d -c resistance of any filter choke should

be as low as practicable for a specified value

of inductance. Smaller filter chokes, such as

those used in radio receivers, usually have

an inductance of from 6 to 15 henrys, and a

d -c resistance of from 200 to 400 ohms. A

high d -c resistance will reduce the output volt-

age, due ro the voltage drop across each choke

coil. Large filter choke coils for radio trans-

mitters and Class B amplifiers usually have

less than 100 ohms d -c resistance.

31 -9 Special Power

Supplies

A complete transmitter usually includes one

or more power supplies such as grid -bias

packs, voltage -regulated supplies, or trans-

formerless supplies having some special char-

acteristic.

Regulated Where it is desired in a circuit

Supplies- to stabilize the voltage supply

V -R Tubes to a load requiring not more than

perhaps 20 to 25 ma., the glow -

discharge type of voltage -regulator tube can

be used to great advantage. Examples of such

circuits are the local oscillator circuit in a

receiver, the tuned oscillator in a v -f -o -, the

oscillator in a frequency meter, or the bridge

circuit in a vacuum -tube voltmeter. A num-

ber of tubes are available for this applica-

tion including the 0A3 /VR75, OB3 /VR90,

0C3 /VR105, OD3 /VR150, and the 0A2

and OB2 miniature types. These tubes stabil-

ize the voltage across their terminals to 75,

90, 105, or 150 volts. The minature types

0A2 stabilize to 150 volts and OB2 to 108

volts. The types 0A2, OB2 and OB3 /VR90

www.americanradiohistory.com

710 Power Supplies T H E R A D I O

have a maximum current rating of 30 ma.

and the other three types have a maximum

current rating of 40 ma. The minimum cur-

rent required by all six types to sustain a

constant discharge is 5 ma.

A VR tube (common term applied to all

glow- discharge voltage regulator tubes) may

be used to stabilize the voltage across a vari-

able load or the voltage across a constant load

fed from a varying voltage. Two or more VR

tubes may be connected in series to provide

exactly 180, 210, 255 volts or other combi-

nations of the voltage ratings of the tubes.

It is not recommended, however, that VR

tubes be connected in parallel since both the

striking and the regulated voltage of the

paralleled tubes normally will be sufficiently

different so that only one of the tubes will

light. The remarks following apply generally

to all the VR types although some examples

apply specifically to the OD3 /VR150 type.

A device requiring say, only 50 volts can

be stabilized against supply voltage variations

by means of a VR -105 simply by putting a

suitable resistor in series with the regulated

voltage and the load, dropping the voltage

from 105 to 50 volts. However, it should be

borne in mind that under these conditions

the device will not be regulated for varying

load; in other words, if the load resistance

varies, the voltage across the load will vary,

even though the regulated voltage remains at

105 volts.

To maintain constant voltage across a vary-

ing load resistance there must be no series

resistance between the regulator tube and the

load. This means that the device must be

operated exactly at one of the voltages ob-

tainable by seriesing two or more similar or

different VR tubes.

A VR -150 may be considered as a stubborn

variable resistor having a range of from 30,000

to 5000 ohms and so intent upon maintaining

a fixed voltage of 150 volts across its ter-

minals that when connected across a voltage

source having very poor regulation it will in-

stantly vary its own resistance within the lim-

its of 5000 and 30,000 ohms in an attempt

to maintain the same 150 volt drop across its

terminals when the supply voltage is varied.

The theory upon which a VR tube operates is

covered in an earlier chapter.

It is paradoxical that in order to do a good

job of regulating, the regulator tube must be

fed from a voltage source having poor regula-

tion (high series resistance) . The reason for

this presently will become apparent.

If a high resistance is connected across the

VR tube, it will not impair its ability to main-

tain a fixed voltage drop. However, if the load

is made too low, a variable 5000 to 30,000

ohm shunt resistance (the VR -150) will not

exert sufficient effect upon the resulting re-

sistance to provide constant voltage except

over a very limited change in supply voltage

or load resistance. The tube will supply maxi-

mum regulation, or regulate the largest load,

when the source of supply voltage has high

internal cr high series resistance, because a

variation in the effective internal resistance

of the VR tube will theti have more control-

ling effect upon the load shunted across it.

In order to provide greatest range of regu-

lation, a VR tube (or two in series) should be

used with a series resistor (to effect a poorly

regulated voltage source) of such a value that

it will permit the VR tube to draw from 8 to

20 ma. under normal or average conditions

of supply voltage and load impedance. For

maximum control range, the series resistance

should be not less than approximately 20,000

ohms, which will necessitate a source of volt-

age considerably in excess of 150 volts. How-

ever, where the supply voltage is limited, good

control over a limited range can be obtained

with as little as 3000 ohms series resistance.

If it takes less than 3000 ohms series resist-

ance to make the VR tube draw 15 to 20

ma. when the VR tube is connected to the

load, then the supply voltage is not high

enough for proper operation.

Should the current through a VR -150, VR-

105, or VR -75 be allowed to exceed 40 ma.,

the life of the tube will be shortened. If the

current falls below 5 ma., operation will be-

come unstable. Therefore, the tube must op-

erate within this range, and within the two

extremes will maintain the voltage within 1.5

per cent. It takes a voltage excess of at least

10 to 15 per cent to "start" a VR type regu-

lator; and to insure positive starting each time

the voltage supply should preferably exceed

the regulated output voltage rating about 20

per cent or more. This usually is automatically

taken care of by the fact that if sufficient

series resistance for good regulation is em-

ployed, the voltage impressed across the VR

tube before the VR tube ionizes and starts

passing current is quite a bit higher than the

starting voltage of the tube.

When a VR tube is to be used to regulate

the voltage applied to a circuit drawing less

than 15 ma. normal or average current, the

simplest method of adjusting the series resist-

www.americanradiohistory.com

HANDBOOK Special Supplies 711

Es

D C. SUPPLY

VOLTAGE

o

Rs, SERIES RESISTOR

VR TUBE

RL

LOAD

Figure 32

STANDARD VR -TUBE REGULATOR

CIRCUIT

The VR -tube regulator will maintain the volt-

age ccross its terminals constant within a

few volts for modcrcte variations in R, or E,.

See text for discussion of the use of VR

tubes in various circuit applications.

ance is to remove the load and vary the series

resistor until the VR tube draws about 30 ma.

Then connect the load, and that is all there is

to it. This method is particularly recommended

when the load is a heater type vacuum tube,

which may not draw current for several sec-

onds after the power supply is turned on.

Under these conditions, the current through

the VR tube will never exceed 40 ma. even

when it is running unloaded (while the heater

tube is warming up and the power supply

rectifier has already reached operating tem-

perature) .

Figure 32 illustrates the standard glow dis-

charge regulator tube circuit. The tube will

maintain the voltage across RI. constant to

within 1 or 2 volts for moderate variations in

Rs. or Es.

Voltage Regulated When it is desired to sta-

Power Supplies bilize the potential across

a circuit drawing more

than a kw milliamperes it is advisable to

use a voltage -regulated power supply of the

type illustrated in figure 33 rather than glow

discharge tubes.

A 6AS7 -G is employed as the series control

element, and type 816 mercury vapor rectifiers

are used in the power supply section. The

6AS7 -G acts as a variable series resistance

which is controlled by a separate regulator

tube much in the manner of a -v -c circuits or

inverse feedback as used in receivers and a -f

amplifiers. A 6SH7 controls the operating bias

on the 6AS7 -G, and therefore controls the in-

ternal resistance of the 6AS7 -G. This, in turn,

controls the output voltage of the supply, which

controls the plate current of the 6SH7, thus

completing the cycle of regulation. It is ap-

parent that under these conditions any change

in the output voltage will tend to "resist it-

self,' much as the a -v -c system of a receiver

resists any change in signal strength delivered

to the detector.

Because it is necessary that there always be

a moderate voltage drop through the 6AS7-G

in order for it to have proper control, the rest

of the power supply is designed to deliver as

much output voltage as possible. This calls

for a low resistance full -wave rectifier, a high

capacitance output capacitor in the filter sys-

tem and a low resistance choke.

Reference voltage in the power supply is

obtained from a VR -150 gaseous regulator.

Note that the 6.3 -volt heater winding for the

6SH7 and the 6AS7 -G tubes is operated at

a potential of plus 150 volts by connecting

the winding to the plate of the VR -150. This

procedure causes the heater -cathode voltage

of the 6SH7 to be zero, and permits an out-

put voltage of up to 450 since the 300 -volt

heater -to- cathode rating of the 6AS7 -G is not

exceeded with an output voltage of 450 from

the power supply.

The 6SH7 tube was used in place of the

more standard 6SJ7 after it was found that

the regulation of the power supply could be

improved by a factor of two with the 6SH7

in place of the 6SJ7. The original version of

the power supply used a 5R4 -GY rectifier

tube in place of the 816's which now are

used. The excessive drop of the 5R4 -GY re-

sulted in loss of control by the regulator por-

JUMPER IN

$W. -VR TUBE

115 V. PLATE 115 V. FILS O Oi

+325 TO GN0.1

450 V.

I _ J

Figure 33

SCHEMATIC OF VOLTAGE

REGULATED POWER SUPPLY

T -615 or 520 volts each side of c.t., 300

ma. Stancor P -8041.

T : -5 volts at 3 amp., 6.3 volts at 6 amp.

Stancor P -5009.

CH -4 henry at 250 ma. Stancor C -1412.

www.americanradiohistory.com

712 Power Supplies THE RADIO

TI

6H

SO MA.

5Y3-GT VR-150 6SJ7 6AS7 -G

+ 20

£X

3.3 V. TO HEATERS

X

TI=350-0-350 V.

AT 50 MA.

5V AT 2A.

6.3 V. AT 3A,

ALL RESISTORS 7 -WATT UNLESS OTHERWISE SPECIFIED.

VR-150

Figure 34.

SCHEMATIC, LOW VOLTAGE REGULATED BIAS SUPPLY.

ISO

11F -0060

tion with an output voltage of about 390

with a 225 -ma. drain. Satisfactory regulation

can be obtained, however, at up to 450 volts

if the maximum current drain is limited to

150 ma. when using a 5R4 -GY rectifier. If

the power transformer is used with the taps

giving 520 volts each side of center, and if

the maximum drain is limited to 225 ma.,

a type 83 rectifier may be used as the power

supply rectifier. The 615 -volt taps on the

power transformer deliver a voltage in excess

of the maximum ratings of the 83 tube. With

the 83 in the power supply, excellent regu-

lation may be obtained with up to about 420

volts output if the output current is limited

to 225 ma. But with the 816's as rectifiers

the full capabilities of all the components

in the power supply may be utilized.

If the power supply is to be used with an

output voltage of 400 to 450 volts, the full

615 volts each side of center should be ap-

plied to the 816's. However, the maximum

plate dissipation rating of the 6AS7 -G will

be exceeded, due to the voltage drop across

the tube, if the full current rating of 250 ma.

is used with an output voltage below 400

volts. If the power supply is to be used with

full output current at voltages below 400 volts

the 520 -volt taps on the plate transformer

should be connected to the 816's. Some varia-

tion in the output range of the power supply

may be obtained by varying the values of the

resistors and the potentiometer across the out-

put. However, be sure that the total plate

dissipation rating of 26 watts on the 6AS7 -G

series regulator is not exceeded at maximum

current output from the supply. The total dis-

sipation in the 6A57 -G is equal to the cur-

rent through it (output current plus the cur-

rent passing through the two bleeder strings)

multiplied by the drop through the tube (volt-

age across the filter capacitor minus the out-

put voltage of the supply).

A Shunt Regulated Many of the popular

Bias Supply class B modulator and

(20 to 80 V.) grounded grid linear am-

plifier tubes require a

few volts of well regulated, negative bias.

Shown in figures 34 and 35 is an electronic

bias supply which will provide a regulated bias

voltage variable over the range of 20 to 80

volts. Regulation is 0.001 volt /ma., which is

remarkable for a supply as simple as this. Be-

tween 30 and 80 volts, the supply will regu-

late grid current up to 200 ma. Between 20

and 30 volts, maximum grid current is re-

stricted to 100 ma.

Figure 35.

Regulated bias supply may be built upon

small steel chassis. "Adjust bias" control is on

front apron, and current adjustment poten-

tiometer for regulator is next to power trans-

former.

www.americanradiohistory.com

HANDBOOK Special Supplies 713

ADJ. DIAS

100 N

TI

115v .N

T 1 ° 460 V. C.T. AT 40 MA.

5V., 2A. 6.3V, 2A.

6CK4 OR

TRIODE -CONNECTED

6L6 -GC

ALL RES /STOPS 7 -WATT.

Figure 36.

SCHEMATIC, HIGH VOLTAGE

REGULATED BIAS SUPPLY.

Basically, the regulated supply consists of a

small power supply which delivers plate volt-

age to a low -µ 6AS7 -G triode. The voltage

drop across the triode is used as the regulated

bias voltage. Associated with the triode is a

d.c. amplifier and a voltage regulator tube

which serve to vary the grid voltage of the

triode regulator tube so that a constant volt-

age is maintained across it. The 5K variable

potentiometer is adjusted to produce about 20

ma. current through the first regulator tube.

A Shunt Regulated Series regulated power

Bias Supply supplies are usually not

(100 to 600 V.) suited for bias units as

the direction of load cur-

rent flow is opposite from that of a regular

supply. In the supply shown in figure 36, the

regulator tube (6CK4) acts as a variable

bleeder resistor which automatically adjusts its

resistance to a value such that the grid current

flowing through it will develop a constant

voltage across the supply terminals. The tap

switch of this supply permits rough bias ad-

justment over the range of about 100 to 600

volts, while the potentiometer permits a fine

adjustment to be made. Maximum permissible

grid current runs from about 100 ma, in the

vicinity of 100 volts to about 25 ma. in the

600 volt region.

31 -10 Power Supply

Design

Power supplies may either be of the choke

input type illustrated in figure 37, or the ca-

pacitor input type, illustrated in figure 38. Ca-

pacitor input filter systems are characterized

115V

50 -60 L

V1

O

o 63V.

COMPONENTS APPROXIMATE

OUTPUT VOLTAGE MAX.

CURRENT 6.3V.

FILAMENT T1 V 1 CH 1 CH 2 C 1 C2 R 1

NO

LOAD FULL

LOAD

350 -0 -350

STANCORPC i 109

MERIT P -3732 573-GT

10 M.

STANCOR

C -7001

MERIT

C -2993

10 N.

STANCOR

C -1001

MERIT

C-2993

1OUF, 450 V.

CORNELL-

DUO /L /ER

OR -POS

20uF, 450 V.

CORNELL-

DUOIL /ER

OR -2045 3515,10W 310 240 ROMA. 3A.

375-0 -373

STANCOR PC-8411

MERIT P -2934 573 -GT

3 -1384

STANCOR

C -1776

MERIT

C -3167

711.

STANCOR

C -1421

MERIT

C -3760

101F,450 V.

CORNELL-

DUO /L /ER

8R -1017

201F,450 V.

CORNELL-

DUO /LIER

8R -203 356,10W 330 230 140MA. 4.5A.

400 -0 -400

STANCOR PC -0473 SU4 -G

2-12 M.

STANCOR

C -102

MERIT

C -3169

4 M.

STANCOR

C -112

MERIT

C -3162

1011F 450 V.

CORNELL-

Of/OILIER

OR -705

10 LIF,430 V.

CORNELL-

DUO /LIER

OR -105 356,10W 360 270 250MA. 3A.

525 -0 -525

UTC S-40 5U4 -G111 5 -2511

OTC 5 -32 20 H

UTC S -37

1010,600V.

MALLORY

TC -92

1010,600 V.

MALLORY

TC -92 356,10W 460 375 240 MA. 4A.

600 -0 -600

UTC S -/ 5114 -GT 5 -25 M.

UTC S -32 20 H.

UTC 5-31

61.1F, 600 V.

SPRAGUE

CR -96

8113,600V

SPRAGUE

CR -113 3511,23W 540 410 200MA. 4A.

900-0 -900

UTC S-45 SR { -GY 3 -25 N.

UTC S -32 20 N.

UTC S -31 0F, 1RV

SPRAGUE

CR -4I Sur, 16V.

SPRAGUE

CR -111 506,25W 1130 650 17SMA. -

Figure 37.

DESIGN CHART FOR CHOKE -INPUT POWER SUPPLIES

www.americanradiohistory.com

714 Power Supplies THE RADIO

5 V.

50 -00

T1 V1

o

O

6.3 V.

COMPONENTS APPROXIMATE

OUTPUT VOLTAGE MAX.

CURRENT 6.3V.

FILAMENT

T 1 V 1 CHI C 1 C 2 R 1

NO

LOAD FULL

LOAD

260 -0 -260

STANCORPCd104

MERIT P -3148 503 -GT

I0 H.

STANCOR

0-100/

MERIT

C-2993

20111%450 V.

CORNELL-

OUDILIER

8R -2043

2011F, 450 V.

CORNELL-

OU8IL /ER

8R -2045 3510,10W 340 240 60 MA. 3A.

375 -0 -375

STANCOR PC -8411

MERIT P -2934 503-GT

7 H.

STANCOR

C-121

MERIT

C -3180

IO11F, 600 V.

MALLORY

rC -92

10LF 600 V.

MALLORY

TC -92 35K,IOW 460 350 123 MA. 4.5A.

435-0 -435

MERIT P-3/36 5U4 -G

4H.

STANCOR

C-1412

MERIT

C -3182

810,600 V.

SPRAGUE

CR -06

610,600 V.

SPRAGUE

CR-U 351,25W 600 400 225 MA. 6 A.

600 -0 -600

STANCOR PC-8474

4 H.

STANCOR

MERIT

C-7412

MERIT

C -3182

411F, 1 KV

SPRAGUE

CR -47

6 LF, 1 KV.

SPRAGUE

CR -81 50 K,25W 800 600 200 MA. 6A.

900-0 -900

UTC S-43 5R4-GY 20 H.

¡ITC S-31

4Lf 1.5 KV.

SPRAGUE

CR -415

8111, 1.5 KV.

SPRAGUE

CR -813 75K 25W 1200 910 150 MA -

Figure 38

DESIGN CHART FOR CAPACITOR -INPUT PCWER SUPPLIES

by a d -c supply output voltage that runs from

0.9 to about 1.3 times the r.m.s. voltage of

one -half of the high voltage secondary wind-

ing of the transformer. The approximate regu-

lation of a capacitor input filter system is

shown in figure 39. Capacitor input filter

systems are not recommended for use with

mercury vapor rectifier tubes, as the peak rec-

tifier current may run as high as five or six

times the d -c load current of the power sup-

ply. It is possible, however, to employ type

872 -A mercury vapor rectifier tubes in ca-

pacitor input circuits wherein the load cur-

rent is less than 600 milliamperes or so, and

where a low resistance bleeder is used to hold

the minimum current drain of the supply to

a value greater than 50 milliamperes or so.

Under these conditions the peak plate current

of the 872 -A mercury vapor tubes will not

be exceeded if the input filter capacitor is

4 µfd. or less.

Choke input filter systems are characterized

by lower peak load currents (1.1 to 1.3 times

the average load current) than the capacitor

input filter, and by better voltage regulation.

Design Charts for capacitor and choke input

filter supplies for various voltages and load

currents are shown in figures 37, 38, and 40.

The construction of power supplies for trans-

mitters, receivers and accessory equipment is

a relatively simple matter electrically since

lead lengths and placement of parts are of

minor importance and since the circuits them-

selves are quite simple. Under -chassis wiring

of a heavy -duty supply is shown in figure 41.

Bridge Supplies Some practical variations of

the common bridge rectifier

circuit of figure 6 are illustrated in figures 42

and 43. In many instances a transmitter or

modulator requires two different supply volt-

ages, differing by a ratio of about 2:1. A

simple bridge supply such as shown in figure

42 will provide both of these voltages from

a simple broadcast "replacement -type" power

transformer. The first supply of figure 42 is

ample to power a transmitter of the 6CL6 -807

type to an input of 60 watts. The second sup-

ply will run a transmitter running up to 120

watts, such as one employing a pair of 6146

tetrodes in the power amplifier stage. It is

to be noted that separate filament transform-

ers are used for rectifier tubes V1 and V_, and

that one leg of each filament is connected to

the cathode of the respective tube, which is

at a high potential with respect to ground.

The choke CH1 in the negative lead of the

supply serves as a common filter choke for

www.americanradiohistory.com

HANDBOOK Special Supplies 715

MAO

MIN 0.9 1.0 1 I 1.2 1.3 l.a

RATIO OF D C OUTPUT VOLTAGE TO R.0 3 VOLTAGE

OF k SECONDARY WINDING OF PLATE TRANSFORMER.

Figure 39

APPROXIMATE REGULATION OF

CAPACITOR -INPUT FILTER SYSTEM

both output voltages. Each portion of the sup-

ply may be considered as having a choke in-

put filter system. Filaments of V1 should be

energized before the primary voltage is ap-

plied to L.

Bridge supplies may also be used to ad-

vantage to obtain relatively high plate volt-

ages for high powered transmitting equip-

ment. Type 866 -A and 872 -A rectifier tubes

can only serve in a supply delivering under

3500 volts in a full -wave circuit. Above this

voltage, the peak inverse voltage rating of the

rectifier tube will be exceeded, and danger

of flash -back within the rectifier tube will be

present. However, with bridge circuits, the

same tubes may deliver up to as much as 7000

volts d.c. without exceeding the peak inverse

voltage rating.

The bridge circuit also permits the use of

the so- called "pole transformer" in high volt-

age power supplies. Two KVA transformers of

this type having a 110/220 volt secondary

winding and a split 2200 volt primary wind-

ing may often be picked up in salvage yards

for a dollar or two. If reversed, and either

110 or 220 volts applied to the "primary"

winding approximately 2200 volts r.m.s. will

be developed across the new "secondary" wind-

ing. If used in a bridge circuit as shown in

figure 43, a d -c supply voltage of about 1900

volts at a current of 500 milliamperes may be

drawn from such a transformer. Do not at-

tempt to use a smaller transformer than the

2 -KVA rating, as the voltage regulation of

the unit will be too poor for practical pur-

poses.

For higher voltages, a pole transformer with

a 4400 volt primary and a 110/220 volt sec-

ondary may be reversed to provide a d -c plate

supply of about 3800 volts.

Commercial plate transformers intended for

full wave rectifier service may also be used

in bridge service provided that the insulation

at the center -tap point of the high voltage

winding is sufficient to withstand one -half of

the r.m.s. voltage of the secondary winding.

Many high voltage transformers are specifical-

ly designed for operation with the center -tap

T2

COMPONENTS APPROXIMATE

OUTPUT VOLTAGE MAX.

CURRENT

(

T1 Ta V1-V2 CHI Chia CI C2 RI NO

LOAD FULL

LOAD

1150-0-1150

CHICAGO TRANS.

P-l07

2.5V,10A.

CHI TRAN.

F-270 666-A

666-A

6H.

CN/.TRAN.

R-6!

ION.

CN/.TRAN.

R-/O7

411,1.5KV.

SANiAMO

7773-4

6UF,1.3KV

SAN0AM0

7113-6 40K,75W 1150 1000 350 NA.

1710-0-1710

CHICAGO TRANS.

P-I312

2.SV.,10A

CHI. TRAN.

f-2fON 666-A

666-A

6H.

CHI. TRAN.

R-63

ION.

CHI. TRAN

R-IOS

411F, 2KV

SANOAMO

7720-4

6LF,2KV.

SANGAMO

7120-6 50K,75W 1700 1500 425 MA.

2900-0-2900

CHICAGO TRANS.

P-t/21

SV., 10A.

CN/.7RAN.

F-S/ON 672A

672-A

6H.

CN/ 7RAN

R-67

6N.

CN/. TRAN.

R-67

4UF,3KV.

SANGAMO

7/30-4

4LF,3KV.

SANGAMO

7130-I 200VY 2750 2500 700 MA.

3500-0-3500

um CG-309

5 V.,10A.

UTC

LS-62

872-A

10H.

UTC

CL-IS

10H.

UTC

CG-75

4UF,4KV.

CORNELL-

7IOOIOEA

41JF 4KV.

CORÁELL-

TI00/IOSA

100K

200W 3400 3000 1000 MA.

4600-0-4600

UTC CC-!10

CHICAGO TRANS.

P-4]33

5V., 20A.

UTC

LS-63

575-A

573-A

10H.

(ITC

Ci- IS

IOU.

UTC

CE-IS

11F, SKV.

AEROVOX

JP-09

4UF,SKV

AEROVOX

JP-O9

100K

300 W 4400 4000 600 MA.

Figure 40

DESIGN CHART FOR CHOKE -INPUT HIGH VOLTAGE SUPPLIES

www.americanradiohistory.com

-

'. i t 17 i 4 ¡

;c)

RC Ú O

C;, U Ú

7 I

r. 0'2 4

s,4c-' -;

U

_

t .6.41

of the secondary winding at ground potential;

consequently the insulation of the winding at

this point is not designed to withstand high

voltage. It is best to check with the manufac-

turer of the transformer and find out if the

insulation will withstand the increased volt-

age before a full wave -type transfotmer is

utilized in bridge rectifier service.

31 -11 300 Volt,

50 Ma. Power Supply

There are many applications in the labora-

tory and amateur station for a simple low

drain power supply. The most common appli-

cation in the amateur station for such a sup-

ply is for items of test equipment such as the

type LM and BC -221 frequency meters, for

frequency converters to be used in conjunction

with the station receiver, and for auxiliary

equipment such as high selectivity i -f strips

or variable frequency oscillators. Equipments

Figure 41

UNDER -CHASSIS

POWER SUPPLY ASSEMBLY

All components are firmly mounted to the

steel chassis and all wiring is cabled. High

voltage leads are run in automobile ignition

cables. Heavy -duty terminal strips are mounted

along the rear edge of the chassis. The con-

trol panel of this supply is shown in figure I

of this chapter.

such as these may be operated from a supply

delivering 250 to 300 volts at up to 50 milli-

amperes of plate current. A filament source of

6.3 or 12 volts may also be required, as will

a source of regulated voltage.

The simple power supply illustrated in fig-

ures 44 and 45 is capable of meeting these

requirements. A two section capacitor input

filter system is employed to provide mini-

mum ripple content and a switch (S,) is

provided to insert a VR -150 regulator tube

in the circuit to provide 150 volts at ap-

proximately 35 milliamperes.

A separate 6.3 volt filament transformer is

connected in series with the 6.3 volt wind-

ing of power transformer T1 to provide 12.6

volts at 1.2 amperes for operating "12 volt"

tubes or a string of two "6 volt" tubes con-

nected in series. The secondary winding of

T_ must be polarized correctly to provide 12.6

volts across the two windings.

Resistor R1 must be adjusted to "fire" the

voltage regulator tube. It should be adjusted

so that approximately 15 milliamperes pass

through the tube. For maximum permissible

current drain from the high voltage tap of

the supply the VR tube should be switched

out of the circuit.

Ti

CH 2

V1

T2

ra

Tz,Ta=e.a v., 1 A.

STANCOR P-013

V2

ISV.

So-60 %.

+N.V. 2

+ N. V. i I

Figure 42

DUAL VOLTAGE

BRIDGE POWER SUPPLIES

COMPONENTS FULL LOAD VOLT. MAX. CURRENT

T/ V1-V2. V3 C1-C2 C3-C4 CH CH2 R1-R2 Ra MVO, Nvs2 s/ *2

a60-0-ae0 67(5-GT 5V4,0 1611F 1e1JF. 10H. 8H. 20R.10w. 100R 800 240 COMA. 40MA.

STANCORPC-Ni0 430 V. 430 V. 120 MA. SOMA. 1W

400-0-400 BAXSiT 554-G6 18LF. 101I e N. 20610W 2oR 10W 100 R 625 280 150 MA. So MA.

STANCORPC-84l2 450 V. 450 V. 22SMA. 75 MA. 1W

www.americanradiohistory.com

Ti 5Y3 -GT

2

CH CH

2011F 40ÚF

001

I

Sa

5

+325-

275 V.

20 K

20W

RI

+150V.

IA. S

115 V. 1,

RISO

6.3 V.

126V.

Figure 45

SCHEMATIC, LIGHT DUTY SUPPLY

T,-350 - 0 - 350 volts at 50 milliamperes, 5

volts at 2 amperes, 6.3 volts at 3 amperes,

"replacement" transformer.

T:-6.3 volts at 1.2 amp. Stancor P -6134

CH -4.5 henry at 50 ma. Stancor C -1706

31 -12 1500 Volt,

425 Milliampere

Power Supply

One of the most popular and also one of

the most convenient power ranges for ama-

teur equipment is that which can be supplied

from a 1500 volt power unit with a current

capability of about 400 ma. The r -f amplifier

of an A -M phone transmitter (1500 volts

at 250 ma.) capable of 375 watts input and

its companion modulator (1500 volts at 20-

200 ma.) can both be run from a supply of

this rating. The use of this supply for SSB

work will permit a p.e.p. of about 600 watts

(1500 volts at 400 ma.) with the new low

voltage, high current RCA 7094 tetrodes.

This voltage will be found to be very eco-

nomical when the cost of power supply com-

ponents is computed. A jump in supply volt-

age to 2000 will almost double the cost of

the various components. Unless full kilowatt

operation is intended, 1500 volts is a very

convenient and relatively economical compro-

mise voltage.

The schematic of a typical supply is shown

in figure 46. Primary power source may be

Figure 43

HIGH -VOLTAGE

POWER SUPPLY

Figure 44

TYPICAL LIGHT DUTY

POWER SUPPLY

This 300 volt, 50 milliampere power supply

may be used to run signal generators, fre-

quency meters, small receivers, etc. Switch S:

(see schematic) is placed on the rear of the

chassis near the line cord.

either 115 or 230 volts, the latter providing

slightly better power supply regulation. The

two transformer primaries are connected in

series for 230 volt operation, or in parallel

for 115 volt operation. In addition the pri-

maries may be connected in series for half -

voltage operation on 115 volts as shown. The

supply provides 750 volts at 400 ma. under

this operating condition.

For optimum dynamic voltage regulation

under varying loads such as imposed by side -

band or class B modulator equipment the

output filter capacitor of the supply should.

rl V,

T2

COMPONENTS PULL LOAD

VOLTAGE

PI/LL LOAD

CURRENT

OCAS)

Ti T2 V+-V4 CHI C1 R1

2200-VOLT

'POLE TRANSFORMER.

2 NVA

OTC 666-A 20K.

500 MA.

uTCS-37 2600 V. 200 W. 1900 500 MA.

3500-0-3500

OTC CG-306 OTC

Ls-Il -Y 72-A 10 K'

5O0 MA

OTC 0040e a."

6600 V. 200 K

300 W. 6000 500 MA.

www.americanradiohistory.com

718 Power Supplies THE RADIO

R Yti 866

1500/750V

AT 425 MA.

35K

50W

Figure 46

SCHEMATIC, 1500 VOLT, 425 MILLIAMPERE SUPPLY

T: -1710 - 0 - 1710 volts, 425 ma. 115 230 volt primary. Chicago P -1512

T/-2.5 volt, 10 ampere, 10 KV insulation. Stancor P -3060

CH, -6 henry at 300 ma., CCS. Chicago R -63

RFC -"Nosh" suppression choke. Millen 77866

be as large as is practical. Occasionally 60

µíd., 2000 volt capacitors can be picked up

on the "surplus" market for a few dollars,

although their new price would give most

amateurs pause for thought. An inexpensive

and reliable substitute may be made up of a

group of replacement -type tubular electrolytic

capacitors connected in series -parallel as shown

in the schematic. Eight 30 pfd., 450 volt

capacitors connected in series parallel will

provide an effective value of 15 pfd., at a

working voltage of 1800. This is the mini-

mum value suitable for sideband operation.

Sixteen capacitors will provide 30 pfd., at

1800 volts.

A power supply of this type should be

built upon a heavy steel chassis, and all wir-

ing must be done with 10,000 volt TV -type

plastic insulated wire. R -F chokes should be

placed in the plate leads of the mercury vapor

rectifiers as shown, to reduce the tendency

these tubes have of breaking into oscillation

over a portion of the operating cycle. Oscil-

lation of this type will produce a 120 cycle

"buzz" on the sidebands of the signal. The

parasitic is eliminated by the use of the

chokes.

31 -13 A Dual Voltage

Transmitter Supply

The majority of high voltage transformers

have tapped secondary windings, similar to

the transformer shown in the schematic of

figure 47. Separate rectifier and filter systems

may be used with the transformer to provide

two different output voltages provided the

total wattage drain from the supply does not

exceed the wattage rating of the transformer.

The drain may be divided between the two

supply systems in any manner desired. The

intermittent rating of T_ (figure 47) is 750

watts and the continuous duty rating is 600

watts. Under CCS rating, the supply can pro-

vide (for example) 2000 volts at 160 ma.

for the operation of an 813 r -f amplifier at

320 watts input, and 1750 volts at 20 -150

ma. for the operation of 811 -A class B modu-

lators. Under intermittent duty rating, the

813 amplifier can run at 400 watts input

(phone) and 500 watts (c -w) without over-

loading the supply.

A remote switch is used to energize the

plate circuit relay of the supply. An auxiliary

antenna relay is also operated by the "trans-

mit" switch.

31 -14 A Kilowatt

Power Supply

Shown in figure 48 is the schematic of a

power supply capable of delivering 2500 volts

at a continuous current drain of 500 milli-

amperes, or 700 milliamperes with an inter-

mittent load. The supply is designed to power

a kilowatt amplifier operating at 2500 volts

and 400 ma., in conjunction with a 500 watt

modulator operating at 2500 volts at a vary-

www.americanradiohistory.com

HANDBOOK Special Supplies 719

IIOV1,

i0 Fit_ ON PL. ON , r

COIL

TRANSMIT

11111

M RY1

RYI

SWITCH

ANTENNA

RELAY

CHI

2000

CH2

R.F. FIL. MOD. FIL.

2000 V.

Figure 47

DUAL VOLTAGE POWER SUPPLY

T,, T,-2.5 volts at 5 amperes. Stancor P -6133

Te- 2400 -2100 volts each side center top at 375 mo. !CAS. Stancor P -8032

CH: -3 to 17 henry, 300 ma. Stancor C -1403

CH:-8 henry, 300 ma., Stancor C -1413

R1, R: -70K., 100 watt

RFC,, RFC. -"Hasa" suppression choke. J. W. Miller 7865 twin chokes (2 req.)

RY, -SPST relay, 115 volt coil.

( MOD. )

+ 1150 V.

AT ISO MA.

(R.F.)

2000 V.

AT 100 MA

2

ing current drain of 50 -300 ma. Specifically,

the supply is employed with a transmitter

having a pair of 4 -250A tetrode tubes in the

class C stage, and a pair of 810 modulator

tubes. For sideband work, the supply may be

used to power a 1750 watt p.e.p. linear ampli-

fier, such as the 4CX -1000A amplifier shown

in an earlier chapter.

Because the total weight of the components

is over 150 pounds, the supply should be

built directly on the bottom of a relay rack

instead of upon a steel chassis.

The r -f hash suppression chokes RFC' and

RFC2 are fastened directly to the high voltage

terminals of the plate transformer. The two

872 -A rectifier tubes are so located that the

leads from the r -f chokes to the plate caps

are only about three inches long.

A 0.15 ofd., 5000 volt paper capacitor is

used to resonate the filter choke to approxi-

mately 120 cycles at a bleeder current of 25

milliamperes. When full load current is

drawn, the inductance of the filter choke

drops, detuning the parallel resonant circuit.

Improved voltage regulation is gained by this

action; the no load voltage increases only

200 volts over the full load voltage.

RYI

o

TO EXCITER

CONTROL CIRCUIT

T1- 2900 -0 -2900 volts at 700 ma., !CAS,

Chicago P -2126

T2 -5 volts, 20 amp., Chicago F -520118

CH1 -6 henries, 700 ma., Chicago R -67

C1 -0.15 µtd., 5000 -volt

C2 -Three 4 -0d. 3000 -volt

R,- 100,000 ohms, 200 -watt

R2- Eleven 0.5 megohm 2 -watt resistors in

series

RYI -DPST relay, 110 v. coil, 20 a. contacts,

Potter 8 Brumfield PR7A

RFC?, 2 -Hash filter, J. W. Miller Co. No. 7868

Figure 48

HIGH VOLTAGE POWER SUPPLY

www.americanradiohistory.com

CHAPTER THIRTY -TWO

With a few possible exceptions, such as

fixed air capacitors, neutralizing capacitors

and transmitting coils, it hardly pays one to

attempt to build the components required for

the construction of an amateur transmitter.

This is especially true when the parts are of

the type used in construction and replacement

work on broadcast receivers, as mass produc-

tion has made these parts very inexpensive.

Transmitters Those who have and wish to

spend the necessary time can

effect considerable monetary saving in their

transmitters by building them from the com-

ponent parts. The necessary data is given

in the construction chapters of this handbook.

To many builders, the construction is as

fascinating as the operation of the finished

transmitter; in fact, many amateurs get so

much satisfaction out of building a well -per-

forming piece of equipment that they spend

more time constructing and rebuilding equip-

ment than they do operating the equipment

on the air.

32 -1 Tools

Beautiful work can be done with metal

chassis and panels with the help of only a

few inexpensive tools. The time required for

construction, however, will be greatly reduced

if a fairly complete assortment of metal -work-

ing tools is available. Thus, while an array of

tools will speed up the work, excellent results

may be accomplished with few tools, if one

has the time and patience.

The investment one is justified in making

in tools is dependent upon several factors. If

you like to tinker, there are many tools use-

ful in radio construction that you would prob-

ably buy anyway, or perhaps already have,

such as screwdrivers, hammer, saws, square,

vise, files, etc. This means that the money

taken for tools from your radio budget can be

used to buy the more specialized tools, such

as socket punches or hole saws, taps and dies,

etc. The amount of construction work one does

determines whether buying a large assortment

720

www.americanradiohistory.com

Tools 721

of tools is an economical move. It also deter-

mines if one should buy the less expensive

type offered at surprisingly low prices by the

familiar mail order houses, "five and ten"

stores and chain auto -supply stores, or whether

one should spend more money and get first -

grade tools. The latter cost considerably more

and work but little better when new, but will

outlast several sets of the cheaper tools.

Therefore they are a wise investment for the

experimenter who does lots of construction

work (if he can afford the initial cash outlay).

The amateur who constructs only an occasional

piece of apparatus need not be so concerned

with tool life, as even the cheaper grade tools

will last him several years, if they are given

proper care.

The hand tools and materials in the accom-

panying lists will be found very useful around

the home workshop. Materials not listed but

ordinarily used, such as paint, can best be

purchased as required for each individual job.

ESSENTIAL HAND TOOLS AND

MATERIALS

1 Good electric soldering iron, about 100

watts,

1 Spool rosin -core wire solder

1 Each large, medium, small, and midget

screwdrivers

1 Good hand drill (eggbeater type), prefer-

ably two -speed

1 Pair regular pliers, 6 inch

1 Pair long nose pliers, 6 inch

1 Pair cutting pliers (diagonals), 5 inch or

6 inch

1 11/8-inch tube -socket punch

1 "Boy Scout" knife

1 Combination square and steel rule, 1 foot

1 Yardstick or steel pushrule

1 Scratch awl or ice pick scribe

1 Center punch

1 Dozen or more assorted round shank drills

(as many as you can afford between no.

50 and 11/4 or 3/8 inch, depending upon

size of hand drill chuck)

1 Combination oil stone

Light machine oil (in squirt can)

Friction tape

1 Hacksaw and blades

1 Medium file and handle

1 Cold chisel (1 / inch tip)

1 Wrench for socket punch

1 Hammer

HIGHLY DESIRABLE HAND TOOLS

AND MATERIALS

1 Bench vise (jaws at least 3 inch)

1 Spool plain wire solder

1 Carpenter's brace, ratchet type

1 Square -shank countersink bit

1 Square -shank taper reamer, small

1 Square -shank taper reamer, large (the two

reamers should overlap; 1/z inch and /8

inch size will usually be suitable)

1 %8 inch tube- socket punch (for electrolytic

capacitors)

1 1 -3/16 inch tube- socket punch

1 3/s-inch tube- socket punch

1 Adjustable circle cutter for holes to 3 inch

1 Set small, inexpensive, open -end wrenches

1 Pair tin shears, 10 or 12 inch

1 Wood chisel (1 /2 inch tip)

1 Pair wing dividers

1 Coarse mill file, flat 12 inch

1 Coarse bastard file, round, 1/2 or 3/4 inch

1 Set allen and spline -head wrenches

6 or 8 Assorted small files; round, half -round

or triangular, flat, square, rat -tail

4 Small "C" clamps

Steel wool, coarse and fine

Sandpaper and emery cloth, coarse, medium,

and fine

Duco cement

File brush

USEFUL BUT NOT ESSENTIAL TOOLS

AND MATERIALS

1 Jig or scroll saw (small) with metal -cut-

ting blades

1 Small wood saw (crosscut teeth)

1 Each square -shank drills: 3/8, 7/16, and 1V2

inch

1 Tap and die outfit for 6 -32, 8 -32, 10 -32

and 10 -24 machine screw threads

4 Medium size "C" clamps

Lard oil (in squirt can)

Kerosene

Empire cloth

Clear lacquer ( "industrial" grade)

Lacquer thinner

Dusting brush

Paint brushes

Sheet celluloid, Lucite, or polystyrene

1 Carpenter's plane

1 Each "Spintite" wrenches, 1/4, 5/16, 11/32

to fit the standard 6 -32 and 8 -32 nuts

used in radio work

1 Screwdriver for recessed head type screws

The foregoing assortment assumes that the

constructor does not want to invest in the

more expensive power tools, such as drill press,

www.americanradiohistory.com

THE RADIO

Figure 1

SOFT ALUMINUM

SHEET MAY BE CUT

WITH HEAVY

KITCHEN SHEARS

grinding head, etc. If power equipment is pur-

chased, obviously some of the hand tools and

accessories listed will be superfluous. A drill

press greatly facilitates construction work,

and it is unfortunate that a good one costs as

much as a small transmitter.

Not listed in the table are several special -

purpose radio tools which are somewhat of a

luxury, but are nevertheless quite handy, such

as various around -the -corner screwdrivers and

wrenches, special soldering iron tips, etc.

These can be found in the larger radio parts

stores and are usually listed in their mail or-

der catalogs.

If it is contemplated to use the newer and

very popular miniature series of tubes (6AK5,

6C4, 6BÁ6, etc.) in the construction of equip-

ment certain additional tools will be required

to mount the smaller components. Miniature

tube sockets mount in a 5/s -inch hole, while

9 -pin sockets mount in a 3/4-inch hole. Green-

lee socket punches can be obtained in these

sizes, or a smaller hole may be reamed to the

proper size. Needless to say, the punch is

much the more satisfactory solution. Mounting

screws for miniature sockets are usually of

the 4 -40 size.

Metal Chassis Though quite a few more tools

and considerably more time

will be required for metal chassis construction,

much neater and more satisfactory equipment

can be built by mounting the parts on sheet

metal chassis instead of breadboards. This type

of construction is necessary when shielding of

the appartus is required. A front panel and a

back shield minimize the danger of shock and

complete the shielding of the enclosure.

Figure 2

CONVENTIONAL

WOOD EXPANSION

BIT IS EFFECTIVE IN

DRILLING SOCKET

HOLES IN REYNOLDS

DO -IT- YOURSELF

ALUMINUM

www.americanradiohistory.com

HANDBOOK

Figure 3

SOFT ALUMINUM

TUBING MAY BE

BENT AROUND

WOODEN FORM

BLOCKS. TO PREVENT

THE TUBE FROM

COLLAPSING ON

SHARP BENDS, IT IS

PACKED WITH

WET SAND.

Material 723

32 -2 The Material

Electronic equipment may be built upon

foundation of wood, steel or aluminum. The

choice of foundation material is governed by the

requirements of the electrical circuit, the weight

of the components of the assembly, and the fin-

ancial cost of the project when balanced against

the pocketbook contents of the constructor.

Breadboard The simp!est method of con-

structing equipment is to lay it

out in breadboard fashion, which consists of

fastening the various components to a board

of suitable size with wood screws or machine

bolts, arranging the parts so that important

leads will be as short as possible.

Breadboard construction is suitable for test-

ing an experimental layout, or sometimes for

assembling an experimental unit of test equip-

ment. But no permanent item of station equip-

ment should be left in the breadboard form.

Breadboard construction is dangerous, since

components carrying dangerous voltages are

left exposed. Also, breadboard construction is

never suitable for any r -f portion of a trans-

mitter, since it would be substantially impos-

sible to shield such an item of equipment for

the elimination of TVI resulting from har-

monic radiation.

Figure 4

A WOODWORKING

PLANE MAY BE USED

TO SMOOTH OR

TRIM THE EDGES OF

REYNOLDS

DO- IT- YOURSELF

ALUMINUM STOCK.

Dish type corstruction is practically the

same as metal chassis construction, the main

difference lying in the manner in which the

chassis is fastened to the panel.

Special For high -powered r -f stages,

Frameworks many amateur constructors pre-

fer to discard the more conven-

tional types of construction and employ in-

stead special metal frameworks and brackets

which they design specially for the parts which

they intend to use. These are usually arranged

to give the shortest possible r -f leads and to

fasten directly behind a relay rack panel by

means of a few bolts, with the control shafts

www.americanradiohistory.com

724 Workshop Practice THE RADIO

DESK

COMPONENT PARTS

1 Legs and stringers -

aluminum angle f4 "afS'

2 Top -flush door

3 Shelves -f5" plywood

6'a' - ,

13 - 3'9" -

5'a' --I

a4ss'

T

Figure 5

INEXPENSIVE OPERATING DESK MADE

FROM ALUMINUM ANGLE STOCK, PLY-

WOOD AND A FLUSH -TYPE DOOR

projecting through corresponding holes in the

panel.

Working with The necessity of employing

Aluminum "electrically tight enclosures"

for the containment of TVI -

producing harmonics has led to the general

use of aluminum for chassis, panel, and en-

closure construction. If the proper type of

aluminum material is used, it may be cut and

worked with the usual woodworking tools

found in the home shop. Hard, brittle alumi-

num alloys such as 24ST and 61ST should be

avoided, and the softer materials such as 2S

or 1/2H should be employed.

A new market product is Reynold's Do -it-

yourself aluminum, which is being distributed

on a nationwide basis through hardware stores,

lumber yards and building material outlets.

This material is an alloy which is temper se-

lected for easy working with ordinary tools.

Aluminum sheet, bar and angle stock may be

obtained, as well as perforated sheets for ven-

tilated enclosures.

Figures 1 through 4 illustrate how this soft

material may be cut and worked with ordinary

shop tools, and fig. 5 shows a simple operating

desk that may be made from aluminum angle

stock, plywood and a flush -type six foot door.

Figure 6

TVI ENCLOSURE MADE FROM

SINGLE SHEET OF

PERFORATED ALUMINUM

Reynolds Metal Co. "Do- it- yourself" aluminum

sheet may be cut and folded to form nil-

proof enclosure. One -half inch lip on edges Is

bolted to center section with 6 -32 machine

screws.

32 -3 TVI -Proof

Enclosures

Armed with a right -angle square, a tin -snips

and a straight edge, the home constructor will

find the assembly of aluminum enclosures an

easy task. This section will show simple con-

struction methods, and short cuts in producing

enclosures.

The simplest type of aluminum enclosure

is that formed from a single sheet of per-

forated material as shown in figure 6. The

top, sides, and back of the enclosure are of

one piece, complete with folds that permit

the formed enclosure to be bolted together

along the edges. The top area of the enclosure

should match the area of the chassis to en-

sure a close fit. The front edge of the en-

closure is attached to aluminum angle strips

that are bolted to the front panel of the

unit; the sides and back can either be bolted

to matching angle strips affixed to the chassis,

or may simply be attached to the edge of the

chassis with self -tapping sheet metal screws.

Enclosures of this type are used on the all -

band transmitter described in chapter 31.

A more sophisticated enclosure is shown

in figure 7. In this assembly aluminum angle

stock is cut to length to form a framework

upon which the individual sides, back, and

top of the enclosure are bolted. For greatest

strength, small aluminum gusset plates should

be affixed in each corner of the enclosure.

The complete assembly may be held together

by no. 6 sheet metal screws.

www.americanradiohistory.com

HANDBOOK Openings 725

Regardless of the type of enclosure to be

made, care should be taken to ensure that all

joints are square. Do not assume that all pre-

fabricated chassis and panel are absolutely true

and square. Check them before you start to

form your shield as any dimensional errors in

the foundation will cause endless patching and

cutting after your enclosure is bolted together.

Finally, be sure that paint is removed from

the panel and chassis at the point the en-

closure attaches to the foundation. A clean,

metallic contact along the seam is required

for maximum harmonic suppression.

32 -4 Enclosure

Openings

Openings into shielded enclosures may be

made simply by covering them by a piece of

shielding held in place by sheet metal screws.

Openings through vertical panels, however,

usually require a bit more attention to pre-

vent leakage of harmonic energy through the

crack of the door which is supposed to seal

the opening. A simple way to provide a panel

opening is to employ the Bud ventilated door

rack panel model PS -814 or 815. The grille

opening in this panel has holes small enough

in area to prevent serious harmonic leakage.

The actual door opening, however, does not

seal tightly enough to be called TVI- proof.

In areas of high TV signal strength where

a minimum of operation on 28 Mc. is con-

templated, the door is satisfactory as is. To

accomplish more complete harmonic suppres-

sion, the edges of the opening should be lined

with preformed contact finger stock manufac-

tured by Eitel- McCullough, Inc., of San Bruno,

Calif. Eimac finger stock is an excellent means

of providing good contact continuity when

using components with adjustable or moving

contact surfaces, or in acting as electrical

"weatherstrip" around access doors in enclo-

sures. Harmonic leakage through such a sealed

opening is reduced to a negligible level. The

mating surface to the finger stock should be

free of paint, and should provide a good elec-

trical connection to the stock.

A second method of re- establishing elec-

trical continuity across an access port is to

employ Metex shielding around the mating

edges of the opening. Metex is a flexible knit-

ted wire mesh which may be obtained in

various sizes and shapes. This r -f gasket ma-

terial is produced by Metal Textile Corp.,

Roselle, N.J. Metex is both flexible and resili-

ent and conforms to irregularities in mating

surfaces with a minimum of closing pressure.

i

.._...: ...

--......;.r .

Figure 7

TVI-PROOF ENCLOSURE BUILT OF

PERFORATED ALUMINUM SHEET

AND ANGLE STOCK

32 -5 Summation

of the Problem

The creation of r -f energy is accompanied

by harmonic generation and leakage of funda-

mental and harmonic energy from the generator

source. For practical purposes, radio frequen-

cy power may be considered as a form of both

electrical and r -f energy. As electrical energy,

it will travel along any convenient conductor.

As r -f energy, it will radiate directly from the

source or from any conductor connected to the

source. In view of this "dual personality" of

r -f emanations, there is no panacea for all

forms of r -f energy leakage. The cure involves

both filtering and shielding: one to block the

paths of conducted energy, the other to pre-

vent the leakage of radiated energy. The proper

combination of filtering and shielding can re-

duce the radiation of harmonic energy from a

signal source some 80 decibels. In most cases,

this is sufficient to eliminate interference caused

by the generation of undesirable harmonics.

www.americanradiohistory.com

726 Workshop Practice THE RADIO

32 -6 Construction

Practice

The chassis first should be covered

with a layer of wrapping paper,

which is drawn tightly down on

all sides and fastened with scotch tape. This

allows any number of measurement lines and

hole centers to be spotted in the correct po-

sitions without making any marks on the

chassis itself. Place on it the parts to be

mounted and play a game of chess with them,

trying different arrangements until all the grid

and plate leads are made as short as possible,

tubes are clear of coil fields, r -f chokes are in

safe positions, etc. Remember, especially if

you are going to use a panel, that a good me-

chanical layout often can accompany sound

electrical design, but that the electrical de-

sign should be given first consideration.

All too often parts are grouped to give a

symmetrical panel, irrespective of the arrange-

ment behind. When a satisfactory arrangement

has been reached, the mounting holes may be

marked. The same procedure now must be

followed for the underside, always being care-

ful to see that there are no clashes between

the two (that no top mounting screws come

down into the middle of a paper capacitor on

the underside, that the variable capacitor rotors

do not hit anything when turned, etc.).

When all the holes have been spotted, they

should be center -punched through the paper

into the chassis. Don't forget to spot holes for

leads which must also come through the

chassis.

For transformers which have lugs on the

bottoms, the clearance holes may be spotted

by pressing the transformer on a piece of pa-

per to obtain impressions, which may then

be transferred to the chassis.

Punching In cutting socket holes, one can

use either a fly- cutter or socket

punches. These punches are easy to operate

and only a few precautions are necessary. The

guide pin should fit snugly in the guide hole.

This increases the accuracy of location of the

socket. If this is not of great importance, one

may well use a drill of 1/32 inch larger di-

ameter than the guide pin. Some of the punches

will operate without guide holes, but the latter

always make the punching operations simpler

and easier. The only other precaution is to be

sure the work is properly lined up before ap-

plying the hammer. If this is not done, the

punch may slide sideways when you strike and

thus not only shear the chassis but also take

Chassis

Layout

DRILL HOLES SLIGHTLY

INSIDE DASHED OUTLINE

OF DESIRED HOLE.

BREAK OUT FILE

PIECE INSIDE SMOOTH

DRILL HOLES.

MAKING RECTANGULAR CUTOUT

Figure 8

off part of the die. This is easily avoided by

always making sure that the piece is parallel

to the faces of the punch, the die, and the

base. The latter should be an anvil or other

solid base of heavy material.

A punch by Greenlee forces socket holes

through the chassis by means of a screw turned

with a wrench. It is noiseless, and works much

more easily and accurately than most others.

The male part of the punch should be placed

in the vise, cutting edge up and the female

portion forced against the metal with a wrench.

These punches can be obtained in sizes to

accommodate all tube sockets and even large

enough to be used for meter holes. In the

octal socket sizes they require the use of a 3/8

inch center hole to accommodate the bolt.

Transformer Cutouts for transformers and

Cutouts chokes are not so simply han-

dled. After marking off the part

to be cut, drill about a 1/4-inch hole on each

of the inside corners and tangential to the

edges. After burring the holes, clamp the piece

and a block of cast iron or steel in the vise.

Then, take your burring chisel and insert it in

one of the corner holes. Cut out the metal by

hitting the chisel with a hammer. The blows

should be light and numerous. The chisel acts

against the block in the same way that the two

blades of a pair of scissors work against each

other. This same process is repeated for the

other sides. A file is used to trim up the com-

pleted cutout.

Another method is to drill the four corner

holes large enough to take a hack saw blade,

then saw instead of chisel. The four holes per-

mit nice looking corners.

Still another method is shown in figure 8.

When heavy panel steel is used and a drill

press or electric drill is available, this is the

most satisfactory method.

Removing

Burrs In both drilling and punching, a

burr is usually left on the work.

There are three simple ways of

www.americanradiohistory.com

HANDBOOK Construction Practice 727

removing these. Perhaps the best is to take a

chisel (be sure it is one for use on metal) and

set it so that its bottom face is parallel to the

piece. Then gently tap it with a hammer. This

usually will make a clean job with a little

practice. If one has access to a counterbore,

this will also do a nice job. A countersink will

work, although it bevels the edges. A drill of

several sizes larger is a much used arrange-

ment. The third method is by filing off the

burr, which does a good job but scratches the

adjacent metal surfaces badly.

Mounting

Components There are two methods in gen-

eral use for the fastening of

transformers, chokes, and similar

NUMBERED DRILL SIZES

Correct for

DI- Tapping

DRILL cruets, Clears Steel or

NUMBER (in.) Screw Brass?

1 .22e - -

2 .221 12 -24 -

3 .. .213 - 14 -24

4 .209 12 -20 -

5 .205 - -

6 .204 -

7 .. .201

8 .199 -

9 .196 -

10* .193 10 -32

11 .. _ .191 10 -24

12_. .189 -

13 .. .185 - -

14 .182 -

15 _. .180 -

16 .177 - 12 -24

17 .173 - -

18* .169 8 -32 -

19 _. .166 - 12 -20

20 .161 - -

21 .. .159 10 -32

22 .157 -

23 .154 -

24 .152

25 .. .149 10 -24

26 .147 - -

27 .144 - -

28* .140 6 -32 -

29* .136 - 8 -32

30 .128 -

31 .120 - -

32 .. ... .116 - -

33* .113 4 -36 4 -40 -

34 .. .111 - -

35* .110 - 6 -32

36 .106 - -

37 _. .104 - -

38 _.._ .102 -

39__.. _.._. .100 3 -48 -

40 _. .098 - -

41 .. .096 - -

42* __... _ .093 - 4 -36 4 -40

43 .089 2 -56 -

44 .086 - -

45 .082 - 3 -48

t Use next size larger drill for tapping

bak elite and similar composition materials

(plastics, etc.).

Sizes most commonly used in radio con-

struction.

Figure 9

pieces of apparatus to chassis or breadboards.

The first, using nuts and machine screws, is

slow, and the commercial manufacturing prac-

tice of using self -tapping screws is gaining

favor. For the mounting of small parts such

as resistors and capacitors, "tie points" are

very useful to gain rigidity. They also con-

tribute materially to the appearance of finished

apparatus.

Rubber grommets of the proper size, placed

in all chassis holes through which wires are

to be passed, will give a neater appearing job

and also will reduce the possibility of short

circuits.

Soldering Making a strong, low -resistance

solder joint does not mean just

dropping a blob of solder on the two parts to

be joined and then hoping that they'll stick.

There are several definite rules that must be

observed.

All parts to be soldered must be absolutely

clean. To clean a wire, lug, or whatever it may

be, take your pocket knife and scrape it thor-

oughly, until fresh metal is laid bare. It is not

enough to make a few streaks; scrape until the

part to be soldered is bright.

Make a good mechanical joint before apply-

ing any solder. Solder is intended primarily

to make a good electrical connection; mechani-

cal rigidity should be obtained by bending the

wire into a small hook at the end and nipping

it firmly around the other part, so that it will

hold well even before the solder is applied.

Keep your iron properly tinned. It is im-

possible to get the work hot enough to take

the solder properly if the iron is dirty. To tin

your iron, file it, while hot, on one side until

a full surface of clean metal is exposed. Im-

mediately apply rosin core solder until a thin

layer flows completely over the exposed sur-

face. Repeat for the other faces. Then take a

clean rag and wipe off all excess solder and

rosin. The iron should also be wiped frequently

while the actual construction is going on; it

helps prevent pitting the tip.

Apply the solder to the work, not to the

iron. The iron should be held against the parts

to be joined until they are thoroughly heated.

The solder should then be applied against the

parts, and the iron should be held in place

until the solder flows smoothly and envelopes

the work. If it acts like water on a greasy

plate, and forms a ball, the work is not suf-

ficiently clean.

The completed joint must be held perfectly

still until the solder has had time to solidify.

If the work is moved before the solder has be-

www.americanradiohistory.com

728 Workshop Practice

START

POOL OF WIRE

FINISH

FORM OF BAKELITE OR

OTHER GOOD INSULATING

MATERIAL.

FINISHED

COIL

WINDING COIL ON INSULATING FORM

Figure 10

- ,;,Iø

HOLD TIGHTLY

I

PIPE OR ROO USED AS

TEMPORARY FORM

WIND TURNS CLOSE TOGETHER

AND SPACE LATER.

WINDING `AIR- SUPPORTED'' COIL

Figure 11

come completely solid, a "cold" joint will re-

sult. This can be identified immediately, be-

cause the solder will have a dull "white" ap-

pearance rather than one of shiny "silver."

Such joints tend to be of high resistance and

will very likely have a bad effect upon a cir-

cuit. The cure is simple, merely reheat the

joint and do the job correctly.

Wipe away all surplus flux when the joint

has cooled if you are using a paste type flux.

Be sure it is non -corrosive, and use it with

plain (not rosin core) solder.

Finishes If the apparatus is constructed on

a painted chassis (commonly avail-

able in black wrinkle and gray wrinkle), there

is no need for application of a protective coat-

ing when the equipment is finished, assuming

that you are careful not to scratch or mar the

finish while drilling holes and mounting parts.

However, many amateurs prefer to use un-

painted (zinc or cadmium plated) chassis, be-

cause it is much simpler to make a chassis

ground connection with this type of chassis.

A thin coat of clear "linoleum" lacquer may

be applied to the whole chassis after the wir-

ing is completed to retard rusting. In localities

near the sea coast it is a good idea to lacquer

the various chassis cutouts even on a painted

chassis, as rust will get a good start at these

points unless the metal is protected where the

drill or saw has exposed it. If too thick a coat

is applied, the lacquer will tend to peel. It

may be thinned with lacquer thinner to permit

application of a light coat. A thin coat will

adhere to any clean metal surface that is not

too shiny.

An attractive dull gloss finish, almost vel-

vety can be put on aluminum by sand -blasting

it with a very weak blast and fine particles

and then lacquering it. Soaking the aluminum

in a solution of lye produces somewhat the

same effect as a fine grain sand blast.

There are also several brands of dull gloss

black enamels on the market which adhere

well to metals and make a nice appearance.

Airdrying wrinkle finishes are sometimes suc-

cessful, but a bake job is usually far better.

Wrinkle finishes, properly applied, are very

durable and are pleasing to the eye. If you

live in a large community, there is probably

an enameling concern which can wrinkle your

work for you at a reasonable cost. A very at-

tractive finish, for panels especially, is to

spray a wrinkle finish with aluminum paint.

In any painting operation (or plating, either,

for that matter) , the work should be very

thoroughly cleaned of all greases and oils.

To protect brass from tarnish, thoroughly

cleanse and remove the last trace of grease by

the use of potash and water. The brass must

be carefully rinsed with water and dried; but

in doing it, care must be taken not to handle

any portion with the bare hands or anything

else that is greasy. Then lacquer.

Winding Coils Coils are of two general types,

those using a form and "air -

wound" types. Neither type offers any particu-

lar constructional difficulties. Figure 10 il-

lustrates the procedure used in form winding

a coil. If the winding is to be spaced, the

spacing can be done either by eye or a string

or another piece of wire may be wound simul-

taneously with the coil wire and removed after

the winding is in place. The usual procedure

is to clamp one end of the wire in a vise, at-

taching the other end to the coil form and with

the coil form in hand, walk slowly towards the

vise winding the wire but at the same time

keeping a strong tension on the wire as the

form is rotated. After the coil is wound, if

there is any possibility of the turns slipping,

the completed coil is either entirely coated

with a coil or Duco cement or cemented in

those spots where slippage might occur.

www.americanradiohistory.com

Figure 12

GOOD SHOP

LAYOUT AIDS

CAREFUL

WORKMANSHIP

Built in a corner of a

garage, this shop has all

features necessary for

electronic work. Test in-

struments are arranged

on shelves above bench.

Numerous outlets reduce

"haywire" produced by

tangled line cords. Not

shown in picture are drill

press and sander at end

of left bench

V -h -f and u -h -f coils are commonly wound

of heavy enameled wire on a form and then

removed from the form as in figure 11. If

the coil is long or has a tendency to buckle,

strips of polystyrene or a similar material may

be cemented longitudinally inside the coil. Due

allowance must be made for the coil springing

out when removed from the form, when select-

ing the diameter of the form.

On air wound coils of this type, spacing be-

tween turns is accomplished after removal

from the form, by running a pencil, the shank

of a screwdriver or other round object spirally

between the turns from one end of the coil to

the other, again making due allowance for

spring.

Air -wound coils, approaching the appearance

of commercially manufactured ones, can be

constructed by using a round wooden form

which has been sawed diagonally from end

to end. Strips of insulating material are tem-

porarily attached to this mandrel, the wire

then being wound over these strips with the

desired separation between turns and cemented

to the strips. When dry, the split mandrel may

be removed by unwedging it.

32 -7 Shop Layout

The size of your workshop is relatively un-

important since the shop layout will deter-

mine its efficiency and the ease with which

you may complete your work.

Shown in figure 12 is a workshop built

into a 10'x10' area in the corner of a garage.

The workbench is 32" wide, made up of four

strips of 2 "x8" lumber supported on a solid

framework made of 2 "x4" lumber. The top

of the workbench is covered with hard -sur-

face Masonite. The edge of the surface is pro-

tected with aluminum "counter edging" strip,

obtainable at large hardware stores. Two wood-

en shelves 12" wide are placed above the

bench to hold the various items of test equip-

ment. The shelves are bolted to the wall studs

with large angle brackets and have wooden

end pieces. Along the edge of the lower shelf

a metal "outlet strip" is placed that has an

115 -volt outlet every six inches along its

length. A similar strip is run along the back

of the lower shelf. The front strip is used

for equipment that is being bench -tested, and

the rear strip powers the various items of test

equipment placed on the shelves.

At the left of the bench is a storage bin

for small components. A file cabinet can be

seen at the right of the photograph. This nec-

essary item holds schematics, transformer data

sheets, and other papers that normally are

lost in the usual clutter and confusion.

The area below the workbench has two

storage shelves which are concealed by sliding

doors made of V4 -inch Masonite. Heavier tools,

and large components are stored in this area.

On the floor and not shown in the photo-

graph is a very necessary item of shop equip-

ment: a large trash receptacle.

A compact and efficient shop built in one-

half of a wardrobe closet is shown in figure

13. The workbench length is four feet. The

top is made of 4 "x6" lumber sheathed with

hard surface Masonite and trimmed with

"counter edging" strip. The supporting struc-

729

www.americanradiohistory.com

730 Workshop Practice

Figure 13

COMPLETE WORKSHOP IN A CLOSET!

Careful layout permits complete electronic

workshop to be placed in one -half of a ward-

robe closet. Work bench is built atop an un-

painted three -shelf bookcase.

cure is made from an unpainted three -shelf

bookcase. A 2 "x2" leg is placed under the

front corners of the bench to provide maxi-

mum stability.

Atop the bench, a small wooden framework

supports needed items of test equipment and

a single shelf contains a 115 -volt "outlet strip."

The instruments at the top of the photo are

placed on the wardrobe shelf.

When not in use, the doors of the ward-

robe are closed, concealing the workshop com-

pletely from view.

www.americanradiohistory.com

CHAPTER THIRTY -THREE

Electronic

Test Equipment

All amateur stations are required by law to

have certain items of test equipment available

within the station. A c -w station is required

to have a frequency meter or other means in

addition to the transmitter frequency control

for insuring that the transmitted signal is on

a frequency within one of the frequency bands

assigned for such use. A radiophone station is

required in addition to have a means of deter-

mining that the transmitter is not being modu-

lated in excess of its modulation capability,

and in any event not more than 100 per cent.

Further, any station operating with a power

input greater than 900 watts is required to

have a means of determining the exact input

to the final stage of the transmitter, so as to

insure that the power input to the plate circuit

of the output stage does not exceed 1000

watts. The additional test and measurement equip-

ment required by a station will be determined

by the type of operation contemplated. It is

desirable that all stations have an accurately

calibrated volt -ohmmeter for routine transmit-

ter and receiver checking and as an assistance

in getting new pieces of equipment into opera-

tion. An oscilloscope and an audio oscillator

make a very desirable adjunct to a phone sta-

tion using AM or FM transmission, and are a

necessity if single -sideband operation is con-

templated. A calibrated signal generator is al-

most a necessity if much receiver work is con-

templated, although a frequency meter of LM

or BC -221 type, particularly if it includes in-

ternal modulation, will serve in place of the

signal generator. Extensive antenna work in-

731

variably requires the use of some type of field -

strength meter, and a standing -wave meter of

some type is very helpful. Lastly, if much v -h -f

work is to be done, a simple grid -dip meter

will be found to be one of the most used items

of test equipment in the station.

33 -1 Voltage,

Current and Power

The measurement of voltage and current in

radio circuits is very important in proper main-

tenance of equipment. Vacuum tubes of the

types used in communications work must be

operated within rather narrow limits in regard

to filament or heater voltage, and they must

be operated within certain maximum limits in

regard to the voltage and current on other

electrodes.

Both direct current and voltage are most

commonly measured with the aid of an instru-

ment consisting of a coil that is free to rotate

in a constant magnetic field (d'Arsonval type

instrument). If the instrument is to be used for

the measurement of current it is called an am-

meter or milliammeter. The current flowing

through the circuit is caused to flow through

the moving coil of this type of instrument. If

the current to be measured is greater than 10

milliamperes or so it is the usual practice to

cause the majority of the current to flow

through a by -pass resistor called a shunt, only

a specified portion of the current flowing

through the moving coil of the instrument.

The calculation of shunts for extending the

range of d -c milliammeters and ammeters is

discussed in Chapter Two.

www.americanradiohistory.com

732 Test Equipment THE RADIO

+10 +100

1001

+200

2500

+1000

1MEG

Figure 1

MULTI -VOLTMETER CIRCUITS

(A) shows a circuit whereby individual multi-

plier resistors are used for each range. (B) is

the more economical "series multiplier" cir-

cuit. The same number of resistors is re-

quired, but those for the higher ranges have

less resistance, and hence are less expen-

sive when precision wirewound resistors are

to be used. (C) shows a circuit essentially

the same as at (A), except that a range

switch is used. With a 0 -500 d -c microam-

meter substituted for the 0 -1 milliammeter

shown above, all resistor values would be

multiplied by two and the voltmeter would

have a "2000- ohm - per -volt" sensitivity. Simi-

larly, if a 0 -50 d -c microammeter were to be

used, all resistance values would be multi-

plied by twenty, and the voltmeter would

have a sensitivity of 20,000 ohms per volt.

A direct current voltmeter is merely a d -c

milliammeter with a multiplier resistor in se-

ries with it. If it is desired to use a low -range

milliammeter as a voltmeter the value of the

multiplier resistor for any voltage range may

be determined from the following formula:

1000 E

R

where: R = multiplier resistor in ohms

E = desired full scale voltage

I = full scale current of meter in ma.

The sensitivity of a voltmeter is commonly

expressed in ohms per volt. The higher the

ohms per volt of a voltmeter the greater its

sensitivity. When the full -scale current drain

of a voltmeter is known, its sensitivity rating

in ohms per volt may be determined by:

1000

Ohms per volt = I

25015 20015 AOK 1015

PIN JACKS

Figure 2

VOLT- OHMMETER CIRCUIT

With the switch In position 1, the 0 -1 milliam-

meter would be connected directly to the

terminals. In position 2 the meter would read

from 0- 100,000 ohms, approximately, with a

resistance value of 4500 ohms at half scale.

(Note: The half -scale resistance value of an

ohmmeter using this circuit is equal to the

resistance in series with the battery inside

the instrument.) The other four taps are volt-

age ranges with 10, 50, 250, and 500 volts

full scale.

Where I is the full -scale current drain of the

indicating instrument in milliamperes.

Multi -Range It is common practice to connect

Meters a group of multiplier resistors

in the circuit with a single in-

dicating instrument to obtain a multi -range

voltmeter. There are several ways of wiring

such a meter, the most common ones of which

are indicated in figure 1. With all these meth-

ods of connection, the sensitivity of the meter

in ohms per volt is the same on all scales.

With a 0 -1 milliammeter as shown the sensi-

tivity is 1000 ohms per volt.

Volt- Ohmmeters An extremely useful piece of

test equipment which should

be found in every laboratory or radio station

is the volt- ohmmeter. It consists of a multi -

range voltmeter with an additional fixed resis-

tor, a variable resistor, and a battery. A typical

example of such an instrument is diagrammed

in figure 2. Tap 1 is used to permit use of

the instrument as an 0 -1 d -c milliammeter.

Tap 2 permits accurate reading of resistors up

to 100,000 ohms; taps 3, 4, 5, and 6 are

for making voltage measurements, the full

scale voltages being 10, 50, 250, and 500

volts respectively.

The 1000 -ohm potentiometer is used to

bring the needle to zero ohms when the ter-

minals are shorted; this adjustment should

always be made before a resistance measure-

ment is taken. Higher voltages than 500 can

be read if a higher value of multiplier resistor

is added to an additional tap on the switch.

The proper value for a given full scale read-

ing can be determined from Ohm's law.

Resistances higher than 100,000 ohms can-

www.americanradiohistory.com

HANDBOOK Ohmmeters 733

not be measured accurately with the circuit

constants shown; however, by increasing the

ohmmeter battery to 45 volts and multiplying

the 4000 -ohm resistor and 1000 -ohm poten-

tiometer by 10, the ohms scale also will be

multiplied by 10. This would permit accurate

measurements up to 1 megohm.

0 -1 d -c milliammeters are available with

special volt -ohmmeter scales which make in-

dividual calibration unnecessary. Or, special

scales can be purchased separately and sub-

stituted for the original scale on the milliam-

meter.

Obviously, the accuracy of the instrument

either as a voltmeter or as an ammeter can be

no better than the accuracy of the milliam-

meter and the resistors.

Because volt -ohmmeters are so widely used

and because the circuit is standardized to a

considerable extent, it is possible to purchase

a factory -built volt -ohmmeter for no more than

the component parts would cost if purchased

individually. For this reason no construction

details are given. However, anyone already

possessing a suitable milliammeter and de-

sirous of incorporating it in a simple volt -

ohmmeter should be able to build one from the

schematic diagram and design data given here.

Special, precision (accurately calibrated) mul-

tiplier resistors are available if a high degree

of accuracy is desired. Alternatively, good

quality carbon resistors whose actual resist-

ance has been checked may be used as multi-

pliers where less accuracy is required.

Medium- and Most ohmmeters, including the

Low -Range one just described, are not

Ohmmeter adapted for accurate measure-

ment of low- resistances -in the

neighborhood of 100 ohms, for instance.

The ohmmeter diagrammed in figure 3 was

especially designed for the reasonably accu-

rate reading of resistances down to 1 ohm. Two

scales are provided, one going in one direc-

tion and the other scale going in the other

direction because of the different manner in

which the milliammeter is used in each case.

The low scale covers from 1 to 100 ohms and

the high scale from 100 to 10,000 ohms. The

high scale is in reality a medium -range scale.

For accurate reading of resistances over 10,000

ohms, an ohmmeter of the type previously

described should be used.

The 1 -100 ohm scale is useful for checking

transformers, chokes, r -f coils, etc., which often

have a resistance of only a few ohms.

The calibration scale will depend upon the

internal resistance of the particular make of

D P.D.T

SWITCH

Figure 3

SCHEMATIC OF A LOW -RANGE

OHMMETER

A description of the operation of this circuit

is given in the text. With the switch in the

left position the half -scale reading of the

meter will occur with an external resistance

of 1000 ohms. With the switch in the right

position, half -scale deflection will be ob-

tained with an external resistance equal to

the d -c resistance of the milliammeter (20 to

SO ohms depending upon the make of in-

strument).

1.5 -ma. meter used. The instrument can be

calibrated by means of a Wheatstone bridge or

a few resistors of known accuracy. The latter

can be series -connected and parallel -connected

to give sufficient calibration points. A hand -

drawn hand -calibrated scale can be cemented

over the regular meter scale to give a direct

reading in ohms.

Before calibrating the instrument or using

it, the test prods should always be touched to-

üether and the zero adjuster set accurately.

Measurement of The measurement of al-

Alternating Current ternating current and

and Voltage voltage is complicated

by two factors; first, the

frequency range covered in ordinary communi-

cation channels is so great that calibration of

an instrument becomes extremely difficult; sec-

ond, there is no single type of instrument

which is suitable for all a -c measurements-as

the d'Arsonval type of movement is suitable

for d -c. The d'Arsonval movement will not

operate on a -c since it indicates the average

value of current flow, and the average value

of an a -c wave is zero.

As a result of the inability of the reliable

d'Arsonval type of movement to record an al-

ternating current, either this current must be

rectified and then fed to the movement, or a

special type of movement which will operate

from the effective value of the current can be

used. For the usual measurements of power fre-

quency a.c. (25 -60 cycles) the iron -vane in-

strument is commonly used. For audio fre-

quency a.c. (50- 20,000 cycles) a d'Arsonval

www.americanradiohistory.com

734 Test Equipment THE RADIO

1 G4-G

Figure 4

SLIDE -BACK V -T VOLTMETER

By connecting a variable source of voltage

in series with the input to a conventional

v -t voltmeter, or in series with the simple

triode voltmeter shown above, a slide -back

a -c voltmeter for peak voltage measurement

con be constructed. Resistor R should be

about 1000 ohms per volt used at battery B.

This type of v -t voltmeter has the advantage

that it can give a reading of the actual peak

voltage of the wave being measured, without

any current drain from the source of voltage.

instrument having an integral copper oxide

or selenium rectifier is usually used. Radio fre-

quency voltage measurements are usually made

with some type of vacuum -tube voltmeter,

while r -f current measurements are almost in-

variably made with an instrument containing

a thermo- couple to convert the r.f. into d.c.

for the movement.

Since an alternating current wave can have

an almost infinite variety of shapes, it can

easily be seen that the ratios between the

three fundamental quantities of the wave

(peak, r.m.s. effective, and average after rec-

tification) can also vary widely. So it becomes

necessary to know beforehand just which qual-

ity of the wave under measurement our in-

strument is going to indicate. For the purpose

of simplicity we can list the usual types of

a -c meters in a table along with the charac-

teristic of an a -c wave which they will indi-

cate: Iron -vane, thermocouple- r.m.s.

Rectifier type (copper oxide or selenium)

-average after rectification.

V.t.v.m.- r.m.s., average or peak, depend-

ing upon design and calibration.

Vacuum -Tube A vacuum -tube voltmeter is es-

Voltmeters sentially a detector in which

a change in the signal placed

upon the input will produce a change in the

indicating instrument (usually a d'Arsonval

meter) placed in the output circuit. A vacuum -

tube voltmeter may use a diode, a triode, or a

multi -element tube, and it may be used either

for the measurement of a.c. or d.c.

When a v.t.v.m. is used in d -c measurement

it is used for this purpose primarily because

of the very great input resistance of the device.

This means that a v.t.v.m. may be used for

the measurement of a -v -c, a -f -c, and discrimina-

tor output voltages where no loading of the

circuit can be tolerated.

A -C V -T There are many different types

Voltmeters of a -c vacuum -tube voltmeters,

all of which operate as some

type of rectifier to give an indication on a d -c

instrument. There are two general types: those

which give an indication of the r -m -s value of

the wave (or approximately this value of a

complex wave) , and those which give an in-

dication of the peak or crest value of the

wave. Since the setting up and calibration of a

wide -range vacuum -tube voltmeter is rather

tedious, in most cases it will be best to pur-

chase a commercially manufactured unit. Sev-

eral excellent commercial units are on the

market at the present time; also kits for home

construction of a quite satisfactory v.t.v.m.

are available from several manufacturers.

These feature a wide range of a -c and d -c volt-

age scales at high sensitivity, and in addition

several feature a built -in vacuum -tube ohm-

meter which will give indications up to 500

or 1000 megohms.

Peak A -C V -T

Voltmeters There are two common types

of peak- indicating vacuum -

tube voltmeters. The first is

the so- called slide -back type in which a sim-

ple v.t.v.m. is used along with a conventional

d -c voltmeter and a source of bucking bias in

series with the input. With this type of ar-

rangement (figure 4) leads are connected to

the voltage to be measured and the slider re-

sistor R across the bucking voltage is backed

down until an indication on the meter (called

a false zero) equal to that value given with

the prods shorted and the bucking voltage

reduced to zero, is obtained. Then the value

of the bucking voltage (read on V) is equal

to the peak value of the voltage under meas-

urement. The slide.back voltmeter has the

disadvantage that it is not instantaneous in its

indication -adjustments must be made for

every voltage measurement. For this reason the

slide -back v.t.v.m. is not commonly used, be-

ing supplanted by the diode -rectifier type of

peak v.t.v.m. for most applications.

High -Voltage

Diode Peak

Voltmeter

the constants

A diode vacuum -tube voltmeter

suitable for the measurement

of high values of a -c voltage is

diagrammed in figure 5. With

shown, the voltmeter has two

www.americanradiohistory.com

HANDBOOK Power Measurement 735

A C

E

FROM

SOURCE

WITH

RETURN

2X2/879

PATH S T

+ I 15 V.

Figure 5

SCHEMATIC OF A HIGH -VOLTAGE PEAK

VOLTMETER

A peak voltmeter such as diagrammed

above is convenient for the measure-

ment of peak voltages at fairly high

power levels from a source of moder-

ately low impedance.

C,- .001 -,,fd. high -voltage mica

C.-- 1.01_fd. high -voltage paper

L-500,000 ohms (two 0.25 -megohm Vs-watt

in series)

R2-1.0 megohm (four 0.25 -megohm 1/2-watt

in series)

T -2.5 v., 1.75 a. filament transformer

M -0-1 d -c milliammeter

S,,,_, ,- S -p -d -t toggle switch

S- S -p -s -t toggle switch

(Note: C, is a by -pass around C:, the induc-

tive reactance of which may be appreciable

at high frequencies.)

ranges: 500 and 1500 volts peak full scale.

Capacitors C, and C_ should be able to with-

stand a voltage in excess of the highest peak

voltage to be measured. Likewise, R, and R.

should be able to withstand the same amount

of voltage. The easiest and least expensive

way of obtaining such resistors is to use sev-

eral low- voltage resistors in series, as shown

in figure 5. Other voltage ranges can be ob-

tained by changing the value of these re-

sistors, but for voltages less than several hun-

dred volts a more linear calibration can be ob-

tained by using a receiving -type diode. A cali-

bration curve should be run to eliminate the

appreciable error due to the high internal re-

sistance of the diode, preventing the capacitor

from charging to the full peak value of the

voltage being measured.

A direct reading diode peak voltmeter of

the type shown in figure 5 will load the source

of voltage by approximately one -half the value

of the load resistance in the circuit (R,, or RI

plus 112, in this case). Also, the peak voltage

reading on the meter will be slightly less than

the actual peak voltage being measured. The

amount of lowering of the reading is deter-

mined by the ratio of the reactance of the

storage capacitance to the load resistance. If

a cathode -ray oscilloscope is placed across the

terminals of the v.t.v.m. when a voltage is be-

ing measured, the actual amount of the lower-

ing in voltage may be determined by inspection

A.G.

VOLTAGE

FROM

SOURCE

WITH

D.C.

RETURN

PATH

2x2/879

I 1 V.A.C.

CONVENTIONAL

NIGH- SENSITIVITY

VOLTMETER

Figure 6

PEAK -VOLTAGE MEASUREMENT

CIRCUIT

Through use of the arrangement shown above

it is possible to make accurate measurements

of peak c -c voltages, such as across the sec-

ondary of a modulation transformer, with a

conventional d -c multi -voltmeter. Capacitor

C and transformer T should, of course, be in-

sulated for the highest peak voltage likely to

bo encountered. A capacitance of 0.25 -. fd.

at C has been found to be adequate. The

higher the sensitivity of the indicating d -c

voltmeter, the smaller will be the error be-

tween the indication on the meter and the ac-

tual peak voltage being measured.

of the trace on the c -r tube screen. The peak

positive excursion of the wave will be slightly

flattened by the action of the v.t.v.m. Usually

this flattening will be so small as to be negli-

gible.

An alternative arrangement, shown in figure

6, is quite convenient for the measurement of

high a -c voltages such as are encountered in

the adjustment and testing of high -power

audio amplifiers and modulators. The arrange-

ment consists simply of a 2X2 rectifier tube

and a filter capacitor of perhaps 0.25 -µfd. ca-

pacitance, but with a voltage rating high

enough that it is not likely to be punctured

as a result of any tests made. Cathode -ray

oscilloscope capacitors, and those for electro-

static- deflection TV tubes often have ratings

as high as 0.25 tad. at 7500 to 10,000 volts.

The indicating instrument is a conventional

multi -scale d -c voltmeter of the high- sensitivity

type, preferably with a sensitivity of 20,000 or

50,000 ohms per volt. The higher the sen-

sitivity of the d -c voltmeter used with the

rectifier, the smaller will be the amount of

flattening of the a -c wave as a result of the

rectifier action.

Measurement

of Power Audio frequency or radio fre-

quency power in a resistive

circuit is most commonly and

most easily determined by the indirect method,

i.e., through the use of one of the following

formulas: P =EI P =E1 /R P =hR

These three formulas mean that if any two of

the three factors determining power are known

www.americanradiohistory.com

736 Test Equipment THE RADIO

Figure 7.

2- KILOWATT DUMMY LOAD FOR

3 -30 MC.

Load is built in case measuring 22" deep, 11"

wide and S" high. Meter is calibrated in watts

against microampere scale as follows: (1),

22.3 pa. (5) SO pa. (10), 70.5 pa. (15),

86.5 pa. (20), 100 pa. Scale may be marked

off as shown in photograph. Calibration tech-

nique is discussed in text. Alternatively, a

standing wave bridge (calibrated in watts)

such as "Micromatch" may be used to deter-

mine power input to bridge.

Vents in top of case, and 1/4-inch holes in

chassis permit circulation of air about re-

sistors. Unit should be fan cooled for con-

tinuous dissipation.

(resistance, current, voltage) the power being

dissipated may be determined. In an ordinary

120 -volt a -c line circuit the above formulas

are not strictly true since the power factor of

the load must be multiplied into the result -

or a direct method of determining power such

as a wattmeter may be used. But in a resistive

a -f circuit and in a resonant r -f circuit the

power factor of the load is taken as being

unity. For accurate measurement of a -f and r -f

power, a thermogalvanometer or thermocouple

ammeter in series with a non -inductive resis-

tor of known resistance can be used. The me-

ter should have good accuracy, and the exact

value of resistance should be known with ac-

curacy. Suitable dummy load resistors are

available in various resistances in both 100

and 250 -watt ratings. These are virtually non-

inductive, and may be considered as a pure

resistance up to 30 Mc. The resistance of

these units is substantially constant for all

values of current up to the maximum dissipa-

tion rating, but where extreme accuracy is re-

quired, a correction chart of the dissipation

coefficient of resistance (supplied by the manu-

facturer) may be employed. This chart shows

the exact resistance for different values of

current through the resistor.

Sine -wave power measurements (r -f or sin-

gle- frequency audio) may also be made

through the use of a v.t.v.m. and a resistor of

known value. In fact a v.t.v.m. of the type

shown in figure 6 is particularly suited to this

work. The formula, P = E' /R is used in this

case. However, it must be remembered that

a v.t.v.m. of the type shown in figure 6 indi-

cates the peak value of the a -c wave. This

reading must be converted to the r -m -s or

heating value of the wave by multiplying it

by 0.707 before substituting the voltage value

in the formula. The same result can be ob-

tained by using the formula P = E' /2R.

Thus all three methods of determining pow-

er, ammeter - resistor, voltmeter- resistor, and

voltmeter- ammeter, give an excellent cross-

check upon the accuracy of the determination

and upon the accuracy of the standards.

Power may also be measured through the

use of a calorimeter, by actually measuring the

amount of heat being dissipated. Through the

use of a water- cooled dummy load resistor this

method of power output determination is being

used by some of the most modern broadcast

stations. But the method is too cumbersome

for ordinary power determinations.

Power may also be determined photometri-

cally through the use of a voltmeter, ammeter,

incandescent lamp used as a load resistor, and

a photographic exposure meter. With this

method the exposure meter is used to deter-

mine the relative visual output of the lamp

running as a dummy load resistor and of the

lamp running from the 120 -volt a -c line. A

rheostat in series with the lead from the a -c

line to the lamp is used to vary its light inten-

sity to the same value (as indicated by the ex-

posure meter) as it was putting out as a dum-

my load. The a -c voltmeter in parallel with the

lamp and ammeter in series with it is then

used to determine lamp power input by:

P = EI. This method of power determination

is satisfactory for audio and low frequency

r.f. but is not satisfactory for v -h -f work be-

cause of variations in lamp efficiency due to

uneven heating of the filament.

Dummy Loads Lamp bulbs make poor dummy

loads for r -f work, in general,

as they have considerable reactance above 2

Mc., and the resistance of the lamp varies with

the amount of current passing through it.

A suitable r -f load for powers up to a few

watts may be made by paralleling 2 -watt com-

position resistors of suitable value to make

www.americanradiohistory.com

HANDBOOK Measurement of Constants 737

a 50 ohm resistor of adequate dissipation. De-

vices such as this are discussed in chapter 29

(4CX 1000A amplifier) .

A 2 kw. dummy load having a s.w.r. of less

than 1.05 /1 at 30 Mc. is shown in figures 7,

8 and 9. The load consists of twelve 600 ohm,

120 watt Globar type CX non -inductive resis-

tors connected in parallel. A frequency com-

pensation circuit is used to balance out the

slight capacitive reactance of the resistors. The

compensation circuit is mounted in an alu-

minum tube 1" in diameter and 25/8" long. The

tube is plugged at the ends by metal discs,

and is mounted to the front panel of the box.

The resistors are mounted on aluminum

T -bar stock and are grounded to the case at

the rear of the assembly. Connection to the

coaxial receptacle is made via copper strap.

The power meter is calibrated using a

VTVM and r.f. probe. Power is applied to the

load at 3.5 Mc. and the level is adjusted to

provide 17.6 volts at "Calibration point."

With the Watts Switch in the 200 watt posi-

tion, the potentiometer is adjusted to provide

a reading of 100 watts on the meter. In the

2000 watt position, the other potentiometer is

adjusted for a meter reading of 200 watts. The

excitation frequency is now changed to 29.7

Mc. and the 17.6 volt level re- established.

Adjust the frequency compensating capacitor

until meter again reads 100 watts. Recheck at

3.5 Mc. and repeat until meter reads 100 watts

at each frequency when 17.6 volt level is

maintained.

33 -2 Measurement

of Circuit Constants

The measurement of the resistance, capaci-

tance, inductance, and Q ( figure of merit) of

the components used in communications work

VENTILATED SHIELD COVER

FR.F.IN 1 00

C

X PROBE WATTS

CYLINDER SWIT H

33K .005

1.5 -T

ERIE M55?

CHASSIS

NOTE FIXED RESISTORS ARE ONM /TE -L ITTLE DEVIL

COMPOSITION UNITS.

Figure 8.

SCHEMATIC, KILOWATT DUMMY

LOAD.

can be divided into three general methods:

the impedance method, the substitution or reso-

nance method, and the bridge method.

The Impedance The impedance method of

Method measuring inductance and

capacitance can be likened to

the ohmmeter method for measuring resistance.

An a -c voltmeter, or milliammeter in series

with a resistor, is connected in series with

the inductance or capacitance to be measured

and the a -c line. The reading of the meter will

be inversely proportional to the impedance of

the component being measured. After the me-

ter has been calibrated it will be possible to

obtain the approximate value of the impedance

directly from the scale of the meter. If the

component is a capacitor, the value of im-

pedance may be taken as its reactance at the

measurement frequency and the capacitance

determined accordingly. But the d-c resistance

of an inductor must also be taken into con-

sideration in determining its inductance. After

the d -c resistance and the impedance have

been determined, the reactance may be deter-

mined from the formula: XI. = V Z' -R'.

Then the inductance may be determined from:

L = XL /2Trf.

Figure 9.

DUMMY LOAD

ASSEMBLY.

Twelve Global' resistors

(surplus) are mounted to

aluminum "Tee" stock,

six to a side, in fuse

clips. Right end is sup-

ported by ceramic pillars

from front panel. Probe,

meter, and potentiom-

eters are at right.

www.americanradiohistory.com

738 Test Equipment THE RADIO

The Substitution The substitution method is

Method a satisfactory system for

obtaining the inductance or

capacitance of high -frequency components. A

large variable capacitor with a good dial hav-

ing an accurate calibration curve is a neces-

sity for making determinations by this method.

If an unknown inductor is to be measured, it is

connected in parallel with the standard capaci-

tor and the combination tuned accurately to

some known frequency. This tuning may be

accomplished either by using the tuned circuit

as a wavemeter and coupling it to the tuned

circuit of a reference oscillator, or by using

the tuned circuit in the controlling position of

a two terminal oscillator such as a dynatron

or transitron. The capacitance required to tune

this first frequency is then noted as G. The

circuit or the oscillator is then tuned to the

second harmonic of this first frequency and

the amount of capacitance again noted, this

tame as G. Then the distributed capacitance

across the coil (including all stray capaci-

tances) is equal to: Co= (C1- 4Cr)/3.

This value of distributed capacitance is

then substituted in the following formula along

with the value of the standard capacitance for

either of the two frequencies of measurement:

1

L- 4trsfi °(C2 -I- Co)

The determination of an unknown capaci-

tance is somewhat less complicated than the

above. A tuned circuit including a coil, the

unknown capacitor and the standard capacitor,

all in parallel, is resonated to some conveni-

ent frequency. The capacitance of the stand-

ard capacitor is noted. Then the unknown ca-

pacitor is removed and the circuit re- resonated

by means of the standard capacitor. The dif-

ference between the two readings of the stand-

ard capacitor is then equal to the capacitance

of the unknown capacitor.

33 -3 Measurements

with a Bridge

Experience has shown that one of the most

satisfactory methods for measuring circuit con-

stants (resistance, capacitance, and inductance)

at audio frequencies is by means of the a-c

bridge. The Wheatstone (d -c) bridge is also

one of the most accurate methods for the

measurement of d -c resistance. With a simple

bridge of the type shown at figure 9A it is

entirely practical to obtain d -c resistance de-

52

Rx= RA RS

RB

Figure 10.

TWO WHEATSTONE BRIDGE CIRCUITS

These circuits are used for the measurement

of d -c resistance. In (A) the "ratio arms" R,,

end R are fixed and balancing of the bridge

is accomplished by variation of the standard

R,. The standard in this case usually con-

sists of a decade box giving resistance in

1 -ohm steps from 0 to 1110 or to 11,110 ohms.

In (B) a fixed standard is used for each range

end the ratio arm is varied to obtain balance,

A calibrated slide -wire or potentiometer cali-

brated by resistance in terms of degrees is

usually employed as R, and R1,. It will be

noticed that the formulo for determining the

unknown resistance from the known is the

some in either case.

terminations accurate to four significant fig-

ures. With an a -c bridge operating within its

normal rating as to frequency and range of

measurement it is possible to obtain results

accurate to three significant figures.

Both the a -c and the d -c bridges consist of

a source of energy, a standard or reference of

measurement, a means of balancing this stand-

ard against the unknown, and a means of in-

dicating when this balance has been reached.

The source of energy in the d -c bridge is a

battery; the indicator is a sensitive galvanome-

ter. In the a -c bridge the source of energy

is an audio oscillator (usually in the vicinity

of 1000 cycles), and the indicator is usually

a pair of headphones. The standard for the d-c

bridge is a resistance, usually in the form of

a decade box. Standards for the a -c bridge can

be resistance, capacitance, and inductance in

varying forms.

Figure 10 shows two general types of the

Wheatstone or d -c bridge. In (A) the so- called

"ratio arms" RA and RA are fixed (usually in

a ratio of 1 -to -1, 1- to -10, 1 -to -100, or 1 -to-

1,000) and the standard resistor Rs is varied

until the bridge is in balance. In commercially

manufactured bridges there are usually two or

www.americanradiohistory.com

HANDBOOK Frequency Measurement 739

more buttons on the galvanometer for progres-

sively increasing its sensitivity as balance is

approached. Figure 10B is the slide wire type

of bridge in which fixed standards are used

and the ratio arm is continuously variable.

The slide wire may actually consist of a mov-

ing contact along a length of wire of uni-

form cross section in which case the ratio of

RA to RD may be read off directly in centi-

meters or inches, or in degrees of rotation if

the slide wire is bent around a circular former.

Alternatively, the slide wire may consist of

linear -wound potentiometer with its dial cali-

brated in degrees or in resistance from each

end. Figure 11A shows a simple type of a -c

bridge for the measurement of capacitance and

inductance. It can also, if desired, be used

for the measurement of resistance. It is neces-

sary with this [type ofl bridge to use a standard

which presents the same type of impedance as

the unknown being measured: resistance stan-

dard for a resistance measurement, capacitance

standard for capacitance, and inductance stan-

dard for inductance determination.

For measurement of capacitances from a few

micro -microfarads to about 0.001 µfd. a Wag-

ner grounded substitution capacitance bridge of

the type shown in figure 11B will be found

satisfactory. The ratio arms RA and RR should

be of the same value within 1 per cent; any

value between 2500 and 10,000 ohms for

both will be satisfactory. The two resistors

RI and R,I should be 1000 -ohm wire -wound

potentiometers. Cs should be a straight -line

capacitance capacitor with an accurate vernier

dial; 500 to 1000 µµtd. will be satisfactory.

Cc can be a two or three gang broadcast ca-

pacitor from 700 to 1000 µµfd. maximum ca-

pacitance.

The procedure for making a measurement

is as follows: The unknown capacitor Cx is

placed in parallel with the standard capacitor

Cs. The Wagner ground RIM is varied back and

forth a small amount from the center of its

range until no signal is heard in the phones

with the switch S in the center position. Then

the switch S is placed in either of the two out-

side positions, CIS is adjusted to a capaci-

tance somewhat greater than the assumed value

of the unknown Cx, and the bridge is brought

into balance by variation of the standard ca-

pacitor G. It may be necessary to cut some

resistance in at Rc and to switch to the other

outside position of S before an exact balance

can be obtained. The setting of Cs is then

noted, Cx is removed from the circuit ( but the

Zx=- ZS Xx= - RA X5 Rx= RA Rs

Z5= IMPEDANCE BEING MEASURED, Rs = RESISTANCE COMPONENT OF Z 5

Zs= IMPEDANCE Of STANDARD, Xx=REACTANCE COMPONENT OF Zx

Rx= RESISTANCE COMPONENT OF Z5, X5= REACTANCE COMPONENT OF ZS

O

Figure 11

TWO A -C BRIDGE CIRCUITS

The operation of these bridges is essentially

the same as those of figure 10 except that

o.c. is fed into the bridge instead of d.c. and

a pair of phones is used as the indicator in-

stead of the galvanometer. The bridge shown

at (A) con be used for the measurement of

resistance, but it is usually used for the

measurement of the impedance and reactance

of coils and capacitors at frequencies from

200 to 1000 cycles. The bridge shown at (B)

is used for the measurement of small values

of capacitance by the substitution method.

Full description of the operation of both

bridges is given in the accompanying text.

leads which went to it are not changed in any

way which would alter their mutual capaci-

tance), and C. is readjusted until balance is

again obtained. The difference in the two set-

tings of Cs is equal to the capacitance of the

unknown capacitor C.

33 -4 Frequency

Measurements

All frequency measurement within the

United States is based on the transmissions of

Station WWV of the National Bureau of

Standards. This station operates continuously

on frequencies of 2.5, 5, 10, 15, 20, and

25 Mc. The carriers of those frequencies

below 25 Mc. are modulated alternately by

a 440 -cycle tone or a 600 -cycle tone for pe-

riods of four minutes each. This tone is in-

terrupted at the beginning of the 59th minute

www.americanradiohistory.com

740 Test Equipment THE RADIO

SCHEMATIC

C- 100 -µpfd. air trimmer

C!,C,- 0.0003 -p/d. midget mica

C4- 50 -ppfd. midget mica

C, -0.002 -pfd. midget mica

R,, R -- 100,000 ohms /2 watt

L,- 10 -mh. shielded r -f choke

Lr- 2.1 -mh. r -f choke

X- 100 -kc. crystal

Figure 12

OF A 100 -KC. FREQUENCY SPOTTER

of each hour and each five minutes thereafter

for a period of precisely one minute. Green-

wich Civil Time is given in code during these

one -minute intervals, followed by a voice an-

nouncement giving Eastern Standard Time.

The accuracy of all radio and audio frequen-

cies is better than one part in 100,000,000.

A 5000 microsecond pulse (5 cycles of a 1000 -

cycle wave) may be heard as a tick for every

second except the 59th second of each minute.

These standard -frequency transmissions of

station WWV may be used for accurately de-

termining the limits of the various amateur

bands with the aid of the station communica-

tions receiver and a 50 -kc., 100 -kc., or 200 -

kc. band -edge spotter. The low frequency oscil-

lator may be self- excited if desired, but low

frequency standard crystals have become so

relatively inexpensive that a reference crystal

may be purchased for very little more than

the cost of the components for a self -excited

oscillator. The crystal has the additional advan-

tage that it may be once set so that its har-

monics are at zero beat with WWV and then

left with only an occasional check to see that

the frequency has not drifted more than a

few cycles. The self- excited oscillator, on the

other hand, must be monitored very frequent-

ly to insure that it is on frequency.

Using o To use a frequency spotter it is

Frequency only necessary to couple the out -

Spotter put of the oscillator unit to the

antenna terminal of the receiver

through a very small capacitance such as might

be made by twisting two pieces of insulated

hookup wire together. Station WWV is then

tuned in on one of its harmonics, 15 Mc.

will usually be best in the daytime and 5 or

10 Mc. at night, and the trimmer adjustment

on the oscillator is varied until zero beat is

obtained between the harmonic of the oscil-

lator and WWV. With a crystal reference

oscillator no difficulty will be had with using

the wrong harmonic of the oscillator to ob-

tain the beat, but with a self- excited oscillator

it will be wise to insure that the reference

oscillator is operating exactly on 50, 100, or

200 kc. (whichever frequency has been

chosen) by making sure that zero beat is ob-

tained simultaneously on all the frequencies

of WWV that can be heard, and by noting

whether or not the harmonics of the oscillator

in the amateur bands fall on the approximate

calibration marks of the receiver.

A simple frequency spotter is diagrammed

in figure 12.

33 -5 Antenna and

Transmission

Line Measurements

The degree of adjustment of any amateur

antenna can be judged by the study e4 the

standing -wave ratio on the transmission line

feeding the antenna. Various types I nstru-

ments have been designed to measure the

s.r.w. present on the transmission line, or to

measure the actual radiation resistance of the

antenna in question. The most important of

these instruments are the slotted line, the

bridge -type s -w -r meter, and the antennascope.

The Slotted Line It is obviously impractical

to measure the voltage -

standing- wave ratio in a length of coaxial line

since the voltages and currents inside the line

are completely shielded by the outer conductor

of the cable. Hence it is necessary to insert

some type of instrument into a section of the

line in order to be able to ascertain the con-

ditions which are taking place inside the

shielded line. Where measurements of a high

degree of accuracy are required, the slotted

line is the instrument most frequently used.

Such an instrument, diagrammed in figure 13,

is an item of test equipment which could be

constructed in a home workshop which in-

cluded a lathe and other metal working tools.

www.americanradiohistory.com

HANDBOOK Antenna Measurements 741

iCOAX TAPER PROBE INNER CONDUCTOR TAPER COAx

FITTING

CARRIER SLOT IN OUTER

FOR PROBE SLIDER CONDUCTOR

CARRYING

PROBE

Figure 13

DIAGRAMMATIC REPRESENTATION

OF A SLOTTED LINE

The conductor ratios in the slotted line, in-

cluding the tapered end sections should be

such that the characteristic impedance of the

equipment is the same as that of the trans-

mission line with which the equipment is to

be used. The indicating instrument may be

operated by the d -c output of the rectifier

coupled to the probe, or it may be operated

by the o -c components of the rectified signal

if the signal generator or transmitter is am-

plitude modulated by a constant percentage.

Commercially built slotted lines are very ex-

pensive since they are constructed with a high

degree of accuracy for precise laboratory work.

The slotted line consists essentially of a

section of air -dielectric line having the same

characteristic impedance as the transmission

line into which it is inserted. Tapered fittings

for the transmission line connectors at each

end of the slotted line usually are required due

to differences in the diameters of the slotted

line and the line into which it is inserted. A

narrow slot from 1/ -inch to 1/4 -inch in width

is cut into the outer conductor of the line. A

probe then is inserted into the slot so that it

is coupled to the field inside the line. Some

sort of accurately machined track or lead screw

must be provided to insure that the probe

maintains a constant spacing from the inner

conductor as it is moved from one end of the

slotted line to the other. The probe usually

includes some type of rectifying element

whose output is fed to an indicating instru-

ment alongside the slotted line.

The unfortunate part of the slotted -line sys-

tem of measurement is that the line must be

somewhat over one -half wavelength long at

the test frequency, and for best results should

be a full wavelength long. This requirement is

easily met at frequencies of 420 Mc. and above

where a full wavelength is 28 inches or less.

But for the lower frequencies such an instru-

ment is mechanically impracticable.

Bridge -Type

Standing -Wove

Indicators

The bridge type of standing -

wave indicator is used quite

generally for making meas-

urements on commercial co-

Tc J

Figure 14

RESISTOR -BRIDGE

STANDING -WAVE INDICATOR

This type of test equipment is suitable

for use with coaxial feed lines.

C,- 0.001 -pfd. midget ceramic capacitor

C: C, -.001 disc ceramic

R., R. -22 -ohm 2 -watt carbon resistors

R,- Resistor equal in resistance to the charac-

teristic impedance of the coaxial transmis-

sion line to be used (I watt)

R. -5000 -ohm wire -wound potentiometer

R - 10,000 -ohm 1 -watt resistor

RFC -R -f choke suitable for operation at the

measurement frequency

OUTPUT

TO

ANTENNA

FEED LINE

axial transmission lines. A simplified version

is available from M. C. Jones Electronics Co.,

Bristol, Conn. ("Micro -Match ") .

One type of bridge standing -wave indicator

is diagrammed in figure 14, This type of in-

strument compares the electrical impedance of

the transmission line with that of the resistor

R3 which is included within the unit. Experi-

ence with such units has shown that the re-

sistor R3 should be a good grade of non- induc-

tive carbon type. The Ohmite "Little Devil"

type resistor in the 2 -watt rating has given

good performance. The resistance at 113 should

be equal to the characteristic impedance of

the antenna transmission line. In other words,

this resistor should have a value of 52 ohms

for lines having this characteristic impedance

such as RG -8 /U and RG -58/U. For use with

lines having a nominal characteristic im-

pedance of 70 ohms, a selected "68 ohm" re-

sistor having an actual resistance of 70 ohms

may be used.

Balance within the equipment is checked by

mounting a resistor, equal in value to the

nominal characteristic impedance of the line

to be used, on a coaxial plug of the type used

on the end of the antenna feed line. Then

this plug is inserted into the input receptacle

of the instrument and a power of 2 to 4 watts

applied to the output receptacle on the desired

frequency of operation. Note that the signal

is passed through the bridge in the direction

www.americanradiohistory.com

742 Test Equipment THE RADIO

Ole

25

23

20 te

1S.

12

10

9

e

7

e

- 5

1.5 4

LS

3

63

A

S W

ZR - ZO

2

I.S

= ZR

R_ 1 t IAI

1 - IAI

25

0 01 02 03 0. 0.5 0.9 0.7 0.e 09 1 O

READING ON 0 -1 INSTRUMENT, OR FACTOR TIMES FULL SCALE

(MAGNITUDE OF REFLECTION COEFFICIENT, A)

Figure 15

RELATION BETWEEN STANDING -

WAVE RATIO AND REFLECTION

COEFFICIENT

This chart may be used to convert reflection -

coefficient indications such as are obtained

with a bridge -type standing -wave indicator

or an indicating twin lamp into values of

standing -wave ratio.

INP.

Ji

REFLECTOMETER

OUTER SHELL

INNER CONDUCTOR

1 D2 p,

1 T k0 270

L 1 R,

01

c

S1

I FORWARD REFLECTED!

POWER POWER

R2 101(

INDICATOR

.01

r--

I

OUT.

2

COMPONENT PARTS

END DISC = 2 3/6" DIAMETER X 1/4' (2 REQ)

OUTER SHELL= 2 3/6" I.D. X 6' (! RED-

ALIGNMENT ROD= 1/4 DIAMETER X 5 1/2" (2 REQ. )

INNER CONDUCTOR= 1/2 " DIAMETER X S 1/6 ", TAPER ENDS

TO SOLDER TO RECEPTACLES (2 4E0. )

RECEPTACLES = SO -239 (2 REQ.) (J 1,-12 )

BINDING POSTS= (3 REQ.)

Figure 16

SCHEMATIC, REFLECTOMETER

D:, D-- Crystal diode, 1N34A or 1N82

R1 -270 ohm, 1 watt composition resistor.

IRC type BTA, matched pair.

M -0 -1 d.c. milliammeter

J,, J_- Coaxial receptacle, SO -239.

opposite to normal for this test. The resistor

R. is adjusted for full -scale deflection on the

0 -100 microammeter. Then the plugs are re-

versed so that the test signal passes through

the instrument in the direction indicated by

the arrow on figure 14, and the power level

is maintained the same as before. If the test

resistor is matched to R3, and stray capacitances

have been held to low values, the indication

on the milliammeter will be very small. The

test plug with its resistor is removed and the

plug for the antenna transmission line is in-

serted. The meter indication now will read

the reflection coefficient which exists on the

antenna transmission line at the point where

the indicator has been inserted. From this

reading of reflection coefficient the actual

standing -wave ratio on the transmission line

may be determined by reference to the chart

of figure 15.

Measurements of this type are quite helpful

in determining whether or not the antenna is

presenting a good impedance match to the

transmission line being used to feed it. How-

ever, a test instrument of the type shown in

figure 14 must be inserted into the line for a

measurement, and then removed from the line

when the equipment is to be operated. Also,

the power input to the line feeding the input

terminal of the standing -wave indicator must

not exceed 4 watts. The power level which the

unit can accept is determined by the dissipa-

tion limitation of resistors R1 plus R.

It is also important, for satisfactory opera-

tion of the test unit, that resistors R1 and R2

be exactly equal in value. The actual resist-

ance of these two is not critically important,

and deviations up to 10 per cent from the

value given in figure 14 will be satisfactory.

But the two resistors must have the same value,

whether they are both 21 ohms or 24 ohms,

or some value in between.

33 -6 A Simple

Coaxial Reflectometer

The reflectometer is a short section of co-

axial transmission line containing two r -f volt-

meters. One voltmeter reads the incident com-

ponent of voltage in the line, and the other

www.americanradiohistory.com

HANDBOOK Reflectometer 743

Figure 17

INTERIOR VIEW OF COAXIAL REFLECTOMETER

The Reflectometer is a short section 'f transmission line containing two r -f voltmeters. Center conductor

of line is a section of brass rod soldered to center pins of input and output receptacles. At either end

of unit are the crystal diodes, bypass capacitors and terminals. Diode load resistors are at center of

instrument, grounded to brass alignment rod.

reads the reflected component. The magnitude

of standing wave ratio on the transmission

line is the ratio of the incident component

to the reflected component, as shown in fig-

ure 15. In actual use, calibration of the re-

flectometer is not required since the relative

reading of reflected power indicates the de-

gree of match or mis -match and all antenna

and transmission line adjustments should be

conducted so as to make this reading as low

as possible, regardless of its absolute value.

The actual meter readings obtained from

the device are a function of the operating fre-

quency, the sensitivity of the instrument being

a function of transmitter power, increasing

rapidly as the frequency of operation is in-

creased. However, the reflectometer is inval-

uable in that it may be left permanently in

the transmission line, regardless of the power

output level of the transmitter. It will indi-

cate the degree of reflected power in the an-

tenna system, and at the same time provide

a visual indication of the power output of

the transmitter.

Reflectometer

Circuit The circuit and assembly in-

formation for the reflectometer

are given in figure 16. Two

diode voltmeters are coupled back -to -back to

a short length of transmission line. The com-

bined inductive and capacative pickup between

each voltmeter and the line is such that the

incident component of the line voltage is bal-

anced out in one case and the inductive com-

ponent is balanced out in the other case. Each

voltmeter, therefore, reads only one wave -com-

ponent. Careful attention to physical symmetry

of the assembly insures accurate and complete

separation of the voltage components by the

two voltmeters. The outputs of the two volt-

meters may be selected and read on an ex-

ternal meter connected to the terminal posts

of the reflectometer.

Each r -f voltmeter is composed of a load

resistor and a pickup loop. The pickup loop

is positioned parallel to a section of transmis-

sion line permitting both inductive and ca-

pacative coupling to exist between the center

conductor of the line and the loop. The 'di-

mensions of the center conductor and the

outer shield of the reflectometer are chosen

so that the instrument impedance closely

matches that of the transmission line.

Reflectometer

Construction A view of the interior of the

reflectometer is shown in fig-

ure 17. The coaxial input and

output connectors of the instrument are

mounted on machined brass discs that are

held in place by brass alignment rods, tapped

at each end. The center conductor is machined

from a short section of brass rod, tapered and

drilled at each end to fit over the center pin

of each coaxial receptacle. The end discs, the

rods, and the center conductor should be sil-

ver plated before assembly. When the center

conductor is placed in position, it is soldered

at each end to the center pin of the coaxial

receptacles.

One of the alignment rods is drilled and

tapped for a 6 -32 bolt at the mid -point, and

the end discs are drilled to hold 1/2-inch

www.americanradiohistory.com

744 Test Equipment THE RADIO

ceramic insulators and binding posts, as shown

in the photograph. The load resistors, crystal

diodes, and bypass capacitors are finally

mounted in the assembly as the last step.

The two load resistors should be measured

on an ohmmeter to ensure that the resistance

values are equal. The exact value of resistance

is unimportant as long as the two resistors

are equal. The diodes should also be checked

on an ohmmeter to make sure that the front

resistances and back resistances are balanced

between the units. Care should be taken dur-

ing soldering to ensure the diodes and resis-

tors are not overheated. Observe that the re-

sistor leads are of equal length and that each

half of the assembly is a mirror -image of the

other half. The body of the resistor is spaced

about 1,18 -inch away from the center conduc-

tor.

Testing the

Reflcctometer The instrument can be ad-

justed on the 28 Mc. band.

An r -f source of a few watts

and nonreactive load are required. The con-

struction of the reflectometer is such that it

will work well with either 52- or 72 -ohm

coaxial transmission lines. A suitable dummy

load for the 52 -ohm line can be made of four

220 ohm, 2 watt composition resistors (Ohm -

ite "Little Devil ") connected in parallel. Clip

the leads of the resistors short and mount

them on a coaxial plug. This assembly pro-

vides an eight watt, 55 ohm load, suitable

for use at 30 Mc. If an accurate ohmmeter

is at hand, the resistors may be hand picked

to obtain four 208 ohm units, thus making

the dummy load resistors exactly 52 ohms.

For all practical purposes, the 55 ohm load

is satisfactory. A 75 ohm, eight watt load

resistor may be made of four 300 ohm, 2

watt composition resistors connected in paral-

lel. R -f power is coupled to the reflectometer

and the dummy load is placed in the "output"

receptacle. The indicator meter is switched to

the "reflected power" position. The meter read-

ing should be almost zero. It may be brought

to zero by removing the case of the instru-

ment and. adjusting the position of the load

resistor. The actual length of wire in the resis-

tor lead and its positioning determine the meter

null. Replace the case before power is applied

to the reflectometer. The reflectometer is now

reversed and power is applied to the "output"

receptacle, with a dummy load attached to

the "input" receptacle. The second voltmeter

(forward power) is adjusted for a null read-

ing of the meter in the same manner.

If a reflected reading of zero is not obtain-

able, the harmonic content of the r -f source

might be causing a slight residual meter read-

ing. Coupling the reflectometer to the r -f

source through a tuned circuit ( "antenna

tuner ") will remove the offending harmonic

and permit an accurate null indication. Be

sure to hold the r -f input power to a low

value to prevent overheating the dummy load

resistors.

Using the The bridge may be used up to

Ref'ectometer 150 Mc. It is placed in the

transmission line at a conveni-

ent point, preferably before any tuner, balun,

or TVI filter. The indicator should be set to

read forward power, with a maximum of re-

sistance in the circuit. Power is applied and

the indicator resistor is adjusted for a full

scale reading. The switch is then thrown to

read reflected power (indicated as A, figure

15) . Assume that the forward power meter

reading is 1.0 and the reflected power read-

ing is 0.5. Substituting these values in the

SWR formula of figure 15 shows the SWR

to be 3. If forward power is always set to

1.0 on the meter, the reflected power (A)

can be read directly from the curve of figure

15 with little error.

If the meter is adjusted so as to provide a

half -scale reading of the forward power, the

reflectometer may be used as a transmitter

power output meter. Tuning adjustments may

then be undertaken to provide greatest meter

reading.

33 -7 Measurements

on Balanced

Transmission Lines

Measurements made on balanced transmis-

sion lines may be conducted in the same man-

ner as those made on coaxial lines. In the

case of the coaxial lines, care must be taken

to prevent flow of r -f current on the outer

surface of the line as this unwanted compo-

nent will introduce errors in measurements

made on the line. In like fashion, the cur-

rents in a balanced transmission line must

be 180 degrees out of phase and balanced

with respect to ground in order to obtain a

realistic relationship between incident and re-

flected power. This situation is not always

easy to obtain in practice because of the prox-

imity effects of metallic objects or the earth

to the transmission line. All transmission line

measurements, therefore, should be conducted

with the realization of the physical limitations

www.americanradiohistory.com

HANDBOOK

TWIN LEAD TRANSMISSION

LINE TO ANTENNA

Figure 18

SKETCH OF THE "TWIN- LAMP"

TYPE OF S -W -R INDICATOR

The short section of line with lamps at each

end usually is taped to the main transmission

tine with plastic electrical tape.

of the equipment and the measuring technique

that is being used.

Measurements on One of the most satisfactory

Molded Parallel- and least expensive devices

Wire Lines for obtaining a rough idea

of the standing -wave ratio

on a transmission line of the molded parallel -

wire type is the twin -lamp. This ingenious in-

strument may be constructed of new compon-

ents for a total cost of about 25 cents; this

fact alone places the twin -lamp in a class by

itself as far as test instruments are concerned.

Figure 18 shows a sketch of a twin -lamp in-

dicator. The indicating portion of the system

consists merely of a length of 300 -ohm Twin -

Lead about 10 inches long with a dial lamp at

each end. In the unit illustrated the dial lamps

are standard 6.3 -volt 150 -ma. bayonet -base

lamps. The lamps are soldered to the two leads

at each end of the short section of Twin -Lead.

To make a measurement the short section

of line with the lamps at each end is merely

taped to the section of Twin -Lead (or other

similar transmission line) running from the

transmitter or from the antenna changeover re-

lay to the antenna system. When there are no

standing waves on the antenna transmission

line the lamp toward the transmitter will light

while the one toward the antenna will not

light. With 300 -ohm Twin -Lead running from

the antenna changeover relay to the antenna,

and with about 200 watts input on the 28 -Mc.

band, the dial lamp toward the transmitter will

light nearly to full brilliancy. With a standing-

Figure 20

SWR BRIDGE FOR BALANCED

TRANSMISSION LINE

A double bridge can be used for two wire

transmission lines. Bridge is inserted in line

and may be driven with grid -dip oscillator or

other low power r -f source.

S.W.R. Indicator 745

f I L I L! t Ic

Figure 19

OPERATION OF THE "TWIN LAMP"

INDICATOR

Showing current flow resulting from inductive

and capacitive fields in a "twin lamp" attached

to a line with a low standing -wave ratio.

wave ratio of about 1.5 to 1 on the transmis-

sion line to the antenna the lamp toward the

antenna will just begin to light. With a high

standing -wave ratio on the antenna feed line

both lamps will light nearly to full brilliancy.

Hence the instrument gives an indication of

relatively low standing waves, but when the

standing -wave ratio is high the twin -lamp

merely indicates that they are high without

giving any idea of the actual magnitude.

33 -8 A "Balanced"

SWR Bridge

Two resistor -type standing wave indicators

may be placed "back -to- back" to form a SWR

bridge capable of being used on two wire

balanced transmission lines. Such a bridge is

shown in figures 20 and 22. The schematic of

such an instrument (figure 21) may be com-

pared to two of the simple bridges shown in

figure 14. When the dual bridge circuit is

balanced the meter reading is zero. This state

is reached when the line currents are equal

and exactly 180 degrees out of phase and the

SWR is unity.

As the condition of the line departs from

the optimum, the meter of the bridge will

show the degree of departure. When the line

currents are balanced and 180 degrees out of

phase, the meter will read the true value of

standing wave ratio on the line. If these con-

ditions are not met, the reading is not abso-

lute, merely giving an indication of the degree

of mis -match in the line. This handicap is not

SET FULL

SCALE

CALIBRATE

SWR. CAL.

SELECTOR

www.americanradiohistory.com

746 Test Equipment T H E R A D I O

0--T"

aurrur¡ Si

Ri

250

CAL. RFC - 5N .Ot R 250

s*+wR .00i 1N54 Ri 250

250 Ri

Figure 21

SCHEMATIC OF BRIDGE FOR

BALANCED LINES

M-0 -200 d -c microammeter

R, -Note: Six 250 ohm resistors are composi-

tion, non -inductive units. IRC type BT, or

Ohmite "Little Devil" 1 -watt resistors may

be used. (see text)

Si -DPDT rotary switch. Centralab type 1464

important, since the relative, not the absolute,

degree of mis -match is sufficient for transmis-

sion line adjustments to be made.

Bridge

Construction A suggested method of con-

struction of the balanced bridge

is illustrated in figures 20 and

22. The unit is constructed within a box mea-

suring 4" x 6" x 2" in size. The 0 - 200 d.c.

microammeter is placed in the center of the

4" x 6" side of the case. The input and output

connectors of the instrument are placed on

each end of the box and the internal wiring

is arranged so that the transmission line, in

effect, passes in one side of the box and out

the other with as little discontinuity as possi-

ble. The input and output terminals are mount-

ed on phenolic plates placed over large cut-

outs in the ends of the box, thus reducing

circuit capacity to ground to a minimum value.

The "SWR -CAL" switch Si is located on one

side of the meter and the "Calibrate" poten-

tiometer R1 is placed on the opposite side.

The transmission line within the unit is

broken by two 250 ohm composition resistors

and switch Si. The line segments are made of

short pieces of #10 copper wire, running be-

tween the various components. Spacing be-

tween the wires is held close to three inches

to approximate a 500 -600 ohm line.

The small components of the bridge are

placed symmetrically about the 250 ohm series

line resistors and the calibrating potentiometer,

as can be seen in figure 22. Exact parts place-

ment is not critical, except that the crystal

diodes should be placed at right angles to

the wires of the transmission line to reduce

capacative pickup. The two r -f chokes should

then be placed at right angles to the diodes.

The six 250 ohm resistors are checked on an

ohmmeter and should be hand -picked to ob-

tain units that are reasonably close in value.

If it is desired to use the bridge with a 600

ohm line, the value of these resistors should

be increased to 300 ohms each. Excessive heat

should not be used in soldering either the re-

sistors or the diodes to ensure that their char-

acteristics will not be altered by application of

high temperatures over an extended period.

Testing the When the instrument is

Balanced Bridge completed, a grid -dip meter

may be coupled to the in-

put terminals via a two turn link. Be careful

not to pin the bridge meter. Place switch Si in

the "Calibrate" (open) position. Set the grid

dip meter in the 10 -Mc. to 20 -Mc. range and

adjust the link and calibration control Ri for

full scale meter reading. A 500 ohm carbon

resistor placed across the output terminals of

the unit should produce a zero meter reading

when Si is set to the "SWR" position. Various

values of resistance may now be placed across

the meter terminals to obtain calibration points

for the meter scale. The ratio of the external

Figure 22

INTERIOR VIEW OF

BALANCED BRIDGE

SHOWING PARTS

PLACEMENT

Diode rectifiers are placed

at right angles to the

short section of trans-

mission line. Both sides

of bridge are balanced

to ground by virtue of

symmetrical construction

of unit.

www.americanradiohistory.com

HANDBOOK 747

Figure 23

THE ANTENNASCOPE

The radiation resistance of r -f loads connected

across the output receptacle may be quickly

determined by a direct dial reading. The An-

tennascope may be driven with a grid -dip

oscillator, covering r -f impedance range of S

to 1000 ohms.

resistor to the design value of the bridge will

provide the SWR value for any given meter

reading. For example, a 1000 ohm resistor

has a ratio of 1000/500, and will give an indi-

cated SWR reading of 2. A 1500 ohm resistor

will give an indicated reading of 3, a 2000

ohm resistor will provide a reading of 4, and

so on. Before each measurement is recorded,

the calibrate control should be set to a full

scale reading with Si open.

This simple system of calibration will lead

to slight errors in calibration if the regulation

of the r -f source is poor. That is, a change in

the external calibrating resistance will produce

a varying load on the r -f generator which could

easily cause a change in the power applied to

the bridge. A separate diode r -f voltmeter

placed across the pickup loop will enable the

input voltage to be held to a constant value

and will provide somewhat more accurate

bridge calibration.

Using the The bridge is placed in the trans -

Bridge mission line and driven from a

low powered source having a min-

imum of harmonic content. Several measure-

ments should be made at various frequencies

within the range of antenna operation. The

selector switch is set to the "Cal." position and

the "Calibrate" potentiometer is adjusted for

full scale meter reading. The switch is then

set to the "SWR" position and a reading is

taken. This reading and others taken at various

frequencies may be plotted on a graph to pro-

vide a "SWR curve" for the particular antenna

and transmission line. Antenna adjustments

and line balancing operations may now be

conducted to provide a smooth SWR curve,

with the point of minimum SWR occuring at

the chosen design frequency of the antenna in-

stallation. The complete adjustment and check-

out procedure for an antenna and transmission

line system is covered in the Beam Antenna

Handbook, published by Radio Publications,

Inc., Wilton, Conn.

33 -9 The Antennascope

The Antennascope is a modified SWR

bridge in which one leg of the bridge is com-

posed of a non -inductive variable resistor. This

resistor is calibrated in ohms, and when its

setting is equal to the radiation resistance of

the antenna under test the bridge is in a bal-

anced state. If a sensitive voltmeter is con-

GuT I

Figure 24

SCHEMATIC, ANTENNASCOPE

R; -1000 ohm composition potentiometer ohm -

ite type AB or Allen Bradley type J,

linear taper

R,-50 ohm, 1 -watt composition resistor, IRC

type BT, or ohmite "Little Devil" (see

text)

M -0 -200 d -c microammeter

www.americanradiohistory.com

748 Test Equipment THE RADIO

netted across the bridge, it will indicate a

voltage null at bridge balance. The radiation

resistance of the antenna may then be read di-

rectly from the calibrated resistor of the in-

strument.

When the antenna under test is in a non -

resonant or reactive state, the null indication

on the meter of the Antennascope will be in-

complete. The frequency of the exciting sig-

nal must then be moved to the resonant fre-

quency of the antenna to obtain accurate read-

ings of radiation resistance from the dial of

the instrument.

A typical Antennascope is shown in figures

23 and 25, and the schematic is shown in fig-

ure 24. A 1000 ohm non -inductive carbon po-

tentiometer serves as the variable leg of the

bridge. The other legs are composed of the 50

ohm composition resistor and the radiation re-

sistance of the antenna. If the radiation resis-

tance of the external load or antenna is 50

ohms and the potentiometer is set at mid -scale

the bridge is in balance and the diode volt-

meter will read zero. If the radiation resistance

of the antenna is any value other than 50

ohms, the bridge may be balanced to this new

value by varying the position of the potentio-

meter. Bridge balance may be obtained with

non - reactive loads in the range of 5 ohms to

1000 ohms with this simple circuit. When

measurements are conducted at the resonant

frequency of the antenna system the radiation

resistance of the installation may be read di-

rectly from the calibrated dial of the Anten-

nascope. Conversely, a null reading of the in-

strument will occur at the resonant frequency,

which may easily be found with the aid of a

calibrated receiver or frequency meter.

Constructing the The Antennascope is built

Antennascope within a sheet metal case

measuring 3" x 6" x 2 ".

The indicating meter is placed at the top of

the case, and the r -f bridge occupies the lower

portion of the box. The input and output co-

axial fittings are mounted on each side of the

box and the non -inductive 50 ohm resistor is

soldered between the center terminals of the

receptacles.

The calibrating potentiometer (R2) is

mounted upon a phenolic plate placed over a

3/4-inch hole drilled in the front of the box.

This reduces the capacity to ground of the po-

tentiometer to a minimum. Placement of the

small components within the box may be seen

in figure 25. Care should be taken to mount

the crystal diode at right angles to the 50 ohm

resistor to reduce capacity coupling between

the components.

The upper frequency limit of accuracy of

the Antennascope is determined by the as-

sembly technique. The unit shown will work

with good accuracy to approximately 100 Mc.

Above this frequency, the self- inductance of the

leads prevents a perfect null from being ob-

tained. For operation in the VHF region, it

would be wise to rearrange the components to

reduce lead length to an absolute minimum,

and to use 1/4-inch copper strap for the r -f

leads instead of wire.

Testing the When the instrument is corn-

Antennascope pleted, a grid -dip meter may

be coupled to the input recep-

tacle of the Antennascope by means of a two

Figure 25

PLACEMENT OF PARTS WITHIN

THE ANTENNASCOPE

With the length of leads shown this model

is useful up to about 100 Mc. Crystal diode

should be placed at right angles to 50 ohm

composition resistor.

www.americanradiohistory.com

HANDBOOK Antennascope 749

Figure 26

SIMPLE SILICON CRYSTAL NOISE

GENERATOR

turn link. The frequency of excitation should

be in the 10 Mc. -20 Mc. region. Coupling

should be adjusted to obtain a half -scale read-

ing of the meter. Various values of 1 -watt

oomposition resistors up to 1000 ohms are

then plugged into the "output" coaxial recep-

tacle and the potentiometer is adjusted for a

null on the meter. The settings of the poten-

tiometer may now be calibrated in terms of

the load resistor, the null position indicating

the value of the test resistor. A calibrated scale

for the potentiometer should be made, as

shown in figure 23.

Using the The antennascope may be

Antennascope driven by a grid -dip oscillator

coupled to it by a two turn

link. Enough coupling should be used to obtain

at least a 3/ scale reading on the meter of

the Antennascope with no load connected to

the measuring terminals. The Antennascope

may be considered to be a low range r -f ohm-

meter and may be employed to determine the

electrical length of quarter -wave lines, surge

impedance of transmission lines, and antenna

resonance and radiation resistance.

In general, the measuring terminals of the

Antennascope are connected in series with the

load at a point of maximum current. This

means the center of a dipole, or the base of a

vertical 1/4-wave ground plane antenna. Exci-

tation is supplied to the Antennascope, and the

frequency of excitation and the resistance con-

trol of the Antennascope are both varied until

a complete null is obtained on the indicating

meter of the Antennascope. The frequency of

the source of excitation is now the resonant

frequency of the load, and the radiation resist-

ance of the load may be read upon the dial of

the Antennascope.

On measurements on 80 and 40 meters, it

might be found that it is impossible to obtain

a complete null on the Antennascope. This is

usually caused by pickup of a nearby broad-

cast station, the rectified signal of the broad-

cast station obscuring the null indication on

the Antennascope. This action is only noticed

when antennas of large size are being checked.

33 -10 A Silicon Crystal

Noise Generator

The limiting factor in signal reception above

25 Mc. is usually the thermal noise generated

in the receiver. At any frequency, however,

the tuned circuits of the receiver must be ac-

curately aligned for best signal -to -noise ratio.

Circuit changes (and even alignment changes)

in the r -f stages of a receiver may do much to

either enhance or degrade the noise figure of

the receiver. It is exceedingly hard to deter-

mine whether changes of either alignment or

circuitry are really providing a boost in signal -

to -noise ratio of the receiver, or are merely in-

creasing the gain (and noise) of the unit.

A simple means of determining the degree

of actual sensitivity of a receiver is to inject

a minute signal in the input circuit and then

measure the amount of this signal that is

needed to overcome the inherent receiver

noise. The less injected signal needed to over-

ride the receiver noise by a certain, fixed

amount, the more sensitive is the receiver.

6 VOLTS

AOi

CERAMIC

WELDED [NCLO URE

A SILICON CRYSTAL NOISE GENERATOR

Figure 27

www.americanradiohistory.com

750 Test Equipment T H E RADIO

NOISE

GENERATOR R CC CI VCR

`TERMINATING

RESISTOR

O

SPAR

O

TEST SET-UP FOR NOISE GENERATOR

Figure 28

A simple source of minute signal may be ob-

tained from a silicon crystal diode. If a small

d -c current is passed through a silicon crystal

in the direction of highest resistance, a small

but constant r -f noise (or hiss) is generated.

The voltage necessary to generate this noise

may be obtained from a few flashlight cells.

The generator is a broad band device and re-

quires no tuning. If built with short leads, it

may be employed for receiver measurements

well above 150 Mc. The noise generator should

be used for comparative measurements only,

since calibration against a high quality com-

mercial noise generator is necessary for ab-

solute measurements.

A Practical Shown in figure 26 is a

Noise Generator simple silicon crystal noise

generator. The schematic of

this unit is illustrated in figure 27. The 1N21

crystal and .001 µfd. ceramic capacitor are

connected in series directly across the output

terminals of the instrument. Three small flash-

light batteries are wired in series and mounted

inside the case, along with the 0 -2 d -c milliam-

meter and the noise level potentiometer.

To prevent heat damage to the 1N21 crystal

during the soldering process, the crystal should

be held with a damp rag, and the connections

soldered to it quickly with a very hot iron.

Across the terminals (and in parallel with the

equipment to be attached to the generator) is

a 1 -watt carbon resistor whose resistance is

equal to the impedance level at which meas-

urements are to be made. This will usually be

either 50 or 300 ohms. If the noise generator

is to be used at one impedance level only, this

resistor may be mounted permanently inside

of the case.

Using the The test setup for use of

Noise Generator the noise generator is shown

in figure 28. The noise gen-

erator is connected to the antenna terminals

of the receiver under test. The receiver is

turned on, the a.v.c. turned off, and the r -f

gain control placed full on. The audio volume

control is adjusted until the output meter ad-

vantes to one -quarter scale. This reading is

the basic receiver noise. The noise generator

is turned on, and the noise level potentiometer

adjusted until the noise output voltage of the

receiver is doubled. The more resistance in

the diode circuit, the better is the signal -to-

noise ratio of the receiver under test. The r -f

circuits of the receiver may be aligned for

maximum signal -to -noise ratio with the noise

generator by aligning for a 2/1 noise ratio at

minimum diode current.

33 -11 A Monitor Scope

for AM and SSB

This miniature monitor scope is designed

to be used with transmitters having a plate

supply of 500 to 3000 volts. The scope draws

its plate power from the transmitter, thus

eliminating the costly and bulky power supply

usually required for an instrument of this type.

The circuit of the scope is shown in figure

30. A 2AP1 tube is used, with electrode volt-

ages obtained from a voltage divider which is

placed across the transmitter power supply. A

60 cycle sweep circuit is used with return trace

blanking derived from a simple phase shift

circuit. This sweep is not ideal, but is satis-

factory for the intended purpose of the scope.

A more sophisticated sweep circuit would re-

Figure 29.

This miniature oscilloscope is designed to be

used with a.m. and s.s.b. transmitters and

draws its anode power from the transmitter

plate supply. A small steel chassis and bottom

plate are used to make the 'scope cabinet.

www.americanradiohistory.com

HANDBOOK Noise Generator 751

1_0

1nv

H

220 11

220/1

3300

50 K

120

1011

5011

IM 104

CENTERING

CONTROLS

+3KV

2API 001 R.F.

1 K 15 INPUT

11

INTENSITY

KV

115V.1,

001

I1

01 RFC

ITV 2.3MH

NOTE

ALL RESISTORS 1 -WATT

TI .150V. AT 25 MA.

6.3 V. AT 0.5 A.

STANCOR P -0181

Figure 30.

SCHEMATIC, MONITOR OSCILLOSCOPE.

quire more circuitry and a low voltage supply,

both of which would increase the size, com-

plexity and cost of the unit.

The cathode circuit is at ground potential

and the centering controls are above ground.

These two potentiometers are mounted on a

phenolic board on the

are adjusted with an

After adjustment they

second phenolic board

from touching them.

side of the unit, and

insulated screwdriver.

are covered with a

to prevent the user

If the scope is to be used with a supply

voltage lower than 2000, one of the 1- megohm

resistors at the "top" of the divider should be

removed. Five hundred to 1000 volts should

be measured across the 1 µfd. filter capacitor.

A VTVM should be used for this measurement.

The monitor scope is built into a steel

chassis measuring 21/2" x 5" x 91/2", and is

designed to sit atop the receiver. R.f. connec-

tion to the transmitter may be made by insert-

ing a coaxial "Tee" in the transmission line

and running a short length of similar line from

the "Tee" to the scope. Operation of the scope

and its uses are covered in chapter 9, "The

Oscilloscope."

Figure 31.

UNDER -CHASSIS

VIEW OF

OSCILLOSCOPE.

Filament transformer is

mounted directly behind

'scope tube so as not to

distort electron beam.

Centering controls are

mounted on phenolic

board on chassis edge.

Controls are covered aft-

er adjustment to elimin-

ate shock hazard.

www.americanradiohistory.com

CHAPTER THIRTY -FOUR

Radio Mathematics

and Calculations

Radiomen often have occasion to cal-

culate sizes and values of required parts. This

requires some knowledge of mathematics. The

following pages contain a review of those parts

of mathematics necessary to understand and

apply the information contained in this book.

It is assumed that the reader has had some

mathematical training; this chapter is not in-

tended to teach those who have never learned

anything of the subject.

Fortunately only a knowledge of fundamen-

tals is necessary, although this knowledge must

include several branches of the subject. Fortu-

nately, too, the majority of practical applica-

tions in radio work reduce to the solution of

equations or formulas or the interpretation of

graphs.

Arithmetic

Notation of In writing numbers in the Ara -

Numbers bic system we employ ten dif-

ferent symbols, digits, or fig-

ures: 1, 2, 3, 4, 5, 6, 7, 8, 9, and 0, and place

them in a definite sequence. If there is more

than one figure in the number the position of

each figure or digit is as important in deter-

mining its value as is the digit itself. When we

deal with whole numbers the righthandmost

digit represents units, the next to the left rep-

resents tens, the next hundreds, the next thou-

sands, from which we derive the rule that ev-

ery time a digit is placed one space further to

the left its value is multiplied by ten.

etw 8 1 4 3

hundreds tens units

It will be seen that any number is actually a

sum. In the example given above it is the sum

of eight tnousands, plus one hundred, plus

four tens, plus three units, which could be

written as follows:

8 thousands 110 x 10 x 101

1 hundreds 1 10 x 10 )

4 tens

3 units

8143

The number in the units position is some-

times referred to as a firit order number, that

in the tens position is of the second order, that

in the hundreds position the third order, etc.

The idea of letting the position of the sym-

bol denote its value is an outcome of the aba-

cus. The abacus had only a limited num-

ber of wires with beads, but it soon became

apparent that the quantity of symbols might

be continued indefinitely towards the left,

each further space multiplying the digit's

value by ten. Thus any quantity, however

large, may readily be indicated.

It has become customary for ease of reading

to divide large numbers into groups of three

digits, separating them by commas.

6,000,000 rather than 6000000

Our system of notation then is characterized

by two things: the use of positions to indicate

the value of each symbol, and the use of ten

symbols, from which we derive the name dec-

imal system.

Retaining the same use of positions, we

might have used a different number of sym-

bols, and displacing a symbol one place to the

left might multiply its value by any other fac-

tor such as 2, 6 or 12. Such other systems have

been in use in history, but will not he discussed

here. There are also systems in which displac-

ing a symbol to the left multiplies its value by

752

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Decimal Fractions 753

O 0.5

I I I I I I I I I I

b

004

3

I I

O5 1

I,,,,1

Figure 1.

AN ILLU`TRATION OF LINEAR FRACTIONS.

varying factors in accordance with complicat-

ed rules. The English system of measurements

is such an inconsistent and inferior system.

Decimal Fractions Since we can extend a

number indefinitely to

the lett to make it bigger, it is a logical step to

extend it towards the right to make it smaller.

Numbers smaller than unity are fractions and

if a displacement one position to the right di-

vides its value by ten, then the number is re-

ferred to as a decimal fraction. Thus a digit

to the right of the units column indicates the

number of tenths, the second digit to the right

represents the number of hundredths, the third,

the number of thousandths, etc. Some distin-

guishing mark must be used to divide unit from

tenths so that one may properly evaluate each

symbol. This mark is the decimal point.

A decimal fraction like four - tenths may be

written .4 or 0.4 as desired, the latter probably

being the clearer. Every time a digit is placed

one space further to the right it represents a

ten times smaller part. This is illustrated in

Figure 1, where each large division represents

a unit; each unit may be divided into ten parts

although in the drawing we have only so di-

vided the first part. The length ab is equal to

seven of these tenth parts and is written as 0.7.

The next smaller divisions, which should be

written in the second column to the right of

the decimal point, are each one -tenth of the

small division, or one one -hundredth each.

They are so small that we can only show them

by imagining a magnifying glass to look at

them, as in Figure 1. Six of these divisions is

to be written as 0.06 (six hundredths). We

need a microscope to see the next smaller divi-

sion, that is those in the third place, which will

be a tenth of one one -hundredth, or a thou-

sandth; four such divisions would be written

as 0.004 (four thousandths).

A

D

0.1

G

0.1

F

0.1 0.1 0.1

0 0 ó 0 ó 0 0

J 0 i 0.1 0 I 0.1

0 _

0

K

c

Figure 2.

IN THIS ILLUSTRATION FRACTIONAL PORTIONS ARE REPRESENTED IN THE

FORM OF RECTANGLES RATHER THAN LINEARLY.

ABCD = 1.0; GFED 0.1; KJEH 0 .01; each small section within KJEN equals 0.001

t

C

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754 Radio Mathematics and Calculations THE RADIO

It should not be thought that such numbers

are merely of academic interest for very small

quantities arc common in radio work.

Possibly the conception of fractions may be

clearer to some students by representing it in

the form of rectangles rather than linearly

(see Figure 2).

Addition When two or more numbers are

to be added we sometimes write

them horizontally with the plus sign between

them. + is the sign or operator indicating ad-

dition. Thus if 7 and 12 are to be added to-

gether we may write 7+12 = 19.

But if larger or more numbers are to be added

together they are almost invariably written one

under another in such a position that the deci-

mal points fall in a vertical line. If a number

has no decimal point, it is still considered as

being just to the right of the units figure; such

a number is a whole number or integer. Ex-

amples:

654 0.654 654

32 3.2 32

53041 53.041 5304.1

53727 56.895 5990.1

The result obtained by adding numbers is

called the rum.

Subtraction Subtraction is the reverse of

addition. Its operator is - (the

minus sign). The number to be subtracted is

called the subtrahend, the number from which

it is subtracted is the minuend, and the result

is called the remainder.

minuend

-subtrahend

Examples: remainder

65.4 65.4

-32 -32.21

33.4 33.19

Multiplication When numbers are to be mul-

tiplied together we use the x

which is known as the multiplication or the

times sign. The number to be multiplied is

known as the multiplicand and that by which

it is to be multiplied is the multiplier, which

may be written in words as follows:

multiplicand

X multiplier

partial product

p r o d u c t

The result of the operation is called the

product.

From the examples to follow it will be obvi-

ous that there are as many partial products

as there are digits in the multiplier. In the fol-

lowing examples note that the righthandmost

digit of each partial product is placed one

space farther to the left than the previous one.

834 834

X 26 X 206

5004 5004

1668 000

1668

21684 171804

In the second example above it will be seen

that the inclusion of the second partial prod-

uct was unnecessary; whenever the multiplier

contains a cipher (zero) the next partial prod-

uct should be moved an additional space to

the left.

Numbers containing decimal fractions may

first be multiplied exactly as if the decimal

point did not occur in the numbers at all; the

position of the decimal point in the product is

determined after all operations have been com-

pleted. It must be so positioned in the product

that the number of digits to its right is equal

to the number of decimal places in the multi-

plicand plus the number of decimal places in

the multiplier.

This rule should be well understood since

many radio calculations contain quantities

which involve very small decimal fractions. In

the examples which follow the explanatory

notations "2 places," etc., are not actually

written down since it is comparatively easy to

determine the decimal point's proper location

mentally.

5.43 2 places

X 0.72 2 places

1086

3 801

3.9096

0.04

X 0.003

0.00012

2+2 =4 places

2 places

3 places

2+3= 5 places

Division Division is the reverse of multi-

plication. Its operator is the =,

which is called the division sign. It is also com-

mon to indicate division by the use of the frac-

tion bar (/) or by writing one number over

the other. The number which is to be divided

is called the dividend and is written before

the division sign or fraction bar or over the

horizontal line indicating a fraction. The num-

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HANDBOOK Division 755

ber by which the dividend is to be divided is

called the divisor and follows the division

sign or fraction bar or comes under the hori-

zontal line of the fraction. The answer or

result is called the quotient.

quotient

divisor dividend

or

Another example: Divide 0.000325 by 0.017.

Here we must move the decimal point three

places to the right in both dividend and di-

visor.

0.019

17 0.325

17

155

153

dividend -divisor

dividend

= quotient

or

= quotient

49

2

In a case where the dividend has fewer deci-

mals than the divisor the same rules still may

be applied by adding ciphers. For example to

divide 0.49 by 0.006 we must move the

decimal point three places to the right. The

0.49 now becomes 490 and we write:

divisor

Examples:

126

834 775373174 49 2436

834 196 81

2168 476 6 laW

1668 441 48

5004 35 remainder 10

5004 6

4

Note that one number often fails to divide

into another evenly. Hence there is often a

quantity left over called the remainder.

The rules for placing the decimal point

are the reverse of those for multiplication.

The number of decimal places in the quotient

is equal to the difference between the number

of decimal places in the dividend and that in

the divisor. It is often simpler and clearer

to remove the decimal point entirely from the

divisor by multiplying both dividend and di-

visor by the necessary factor; that is we move

the decimal point in the divisor as many places

to the right as is necessary to make it a whole

number and then we move the decimal point

in the dividend exactly the same number of

places to the right regardless of whether this

makes the dividend a whole number or not.

When this has been done the decimal point

in the quotient will automatically come di-

rectly above that in the dividend as shown in

the following example.

Example: Divide 10.5084 by 8.34. Move the

decimal point of both dividend and divisor

two places to the right.

1.26

834 1050.84

834

2168

1668

When the division shows a remainder it is

sometimes necessary to continue the work so

as to obtain more figures. In that case ciphers

may be annexed to the dividend, brought down

to the remainder, and the division continued

as long as may be necessary; be sure to place

a decimal point in the dividend before the

ciphers are annexed if the dividend does not

already contain a decimal point. For ex-

ample:

80.33

6 482.00

48

20 18

2

This operation is not very often required

in radio work since the accuracy of the mea-

surements from which our problems start

seldom justifies the use of more than three

significant figures. This point will be cov-

ered further later in this chapter.

Fractions Quantities of less than one

(unity) are called fractions. They

5004 may be expressed by decimal notation as we

5004 have seen, or they may be expressed as vulgar

fractions. Examples of vulgar fractions:

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756 Radio Mathematics and Calculations T H E R A D I O

numerator 3 6 1

denominator 4 7 5

The upper position of a vulgar fraction is

called the numerator and the lower position

the denominator. When the numerator is the

smaller of the two, the fraction is called a

proper fraction; the examples of vulgar frac-

tions given above are proper vulgar fractions.

When the numerator is the larger, the ex-

pression is an improper fraction, which can

be reduced to an integer or whole number

with a proper fraction, the whole being called

a mixed number. In the following examples

improper fractions have been reduced to

their corresponding mixed numbers.

7 3

4 - 1 4 3 = 1

Adding or Subtracting Except when the

Fractions fractions are very

simple it will usual-

ly be found much easier to add and subtract

fractions in the form of decimals. This rule

likewise applies for practically all other oper-

ations with fractions. However, it is occa-

sionally necessary to perform various opera-

tions with vulgar fractions and the rules

should be understood.

When adding or subtracting such fractions

the denominators must be made equal. This

may be done by multiplying both numerator

and denominator of the first fraction by the

denominator of the other fraction, after

which we multiply the numerator and de-

nominator of the second fraction by the de-

nominator of the first fraction. This sounds

more complicated than it usually proves in

practice, as the following examples will show.

1 1_ 1x3 1x2 3 2 S

2+ 3- 2x3+3:2]= 6+ 6= 6

3 2 3xs _2x4 _ 15 0 7

4 ---5-= 4 x 5 5x4- ]- 20 20 = 20

Except in problems involving large numbers

the step shown in brackets above is usually

done in the head and is not written down.

Although in the examples shown above we

have used proper fractions, it is obvious that

the same procedure applies with improper

fractions. In the case of problems involving

mixed numbers it is necessary first to convert

them into improper fractions. Example:

3 2x7+3 17

27 = -7

The numerator of the improper fraction is

equal to the whole number multiplied by the

denominator of the original fraction, to which

the numerator is added. That is in the above

example we multiply 2 by 7 and then add 3

to obtain 17 for the numerator. The denomi-

nator is the same as is the denominator of

the original fraction. In the following ex-

ample we have added two mixed numbers.

3 3_ 17 1s r17x4 15x71

2 7 + 3 4= 7+ 4 =L 7x4 + 4x7

68 105 173 5

2 + 2I _ -28 = 6 2$

Multiplying All vulgar fractions are multi -

Fractions plied by multiplying the nu-

merators together and the de-

nominators together, as shown in the follow-

ing example:

3 2 3x2_ 6 3

4 x 5 - 4x5 ] -20 -10

As above, the step indicated in brackets is

usually not written down since it may easily

be performed mentally. As with addition and

subtraction any mixed numbers should be

first reduced to improper fractions as shown

in the following example:

3 1 3 13 39 13

23 x 4 3 - 23 3- 69 -23

Division of

Fractions

Example:

Fractions may be most easily

divided by inverting the di-

visor and then multiplying.

= 5 x 3=

In the above example it will be seen that to

divide by 3/4 is exactly the same thing as to

multiply by 4/3. Actual division of fractions

is a rather rare operation and if necessary is

usually postponed until the final answer is se-

cured when it is often desired to reduce the

resulting vulgar fraction to a decimal frac-

tion by division. It is more common and

usually results in least overall work to re-

duce vulgar fractions to decimals at the be-

ginning of a problem. Examples:

= 0.375 5 = 0.15625

0.15625

32 5.00000

32

1 80

1 60 200

192

80 64

160

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HANDBOOK Division of Fractions 757

It will be obvious that many vulgar fractions

cannot be reduced to exact decimal equiva-

lents. This fact need not worry us, however,

since the degree of equivalence can always be

as much as the data warrants. For instance,

if we know that one -third of an ampere is

flowing in a given circuit, this can be written

as 0.333 amperes. This is not the exact

equivalent of 1/3 but is close enough since it

shows the value to the nearest thousandth of

an ampere and it is probable that the meter

from which we secured our original data was

not accurate to the nearest thousandth of an

ampere.

Thus in converting vulgar fractions to a

decimal we unhesitatingly stop when we have

reached the number of significant figures war-

ranted by our original data, which is very

seldom more than three places (see section

Signt/icant Figurer later in this chapter).

When the denominator of a vulgar fraction

contains only the factors 2 or 5, division can

be brought to a finish and there will be no

remainder, as shown in the examples above.

When the denominator has other factors

such as 3, ', 11, etc., the division will seldom

come out even no matter how long it is con-

tinued but, as previously stated, this is of

no consequence in practical work since it may

be carried to whatever degree of accuracy is

necessary. The digits in the quotient will

usually repeat either singly or in groups, al-

though there may first occur one or more

digits which do not repeat. Such fractions

are known as repeating fractions. They are

sometimes indicated by an oblique line (frac-

tion bar) through the digit which repeats, or

through the first and last digits of a repeating

group. Example:

t r

= 0.3333 .... =0,3

- = 0.142857142857.... = 0714285 /

The foregoing examples contained only re-

peating digits. In the following example a

non -repeating digit precedes the repeating

digit:

tó =0.2333.... =0.2,

While repeating decimal fractions can be

converted into their vulgar fraction equiva-

lents, this is seldom necessary in practical

work and the rules will be omitted here.

Powers and When a number is to be mul-

Roots tiplied by itself we say that

it is to be squared or to be

rained to the second you er. When it is to be

multipled by itself once again, we say that

it is cubed or faired to the third poet er.

In general terms, when a number is to be

multipled by itself we speak of raising to a

you er or involution; the number of times

which the number is to be multiplied by it-

self is called the order of the power. The

standard notation requires that the order of

the power be indicated by a small number

written after the number and above the line,

called the exponent. Examples:

2' = 2 X 2, or 2 squared, or the second

power of 2

2' = 2 X 2 X 2, or 2 cubed, or the third

power of 2

2' = 2 X 2 X 2 X 2, or the fourth pow-

er of 2

Sometimes it is necessary to perform the

reverse of this operation, that is, it may be

necessary, for instance, to find that number

which multiplied by itself will give a product

of nine. The answer is of course 3. This

process is known as extracting the root or

evolution. The particular example which is

cited would be written:

=3

The sign for extracting the root is yr,

which is known as the radical sign: the order

of the root is indicated by a small number

above the radical as in ` ; which would mean

the fourth root; this number is called the

index. When the radical bears no index, the

square or second root is intended.

Restricting our attention for the moment

to square root, we know that 2 is the square

root of 4, and 3 is the square root of 9. If

we want the square root of a number between

3 and 9, such as the square root of 5, it is

obvious that it must lie between 2 and 3. In

general the square root of such a number can-

not be exactly expressed either by a vulgar

fraction or a decimal fraction. However, the

square root can be carried out decimally as

far as may be necessary for sufficient accur-

acy. In general such a decimal fraction will

contain a never -ending series of digits with-

out repeating groups. Such a number is an

irrational number, such as

= 2.2361 . . . .

The extraction of roots is usually done by

tables or logarithms the use of which will

be described later. There are longhand meth-

ods of extracting various roots, but we shall

give only that for extracting the square root

since the others become so tedious as to make

other methods almost invariably preferable.

Even the longhand method for extracting the

square root will usually be used only if loga-

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758 Radio Mathematics and Calculations THE RADIO

rithm tables, slide rule, or table of roots are

not handy.

Extracting the First divide the number the

Square Root root of which is to be ex-

tracted into groups of two

digits starting at the decimal point and going

in both directions. If the lefthandmost group

proves to have only one digit instead of two,

no harm will be done. The righthandmost

group may be made to have two digits by

annexing a zero if necessary. For example,

let it be required to find the square root of

5678.91. This is to be divided off as follows:

J56' 78.91

The mark used to divide the groups may

be anything convenient, although the prime -

sign (') is most commonly used for the

purpose.

Next find the largest square which is con-

tained in the first group, in this case 56. The

largest square is obviously 49, the square of 7.

Place the 7 above the first group of the num-

ber whose root is to be extracted, which is

sometimes called the dividend from analogy

to ordinary division. Place the square of this

figure, that is 49, under the first group, 56,

and subtract leaving a remainder of 7.

7

56' 78.91

49

7

Bring down the next group and annex it to

the remainder so that we have 778. Now to

the left of this quantity write down twice the

root so far found (2 x 7 or 14 in this ex-

ample), annex a cipher as a trial divisor, and

see how many times the result is contained

in 778. In our example 140 will go into 778

5 times. Replace the cipher with a 5, and

multiply the resulting 145 by 5 to give 725.

Place the 5 directly above the second group

in the dividend and then subtract the 725

from 778.

7 5

J56' 78.91

49

140 7 78

145 X 5 = 7 25

53

The next step is an exact repetition of the

previous step. Bring down the third group

and annex it to the remainder of 53, giving

5391. Write down twice the root already

found and annex the cipher (2 x 75 or 150

plus the cipher, which will give 1500). 1500

will go into 5391 3 times. Replace the last

cipher with a three and multiply 1503 by 3 to

give 4509. Place 3 above the third group.

Subtract to find the remainder of 882. The

quotient 75.3 which has been found so far is

not the exact square root which was desired;

in most cases it will be sufficiently accurate.

However, if greater accuracy is desired groups

of two ciphers can be brought down and the

process carried on as long as necessary.

7 5. 3

V-S-6' 78.91

49

140 7 78

145 x 5 = 7 25

1500 53 91

1503 X 3 = 45 09

8 82

Each digit of the root should be placed di-

rectly above the group of the dividend from

which it was derived; if this is done the

decimal point of the root will come directly

above the decimal point of the dividend.

Sometimes the remainder after a square

has been subtracted (such as the 1 in the fol-

lowing example) will not be sufficiently large

to contain twice the root already found even

after the next group of figures has been

brought down. In this case we write a cipher

above the group just brought down and bring

down another group.

7. 0 8 2

V50.16' 00' 00

49

1400 1 16 00

1408 X 8 = 1 12 64

14160 3 36 00

14162 X 2 = 2 83 24

52 76

In the above example the amount 116 was not

sufficient to contain twice the root already

found with a cipher annexed to it; that is,

it was not sufficient to contain 140. There-

fore we write a zero above 16 and bring down

the next group, which in this example is a

pair of ciphers.

Order of One frequently encounters prob.

Operations lems in which several of the fun-

damental operations of arithme-

tic which have been described are to be per-

formed. The order in which these operations

www.americanradiohistory.com

HANDBOOK Order of Operations 759

must be performed is important. First al: pow-

ers and roots should be calculated; multipli-

cation and division come next; adding and

subtraction come last. In the example

2+3X4'

we must first square the 4 to get 16; then we

multiply 16 by 3, making 48, and to the

product we add 2, giving a result of 50.

If a different order of operations were fol-

lowed, a different result would be obtained.

For instance, if we add 2 to 3 we would ob-

tain 5, and then multiplying this by the square

of 4 or 16, we would obtain a result of 80,

which is incorrect.

In more complicated forms such as frac-

tions whose numerators and denominators may

both be in complicated forms, the numerator

and denominator are first found separately

before the division is made, such as in the

following example:

3X4+5X2_ 12±10 22

-

2 X 3+ 2+ 3- 6+ 2+ 3- 11

Problems of this type are very common in

dealing with circuits containing several in-

ductances, capacities, or resistances.

The order of operations specified above does

not always meet all possible conditions; if a

series of operations should be performed in a

different order, this is always indicated by

pareathr e( or braiketr, for example:

2+3 X4'=2+3 X 16=2+48=50

(2 + 3) X4'=5 X4'=5 X 16 = 80

2 + (3 X 4r= 2 + 12' = 2 + 144 = 146

In connection with the radical sign, brackets

may be used or the "hat" of the radical may

be extended over the entire quantity whose

root is to be extracted. Example:

V4 + 5 = V%4 + 5 = 2 + 5 = 7

V(4 +5) = V4 +5= Vi =3

It is recommended that the radical always be

extended over the quantity whose root is to be

extracted to avoid any ambiguity.

Cancellation In a fraction in which the

numerator and denominator

consist of several factors to be multiplied, con-

siderable labor can often be saved if it is

found that the same factor occurs in both

numerator and denominator. These factors

cancel each other and can he removed. Ex-

ample:

In the foregoing example it is obvious that the

3 in the numerator goes into the 6 in the de-

nominator twice. We may thus cross out

the three and replace the 6 by a 2. The 2

which we have just placed in the denominator

cancels the 2 in the numerator. Next the 5

in the denominator will go into the 25 in the

numerator leaving a result of 5. Now we

have left only a 5 in the numerator and a 7

in the denominator, so our final result is 5/7.

If we had multiplied 2 x 3 x 25 to obtain

150 and then had divided this by 6 x 5 x 7

or 210, we would have obtained the same re-

sult but, with considerably more work.

Algebra

Algebra is not a separate branch of mathe-

matics but is merely a form of generalized

arithmetic in which letters of the alphabet and

occasional other symbols are substituted for

numbers, from which it is often referred to as

literal notation. It is simply a shorthand meth-

od of writing operations which could be spelled

out. The laws of most common electrical phe-

nomena and circuits (including of course ra-

dio phenomena and circuits) lend themselves

particularly well to representation by literal no-

tation and solution by algebraic equations or

formulas.

While we may write a particular problem in

Ohm's Law as an ordinary division or multi-

plication, the general statement of all such

problems calls for the replacement of the num-

bers by symbols. We might be explicit and

write out the names of the units and use these

names as symbols:

volts = amperes X ohms

Such a procedure becomes too clumsy when

the expression is more involved and would be

unusually cumbersome if any operations like

multiplication were required. Therefore as a

short way of writing these generalized rela-

tions the numbers are represented by letters.

Ohm's Law then becomes

E = I X R

In the statement of any particular problem

the significance of the letters is usually indi-

cated directly below the equation or formula

using them unless there can be no ambiguity.

Thus the above form of Ohm's Law would he

more completely written as:

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760 Radio Mathematics and Calculations THE RADIO

E = I X R

where E = e.m.f. in volts

I = current in amperes

R = resistance in ohms

Letters therefore represent numbers, and for

any letter we can read "any number." When

the same letter occurs again in the same ex-

pression we would mentally read "the same

number," and for another letter "another num-

ber of any value."

These letters are connected by the usual op-

erational symbols of arithmetic, +, -, x

and so forth. In algebra, the sign for division

is seldom used, a division being usually written

as a fraction. The multiplication sign, x, is

usually omitted or one may write a period

only. Examples:

2XaXb =tab

2.3.4.5a = 2 X 3 X 4 X 5 X a

In practical applications of algebra, an ex-

pression usually states some physical law and

each letter represents a variable quantity which

is therefore called a variable. A fixed number

in front of such a quantity (by which it is to

be multiplied) is known as the coefficient.

Sometimes the coefficient may be unknown, yet

to be determined; it is then also written as a

letter; k is most commonly used for this pur-

pose.

The Negative In ordinary arithmetic we

Sign seldom work with negative

numbers, although we may

be "short" in a subtraction. In algebra, how-

ever, a number may be either negative or pos-

itive. Such a thing may seem academic but a

negative quantity can have a real existence.

We need only refer to a debt being considered

a negative possession. In electrical work, how-

ever, a result of a problem might be a negative

number of amperes or volts, indicating that the

direction of the current is opposite to the di-

rection chosen as positive. We shall have il-

lustrations of this shortly.

Having established the existence of negative

quantities, we must now learn how to work

with these negative quantities in addition, sub-

traction, multiplication and so forth.

In addition, a negative number added to a

positive number is the same as subtracting a

positive number from it.

7 7

-3 (add) is the same as 3

4 4

or we might write it

(subtract)

7 + (- 3) = 7 - 3 = 4

Similarly, we have:

o + ( -b) =a -b

When a minus sign is in front of an expres-

sion in brackets, this minus sign has the effect

of reversing the signs of every term within the

brackets: - (a - b) = - o + b

- (2a +3b -5c) = -2a -3b +5c

Multiplication. When both the multiplicand

and the multiplier are negative, the product is

positive. When only one (either one) is nega-

tive the product is negative. The four possible

cases are illustrated below:

+ X + _ + + X - _ -

- X + _ - - X - _ +

Division. Since division is but the reverse of

multiplication, similar rules apply for the sign

of the quotient. When both the dividend and

the divisor have the same sign (both negative

or both positive) the quotient is positive. If

they have unlike signs (one positive and one

negative) the quotient is negative.

Pou ers. Even powers of negative numbers

are positive and odd powers are negative. Pow-

ers of positive numbers are always positive.

Examples: - 22 = - 2 X - 2 = + 4

- 2' =- 2X- 2X -2 = +4X

- 2 = - 8

Roots. Since the square of a negative num-

ber is positive and the square of a positive

number is also positive, it follows that a posi-

tive number has two square roots. The square

root of 4 can be either +2 or -2 for ( +2)

x ( +2) = +4 and ( -2) x ( -2) = +4.

Addition and Polynomials are quantities

Subtraction like 3ab' + 4ab' - ?a'b'

which have several terms of

different names. When adding polynomials,

only terms of the same name can be taken to-

gether.

7a' +8ob' +3o'b +3

a' - 5 ab' - b'

8a' +3 ab' +3a'b -b' +3

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HANDBOOK Division 761

Collecting terms. When an expression con-

tains more than one term of the same name,

these can be added together and the expression

made simpler:

5 x' + 2 xy + 3 xy' - 3 x' + 7 xy =

5 x' - 3 x' + 2 xy + 7 xy + 3 xy'-

2x'+ 9 xy + 3 xy'

Multiplication Multiplication of single terms

is indicated simply by writing

them together.

o X b Is written os ob

a X b' is written os ob'

Bracketed quantities are multiplied by a

single term by multiplying each term:

o(b+c+d) = ab + ac + od

When two bracketed quantities are multi-

plied, each term of the first bracketed quantity

is to be multiplied by each term of the second

bracketed quantity, thereby making every pos-

sible combination.

(o + b) )c + d) = ac + ad + bc + bd

In this work particular care must be taken

to get the signs correct. Examples:

(o +b) (a - =a +ab-ab-b'=

a' - b`

(o +b )a+b) =a +ab+ob+b-=

a'+2ab+ b

(o - b) ) o- b) =a - ob -ab+b'-

a'- 2 ob + b

Division It is possible to do longhand divi-

sion in algebra, although it is

some" hat more complicated than in arithme-

tic. However, the division will seldom come

out even, and is not often done in this form.

The method is as follows: Write the terms of

the dividend in the order of descending powers

of one variable and do likewise with the di-

visor. Example:

Divide 501b + 21b + 2a" - 26ab' by

2a - 3b

Write time dividend in the order of descending

powers of a and divide in the same way as in

arithmetic.

o' + 4ob - 7b'

2a - 3b í 2a + 5o b - 26 ab + 21b

2a - 3a b

+ 80'6 - 260b'

+ 8a `b - 12ab'

- 1 4ab' + 21133

- 140122 + 21 b'

Another example: Divide x' - y' by x - y

x- y I x'+ o+ a- y' Z x' + x y +?

x' - x'y

+ x'Y

x'Y -ay'

+ xy' - y'

x y' - y'

Factoring Very often it is necessary to sim-

plify expressions by finding a fac-

tor. This is done by collecting two or more

terms having the same factor and bringing the

factor outside the brackets:

6ab + 3ac = 3a (2b + c)

In a four term expression one can take to-

gether two terms at a time; the intention is to

try getting the terms within the brackets the

same after the factor has been removed:

30ac - 18bc + load - 6bd =

6c )50 - 3b) + 2d 15e - 3b' =

5o - 3b) )6c + 2d

Of course, this is not always possible and

the expression may not have any factors. A

similar process can of course be followed when

the expression has six or eight or any even

number of terms.

A special case is a three -term polynomial,

which can sometimes be factored by writing

the middle term as the sum of two terms:

- 7xy + 12y' may be rewritten as

x-3xy-4xy+ 12y'=

x z - 3y1 - 4y a - 3y1 =

x - 4yi ix - 3y)

The middle term should be split into two

in such a way that the sum of the two new

ternis equals the original middle term and that

their product equals the product of the two

outer terms. In the above example these condi-

tions are fulfilled for - 3xy -- .i xv - - 7xy

and (-- 3xy) ( - -lxy 1 12 x'y'. It is not al-

ways possible to do this and there are then no

simple factors.

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762 Radio Mathematics and Calculations THE RADIO

Working with

Powers

and Roots

When two powers of the

same number are to be mul-

tiplied, the exponents are

added.

a'Xo'=aoXoao=

a' a' o°

b'Xb=b'

c°Xc'=c'=

= a° or

Similarly, dividing of powers is done

subtracting the exponents.

a° = aaa = e or

a' oo a=

ieaI a'

by

b° bbbbb b' or b° -b'' - b'

b' bbb b'

Now we are logically led into some impor-

tant new ways of notation. We have seen that

when dividing, the exponents are subtracted.

This can be continued into negative exponents.

In the following series, we successively divide

by a and since this can now be done in two

ways, the two ways of notation must have the

same meaning and be identical.

a'

o'

a'

a' = e

o° = 1

a_,

_, 1

-72

a_3 1

a°

These examples illustrate two rules: (1) any

number raised to "zero' power equals one or

unity; (2) any quantity raised to a negative

power is the inverse or reciprocal of the same

quantity raised to the same positive power.

n ° =1 a °= á°

Roots. The product of the square root of

two quantities equals the square root of their

product. X VT)= ab

Also, the quotient of two roots is equal to the

root of the quotient.

Note, however, that in addition or subtrac-

tion the square root of the sum or difference is

not the same as the sum or difference of the

square roots.

Thus, V- V 4 = 3- 2= 1

but V9 - 4 = = 2.2361

Likewise V.; + 1is not the same as v'o + b

Roots may be written as fractional powers.

Thus V á may be written as a'' because

Vó X fe=a

and, a l/2 X a1/2 = a"*" = a' = a

Any root may be written in this form

= =b" ./b' =bv'

The same notation is also extended in the

negative direction:

V" = t - t

b' -%bb C - - t

h -

Following the previous rules that exponents

add when powers are multiplied,

NY;X =:a'

but also a" X a" = a"`-

therefore a'" = o'

Powers of powers. When a power is again

raised to a power, the exponents are multi-

plied;

(a')'=o'

la'). a" (b-')'

(b-')-'= b'

This same rule also applies to roots of roots

and also powers of roots and roots of powers

because a root can always be written as a frac-

tional power.

á= for (o)' =

Removing radicals. A root or radical in the

denominator of a fraction makes the expres-

sion difficult to handle. If there must be a rad-

ical it should be located in the numerator

rather than in the denominator. The removal

of the radical from the denominator is done

by multiplying both numerator and denomina-

tor by a quantity which will remove the radi-

cal from the denominator, thus rationalizing it:

1 J

VIi= v á X- a V O

Suppose we have to rationalize

3a In this case we must multiply

+ VS-

numerator and denominator by - V b, the

same terms but with the second having the

opposite sign, so that their product will not

contain a root.

3a 30( f -) 3a( f= NA-)

Ve. +Vi; (a+b)(fa-fl a-b

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HANDBOOK Powers, Roots, Imaginaries 763

Imaginary Since the square of a negative

Numbers number is positive and the

square of a positive number is

also positive, the square root of a negative

number can be neither positive nor negative.

Such a number is said to be imaginary; the

most common such number (\ -1) is often

represented by the letter i in mathematical

work or in electrical work.

-1 = iorj and i'orj'= -1

Imaginary numbers do nut exactly corre-

spond to anything in our experience and it is

best not to try to visualize them. Despite this

fact, their :nterest is much more than academ-

ic, for they are extremely useful in many cal-

culations involving alternating currents.

The square root of any other negative num-

ber may be reduced to a product of two roots,

one positive and one negative. For instance:

V7-57 = V - 1 V57 = i 7

or, in general =i

Since i = 51, the powers of i have the

following values:

i' = -1

i' = - 1 X i = -i

i' = + 1

i'=+1 X i = i

Imaginary numbers are different from either

positive or negative numbers; so in addition or

subtraction they must always be accounted for

separately. Numbers which consist of both real

and imaginary parts are called comp /ex num-

bers. Examples of complex numbers:

3 +4i =3 +4

o + bi = a + bV -1

Since an imaginary number can never be

equal to a real number, it follows that in an

equality like a+ bi = c + di

a must equal c and bi must equal di

Complex numbers are handled in algebra

just like any other expression, considering t

as a known quantity. Whenever powers of i

occur, they can be replaced by the equivalents

given above. This idea of having In one equa-

tion twr' separate sets of quantities which must

he accounted for separately, has found a sym-

bolic application in vector notation. These are

covered later in this chapter.

Equations of the Algebraic expressions usu-

First Degree ally come in the form of

equations, that is, one set

of terms equals another set of terms. The sim-

plest example of this is Ohm's Law:

E = IR

One of the three quantities may be unknown

but if the other two are known, the third can

be found readily by substituting the known

values in the equation. This is very easy if it

is E in the above example that is to be found;

hut suppose we wish to find I while E and R

are given. We must then rearrange the equa-

tion so that / comes to stand alone to the left

of the equality sign. This is known as ro /rung

the equation Jo, I.

Solution of the equation in this case is done

simply by transposing. If two things are equal

then they must still he equal if both are multi-

plied or divided by the same number. Dividing

both sides of the equation by R:

-ER-= R = I or I = E

If it were required to solve the equation for

R, we should divide both sides of the equation

by I. = R or R = E

A little more complicated example is the

equation for the reactance of a condenser:

X= 2n'fC

To solve this equation for C. we may multi-

ply both sides of the equation by C and divide

both sides by X

X x X= nlf x X, or

1

2nfX

This equation is one of those which requires

a good knowledge of the placing of the deci-

mal point when solving. Therefore we give a

few examples: What is the reactance of a 25

µpfd. capacitor at IOou kc.? In tilling in the

given values in the equation we must remem-

ber that the units used are farads, cycles, and

ohms. Hence, we must write 25 µpfd. as 25

millionths of a millionth of a farad or 25 x

10 -12 farad; similarly, 1000 kc. must be con -.

verted to I,uoo,ouo cycles Substituting these

values in the original equation, we have

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764 Radio Mathematics and Calculations THE RADIO

X - 2 x 3.14 x 1,000,000 x 25 x 10-19

X = 1 - 10'

6.28 x 10"x2Sx 10-v - 6.28 x 25

= 6360 ohms

1

A bias resistor of 1000 ohms should be by-

passed, so that at the lowest frequency the re-

actance of the condenser is 1 /10th of that of

the resistor. Assume the lowest frequency to be

50 cycles, then the required capacity should

have a reactance of 100 ohms, at 50 cycles:

It is, however, simpler in this case to use

the elimination method. Multiply both sides of

the first equation by two and add it to the

second equation:

6x + l0y = 14

4x- 10y= 3 add

10x = 17 x = 1.7

Substituting this value of x in the first equa-

tion, we have

C = 1

2x3.14x50x100 farads 5.1 + 5y = 7 .'. 5y = 7

y = 0.38 - 5.1 = 1.9 .'.

C = 6.28 x05000 microfarods

C = 32 pfd.

In the third possible case, it may be that the

frequency is the unknown. This happens for

instance in some tone control problems. Sup-

pose it is required to find the frequency which

makes the reactance of a 0.03 pfd. condenser

equal to 100,000 ohms.

First we must solve the equation for f. This

is done by transposition.

X 1

2nfC

Substituting known values

_ 2nCX

1

f -2x3.14x0.03x10x 100,000cycles

f = -0 -01!!4 cycles = 53 cycles

These equations are known as first degree

equations with one unknown. First degree, be-

cause the unknown occurs only as a first power.

Such an equation always has one possible so-

lution or root if all the other values are known.

If there are two unknowns, a single equa-

tion will not suffice, for there are then an infi-

nite number of possible solutions. In the case

of two unknowns we need two independent

simultaneous equations. An example of this is:

3x + 5y = 7 4x - 10y =3

Required, to find x and y.

This type of work is done either by the sub-

stitution method or by the elimination method.

In the substitution method we might write for

the first equation:

3x= 7- 5y.'.x- 7 -5y 3

(The symbol .. means. therefore or hence).

This value of x can then be substituted for x

in the second equation making it a single equa-

tion with but one unknown, y.

C

Figure 3.

In this simple network the current divides

through the 2000 -ohm and 3000 -ohm resistors.

The current through each may be found by

using two simultaneous linear equations. Note

that the arrows indicate the direction of elec-

tron flow as explained on page 18.

An application of two simultaneous linear

equations will now be given. In Figure 3 a

simple network is shown consisting of three re-

sistances; let it be required to find the currents

I, and I, in the two branches.

The general way in which all such prob-

lems can be solved is to assign directions to

the currents through the various resistances.

When these are chosen wrong it will do no

harm for the result of the equations will then

be negative, showing up the error. In this sim-

ple illustration there is, of course, no such dif-

ficulty.

Next we write the equations for the meshes,

in accordance with Kirchhoff's second law. All

voltage drops in the direction of the curved

arrow are considered positive, the reverse ones

negative. Since there are two unknowns we

write two equations.

1000 (I, + I,1 + 2000 1, = 6

-2000 I, + 3000 I, = 0

Expand the first equation

3000 1, + 1000 I, = 6

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HANDBOOK Quadratic Equations 765

A

Figure 4.

A MORE COMPLICATED PROB-

LEM REQUIRING THE SOLUTION

OF CURRENTS IN A NETWORK.

This problem is similar to that in Figure 3 but

requires the use of three simultaneous linear

equations.

Multiply this equation by 3

9000 11 + 3000 12 = 18

Subtracting the second equation from the first

11000 I, = 18

I, = 18/11000 = 0.00164 amp.

Filling in this value in the second equation

3000 I, = 3.28 I, = 0.00109 amp.

A similar problem but requiring three equa-

tions is shown in Figure 4. This consists of an

unbalanced bridge and the problem is to find

the current in the bridge- branch, L. We again

assign directions to the different currents,

guessing at the one marked L. The voltages

around closed loops ABC [eq. (1) ] and BDC

[eq. (2)) equal zero and are assumed to be

positive in a counterclockwise direction; that

from D to A equals IO volts [eq. (3)).

(11

-1000 1, + 2000 12 - 1000 1, = 0

(2)

-1000 (1, -13) +1000 13 +3000 (Is+ 13) =0

(31

1000 I, + 1000 (I, - I3) - 10 = 0

Expand equations (2) and (3)

(2)

-1000 I, + 3000 13 + 5000 13 = 0

(3)

2000 I, - 1000 13 - 10 = 0

Subtract equation (2) from equation (1)

(a)

-1000 Is - 6000 13 = 0

Multiply the second equation by 2 and add

it to the third equation

(b)

6000 12 + 9000 13 - 10 = 0

Now we have but two equations with two

unknowns.

Multiplying equation (a) by 6 and adding

to equation (b) we have

-27000 13 - 10 = 0

13 = - 10/27000 = -0.00037 amp.

Note that now the solution is negative

which means that we have drawn the arrow

for I, in Figure 4 in the wrong direction. The

current is 0.37 ma. in the other direction.

Second Degree or A somewhat similar

Quadratic Equations problem in radio would

be, if power in watts

and resistance in ohms of a circuit are given,

to find the voltage and the current. Example:

When lighted to normal brilliancy, a 100 watt

lamp has a resistance of 49 ohms; for what

line voltage was the lamp designed and what

current would it take.

Here we have to use the simultaneous equa-

tions:

P =EI and E =IR

Filling in the known values:

P = EI = 100 and E = IR = I X 49

Substitute the second equation into the first

equation

P = El = (I) X I X 49 = 49 12 = 100

-04-11 = i7 = 1.43 amp.

Substituting the found value of L43 amp. for

/ in the first equation, we obtain the value of

the line voltage, 70 volts.

Note that this is a second degree equation

for we finally had the second power of /. Also,

since the current in this problem could only be

positive, the negative square root of 100/49

or - 10/7 was not used. Strictly speaking,

however, there are two more values that sat-

isfy both equations, these are -1.43 and -70.

In general, a second degree equation in one

unknown has two roots, a third degree equa-

tion three roots, etc.

The Quadratic

Equation

general forni

Quadratic or second degree

equations with but one un-

known can be reduced to the

ox'+ bx+c=0

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766 Radio Mathematics and Calculations THE RADIO

where x is the unknown and a, b, and c are

constants.

This type of equation can sometimes be

solved by the method of factoring a three -

term expression as follows:

factoring:

2x' +7x +6 =0

2x' +4x +3x +6 =0

2x(x +2) +3 (x +2) =0

(2x + 3) (x + 2) =

There are two possibilities when a product

is zero. Either the one or the other factor

equals zero. Therefore there are two solutions.

2x,+3=0

2x, = -3

x, = -11/2

x,+2=0

x, = -2

Since factoring is not always easy, the fol-

lowing general solution can usually be em-

ployed; in this equation a, b, and c are the co-

efficients referred to above.

X= - btfb° - 4ac

2a

Applying this method of solution to the pre-

vious example:

x....-7±V49 - 8 X 6_ -7th - -7t 1

4 - 4 4

X,- -7 +1

4 - 1 t/2

X, = -74- 1 = -2

A practical example involving quadratics is

the law of impedance in a.c. circuits. However,

this is a simple kind of quadratic equation

which can be solved readily without the use

of the special formula given above.

Z = 1/ R' + (X1. - Xo )'

This equation can always be solved for R,

by squaring both sides of the equation. It

should now be understood that squaring both

sides of an equation as well as multiplying

both sides with a term containing the unknown

may add a new root. Since we know here that

Z and R are positive, when we square the ex-

pression there is no ambiguity.

Z' =R'+ (X1. -X0)'

and R' = Z' - (XL - Xc)'

or R = V Z' (XL - Xa)'

Also: (XL - Xo)' =Z' -R'

and (XL -X0) =

But here we do not know the sign of the so-

lution unless there are other facts which indi-

cate it. To find either X1. or Xc alone it would

have to be known whether the one or the other

is the larger.

Logarithms

Definition A logarithm is the power (or ex-

end Use ponent) to which we must raise

one number to obtain another.

Although the large numbers used in logarith-

mic work may make them seem difficult or

complicated, in reality the principal use of

logarithms is to simplify calculations which

would otherwise be extremely laborious.

We have seen so far that every operation

in arithmetic can be reversed. If we have the

addition:

o + b = c

we can reverse this operation in two ways. It

may be that b is the unknown, and then we

reverse the equation so that it becomes

e - o = b

It is also possible that we wish to know a, and

that b and c are given. The equation then be-

comes

e - b = o

We call both of these reversed operations sub-

traction, and we make no distinction between

the two possible reverses.

Multiplication can also be reversed in two

manners. In the multiplication

ob=e

we may wish to know a, when b and c are

given, or we may wish to know b when a and

c are given. In both cases we speak of division,

and we make again no distinction between the

two. In the case of powers we can also reverse

the operation in two manners, but now they

are not equivalent. Suppose we have the equa-

tion o°=e

If a is the unknown, and b and c are given,

we may reverse the operation by writing

This operation we call taking the root. But

there is a third possibility: that a and c are

given, and that we wish to know b. In other

www.americanradiohistory.com

HANDBOOK

words, the question is "to which power must

we raise a so as to obtain c ?". This operation

is known as taking the logarithm, and b is the

logarithm of c to the base a. We write this

operation as follows:

log. c = b

Consider a numerical example. We know 2' =8.

We can reverse this operation by asking "to

which power must we raise 2 so as to obtain

82" Therefore, the logarithm of 8 to the base

2 is 3, or log, 8 = 3

Taking any single base, such as 2, we might

write a series of all the powers of the base next

to the series of their logarithms:

Number: 2 4 8 16 32 64 128 256 512 1024

Logarithm: 1 2 3 4 5 6 7 8 9 10

We can expand this table by finding terms

between the terms listed above. For instance,

if we let the logarithms increase with 1/2 each

time, successive terms in the upper series

would have to be multiplied by the square root

of 2. Similarly, if we wish to increase the log-

arithm by 1/10 at each term, the ratio between

two consecutive terms in the upper series

would he the tenth root of 2. Now this short

list of numbers constitutes a small logarithm

table. It should be clear that one could find

the logarithm of any number to the base 2.

This logarithm will usually he a number with

many decimals.

Logarithmic The fact that we chose 2 as

Bases a hase for the illustration is

purely arbitrary. Any base

could be used, and therefore there are many

possible systems of logarithms. In practice we

use only two bases: The most frequently used

base is 10, and the system using this, base is

known as the system of common logarithms,

or Briggs' logarithms. The second system em-

ploys as a base an odd number, designated by

the letter e; e = 2.71828. . . . This is known

as the natural logarithmic system, also as the

Napierian system, and the hyperbolic system.

Although different writers may vary on the

subject, the usual notation is simply log a for

the common logarithm of a. and loge a (or

sometimes In a) for the natural logarithm of

a. VC'e shall use the common logarithmic sys-

tem in most cases, and therefore we shall ex-

amine this system more closely.

Common In the system wherein 10 is the

Logarithms base, the logarithm of 10 equals

1; the logarithm of 100 equals 2,

etc., as shown in the following table:

Logarithms 767

log 10 = log 10' = 1

log 100 = log 10' = 2

log 1,000 = log 10' = 3

log 10,000 = log 10' = 4

log 100,000 = log 10° = 5

log 1,000,000 = log 10° = 6

This table can be extended for numbers less

than 10 when we remember the rules of pow-

ers discussed under the subject of algebra.

Numbers less than unity, too, can be written

as powers of ten.

log 1 = log 10" = 0

log 0.1 = log 10 ' = -1

log 0.01 = log 10 = -2

log 0.001 = log 10 -' = -3

log 0.0001 = log 10-' = -4

From these examples follow several rules:

The logarithm of any number between zero

and + 1 is negative; the logarithm of zero is

minus infinity; the logarithm of a number

greater than + 1 is positive. Negative num-

bers have no logarithm. These rules are true

of common logarithms and of logarithms to

any base.

The logarithm of a number between the

powers of ten is an irrational number, that is,

it has a never ending series of decimals. For in-

stance, the logarithm of 20 must be between

1 and 2 because 20 is between 10 and 100; the

value of the logarithm of 20 is 1.30103. . . .

The part of the logarithm to the left of the

decimal point is called the characteristic, while

the decimals are called the mantissa. In the

case of 1.30103 . ., the logarithm of 20, the

characteristic is 1 and the mantissa is .30103. .

Properties of

Logarithms If the base of our system is

ten, then, by definition of a

logarithm:

10'°` ° =a

or, if the base is raised to the power having an

exponent equal to the logarithm of a number,

the result is that number.

The logarithm of a product is equal to the

sum of the logarithms of the two factors.

log ob = log a + log b

This is easily proved to be true because, it

www.americanradiohistory.com

768 Radio Mathematics and Calculations T H E R A D I O

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www.americanradiohistory.com

HANDBOOK Logarithm Tables 769

was shown before that when multiplying to

powers, the exponents are added; therefore,

O X b = 101og . X H01"' = 10 rlog .. rag n'

Similarly, the logarithm of a quotient is the

difference between the logarithm of the divi-

dend and the logarithm of the divisor.

log -b = log o - log b

This is so because by the same rules of ex-

ponents:

a 1010F. 101 IoF n - log nl

b = ioloR n

We have thus established an easier way of

multiplication and division rince these opera-

tion( have been reduced to adding and rub -

tracting.

The logarithm of a power of a number is

equal to the logarithm of that number, multi-

plied hy the exponent of the power.

log a- = 2 logo and log o' = 3 log o

or, in general:

log a" = n log a

Also, the logarithm of a root of a number

Is equal to the logarithm of that number di-

vided by the index of the root:

log VO _ -t log o

It follows from the rules of multiplication,

that numbers having the sanie digits but dif-

ferent locations for the decimal point, have

logarithms with the same mantissa:

log 829 = 2.918555

log 82.9 = 1.918555

log 8.29 = 0.918555

log 0.829 = -1.918555

log 0.0829 = -2.918555

log 829 = log (8.29 X 100) = log 8.29 +

log 100 = 0.918555 + 2

Logarithm tables give the mantissas of log-

arithms only. The characteristic has to be de-

termined by inspection. The characteristic is

equal to the number of digits to the left of the

decimal point minus one. In the case of loga-

rithms of numbers less than unity, the charac-

teristic is negative and is equal to the number

of ciphers to the right of the decimal point

plue ore.

For reasons of convenience in making up

logarithm tables, it has become the rule that

the mantissa should always be positive. Such

notations above as -1.918555 really mean

( +0.918555 -1) and -2.981555 means

(+ 0.918555 - 2). There are also some other

notations in use such as

1.918555 and Y918555

also 9.918555 - 10 8.918555 - 10

7.918555 - 10, etc.

When, after some addition and subtraction

of logarithms a mantissa should come out neg-

ative, one cannot look up its equivalent num-

ber or anti -logarithm in the table. The man-

tissa must first be made positive by adding and

subtracting an appropriate integral number.

Example: Suppose we find that the logarithm

of a number is - 0.3.4569, then we can trans-

form it into the proper form by adding and

subtracting 1

1 -1

- 0.34569

0.65431 -1 or -1.65431

Using Logarithm Tables

Logarithms are used for calculations involv-

ing multiplication, division, powers, and roots.

Especially when the numbers are large and for

higher, or fractional powers and roots, this be-

comes the most convenient way.

Logarithm tables are available giving the

logarithms to three places, some to four places,

others to five and six places. The table to use

depends on the accuracy required in the result

of our calculations. The four place table,

printed in this chapter, permits the finding of

answers to problems to four significant figures

which is good enough for most constructional

purposes. If greater accuracy is required a five

place table should be consulted. The five place

table is perhaps the most popular of all.

Referring now to the four place table, to

find a common logarithm of a number, pro-

ceed as follows. Suppose the number is 5576.

First, determine the characteristic. An inspec-

tion will show that the characteristic should

be 3. This figure is placed to the left of the

decimal point. The mantissa is now found by

reference to the logarithm table. The first two

numbers are 55; glance down the N column

until coming to these figures. Advance to the

right until coming in line with the column

headed 7; the mantissa will he 7459. (Note that

the column headed 7 corresponds to the third

figure in the number 55,6.) Place the mantissa

7459 to the right of the decimal point, making

the logarithm of 5576 now read 3.,459. Impor-

tant: do not consider the last figure 6 in the

www.americanradiohistory.com

770 Radio Mathematics and Calculations THE RADIO

N L. O' I 2 3 4 5 6 7 8 9 P.P.

250 39 794 811 829 846 863 881 898 915 933 950

251 967 985 002 019 037 054 '071 '088 '106 123 18

252 40 140 157 175 192 209 226 243 261 278 295 11.8

253 312 329 346 364 381 398 415 432 449 466 2 3.6

254 483 500 518 535 552 569 586 603 620 637 3 5.4

4 7.2

255 654 671 688 705 722 739 756 773 790 807 eft.

Figure 6.

A SMALL SECTION OF A FIVE PLACE

LOGARITHM TABLE.

Logarithms may be found with greater accu-

racy with such tables, but they ore only of

use when the accuracy of the original dato

warrants greater precision in the figure work.

Slightly greater accuracy may be obtained for

intermediate points by interpolation, as ex-

plained in the text.

number 5576 when looking for the mantissa

in the accompanying four place tables; in

fact, one may usually disregard all digits be-

yond the first three when determining the man-

tissa. (Interpolation. sometimes used to find a

logarithm more accurately, is unnecessary un-

less warranted by unusual accuracy in the

available data.) However, be doubly sure to

include all figures when ascertaining the mag-

nitude of the characteristic.

To find the anti -logarithm, the table is used

in reverse. As an example, let us find the anti-

logarithm of 1.272 or, in other words, find

the number of which 1.272 is the logarithm.

Look in the table for the mantissa closest to

272. This is found in the first half of the table

and the nearest value is 2718. Write down the

first two significant figures of the anti -loga-

rithm by taking the figures at the beginning of

the line on which 2718 was found. This is 18;

add to this, the digit above the column in

which 2718 was found; this is 7. The anti-log-

arithm is 187 but we have not yet placed the

decimal point. The characteristic is 1, which

means that there should be two digits to the

left of the decimal point. Hence, 18.7 is the

anti -logarithm of 1.272.

For the sake of completeness we shall also

describe the same operation with a five -place

table where interpolation is done by means of

tables of proportional parts (P.P. tables).

Therefore we are reproducing here a small

part of one page of a five -place table.

Finding the logarithm of 0.025013 is done as

follows: We can begin with the characteristic,

which is -2. Next find the first three digits in

the column, headed by N and immediately

after this we see 39, the first two digits of the

mantissa. Then look among the headings of

the other columns for the next digit of the

number, in this case 1. In the column, headed

by i and on the line headed 250, we find the

next three digits of the logarithm, 811. So far,

the logarithm is -2.39811 but this is the loga-

rithm of 0.025010 and we want the logarithm

of 0.025013. Here we can interpolate by ob-

serving that the difference between the log of

0.02501 and 0.02502 is 829 - 811 or 18, in

the last two significant figures. Looking in the

P.P. table marked 18 we find after 3 the num-

ber 5.4 which is to be added to the logarithm.

-2.39811 5.4

-2.39816, the logarithm of 0.025013

Since our table is only good to five places,

we must eliminate the last figure given in the

P.P. table if it is less than 5, otherwise we

must add one to the next to the last figure,

rounding off to a whole number in the P.P.

table.

Finding the anti- logarithm is done the same

way but with the procedure reversed. Suppose

it is required to find the anti -logarithm of

0.40100. Find the first two digits in the column

headed by L. Then one must look for the next

three digits or the ones nearest to it, in the

columns after 40 and on the lines from 40 to

41. Now here we find that numbers in the

neighborhood of 100 occur only with an aster-

isk on the line just before 40 and still after 39.

The asterisk means that instead of the 39 as

the first two digits, these mantissas should

have 40 as the first two digits. The logarithm

0.40100 is between the logs 0.40088 and

0.40106; the anti -logarithm is between 2517

and 2518. The difference between the two

logarithms in the table is again 18 in the

last two figures and our logarithm 0.40100

differs with the lower one 12 in the last

figures. Look in the P.P. table of 18 which

number comes closest to 12. This is found

to be 12.6 for 7 x 1.8 = 12.6. Therefore

we may add the digit 7 to the anti -logarithm

already found; so we have 25177. Next,

place the decimal point according to the rules:

There are as many digits to the left of the

decimal point as indicated in the characteris-

tic plus one. The anti -logarithm of 0.40100 is

2.5177.

In the following examples of the use of log-

arithms we shall use only three places from the

tables printed in this chapter since a greater

degree of precision in our calculations would

not be warranted by the accuracy of the data

given.

In a 375 ohm bias resistor flows a current

of 41.5 milliamperes; how many watts are dis-

sipated by the resistor?

We write the equation for power in watts:

P = PR

www.americanradiohistory.com

HANDBOOK The Decibel 771

and filling in the quantities in question, we

have:

P = 0.0415' X 375

Taking logarithms,

log P = 2 log 0.0415 + log 375

log 0.0415 = -2.618

So 2 X log 0.0415 = -3.236

log 375 = 2.574

log P = -1.810

ontilog = 0.646. Answer = 0.646 watts

Caution: Do not forget that the negative

sign before the characteristic belongs to the

characteristic only and that mantissas are al-

ways positive. Therefore we recommend the

other notation, for it is less likely to lead to

errors. The work is then written:

log 0.0415 = 8.618 -10

2 X log 0.0415 = 17.236 -20 = 7.236-10

log 375 = 2.574

logP =9.810-10

Another example follows which demon-

strates the ease in handling powers and roots.

Assume an all -wave receiver is to be built,

covering from 550 kc. to 60 mc. Can this be

done in five ranges and what will be the re-

quired tuning ratio for each range if no over-

lapping is required? Call the tuning ratio of

one band, x. Then the total tuning ratio for

five such bands is x'. But the total tuning ratio

for all hands is 60/0.55. Therefore:

60 = or :: = ' 60

0.55 N 0.55

Taking logarithms:

10g x log 60 - log 0.55

S

log 60 1.778

log 0.55 -1.740 subtract

2.038

Remember again that the mantissas are posi-

tive and the characteristic alone can be nega-

tive. Subtracting -1 is the same as adding + I.

log x = 2's 9 _ 0.408

x = antilog 0.408 = 2.56

The tuning ratio should be 2.56.

db Power

Ratio

0 1.00

1 1.26

2 1.58

3 2.00

4 2.51

S 3.16

6 3.98

7 5.01

8 6.31

9 7.94

10 10.00

20 100

30 1,000

40 10,000

50 100,000

60 1,000,000

70 10,000,000

80 100,000,000

Figure 7.

A TABLE OF DECIBEL GAINS VERSUS

POWER RATIOS.

The Decibel

The decibel is a unit for the comparison of

power or voltage levels in sound and electrical

work. The sensation of our ears due to sound

waves in the surrounding air is roughly pro-

portional to the logarithm of the energy of the

sound -wave and not proportional to the energy

itself. For this reason a logarithmic unit is used

so as to approach the reaction of the ear.

The decibel represents a ratio of two power

levels, usually connected with gains or loss due

to an amplifier or other network. The decibel

is defined

Nab = 10 log Pi

where P. stands for the output power, P, for

the input power and Nib for the number of

decibels. When the answer is positive, there is

a gain; when the answer is negative, there is

a loss.

The gain of amplifiers is usually given in

decibels. For this purpose both the input power

and output power should be measured. Ex-

ample: Suppose that an intermediate amplifier

is being driven by an input power of 0.2 watt

and after amplification, the output is found to

be 6 watts.

Po a 30

log 30 = 1.48

Therefore the gain is 10 x 1.48 = 14.8

decibels. The decibel is a logarithmic unit;

when the power was multiplied by 30, the

power level in decibels was increased - by 14.8

decibels, or 14.8 decibels added.

www.americanradiohistory.com

772 Radio Mathematics and Calculations THE RADIO

TUBE

GAIN. A STEP -uP

NATIO' 3.5 1

.e

Figure 8.

STAGE GAIN.

The voltage gain in decibels in this stage is

equal to the amplification in the tube plus the

step -up ratio of the transformer, both ex-

pressed in decibels.

When one amplifier is to be followed by

another amplifier, power gains are multiplied

but the decibel gains are added. If a main am-

plifier having a gain of 1,000,000 (power ratio

is 1,000,000) is preceded by a pre -amplifier

with a gain of 1000, the total gain is 1,000,-

000,000. But in decibels, the first amplifier has

a gain of 60 decibels, the second a gain of 30

decibels and the two of them will have a gain

of 90 decibels when connected in cascade.

(This is true only if the two amplifiers are

properly matched at the junction as otherwise

there will be a reflection loss at this point

which must be subtracted from the total.)

Conversion of power ratios to decibels or

vice versa is easy with the small table shown

on these pages. In any case, an ordinary loga-

rithm table will do. Find the logarithm of the

power ratio and multiply by ten to find deci-

bels. Sometimes it is more convenient to figure

decibels f rom voltage or current ratios or gains

rather than from power ratios. This applies

especially to voltage amplifiers. The equation

for this is

Nan = 20 log or 20 log -II",

where the subscript, °, denotes the output volt-

age or current and , the input voltage or cur-

rent. Remember, this equation is true only if

the voltage or current gain in question repre-

sents a power gain which is the square of it

and not if the power gain which results from

this is some other quantity due to impedance

changes. This should be quite clear when we

consider that a matching transformer to con-

nect a speaker to a line or output tube does

not represent a gain or loss; there is a voltage

change and a current change yet the power re-

mains the same for the impedance has changed.

On the other hand, when dealing with volt-

age amplifiers, we can figure the gain in a

stage by finding the voltage ratio from the grid

of the first tube to the grid .of the next tube.

Example: In the circuit of Figure 8, the gain

in the stage is equal to the amplification in the

tube and the step -up ratio of the transformer.

If the amplification in the tube is 10 and the

step -up in the transformer is 3.5, the voltage

gain is 35 and the gain in decibels is:

20 x log 35 = 20 x 1.54 = 30.8 db

Decibels as The original use of the decibel

Power Level was only as a ratio of power

levels -not as an absolute

measure of power. However, one may use the

decibel as such an absolute unit by fixing an

arbitrary "zero" level, and to indicate any

power level by its number of decibels above or

below this arbitrary zero level. This is all very

good so long as we agree on the zero level.

Any power level may then be converted to

decibels by the equation:

Nan = 10 log

where No is the desired power level in deci-

bels, P. the output of the amplifier, Prer the

arbitrary reference level.

The zero level most frequently used (but not

always) is 6 milliwatts or 0.006 watts. For this

zero level, the equation reduces to

Nan = 10 log P°

8 0.006

Example: An amplifier using a 6F6 tube

should be able to deliver an undistorted output

of 3 watts. How much is this in decibels?

P" 3 - 500

-

Prr. .006

10 X log 500 = 10 X 2.70 = 27.0

Therefore the power level at the output of

the 6F6 is 27.0 decibels. When the power level

to be converted is less than 6 milliwatts, the

level is noted as negative. Here we must re-

member all that has been said regarding loga-

rithms of numbers less than unity and the fact

that the characteristic is negative but not the

mantissa.

A preamplifier for a microphone is feeding

1.5 milliwatts into the line going to the regu-

lar speech amplifier. What is this power level

expressed in decibels?

decibels = 10 log P°

0.006

10 log 0.óo6 = 10 log 0.25

Log 0.25 = -1.398 (from table). There-

fore, 10 x -1.398 = (10 x -1 = -10)

+ (10 x .398 = 3.98) ; adding the products

algebraically, gives -6.02 db.

The conversion chart reproduced in this

chapter will be of use in converting decibels to

watts and vice versa.

0.00

www.americanradiohistory.com

HANDBOOK Decibel -Power Conversion 773

40

30

20

IO

-1-40

-50

-60

-70

-80

o

t =MN - .r.RWz

...[.z.Z,...---

M....--

iiit1i

Rtt.t...w!T y_y_

.._.....a.t,.as,- -

es1.- _

EE.- -

....trL i T......

-

_,,.....

I..t.t... ......

.....liaa,

......

- --.i tift....- -

tl.t....lT.r:in/a!

R.....lace -

.... ..i t.

.---

ir:>..

R.t....riarr= =

._._._._..._ -

....... =INN

m.. .rTS1/Jii

t.fiti..tIR11Llbkr.- .

mmilmm.

_..--

=..w!}T71.IIIg1ifJiC

m.m..RL>ai'0.1116 =

Num_u.:

-a-am... - ll'S' .

.i...}7TiI,1------80,....---.

----..s-

__.. _.r

t.elt.t...i , ..ri,ulsu/.a-

----..R1P

I MI

MMI. Ms

19.=.....mol. Il

INIMMIMI..ll.

Rtta.. sltaer:

t.MM..11i\+It61= =

MI M1111111111110

_..--

i=M.iiTT.TÜ,INIGi'yf.

ti=t...Rrlsa -

itit.t...m 1.

_-- it.ii I.t... .e .r.riTr:1SIl - ti

Rtt.t...IR iLLrr..-

.t it... 19 IN

MMt.IIIN..e ee17üI I11

.....R1.1tii

MI MOW

t.t...ttl-twncr.trn - _GEE

t.titit...irdt.la:Jl:.

=:>teiAarli

.t.. -_ aims . MO

9 1 2 3 4

POWER

Bard an .eei reW r are ht

60

50

40

30

20

IO

-20 J W

-70

-80

-90

6

Figure 9.

CONVERSION CHART: POWER TO DECIBELS

Power levels between 6 micromicrowatts and 6000 watts may be referred to corresponding decibel

levels between -90 and 60 db, and vice versa, by means of the above chart. Fifteen ranges are

provided. Each curve begins at the same point where the preceding one ends, enabling uninterrupted

go of the wide db and power ranges with condensed chart. For example: the lowermost curve

ends at -00 db or 60 micromlcrowatts and the next range starts at the same level. Zeno db level is

taken as 6 milliwatts (.006 watt).

www.americanradiohistory.com

774 Radio Mathematics and Calculations THE RADIO

Converting Decibels It is often convenient

to Power to he able to convert a

decibel value to a pow-

er equivalent. The formula used for this oper-

ation is

P = 0.006 X antilog N'ö

where P is the desired level in watts and Nab

the decibels to be converted.

To determine the power level P from a dec-

ibel equivalent, simply divide the decibel value

by 10; then take the number comprising the

antilog and multiply it by 0.006; the product

gives the level in watts.

Note: In problems dealing with the conver-

sion of miner decibels to power, it often hap-

pens that the decibel value -Na, is not

divisible by 10. When this is the case,

N,,n

the numerator in the factor - - -- must be

10

made evenly divisible by 10, the negative signs

must be observed, and the quotient labeled ac-

cordingly.

To make the numerator evenly divisible by

10 proceed as follows: Assume, for example,

that - N,ib is some such value as -38; to

make this figure evenly divisible by 10, we

must add -2 to it, and, since we have added

a negative 2 to it, we must also add a positive

2 so as to keep the net result the same.

Our decibel value now stands, -40 + 2.

Dividing both of these figures by 10, as in the

equation above, we have - 4 and +0.2. Put-

ting the two together we have the logarithm

-4.2 with the negative characteristic and the

positive mantissa as required.

The following examples will show the tech-

nique to be followed in practical problems.

(a) The output of a certain device is rated

at -74 db. What is the power equivalent?

Solution:

NÖ, -74 ( not evenly divisible by 10)

Routine: - 74

- 6 +6

- 80 +6

Nni -80 + 6 8.6

10 - 10 -

ontilog -8.6 = 0.000 000 04

.006 X 0.000 000 04 =

0.000 000 000 24 watt or

240 micro- microwatt

(b) This example differs somewhat from

that of the foregoing one in that the mantissas

are added differently. A low- powered amplifier

has an input signal level of -17.3 db. How

many milliwatts does this value represent?

Solution:

Nan

10

-17.3

- 2.7 + 2.7

- 20

-20

- 10 + 2.7

+ 2.7

-2.27

Antilog -2.27 = 0.0186

0.006 X 0.0186 = 0.000 1 1 16 watt or

0.1116 milliwatt

Input voltages: To determine the required

input voltage, take the peak voltage necessary

to drive the last class A amplifier tube to max-

imum output, and divide this figure by the to-

tal overall voltage gain of the preceding stages.

Computing Specifications: From the preced-

ing explanations the following data can be

computed with any degree of accuracy war-

ranted by the circumstances:

(1) Voltage amplification

(2) Overall gain in db

(3) Output signal level in db

(4) Input signal level in db

(5) Input signal level in watts

(6) Input signal voltage

When a power level is available which must

be brought up to a new power level, the gain

required in the intervening amplifier is equal

to the difference between the two levels in dec-

ibels. If the required input of an amplifier for

full output is -30 decibels and the output

from a device to be used is but -45 decibels,

the pre -amplifier required should have a gain

of the difference, or 15 decibels. Again this is

true only if the two amplifiers are properly

matched and no losses are introduced due to

mismatching.

Push -Pull To double the output of any cas -

Amplifiers cade amplifier, it is only neces-

sary to connect in push -pull the

last amplifying stage, and replace the inter -

stage and output transformers with push -pull

types.

To determine the voltage gain (voltage ra-

tio) of a push -pull amplifier, take the ratio of

one hall of the secondary winding of the push.

pull transformer and multiply it by the a of

one of the output tubes in the push -pull stage;

the product, when doubled, will be the voltage

amplification, or step-up.

Other Units and When working with deci-

Zero Levels bels one should not im-

mediately take for granted

that the zero level is 6 milliwatts for there are

other zero levels in use.

In broadcast stations an entirely new system

is now employed. Measurements made in

www.americanradiohistory.com

HANDBOOK Trigonometry 775

acoustics are now made with the standard zero

level of 10-'° watts per square cm.

Microphones are often rated with reference

to the following zero level: one volt at open

circuit u hen the sound pressure is one millibar.

In any case, the rating of the microphone must

include the loudness of the sound. It is obvious

that this zero level does not lend itself readily

for the calculation of required gain in an am-

plifier.

The VU: So far, the decibel has always re-

ferred to a type of signal which can readily be

measured, that is, a steady signal of a single

frequency. But what would be the power level

of a signal which is constantly varying in vol-

ume and frequency? The measurement of volt-

age would depend on the type of instrument

employed, whether it is measured with a

thermal square law meter or one that shows

average values; also, the inertia of the move-

ment will change its indications at the peaks

and valleys.

After considerable consultation, the broad-

cast chains and the Bell System have agreed

on the VU. The level in VU is the level in

decibels above 1 milltu att zero level and meas-

ured with a carefully defined type of instru-

ment across a 600 ohm line. So long as we

deal with an unvarying sound, the level in VU

is equal to decibels above 1 milliwatt; but

when the sound level varies, the unit is the

VU and the special meter must be used. There

is then no equivalent in decibels.

The Neper: We might have used the natural

logarithm instead of the common logarithm

when defining our logarithmic unit of sound.

This was done in Europe and the unit obtained

is known as the neper or napier. It is still

found in some American literature on filters.

1 neper = 8.686 decibels

1 decibel = 0.1151 neper

AC Meters With Many test instruments

Decibel Scales are now equipped with

scales calibrated in deci-

bels which is very handy when making meas-

urements of frequency characteristics and gain.

These meters are generally calibrated for con-

nection across a 500 ohm line and for a zero

level of 6 milliwatts. When they are connected

across another impedance, the reading on the

meter is no longer correct for the zero level of

6 milliwatts. A correction factor should be

applied consisting in the addition or subtrac-

tion of a steady figure to all readings on the

meter. This figure is given by the equation:

db to be added = 10 log szo

where Z is the impedance of the circuit under

measurement.

SECOND

QUADRANT

FIRST

QUADRANT

THIRD

QUADRANT

FOURTH

QUADRANT

Figure 10.

THE CIRCLE IS DIVIDED INTO

FOUR QUADRANTS BY TWO PER-

PENDICULAR LINES AT RIGHT

ANGLES TO EACH OTHER.

The "northeast" quadrant thus formed is

known os the first quadrant; the others are

numbered consecutively in a counterclockwise

direction.

Trigonometry

Definition Trigonometry is the science of

and Use mensuration of triangles. At first

glance triangles may seem to

have little to do with electrical phenomena;

however, in a.c. work most currents and volt-

ages follow laws equivalent to those of the

various trigonometric relations which we are

about to examine briefly. Examples of their

application to a.c. work will he given in the

section on Vectors.

Angles are measured in degiees or in radi-

ans. The circle has been divided into 360

degrees, each degree into 60 minutes, and each

minute into 60 seconds. A decimal division of

the degree is also in use because it makes cal-

culation easier. Degrees, minutes and seconds

are indicated by the following signs: °, ' and "

Example: 6° 5' 23" means six degrees, five

minutes, twenty -three seconds. In the decimal

notation we simply write 8.47 °, eight and

forty -seven hundredths of a degree.

When a circle is divided into four quadrants

by two perpendicular lines passing through

the center (Figure 10) the angle made by the

two lines is 90 degrees, known as a right angle.

Two right angles, or 180° equals a straight

angle.

The radian: If we take the radius of a circle

and bend it so it can cover a part of the cir-

cumference, the arc it covers subtends an angle

called a radian (Figure 11 ). Since the diam-

eter, of a circle equals 2 times the radius,

there are 27 radians in 360 °. So we have the

following relations:

1 radian =57° 17'45 " = 57.2958° 7= 3.14159

1 degree = 0.01745 radians

yr radians =180 °' /2 radians =90°

7/3 radians =60°

www.americanradiohistory.com

776 Radio Mathematics and Calculations THE RADIO

Figure 11.

THE RADIAN.

A radian is an angle whose arc is exactly equal

to the length of either side. Note that the

angle is constant regardless of the length of

the side and the arc so long as they are equal.

A radian equals 57.2958 .

In trigonometry we consider an angle gen-

erated by two lines, one stationary and the

other rotating as if it were hinged at 0, Figure

12. Angles can be greater than 180 degrees and

even greater than 360 degrees as illustrated in

this figure.

Two angles are complements of each other

when their sumis 90 °, or a right angle. A is

the complement of B and B is the complement

of A when A=(90° -B)

and when

B= (90° -A)

Two angles are supplements of each other

when their sum is equal to'a straight angle, or

180 °. A is the supplement of B and B is the

supplement of A when

A= (180 ° -B)

and

B= (180 ° -A)

In the angle A, Figure 13A, a line is drawn

from P, perpendicular to b. Regardless of the

point selected for P, the ratio a/c will always

be the same for any given angle, A. So will all

the other proportions between a, b, and c re-

main constant regardless of the position of

point P on c. The six possible ratios each are

named and defined as follows:

a

sine A = c

a

tangent A =

C

secant A =

cosine A = b

c

cotangent A =

cosecant A = a

Let us take a special angle as an example.

For instance, let the angle A be 60 degrees as

in Figure 13B. Then the relations between the

sides are as in the figure and the six functions

become:

a

t/2 1(3

sin. 60° _ = = 1/2

cos 60 ° =b c = 1/2 =1/2

a 1/2 V-3

tan 60° = b = 1/2

sec 60° = b

1/2 1

cot 60° = 1/2 - =

1

= =2

1/2

1

csc 60° = a = 1/2

2/j

Another example: Let the angle be 45 °, then

the relations between the lengths of a, b, and

c are as shown in Figure 13C, and the six

functions are:

AC

Figure 12.

AN ANGLE IS GENERATED BY TWO LINES, ONE STATIONARY AND THE OTHER

ROTATING.

The line OX is stationary; the line with the small arrow at the far end rotates in a counterclockwise

direction. At the position illustrated in the lefthandmost section of the drawing it makes an angle,

A, which is less thon 90 and is therefore in the first quadrant. In the position shown in the second

portion of the drawing the angle A has increased to such a value that it now lies in the third

quadrant; note that an angle can be greater than 180 . In the third illustration the angle A is in

the fourth quadrant. In the fourth position the rotating vector has made more than one complete

revolution and is hence in the fifth quadrant; since the fifth quadrant is an exact repetition of the

first quadrant, its values will be the same os in the lefthandmost aortion of the illustration.

www.americanradiohistory.com

H A N D B O O K Trigonometric Relations 777

P

C=e

90"

b=0 O

Figure 13.

THE TRIGONOMETRIC FUNCTIONS.

In the right triangle shown in (A) the side opposite the ongle A is o, while the adjoining sides are

b and c; the trigonometric functions of the ongle A are completely defined by the ratios of the

sides a, b and c. In (B) are shown the lengths of the sides a and b when angle A is 60- and side c

is 1. In (C) angle A is IS ; a and b equal 1, while c equals V-2. In (D) note that c equals a for a

right angle while b equals O.

1

sin 45° _ = 1/, V-2

tan 45° = 1 = 1

sec 45° = V-2-

1

cos 45° _ - = 1z Ari

1

cot 45° = 1 = 1

cosec 45° _ 12

There are some special difficulties when the

angle is zero or 90 degrees. In Figure 13D an

angle of 90 degrees is shown; drawing a line

perpendicular to b from point P makes it fall

on top of c. Therefore in this case a = c and

b = 0. The six ratios are now:

a

sin 90 °= --= 1

tan 90° = a - a

0 b =

c c

sec 90° -=-= co

b

When the angle is zero,

values are then:

sin 0 a -= 0 - 0

c c

cos 90°

cot 90°

b o

= - =0

c C

0

= = o

a

cosec 90° _ -a = 1

a =0 and b =c. The

cos 0° = b

c = 1

tan 0° = b- 0 cot 0° = a 0 CO

Relations Between It follows from the defi-

Functions nitions that

I

sin A = cosec A cos A -

1

and tan A -cot A

1

sec A

From the definitions also follows the relation

cos A =sin (complement of A) =sin (90 ° -A)

because in the right triangle of Figure 15,

cos A =b /c =sin B and B=90° -A or the

complement of A. For the same reason:

cotA = tan (90 ° -A)

csc A = sec

Relations in

Right Triangles

For the same

identities:

tan A = a/b

cot A = b/a

(90 ° -A)

In the right triangle of

Figure 15, sin A =a /c and

by transposition

a =c sin A

reason we have the following

a = b tan A

b = a cot A

In the same triangle we can do the same

functions of the angle B for

c _c

cosec0° = a 0- co

angle, there will be defi-

functions. Conversely,

functions is known, the

have been calculated

functions for

Angle

0

30°

45°

60°

90°

Sin

0

1/2

1/2 13

1

Cos.

1

1/2 \ 3

1/2

0

Tan

0

1/3 V-3-

1

1/3

w

Cot Sec.

m 1

VT 2/3 v,

1 \ri

1/3 \ 2

0 cc

Cosec.

2 V VT

2/3 '

1

c

sec 0° = b = 1

In general, for every

nite values of the six

when any of the six

angle is defined. Tables

the value of the

giving angles.

From the foregoing we can make up a small

table of our own (Figure 14), giving values of

the functions for some common angles.

Figure 14.

Values of trigonometric functions for common

angles in the first quadrant.

www.americanradiohistory.com

778 Radio Mathematics and Calculations THE RADIO

b

Figure 15.

in this figure the sides a, b, and c are used

to define the trigonometric functions of angle

B as well as angle A.

sin B = b/c

cos B = a/c

tan B = b/a

cot B = a/b

b=csin B

a = c cos B

b = a tan B

a = b cot B

Functions of Angles In angles greater than

Greater than 90 degrees, the values

90 Degrees of a and b become neg-

ative on occasion in ac-

cordance with the rules of Cartesian coordi-

nates. When b is measured from 0 towards

the left it is considered negative and similarly,

when a is measured from 0 downwards, it is

negative. Referring to Figure 16, an angle in

the second quadrant (between 90° and 180 °)

has some of its functions negative:

a

sin A = - c = pos.

tan A = a =

-b

sec A = -c

cos A = - - neg.

-b

c =

neg. cot A = --ab neg.

neg. cosec A = c = pos.

a

For an angle in the third quadrant (180° to

270 °), the functions are

-a -b

sin A = b = neg. cos A = = neg.

-a -b

tan A = = pos. cot A = - = pos.

POSITIVE FUNCTIONS

SECOND QUADRANT FIRST QUADRANT

sins, cosec all functions

tan, cot cosine, secant

THIRD QUADRANT FOURTH QUADRANT

Figure 17.

SIGNS OF THE TRIGONOMETRIC

FUNCTIONS.

The functions listed in this diagram are posi-

tive; all other functions are negative.

sec A = = neg. cosec A = - -c a = neg.

And in the fourth quadrant (270° to 360 °):

sin A = - -

c

a =

tan A = b

-a =

c

sec A = -b

b

neg. cos A = = pos.

neg. cot A = -b a = neg.

pos. cosec A =- = neg.

-a

Summarizing, the sign of the functions in

each quadrant can be seen at a glance from

Figure 17, where in each quadrant are written

the names of functions which are positive;

those not mentioned are negative.

SECOND

QUADRANT

THIRD

QUADRANT

tb

FOURTH

QUADRANT

Figure 16.

TRIGONOMETRIC FUNCTIONS IN THE SECOND, THIRD, AND FOURTH QUADRANTS.

The trigonometric functions in these quadrants are similar to first quadrant values, but the

signs of the functions vary as listed in the text and in Figure 17.

www.americanradiohistory.com

HANDBOOK Trigonometric Curves 779

A1

Figure 18.

SINE AND COSINE CURVES.

In (A) we have a sine curve

drawn in Cartesian coordinates.

This is the usual representation

of an alternating current wave

without substantial harmonics. In

(B) we have a cosine wave;

note that it is exactly similar

to a sine wave displaced by

90 or n 2 radians.

(B)

90 180' 270' 360 450 540' 630 720'

2m srr

90" 180" 270" 360" 450" 540 630° 720'

-- -- - C

_ - - - -

S _ - Ado

o1m = = = - - - Now =_ LI

7r

"37r Irr

2

Graphs of Trigono- The sine u ate. When

metric Functions we have the relation

y= sin x. where x is an

angle measured in radians or degrees, we can

draw a curve of y versus x for all values of

the independent variable, and thus get a good

conception how the sine varies with the mag-

nitude of the angle. This has been done in

Figure 18A. We can learn from this curve the

following facts.

1. The sine varies between +1 and -1

2. It is a periodic curve, repeating itself after

every multiple of 27 or 360°

3. Sin x = sin (180 ° -x) or sin (ir -x)

4. Sin x = -sin (180° + x), or

-sin (R + x)

The cosine :cave. Making a curve for the

function y = cos x, we obtain a curve similar

to that for y = sin x except that it is displaced

by 90° or 7/2 radians with respect to the

Y -axis. This curve (Figure 18B) is also peri-

odic but it does not start with zero. We read

from the curve:

1. The value of the cosine never goes be-

yond +1 or -1

2. The curve repeats, after every multiple

of 27 radians or 360°

3. Cos x = -cos (180° -x) or

-cos (7 -x)

4. Cos x r cos (360 ° -x) or cos (tir -x)

The graph of the tangent is illustrated in

Figure 19. This is a discontinuous curve and

illustrates well how the tangent increases from

zero to infinity when the angle increases from

zero to 90 degrees. Then when the angle is

further increased, the tangent starts from

minus infinity going to zero in the second quad-

rant, and to infinity again in the third quadrant.

1. The tangent can have any value between

+m and -

2. The curve repeats and the period is err

radians or 180 °, not 2,r radians

3. Tan x = tan (180° +x) or tan (-rr +x)

4. Tan x = -tan (180° -x) or

-tan (- -x)

The graph of the cotangent is the inverse of

that of the tangent, see Figure 20. It leads us

to the following conclusions:

1. The cotangent can have any value be-

tween + m and -

2. It is a periodic curve, the period being

radians or 180°

3. Cot x cot (180° +x) or cot (^.r +x)

4. Cot x = - cot (180° -x) or

-cot (r -x)

z.r

0° 180° 360° 540° 720'

90' 270° 450 630°

Figure 19.

TANGENT CURVES.

The tangent curve increases from 0 to m with

an angular increase of 90 °. In the next 180'

it increases from -m to + s-

90° 270' 450° 630'

0° 180° 360 540° 720'

en"

Figure 20.

COTANGENT CURVES.

Cotangent curves are the inverse of the tan-

gent curves. They vary from + m to -m in

each pair of quadrants.

www.americanradiohistory.com

780 Rodio Mathematics and Calculations THE RADIO

COSINE

Figure 21.

ANOTHER REPRESENTATION OF

TRIGONOMETRIC FUNCTIONS.

If the radius of a circle is considered as the

unit of measurement, then the lengths of the

various lines shown in this diagram are numer-

ically equal to the functions marked adjacent

to them.

The graphs of the secant and cosecant are

of lesser importance and will not be shown

here. They are the inverse, respectively, of the

cosine and the sine, and therefore they vary

from +1 to infinity and from -1 to - infinity.

Perhaps another useful way of visualizing

the values of the functions is by considering

Figure 21. If the radius of the circle is the unit

of measurement then the lengths of the lines

are equal to the functions marked on them.

Trigonometric Tables There are two kinds of

trigonometric tables.

The first type gives the functions of the angles,

the second the logarithms' of the functions.

The first kind is also known as the table of

natural trigonometric functions.

These tables give the functions of all angles

between 0 and 45 °. This is all that is necessary

for the function of an angle between 45° and

90° can always be written as the co- function

of an angle below 45 °. Example: If we had to

find the sine of 48 °, we might write

sin 48° = cos (90° -48 °) = cos 42°

Tables of the logarithms of trigonometric

functions give the common logarithms (log.)

of these functions. Since many of these logar-

ithms have negative characteristics, one should

add -10 to all logarithms in the table which

have a characteristic of 6 or higher. For in-

stance, the log sin 24° = 9.60931 -10. Log

tan 1° = 8.24192 -10 but log cot 1° =

1.75808. When the characteristic shown is less

than 6, it is supposed to be positive and one

should not add -10.

Vectors

A ¡calar quantity has magnitude only; a

vector quantity has both magnitude and direc-

tion. When we speak of a speed of 50 miles

per hour, we are using a scalar quantity, but

when we say the wind is Northeast and has a

Figure 22.

Vectors may be added as shown in these

sketches. In each case the long vector repre-

sents the vector sum of the smaller vectors.

For many engineering applications sufficient

accuracy can be obtained by this method

which avoids long and laborious calculations.

velocity of 50 miles per hour, we speak of a

vector quantity.

Vectors, representing forces, speeds, dis-

placements, etc., are represented by arrows.

They can be added graphically by well known

methods illustrated in Figure 22. We can make

the parallelogram of forces or we can simply

draw a triangle. The addition of many vectors

can be accomplished graphically as in the same

figure.

In order that we may define vectors algebra-

ically and add, subtract, multiply, or divide

them, we must have a logical notation system

that lends itself to these operations. For this

purpose vectors can be defined by coordinate

systems. Both the Cartesian and the polar co-

ordinates are in use.

Vectors Defined Since we have seen how the

by Cartesian sum of two vectors is ob-

Coordinates tained, it follows from Fig-

ure 23, that the vector Z

equals the sum of the two vectors x and y. In

fact, any vector can be resolved into vectors

along the X- and Y -axis. For convenience in

working with these quantities we need to dis-

yS

o

3

YA5

Figure 23.

RESOLUTION OF VECTORS.

Any vector such as i may be resolved Into

two vectors, x and y, along the X- and Y-

axes. If vectors are to be added, their respec-

tive z and y components may be added to

find the x and y components of the resultant

vector.

www.americanradiohistory.com

HANDBOOK Vectors 781

Figure 24.

ADDITION OR SUBTRACTION OF

VECTORS.

Vectors may be added or subtracted by

adding or subtracting their x or y com-

ponents separately.

tinguish between the X- and Y- component,

and so it has been agreed that the Y- compo-

nent alone shall be marked with the letter j.

Example (Figure 23) :

Z =3 +4j

Note again that the sign of components

along the X -axis is positive when measured

from 0 to the right and negative when meas-

ured from O towards the left. Also, the compo-

nent along the Y -axis is positive when meas-

ured from 0 upwards, and negative when

measured from 0 downwards. So the vector,

R, is described as

R =5 -3j

Vector quantities are usually indicated by

some special typography, especially by using a

point over the letter indicating the vector, as

R.

Absolute Value The absolute or scalar

of a Vector value of vectors such as 2

or R in Figure 23 is easily

found by the theorem of Pythagoras, which

states that in any right -angled triangle the

square of the side opposite the right angle is

equal to the sum of the squares of the sides

adjoining the right angle. In Figure 23, OAB

is a right -angled triangle; therefore, the square

of OB (or Z) is equal to the square of OA

(or x) plus the square of AB (or y). Thus the

absolute values of Z and R may be determined

as follows:

IZI= Vx' +y'

ZI= V3' +4' =5

RI= b +3'= 34 =5.83

The vertical lines indicate that the absolute

or scalar value is meant without regard to sign

or direction.

Addition of Vectors An examination of Fig-

ure 24 will show that

the two vectors

=x,+jY,

Z = x: + j Y=

can be added, if we add the X- components

and the Y- components separately.

R + Z = x, + x: + j (y, + Y=)

For the same reason we can carry out sub-

traction by subtracting the horizontal compo-

nents and subtracting the vertical components

R -Z = x, - x: + j (y, - y2)

Let us consider the operator j. If we have a

vector a along the X -axis and add a j in front

of it (multiplying by j) the result is that the

direction of the vector is rotated forward 90

degrees. If we do this twice (multiplying by

11) the vector is rotated forward by 180 degrees

and now has the value -a. Therefore multi-

plying by f is equivalent to multiplying by -1.

Then

12 = -1 and j = V -1

This is the imaginary number discussed be-

fore under algebra. In electrical engineering

the letter j is used rather than i, because i is

already known as the symbol for current.

Multiplying Vectors When two vectors are

to be multiplied we can

perform the operation just as in algebra, re-

membering that j2 _ -1.

RZ= (x, +j',) (x : +jy2)

=x,x2 4- ¡XI Y:4-jx_yt+j'y,ya

= x, x: - y, Y= + j (x, Ya + x: y,)

Division has to be carried out so as to re-

move the j -term from the denominator. This

can be done by multiplying both denominator

and numerator by a quantity which will elimi-

nate j from the denominator. Example:

R x, +jy,_(x, +jy,) (x: -jxa)

Z x: + hY: (x: + jY:) (x: - jy:)

x,x: + YiY= + j (x:y, - x,y:)

x:' + Ya'

Polar Coordinates A vector can also be de-

fined in polar coordi-

nates by its magnitude and its vectorial angle

with an arbitrary reference axis. In Figure 25

www.americanradiohistory.com

782 Radio Mathematics and Calculations THE RADIO

X

Figure 25.

IN THIS FIGURE A VECTOR HAS

BEEN REPRESENTED IN POLAR

INSTEAD OF CARTESIAN CO-

ORDINATES.

In polar coordinates a vector is defined by

a magnitude and an angle, called the vec-

torial angle, instead of by two magnitudes

as in Cartesian coordinates.

the vector Z has a magnitude 50 and a vector-

ial angle of 60 degrees. This will then be

written i = 5OL6O°

A vector a + jb can be transformed into

polar notation very simply (see Figure 26)

=o +jb= Vo'+ b'Lton'u

In this connection tan -' means the angle of

which the tangent is. Sometimes the notation

arc tan b/a is used. Both have the same mean-

ing. A polar notation of a vector can be trans-

formed into a Cartesian coordinate notation in

the following manner (Figure 27)

i = pLA = p cos A + jp sin A

A sinusoidally alternating voltage or cur-

rent is symbolically represented by a rotating

vector, having a magnitude equal to the peak

voltage or current and rotating with an angular

velocity of 2-,rf radians per second or as many

revolutions per second as there are cycles per

second.

The instantaneous voltage, e, is always equal

to the sine of the vectorial angle of this rotat-

ing vector, multiplied by its magnitude.

e = E sin 2^rft

The alternating voltage therefore varies with

time as the sine varies with the angle. If we

plot time horizontally and instantaneous volt-

age vertically we will get a curve like those

in Figure 18.

In alternating current circuits, the current

Figure 26.

Vectors can be trans formed from Cartesian

into polar notation as shown in this figure.

which flows due to the alternating voltage is

not necessarily in step with it. The rotating

current vector may be ahead or behind the

voltage vector, having a phase difference with

it. For convenience we draw these vectors as

if they were standing still, so that we can indi-

cate the difference in phase or the phase angle.

In Figure 28 the current lags behind the volt-

age by the angle e, or we might say that the

voltage leads the current by the angle B.

Vector diagrams show the phase relations

between two or more vectors (voltages and

currents) in a circuit. They may be added and

subtracted as described; one may add a voltage

vector to another voltage vector or a current

vector to a current vector but not a current

vector to a voltage vector (for the same reason

that one cannot add a force to a speed). Figure

28 illustrates the relations in the simple series

circuit of a coil and resistor. We know that the

current passing through coil and resistor must

he the same and in the same phase, so we draw

this current I along the X -axis. We know also

that the voltage drop IR across the resistor is

in phase with the current, so the vector IR rep-

resenting the voltage drop is also along the

X -axis.

The voltage across the coil is 90 degrees

ahead of the current through it; /X must

therefore be drawn along the Y -axis. E the

applied voltage must be equal to the vectorial

sum of the two voltage drops, IR and IX, and

we have so constructed it in the drawing. Now

expressing the same in algebraic notation, we

have =IR+jIX

IZ = IR + jIX

Dividing by I i =R +jX

Due to the fact that a reactance rotates the

voltage vector ahead or behind the current

vector by 90 degrees, we must mark it with a

j in vector notation. Inductive reactance will

have a plus sign because it shifts the voltage

vector forwards; a capacitive reactance is neg-

www.americanradiohistory.com

HANDBOOK Graphical Representation 783

p COS A

Figure 27.

Vectors can be trans formed from polar into

Cartesian notation as shown in this figure.

ative because the voltage will lag behind the

current. Therefore:

X,. = + j 2 -fL

X =_j 1

2nfC

In Figure 28 the angle 8 is known as the

phase angle between E and I. When calculat-

ing power, only the real components count.

The power in the circuit is then

P = I OR)

but IR = E cos 8

P= EIcost)

This coi 8 is known as the power factor of

the circuit. In many circuits we strive to keep

the angle e as small as possible, making cos e

as near to unity as possible. In tuned circuits,

we use reactances which should have as low a

power factor as possible. The merit of a coil

or condenser, its Q. is defined by the tangent of

this phase angle:

Q = tan 8 = X/R

For an efficient coil or condenser, Q should

be as large as possible; the phase -angle should

then be as close to 90 degrees as possible, mak-

ing the power factor nearly zero. Q is almost

but not quite the inverse of cos e. Note that in

Figure 29

Q =X /R and cos e = R/Z

When Q is more than 5, the power factor is

less than 20%; we can then safely say Q =

1 /cos e with a maximum error of about 21 /2

percent, for in the worst case, when cos 8 =

0.2, Q will equal tan e = 4.89. For higher

values of Q, the error becomes less.

Note that from Figure 29 can be seen the

simple relation: -R+jX,

IZI= R'+Xt'

Figure 28.

VECTOR REPRESENTATION OF A

SIMPLE SERIES CIRCUIT.

The righthand portion of the illustration shows

the vectors representing the voltage drops in

the coil and resistance illustrated at the left.

Note that the voltage drop across the coil Xi.

leads that across the resistance by 90 °.

Graphical Representation

Formulas and physical laws are often pre-

sented in graphical form; this gives us a

"bird's eye view" of various possible conditions

due to the variations of the quantities involved.

In some cases graphs permit us to solve equa-

tions with greater ease than ordinary algebra.

Coordinate Systems All of us have used co-

ordinate systems with-

out realizing it. For instance, in modern cities

we have numbered streets and numbered ave-

nues. By this means we can define the location

of any spot in the city if the nearest street

crossings are named. This is nothing but an

application of Cartesian coordinates.

In the Cartesian coordinate system (named

after Descartes), we define the location of any

point in a plane by giving its distance from

each of two perpendicular lines or axes. Figure

30 illustrates this idea. The vertical axis is

called the Y -axis, the horizontal axis is the

X -axis. The intersection of these two axes is

called the origin, O. The location of a point,

P, (Figure 30) is defined by measuring the

respective distances, x and y along the X -axis

and the Y -axis. In this example the distance

along the X -axis is 2 units and along the Y-

axis is 3 units. Thus we define the point as

0= TAN e.

POWER FACTOR =COS e- -

Figure 29.

The figure of merit of a coil and its resistance

is represented by the ratio of the inductive

reactance to the resistance, which as shown

in this diagram is equal to R' which equals

tan 9. For large values of 0 (the phase angle)

this is approximately equal to the reciprocal

of the cos O.

www.americanradiohistory.com

784 Radio Mathematics and Calculations THE RADIO

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Figure 30.

CARTESIAN COORDINATES.

The location of any point can be defined by

its distance from the X and Y axes.

P 2, 3 or we might say x = 2 and y = 3. The

measurement x is called the abscissa of the

point and the distance y is called its ordinate.

It is arbitrarily agreed that distances measured

from 0 to the right along the X -axis shall be

reckoned positive and to the left negative. Dis-

tances measured along the Y -axis are positive

when measured upwards from 0 and negative

when measured downwards from 0. This is

illustrated in Figure 30. The two axes divide

the plane area into four parts called quadrants.

These four quadrants are numbered as shown

in the figure.

It follows from the foregoing statements,

that points lying within the first quadrant have

both x and y positive, as is the case with the

point P. A point in the second quadrant has a

negative abscissa, x, and a positive ordinate, y.

This is illustrated by the point Q, which has

the coordinates x = -4 and y = +1. Points

in the third quadrant have both x and y nega-

tive. x = -5 and y = -2 illustrates such a

point, R. The point S, in the fourth quadrant

has a negative ordinate, y and a positive ab-

scissa or x.

In practical applications we might draw

only as much of this plane as needed to illus-

trate our equation and therefore, the scales

along the X -axis and Y -axis might not start

with zero and may show only that part of the

scale which interests us.

Representation of In the equation:

Functions 300,000

2000

1800

1600

1400

1200

1000

ROO

600 100 200 300 400 600

Figure 31.

REPRESENTATION OF A SIMPLE

FUNCTION IN CARTESIAN CO-

ORDINATES. 300,000

In this chart of the function fk, = mr

distances along the X axis represent wave-

length in meters, while those along the Y

axis represent frequency in kilocycles. A curve

such as this helps to find values between

those calculated with sufficient accuracy for

most purposes.

f is said to be a function of X. For every value

of f there is a definite value of X. A variable is

said to be a function of another variable when

for every possible value of the latter, or inde-

pendent variable, there is a definite value of

the first or dependent variable. For instance,

if y = 5x', y is a function of x and x is called

the independent variable. When a = 3b' + 513'

-25b + 6 then a is a function of b.

A function can be illustrated in our coordi-

nate system as follows. Let us take the equa-

tion for frequency versus wavelength as an

example. Given different values to the inde-

pendent variable find the corresponding values

of the dependent variable. Then plot the points

represented by the different sets of two values.

600

800

1000

1200

1400

1600

1800

2000

500

375

300

250

214

187

167

150

Plotting these points in Figure 31 and draw-

ing a smooth curve through them gives us the

curve or graph of the equation. This curve

will help us find values of f for other values

of 7, (those in between the points calculated)

and so a curve of an often -used equation may

serve better than a table which always has

gaps. When using the coordinate system described

so far and when measuring linearly along both

axes, there are some definite rules regarding

www.americanradiohistory.com

HANDBOOK Representation of Functions 785

Figure 32.

Only two points are needed to define func-

tions which result in a straight line as shown

in this diagram representing Ohm's Law.

the kind of curve we get for any type of

equation. In fact, an expert can draw the curve

with but a very few plotted points since the

equation has told him what kind of curve to

expect.

First, when the equation can be reduced to

the form y = mx + b, where x and y are the

variables, it is known as a linear or first degree

function and the curve becomes a straight line.

(Mathematicians still speak of a "curve" when

it has become a straight line.)

When the equation is of the second degree,

that is, when it contains terms like x' or y'

or Ay. the graph belongs to a group of curves,

called conic sections. These include the circle,

the ellipse, the parabola and the hyperbola. In

the example given above, our equation is of

the form

xy = c, c being equal to 300,000

which is a second degree equation and in this

case, the graph is a hyperbola.

This type of curve does not lend itself read-

ily for the purpose of calculation except near

the middle, because at the ends a very large

change in represents a small change in f and

vice versa. Before discussing what can be done

about this let us look at some other types of

curves.

Suppose we have a resistance of 2 ohms

and we plot the function represented by Ohm's

Law: E = 21. Measuring E along the X -axis

and amperes along the Y -axis, we plot the

necessary points. Since this is a first degree

equation, of the form y = mx + b (for E -

y, m = 2 and / = x and b = 0) it will be a

straight line so we need only two points to

plot it.

(line passes through origin)

I E

0 0

5 10

The line is shown in Figure 32. It is seen to

be a straight line passing through the origin.

Figure 33.

A TYPICAL GRID - VOLTAGE

PLATE -CURRENT CHARACTER-

ISTIC CURVE.

The equation represented by such a curve Is

so complicated that we do not use it. Data

for such a curve is obtained experimentally,

and intermediate values can be found with

sufficient accuracy from the curve.

If the resistance were 4 ohms, we should get

the equation E = 4I and this also represents

a line which we can plot in the same figure.

As we see, this line also passes through the

origin but has a different slope. In this illus-

tration the slope defines the resistance and we

could make a protractor which would convert

the angle into ohms. This fact may seem incon-

sequential now, but use of this is made in the

drawing of loadlines on tube curves.

Figure 33 shows a typical, grid -voltage,

plate -current static characteristic of a triode.

The equation represented by this curve is

rather complicated so that we prefer to deal

with the curve. Note that this curve extends

through the first and second quadrant.

Families of curves. It has been explained

that curves in a plane can be made to illustrate

the relation between to o variables when one

of them varies independently. However, what

are we going to do when there are three vari-

ables and two of them vary independently. It

is possible to use three dimensions and three

axes but this is not conveniently done. Instead

of this we may use a family of curves. We

have already illustrated this partly with Ohm's

Law. If we wish to make a chart which will

show the current through any resistance with

any voltage applied across it, we must take the

equation E = IR, having three variables.

We can now draw one line representing a

resistance of 1 ohm, another line representing

2 ohms, another representing 3 ohms, etc., or

as many as we wish and the size of our paper

will allow. The whole set of lines is then

applicable to any case of Ohm's Law falling

within the range of the chart. If any two of

the three quantities are given, the third can be

found.

www.americanradiohistory.com

786 Radio Mathematics and Calculations THE RADIO

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A FAMILY OF CURVES.

An equation such as Ohm's Law has three

'variables, but can be represented in Cartesian

coordinates by a family of curves such as

shown here. If any two quantities are given,

the third can be found. Any point in the

chart represents o definite value each of E,

1, and R, which will satisfy the equation of

Ohm's Low. Values of R not situated on an R

line can be found by interpolation.

Figure 34 shows such a family of curves to

solve Ohm's Law. Any point in the chart rep-

resents a definite value each of E, 1, and R

which will satisfy the equation. The value of

R represented by a point that is not situated

on an R line can be found by interpolation.

It is even possible to draw on the same chart

a second family of curves, representing a

fourth variable. But this is not always possible,

for among the four variables there should be

no more than two independent variables. In

our example such a set of lines could represent

power in watts; we have drawn only two of

these but there could of course be as many as

desired. A single point in the plane now indi-

cates the four values of E, I, R, and P which

belong together and the knowledge of any

two of them will give us the other two by

reference to the chart.

Another example of a family of curves is

the dynamic transfer characteristic or plate

family of a tube. Such a chart consists of sev-

eral curves showing the relation between plate

voltage, plate current, and grid bias of a tube.

Since we have again three variables, we must

show several curves, each curve for a fixed

value of one of the variables. It is customary

to plot plate voltage along the X -axis, plate

current along the Y -axis, and to make different

curves for various values of grid bias. Such a

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"PLATE" CURVES FOR A

TYPICAL VACUUM TUBE.

In such we have three variables, plate

voltage, plate current, and grid bias. Each

point on a grid bias line corresponds to the

plate voltage and plate current represented

by its position with respect to the X and Y

axes. Those for other values of grid bias may

be found by interpolation. The loadline shown

in the lower left portion of the chart is ex-

plained in the text.

set of curves is illustrated in Figure 35. Each

point in the plane is defined by three values,

which belong together, plate voltage, plate

current, and grid voltage.

Now consider the diagram of a resistance -

coupled amplifier in Figure 36. Starting with

the B- supply voltage, we know that whatever

plate current flows must pass through the

resistor and will conform to Ohm's Law. The

voltage drop across the resistor is subtracted

from the plate supply voltage and the remain-

der is the actual voltage at the plate, the kind

that is plotted along the X -axis in Figure 35.

We can now plot on the plate family of the

+e

Figure 36.

PARTIAL DIAGRAM OF A RESIS-

TANCE COUPLED AMPLIFIER.

The portion of the supply voltage wasted

across the 50,000 -ohm resistor Is represented

in Figure 35 as the loadline.

www.americanradiohistory.com

HANDBOOK Logarithmic Scales 787

tube the loadline, that is the line showing

which part of the plate supply voltage is across

the resistor and which part across the tube for

any value of plate current. In our example, let

us suppose the plate resistor is 50,000 ohms.

Then, if the plate current were zero, the volt-

age drop across the resistor would be zero and

the full plate supply voltage is across the tube.

Our first point of the loadline is E = 250,

1 = 0. Next, suppose, the plate current were

1 ma., then the voltage drop across the resistor

would be 50 volts, which would leave for the

tube 200 volts. The second point of the load-

line is then E = 200, / = 1. We can continue

like this but it is unnecessary for we shall find

that it is a straight line and two points are

sufficient to determine it.

This loadline shows at a glance what hap-

pens when the grid -bias is changed. Although

there are many possible combinations of plate

voltage, plate current, and grid bias, we are

now restricted to points along this line as long

as the 50,000 ohm plate resistor is in use. This

line therefore shows the voltage drop across

the tube as well as the voltage drop across the

load for every value of grid bias. Therefore, if

we know how much the grid bias varies, we

can calculate the amount of variation in the

plate voltage and plate current, the amplifi-

cation, the power output, and the distortion.

Logarithmic Scales Sometimes it is conven-

ient to measure along

the axes the logarithm) of our variable quan-

tities. Instead of actually calculating the logar-

ithm, special paper is available with logarith-

mic scales, that is, the distances measured

along the axes are proportional to the logar-

ithms of the numbers marked on them rather

than to the numbers themselves.

There is semi -logarithmic paper, having

logarithmic scales along one axis only, the

other scale being linear. We also have full

logarithmic paper where both axes carry log-

arithmic scales. Many curves are greatly sim-

plified and some become straight lines when

plotted on this paper.

As an example let us take the wavelength -

frequency relation, charted before on straight

cross- section paper.

f 300,000

X

Taking logarithms:

log f = log 300,000 - log X

If we plot log f along the Y -axis and log X

along the X -axis, the curve becomes a straight

line. Figure 37 illustrates this graph on full

logarithmic paper. The graph may be read

with the same accuracy at any point in con-

z

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1500

1000

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Figure 37.

A LOGARITHMIC CURVE.

Many functions become greatly simplified and

some become straight lines when plotted to

logarithmic scales such as shown in this

diagram. Here the frequency versus wavelength

curve of Figure 31 has been replotted to con-

form with logarithmic axes. Note that it is

only necessary to calculate two points in

order to determine the "curve" since this type

of function results in a straight line.

tract to the graph made with linear coordi-

nates.

This last fact is a great advantage of logar-

ithmic scales in general. It should be clear that

if we have a linear scale with 100 small divi-

sions numbered from 1 to 100, and if we are

able to read to one tenth of a division, the

possible error we can make near 100, way up

the scale, is only 1/10th of a percent. But near

the beginning of the scale, near 1, one tenth of

a division amounts to 10 percent of 1 and we

are making a 10 percent error.

In any logarithmic scale, our possible error

in measurement or reading might be, say 1/32

of an inch which represents a fixed amount of

the log depending on the scale used. The net

result of adding to the logarithm a fixed quan-

tity, as 0.01, is that the anti -logarithm is mul-

tiplied by 1.025, or the error is 21/2%. No mat-

ter at what part of the scale the 0.01 is added,

the error is always 21/2%.

An example of the advantage due to the use

www.americanradiohistory.com

788 Radio Mathematics and Calculations

1.0

0.9

0.8

0.6

0.4

0.3

o

IS 10 5 0 5

KC. OFF RESONANCE

Figure 38.

A RECEIVER RESONANCE CURVE.

This curve represents the output of a re-

ceiver versus frequency when plotted to linear

coordinates.

10 15

of semi- logarithmic paper is shown in Figures

38 and 39. A resonance curve, when plotted on

linear coordinate paper will look like the curve

in Figure 38. Here we have plotted the output

of a receiver against frequency while the ap-

plied voltage is kept constant. It is the kind of

curve a "wobbulator" will show. The curve

does not give enough information in this form

for one might think that a signal 10 kc. off

resonance would not cause any current at all

and is tuned out. However, we frequently have

off resonance signals which are 1000 times as

strong as the desired signal and one cannot

read on the graph of Figure 38 how much any

signal is attenuated if it is reduced more than

about 20 times.

In comparison look at the curve of Figure

39. Here the response (the current) is plotted

in logarithmic proportion, which allows us to

plot clearly how far off resonance a signal has

to be to be reduced 100, 1,000, or even 10,000

times.

Note that this curve is now "upside down ";

it is therefore called a .telectivily curve. The

reason that it appears upside down is that the

method of measurement is different. In a se-

lectivity curve we plot the increase in signal

voltage necessary to cause a standard output

off resonance. It is also possible to plot this in-

crease along the Y -axis in decibels; the curve

then looks the same although linear paper can

10.000

1000

100

10 9 8

4

3

Ï

-20 -lo o +lo

KC. OFF RESONANCE

+20

Figure 39.

A RECEIVER SELECTIVITY CURVE.

This curve represents the selectivity of a re-

ceiver plotted to logarithmic coordinates for

the output, but linear coordinates for fre-

quency. The reason that this curve appears

inverted from that of Figure 38 is explained

in the text.

be used because nuw our unit is logarithmic.

An example of full logarithmic paper being

used for families of curves is shown in the re-

actance charts of Figures 40 and 41.

Nomograms or An alignment chart con -

Alignment Charts sists of three or more sets

of scales which have been

so laid out that to solve the formula for which

the chart was made, we have but to lay a

straight edge along the two given values on

any two of the scales, to find the third and

unknown value on the third scale. In its sim-

www.americanradiohistory.com

Figure 40.

REACTANCE -FREQUENCY CHART FOR AUDIO FREQUENCIES

See text for applications and instructions for use.

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790 Radio Mathematics and Calculations THE RADIO

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www.americanradiohistory.com

HANDBOOK Polar Coordinates 791

o

Figure 42.

THE SIMPLEST FORM OF NOMOGRAM.

plest form, it is somewhat like the lines in Fig-

ure 42. If the lines a, b, and c are parallel and

equidistant, we know from ordinary geometry,

that b = 1/2 (a + c). Therefore, if we draw a

scale of the same units on all three lines, start-

ing with zero at the bottom, we know that by

laying a straight -edge across the chart at any

place, it will connect values of a, b, and c,

which satisfy the above equation. When any

two quantities are known, the third can be

found.

If, in the same configuration we used loga-

rithmic scales instead of linear scales, the rela-

tion of the quantities would become

log b = 1/2 (log a + log c) orb =

By using different kinds of scales, different

units, and different spacings between the scales,

charts can be made to solve many kinds of

equations.

If there are more than three variables it is

generally necessary to make a double chart,

that is, to make the result from the first chart

serve as the given quantity of the second one.

Such an example is the chart for the design of

coils illustrated in Figure 45. This nomogram

is used to convert the inductance in microhen-

ries to physical dimensions of the coil and vice

versa. A pin and a straight edge are required.

The method is shown under "R. F. Tank Cir-

cuit Calculations" later in this chapter.

Polar Coordinates Instead of the Cartesian

coordinate system there

is also another system for defining algebraical-

ly the location of a point or line in a plane. In

this, the polar coordinate system, a point is de-

termined by its distance from the origin, O,

and by the angle it makes with the axis O -X.

In Figure 43 the point P is defined by the

length of OP, known as the radius vector and

RADIUS /

VECTOR /

/ PNG-

ORIGIN AXIS X

Figure 43.

THE LOCATION OF A POINT BY

POLAR COORDINATES.

In the polar coordinate system any point is

determined by its distance from the origin

and the angle formed by a line drawn from

it to the origin and the O-X axis.

by the angle A the vectorial angle. We give

these data in the following form

P = 3 Lo°

Polar coordinates are used in radio chiefly

for the plotting of directional properties of mi-

crophones and antennas. A typical example of

such a directional characteristic is shown in

Figure 44. The radiation of the antenna rep-

resented here is proportional to the distance of

the characteristic from the origin for every

possible direction.

Figure 44.

THE RADIATION CURVE OF AN

ANTENNA.

Polar coordinates are used principally in radio

work for plotting the directional characteris-

tics of an antenna where the radiation is

represented by the distance of the curve from

the origin for every Possible direction.

www.americanradiohistory.com

792 Radio Mathematics and Calculations

Reactance Calculations

In audio frequency calculations, an accuracy

to better than a few per cent is seldom re-

quired, and when dealing with calculations in-

volving inductance, capacitance, resonant fre-

quency, etc., it is much simpler to make use of

reactance -frequency charts such as those in

figures 40 and 41 rather than to wrestle with a

combination of unwieldy formulas. From these

charts it is possible to determine the reactance of

a condenser or coil if the capacitance or induc-

tance is known, and vice versa. It follows from

this that resonance calculations can be made

directly from the chart, because resonance

simply means that the inductive and capacitive

reactances are equal. The capacity required to

resonate with a given inductance, or the induc-

tance required to resonate with a given capac-

ity, can be taken directly from the chart.

While the chart may look somewhat formid-

able to one not familiar with charts of this

type, its application is really quite simple, and

can be learned in a short while. The following

example should clarify its interpretation.

For instance, following the lines to their in-

tersection, we see that 0.1 hy. and 0.1 pfd. in-

tersect at approximately 1,500 cycles and 1,000

ohms. Thus, the reactance of either the coil or

condenser taken alone is about 1000 ohms, and

the resonant frequency about 1,500 cycles.

To find the reactance of 0.1 hy. at, say,

10,000 cycles, simply follow the inductance

line diagonally up towards the upper left till it

intersects the horizontal 10,000 kc. line. Fol-

lowing vertically downward from the point of

intersection, we see that the reactance at this

frequency is about 6000 ohms.

To facilitate use of the chart and to avoid

errors, simply keep the following in mind: The

vertical lines indicate reactance in ohms, the

horizontal lines always indicate the frequency,

the diagonal lines sloping to the lower right

represent inductance, and the diagonal lines

sloping toward the lower left indicate capaci-

tance. Also remember that the scale is loga-

rithmic. For instance, the next horizontal line

above 1000 cycles is 2000 cycles. Note that

there are 9, not 10, divisions between the heavy

lines. This also should be kept in mind when

interpolating between lines when best possible

accuracy is desired; halfway between the line

representing 200 cycles and the line represent-

ing 300 cycles is not 250 cycles, but approxi-

mately 230 cycles. The 250 cycle point is ap-

proximately 0.7 of the way between the 200

cycle line and the 300 cycle line, rather than

halfway between.

Use of the chart need not be limited by the

physical boundaries of the chart. For instance,

the 10 -µpfd. line can be extended to find where

it intersects the 100 -hy. line, the resonant fre-

quency being determined by projecting the in-

tersection horizontally back on to the chart.

To determine the reactance, the logarithmic

ohms scale must be extended.

R. F. Tank When winding coils for use in

Circuit radio receivers and transmit -

Colculotions ters, it is desirable to be able to

determine in advance the full

coil specifications for a given frequency. Like-

wise, it often is desired to determine how much

capacity is required to resonate a given coil so

that a suitable condenser can be used.

Fortunately, extreme accuracy is not re-

quired, except where fixed capacitors are used

across the tank coil with no provision for trim-

ming the tank to resonance. Thus, even though

it may be necessary to estimate the stray cir-

cuit capacity present in shunt with the tank

capacity, and to take for granted the likelihood

of a small error when using a chart instead of

the formula upon which the chart was based.

the results will be sufficiently accurate in most

cases, and in any case give a reasonably close

point from which to start "pruning."

The inductance required to resonate with a

certain capacitance is given in the chart in

figure 41. By means of the r.f. chart , the

inductance of the coil can be determined,

or the capacitance determined if the induc-

tance is known. When making calculations, be

sure to allow for stray circuit capacity, such as

tube interelectrode capacity, wiring, sockets,

etc. This will normally run from 5 to 25 micro -

microfarads, depending upon the components

and circuit.

To convert the inductance in microhenries

to physical dimensions of the coil, or vice

versa, the nomograph chart in figure 45 is

used. A pin and a straightedge are required.

The inductance of a coil is found as follows:

The straightedge is placed from the correct

point on the turns column to the correct point

on the diameter -to- length ratio column, the

latter simply being the diameter divided by the

length. Place the pin at the point on the plot

axis column where the straightedge crosses it.

From this point lay the straightedge to the cor-

rect point on the diameter column. The point

where the straightedge intersects the induc-

tance column will give the inductance of the

coil.

From the chart, we see that a 30 turn coil

having a diameter -to- length ratio of 0.7 and a

diameter of 1 inch has an inductance of ap-

proximately 12 microhenries. Likewise any one

of the four factors may be determined if the

other three are known. For instance, to deter-

mine the number of turns when the desired in-

www.americanradiohistory.com

Figure 45. COIL CALCULATOR NOMOGRAPH

For single loyer solenoid coils, any wire size. See text for instructions.

N OF PLOT INDUCTANCE IN

TURNS AXIS MICROHENRIES

- 400

- 300

- 200

- 150

-100

- 90

- 80

- 70

- 80

- 50

- 40

- 20

-15

- 10

-5

--4

3

- 20000

-10000

8000

- 6000

- 4000

- 3000

- 2000

1000

- 800

- 600

400

- 300

200

- 100

- 80

60 .8-

-- 40

RATIO DIAMETER

LENGTH

8- 6- 5-

4 -

3-

2-

30 - --

20 --

1Ó

8

r- 6

-4

- 3

- 2

- .6

-.4 -.3 -.2

.8-

DIAMETER

INCHES

-S

-4

-3

-2

-1.5

-.75

-- o

www.americanradiohistory.com

794 Radio Mathematics and Calculations

ductance, the D/L ratio, and the diameter are

known, simply work backwards from the ex-

ample given. In all cases, remember that the

straightedge reads either turns and D/L ratio,

or it reads inductance and diameter. It can

read no other combination.

The actual wire size has negligible effect

upon the calculations for commonly used wire

sizes (no. 10 to no. 30). The number of turns

of insulated wire that can be wound per inch

(solid) will be found in a copper wire table.

Significant Figures

In most radio calculations, numbers repre-

sent quantities which were obtained by meas-

urement. Since no measurement gives absolute

accuracy, such quantities are only approximate

and their value is given only to a few signifi-

cant figures. In calculations, these limitations

must be kept in mind and one should not fin-

ish for instance with a result expressed in more

significant figures than the given quantities at

the beginning. This would imply a greater ac-

curacy than actually was obtained and is there-

fore misleading, if not ridiculous.

An example may make this clear. Many am-

meters and voltmeters do not give results to

closer than 1/4 ampere or 1/4 volt. Thus if we

have 21/4 amperes flowing in a d.c. circuit at

63/4 volts, we can obtain a theoretical answer

by multiplying 2.25 by 6.75 to get 15.1875

watts. But it is misleading to express the an-

swer down to a ten -thousandth of a watt when

the original measurements were only good to

1/4 ampere or volt. The answer should be ex-

pressed as 15 watts, not even 15.0 watts. If

we assume a possible error of 1/8 volt or am-

pere (that is, that our original data are only

correct to the nearest 1/4 volt or ampere) the

true power lies between 14.078 (product of

21/8 and 65/8) and 16.328 (product of 23/8 and

67/8). Therefore, any third significant figure

would be misleading as implying an accuracy

which we do not have.

Conversely, there is also no point to calcu-

lating the value of a part down to 5 or 6 sig-

nificant figures when the actual part to be used

cannot be measured to better than 1 part in

one hundred. For instance, if we are going to

use 1% resistors in some circuit, such as an

ohmmeter, there is no need to calculate the

value of such a resistor to 5 places, such as

1262.5 ohrr>i. Obviously, 1% of this quantity

is over 12 ohms and the value should simply

be written as 1260 ohms.

There is a definite technique in handling

these approximate figures. When giving values

obtained by measurement, no more figures are

given than the accuracy of the measurement

permits. Thus, if the measurement is good to

two places, we would write, for instance, 6.9

which would mean that the true value is

somewhere between 6.85 and 6.95. If the meas-

urement is known to three significant figures,

we might write 6.90 which means that the true

value is somewhere between 6.895 and 6.905.

In dealing with approximate quantities, the

added cipher at the right of the decimal point

has a meaning.

There is unfortunately no standardized sys-

tem of writing approximate figures with many

ciphers to the left of the decimal point. 69000

does not necessarily mean that the quantity is

known to 5 significant figures. Some indicate

the accuracy by writing 69 x 10' or 690 x 10'

etc., but this system is not universally em-

ployed. The reader can use his own system, but

whatever notation is used, the number of sig-

nificant figures should be kept in mind.

Working with approximate figures, one may

obtain an idea of the influence of the doubtful

figures by marking all of them, and products

or sums derived from them. In the following

example, the doubtful figures have been under-

lined. 603 34.6

0.120

637.720 answer: 638

Multiplication

654 654

0.342 0.341

1308 19612

2616

1962

223.668 answer: 224 224

26116

1108

It is recommended that the system at the

right be used and that the figures to the right

of the vertical line be omitted or guessed so as

to save labor. Here the partial products are

written in the reverse order, the most impor-

tant ones first.

In division, labor can be saved when after

each digit of the quotient is obtained, one fig-

ure of the divisor be dropped. Example:

1.28

527 1 673

527

53 146

106

5 40 40

www.americanradiohistory.com

INDEX

795

www.americanradiohistory.com

Absorption, Ionospheric 414

Acceptor, def. of 92

A.C. -D.C. power supply 694

A -C V -T voltmeter 734

Adaptor, F -M 321

Admittance, def. of 52

"A" -frame antenna mast 444

Air capacitor 355

Air dielectric 32

Air -gap, capacitor 262

Alignment - B.f.o. - Filter 234

" Receiver - I -F - R -F 233, 234, 180

All- driven antenna 500

Alpha, def. of 93

Alternating current - Amplitude - Average 45

" Def. of - Effective value 41, 45

Generation 42

Impedance - "J" operator 47

Ohm's law - Phase angle

Pulsating - R.M.S.

Transformer

Transient

Voltage divider

Alternator

Amateur frequency bands - licenses

Amateur Radio

Ammeter

Ampere, def. of

" turns

Amplification factor

Amplifier - audio, 15 -watt

" Audio - Cascode 225,

cascode grounded -grid

constant current curves

Cathode - Cathode driven 1 27,

Cathode follower

Class A - A2 99, 108,

Class AB - AB1 108,

Class B 99, 108, 123, 129,

Class C 108, 124,

coupling - inductive 113, 118,

D.C.

distortion products

Doherty

Driver

Equivalent circuit 110,

feedback

Grid circuit

Grounded cathode, grounded screen

45, 46 45

61 59 52 42

12

11

731

22, 23

36, 62

73, 74

Grounded -grid

Grounded -grid, use of

350 w., grounded -grid

813 grounded -grid

KW -2 grounded -grid

Hi -fi, 25 -watt

Hi -fi - High mu (A)

Horizontal - I -F

Kilowatt

Kilowatt 4 -400A

Linear

Linear, 2kw. P.E.P.

Load line

Loftin -White

Neutralization

Noise factor

Operational

Pentode

Pi- network

Plate efficiency

Power

162, 212, 256,

138,

171,

250,

1 18,

678

212

169

168

256

127

118

137

249

123

117

613

296

126

111

129

121

162

612

617

627

634

146

112

208

649

284

654

166

118

615

152

199

120

609

159

602

A

" Push -pull 121, 604

" Response 112

" R.F., Class AB1 166

" R.F., Class B 168

" R.F., Grounded screen 611, 612

" R -F - RC 109

" R -F - 4CX -1000A 211

" Semi -conductor 104

" Summation 200

" Summing 201

" Terman- Woodyard 296

" Tetrode 606

" Transistor 104

" "Tri- band" 4CX300A 622

" Ultra- linear 142

" Vertical s 138

" Video 112, 113, 128

" V.H.F. "strip -line" 595

" Voltage 110

" Williamson 140

" 3 -10002 661

" 4 -400A all -band 643

" 4CX1000A, 2 kw. 654

Amplitude - A.C. 45

Amplitude, Distortion - Modulation..109, 280, 669

Analog computers - problems 195, 200

"And -or" circuitry 204

Angle of radiation - V.h.f. 407, 456, 474

Anode, def. of - dissipation 67, 72

Antenna - Adjustment 505

" All- driven 500

" Bandwidth 401, 409

796

t.

Beam - design chart - 2- element 494, 491

Bidirectional - Bi- square

Bobtail

Broadside

Bruce

Center -fed

Colinear

Combinations of

Construction

501, 466

468

464

466

434

462

471

444, 502

Corner reflector 481

Couplers - mobile 450, 516, 520

Coupling systems 447

Cubical quad 467

Curtain 464

Delta match 497

Dipole array - broad -band 461, 431

Directivity 400, 406

Discone 437, 477

Doublet 423

Dummy 736

Efficiency 405

Element spacing 464

End -effect 402

End -fed - end -fire 422, 433, 469

Feed systems 424, 494

Franklin array 462

Fuchs 422

Gain, beam 491

Gamma match 499

Ground loss 405

Ground plane 427

Hertz 422

High frequency type 455

Impedance 404

Intercept, v.h.f. 473

Lazy -H 464

length - Length -to- diameter ratio_.401, 402

Long wire 457

www.americanradiohistory.com

Marconi 404, 428

Matching systems 438

Measurements 740

" Multee 436

" Multi -band 432, 510

Mutual coupling 475

Parasitic beam 490

Patterns 400, 456

" Phasing 462

Polarization 400

" Polarization, v.h.f. 475

" Power gain 401

" 0 405

" Reactance - Resonance 404, 400

" Rhombic 460

" Rotary - Rotary match 490, 498

" Rotator 508

" Single wire feed 437

" Six- shooter 468

Sleeve 476

Space conserving 430

Stacked 464, 494

Sterba 465

Stub adjustment - stub match 441, 499

T -match 497

3- element beam 484

Triplex 471

Tuner 452

Turnstile 476

Two -band Marconi 434

V -type 458

Vertical 426

V.h.f. - 8 element, ground plane....484, 477

V.h.f. - helical - horn 479, 482

V.h.f. nondirectional 478

V.h.f. rhombic 482

V.h.f. - Screen array - Yagi 487, 484

W8JK 469

X -array 465

Yoke match 494

"Zepplin" 423, 463

Antennascope 748

Antennascope, SWR meter 747

Anti- resonance 54

Aquadag 85

Area, capacitor plate 33

Arrays, antenna 461, 408

"Assumed" voltage 52

Atom, def. of - atomic number 21

Attenuation, Bass /Treble 139

Audio - Amplifier 225

" 15 watt 678

" Distortion 135

Equalizer 139

Feedback loop 141

Filter, SSB 329

Frequency, def. of 42

Hum 142

limiters 155, 185, 226

Phase Inverter 115

Phasing network - SSB 334

Rectification 379

Wiring technique 142

Autodyne detector 206

Automatic load control, SSB 341

Automatic modulation control 670

Automatic volume control 223

Autotransformer 63, 386

Avalanche voltage 697

to

vt

It

1.

tt

to

ft

el

0.

O.

OS

tt

to

t.

B

Balanced modulator - SSB 327, 334,

Balanced modulator, deflection

Balanced SWR bridge

Balanced transmission line measurements

Bands, amateur

Bandspread tuning

Bandwidth, antenna 401,

Bandwidth, modulation

Base electrode

Bass suppression, audio

Battery bias

Beam - deflection modulator

" Design chart - Dimensions 494,

" Element spacing

Front -back ratio

H -F - 5- element - 2- element 492,

Power tube

Radiation resistance

" Three element HF - 220 Mc.

Beat note

Beat oscillator

B -H curve

Bias - Battery

" Cathode

Def. of - cut -off

Grid 108,

Grid leak 112,

R -F amplifier

Safety

Shift modulator

Supply, modulator

Supply, regulated

Transistor

Bidirectional antenna

Billboard antenna

Binary notation 195,

Bishop noise limiter

Bi- square antenna

Bistable multivibrator

Bit Blanketing interference

Bleeder resistor - safety 27, 709,

Blocked grid keying

Blocking diodes

Blocking oscillator 102, 189,

Bob -tail antenna

Bombardment, cathode

Boost, Bass /treble

Break voltage

Bridge, impedance

" Measurements

" Power supply

" Rectifier 689,

" Slide -wire

" SWR

Bridge -T oscillator

Bridge -type vacuum tube voltmeter

Bridge, Wheatstone

Broad band dipole

Broadcast interference

Broadside antennas - arrays 464,

Bruce antenna

Butterfly circuit

339

350

746

744 12

215

409

281 92

304

267

350

492

493

492

490 78

491

492

206

222 36

267

266 73

265

266

604

266

305, 307

675

71 2

96

501

473

196

226

466

102

196

376

391

394

397

190

468 70

139

203

737

738

716

690

738

454, 746

192

131

738

431

375

408

466

231

C

Calorimeter 736

Capacitance - Calculation - definition 32, 30

" Interelectrode 76, 106

" Neutralization 107

" Tank - Stray 259, 216

797

www.americanradiohistory.com

Capacitive coupling 268

Capacitive reactance 46

Capacitor - A.C. circuit 34

" Air - Characteristics 355, 354

" Breakdown 262

Charge 31

Color code 527

Electrolytic 34, 708

Equalizing 34

Filter 707, 690

Fixed 30

Input filter 714, 690

Leakage 34

Low inductance 611

Parallel 33

Q 55

Series 33

Temperature coefficient 32

Time constant 39

Vacuum - variable vacuum 360

Voltage rating 34

Carrier - Distortion 309

Carrier, Shift 282

Carrier, S.S.B. 323, 327, 328, 330

Cascade amplifier - V.H.F. 111, 212

Cathode - Bias 108, 266

" Coupling - coupled inverter 115, 116

" Current - Definition 79, 67

" Driven amplifier 612, 256

Follower 272

Follower amplifier 127, 164

Follower driver 680, 683

Follower modulator 288

Keying 393

Modulation 295

Ray oscilloscope - tube 170, 84

Tank, grounded -grid 165

Cavity, resonant 230

Cell, Weston 24

Center -fed antenna 434

Ceramic dielectric 32

Charge 22, 23, 30

Chassis layout 726

Choke, filter 709

Choke input filter 713, 715

Choke, R -F 270, 357

Circuit constants, measurement of 737

Circuits - Coupled 153

" D.C. - Equivalent 21, 51

" Input loading 152

" Limiting 185

Magnetic 35

Parallel 25

Parasitic 361

Q 55

Resonant - Series - Tank ....53, 24, 55, 57

Circulating current 56

Clamping circuit 187, 188

Clapp oscillator 239

Class A amplifier, def. of 108

Class A2 amplifier 118

Class AB amplifier, def. of 108

Class ABI amplifiers 108

Class B amplifiers 108, 123, 125, 159

Class B amplifier, grounded -grid 613

Class B modulator 126, 292

Class B, R -F amplifier 249

Class C amplifiers 108, 124, 125, 249

Class C bias 267

Clipper -Amplifier design 678

Clipper filter, speech 302

Clipping circuits 185

1.

Clipping, negative peak 670

Clipping, phase shift 299

Clipping, speech 298, 670

Closed loop feedback 192

Coaxial reflectometer 742

Coaxial transmission line 419

Coaxial tuned circuit 230

Cade 14 -20

Code, practice set 20

Coefficient of coupling 38

Coefficient, temperature 32

Coercive force 37

Coil, core loss 55

Coil, Q 54, 356

Colinear antenna 462

Collector electrode 92

Collins feed system 443

Collins filter 220

Color code, component 528

Color code, standard 527, 528

Colpitts oscillator 239

Complementary symmetry 97

Complex quantity 49

Component color code 528

Component nomenclature graph 526

Compound, def. of 21

Compression, volume 670

Compressor, A.L.C. 341

Computers, electronic 194

Conductance, def. of 52

Conduction 67, 90

Conductivity - Conductor 23

Constant amplitude recording 137

Constant current curve 74, 157, 168

Constant -K filter 64

Constant velocity recording 136

Constants, vacuum tube 107

Control circuit, mobile 521

Control circuit, transmitter 387

Conversion conductance 80

Conversion frequency, SSB 336

Converter stage 209

Converter, transistor 100

Core loss 55

Corner reflector antenna 481

Corona static 525

Coulomb 22, 31

Counter e.m.f 37

Counting circuit 190

Coupling - Capacitive 268

" Cathode 115

Choke 114

Critical 57, 216

Direct 115

Effect of 56

Impedance - Inductive 113, 269

Interstage - r -f 268

Link 270, 603

Magnetic 38

Parasitic 360

R -F feedback 273

Systems, antenna 447

Transformer 113

Unity 269

Critical coupling 216

Critical inductance 686

Cross modulation 377

Crossover point 136

C.R.P.L. Predictions 414

Crystal - Current 244

" Filter - Lattice filter 218, 332

" Harmonic 244

798

www.americanradiohistory.com

" Oscillator - Quartz 242,

" Pickup

Cubical quad antenna

243

137

467

Dissipation, anode

Dissipation, grid

Distortion - Amplitude - Frequency

72

165

109

Current - Alternating 41 " Audio 135

" Amplification (alpha) 93 " Carrier 309

" Cathode 79 " Harmonic 119

Circulating - Closed path 56, 28 Intermodulation 136

Def. of 22 Modulation 305

" Effective - Effective (a.c.) 733, 45 Nonlinear - Phase 109, 135

" Electrode 106 Products 613

" Induced - Instantaneous 42, 44 Products, SSB 340

" Inverse 693 Transient 135

" Measurement 731, 733 Transistor 98

Peak amplifier 125 Divider, frequency 190

" Rating, Power supply 685 Divider, voltage 26

" Saturation - Skin effect 72, 55 Doherty amplifier 296

Curtain antenna 464 Dome audio phasing network 334

Cutoff (alpha) 93 Double conversion circuit 213

Cutoff bias 73 Double sideband, DSB 329, 349

Cutoff, extended 285 Doubler, frequency 256

Cutoff frequency 64 Doubler, push -pull 258

Cycle 41 Doublet, antenna - multi -wire 423, 425

Cycle, sunspot 414 Drift transistor 93

Driver, amplifier 126

D Driver, cathode follower 680, 683

d'Arsonval meter 731 Duct propagation 411

D. C. amplifier 117 Dummy antenna - dummy loads 726

D. C. clamping circuit 187 Dynamic resistance 95

D.C. power supplies 684 Dynamotor, PE -103 522

D.C. restorer - circuit 188 Dynatron oscillator 240

Defector - autodyne - crystal

Deflection, plate 206

171 E

Degeneration, transistor 96 Eddy current 38

Degree, electrical 43 Efficiency, antenna 405

Delta match - antenna - system....497, 426, 438 Electric filters 63

Demodulator 205 Electrical energy 29

Detection, slope 208 Electrical potential 22

Detection, synschronous, DSB 349 Electromagnetic deflection 86

Detector - Diode 222 Electrolytic capacitor 34, 708

" Fremodyne 208 Electrolytic conductor 22, 67

Grid leak - Impedance - Plate 222 Electromagnetism 35

Product 235, 348 Electromotive force (e.m.f.) 22, 37

Ratio 319 Electron coupled oscillator 239

Super -regenerative 207 Electron, def. of - drift - orbit 67, 21

Deviation, FM - Measurement 310, 316 Electronic computers 194

Dielectric, ceramic - Constant (K) 30, 31, 32 Electronic conduction 67

Differential keying 396 Electronic differentiation - integration 198

Differentiation, electronic 198 Electronic Keyer, "910" 597

Differentiator (RC) 59 Electronic multiplication 198

Digital circuits - computers 195 Electrostatics 30

Digital package 204 Electrostatic deflection - energy 83, 30

Diode - A.v.c. 223 Element, def. of - metallic, non -metallic 21, 90

" Blocking 397 Element, reactive, non- reactive 21

Crystal 232 Emission equation 70

Def. of 71 Emission - photoelectric 67

Detector 222 " Secondary 71, 77

1. Gate 204 " Spurious 379

Limiter 1 85 " Thermionic 67

Mixer 210 Emitter electrode 92

Modulator 328 Enclosures 724, 725

Semi -conductor 90 End effect, antenna 402

Storage time 103 End -fed antenna 422, 433

Voltmeter 734 End -fire antennas - arrays 469, 408

Zener 105 Energy - Electrical - Electrostatic 29, 30

Dipoles 399, 432, 461, 494 " Potential 30

Direct current circuits 21 " Storage, capacitor 31

Directive antennas, H -F 455 " Transistor 93

Directivity, antenna 400, 406 ENIAC 195

Discharge of capacitor 30 Envelope, carrier 280

Discone antenna 437, 477 Equalizer, audio 139

Discontinuities, transmission line 421 Equal- tempered scale 135

Discriminator, F -M 318 Equivalent circuit 51, 110, 111

799

www.americanradiohistory.com

Equivalent circuit, transistor 95

Equivalent noise resistance 153

Error signal 193

Excitation, grid 250

Exciters, low power 576

Exciters, SSB 342, 345

Extended -Zepp antenna 463

F

Factor of merit 54

Fading 414

Farad, def. of 31

Feedback amplifier 129

" Audio 141

" Circuits 192

" Control 192

" Error cancellation 193

" Miller 199

" R -F 272, 607

" Transistor 101

Feed systems, antenna 424, 494

Feedthrough power 612

Field, magnetic 35

Filament, def. of 67

Filament reactivation 68

Filter - Capacitor 707, 690

" Capacitor input 714, 690

" Carrier, SSB 330

" Choke 709

Choke input 713, 715

Circuits, mobile 514

Crystal 218

Crystal lattice - SSB 332

Generator, SSB 330

High Pass 64, 368

Inductor input 687

Insertion loss 65

low pass 64, 372

M- derived 64

Mechanical 220, 332

Noise 225

Passband, SSB 332

O- multiplier 234

Resistance- capacitance -Resonance -.687, 688

Ripple factor 688

Sections - Series - Shunt 64

TVI, receiver - TVI -type 65, 372

Wave 63

Filter -type exciter, SSB 345

Fixed bias 108

Flat -top beam antenna 469

Fletcher- Munson curve 140

Floating Paraphase inverter 116

Flux, def. of - density 35, 36

Flywheel effect 57, 257

Folded dipole 440, 425, 494

Forcing function 200

Foster -Seely discriminator 319

Franklin antenna 462

Franklin oscillator 241

Free electrons 21

Free -running multivibrator 189

Fremodyne detector 208

Frequency 41, 58

" Conversion, SSB 336

" Cutoff 64

" Distortion 109

Divider 190

Interruption 207

Maximum useable 413

Measurements 739

Multiplier 256

Pulse repetition 190

Range, hi- fidelity 136

Ratio, Lissajous 176

Resonant 53

Shift keying 322

Sound 134

Spectrum 41

Spotter 740

Standard WWV 739

Sweep 172

Frequency Modulation 308

" Adapter 321

Deviation - ratio - index 310

Discriminator 318

Limiter 320

Linearity 314

Narrow band 311

Pre -emphasis 321

Reception 317

Front -back ratio, beam 492

Fuchs antenna 422

Full -wave limiter 228

Full -wave rectifier 689

Function, Forcing 200

Function generator - non -linear - ramp --202, 203

G

G, conductance, def. of

Gain, antenna - beam

Gain, power - resistance - voltage

Gamma match, antenna - system 499,

Gas tube - generator 87,

Gate, diode - limiter 204,

Gauss, def. of

Generator, function

Generator noise 524,

Generator, sawtooth

Generator, time base

Gilbert, def. of

Grid bias 108,

Grid, def. of

Grid dissipation

Grid excitation

Grid leak bias 109, 112,

Grid leak detector

Grid limiter, limiting

Grid modulation 250,

Grid neutralization

Grid-screen factor

Ground bus

Ground currents

Ground loss, antenna

Ground plane antenna - VHF 427,

Ground resistance

Ground, R -F

Ground termination

Ground, transmitter

Ground, wave

Grounded -cathode amplifier

Grounded -grid amplifier 162, 212, 256,

Grounded -grid cascade amplifier

Grounded -grid, cathode tank

Grounded -grid r.f. amplifier

Guy wires, antenna

52

491, 457 94

440

172

185 36

202

750

172

171 36

265 72

165

250

266

222

187

286

251

78

142

358

405

477

406

358

429

390

410

162

612

169

165

612

445

H

Hairpin coupling 230

Half -wave rectifier 689

Harmonic - B.f.o. 223

" Crystal 244

" Def. of 58

800

www.americanradiohistory.com

" Distortion 119 Initial condition voltage 203

" Music 135 Injection voltage 211

" Oscillator

" Radiation 245 Input loading 152

260, 369 Input resistance 107, 214, 274

Harmoniker TVI filter 375 Insertion loss, filter 65

Hartley oscillator 238 Instability, R -F amplifier 607

Hash rectifier 694 Instability, static 117

Heat cycle, resistor 353 Insulation, VHF 475

Heat sink 99 Insulator, def. of 22, 23

Heat sink, transistor 104, 707 Integration amplifier 200

Heater cathode 70 Integration, electronic 198

Heising modulation 292 Integrator, Miller 199

Helical antenna 479 Integrator (RC) 59

Henry, def. of 37 Interelectrode capacitance 76

Hertz antenna 422 Interference - Broadcast 375

Heterodyne 206 " Harmonic 369

H -F antennas 455 " Hi -fi - image - TV 381, 382, 367

High fidelity 134 -138 Interlock, power 391

High fidelity, Interference 382 Intermodulation distortion 136

High -pass TVI filter 368 Intermodulation test 145

Holes in semi -conductor 91 Internal resistance 25

Horizontal directivity 406 International Morse Code 15

Horn antenna, VHF 482 Interruption frequency 207

Hot cathode phase inverter 116 Interstage coupling 268

Hum, audio 142 Intrinsic semi -conductor 91

Hysteresis loop - loss 36, 38 Inverse current - Voltage 693

Inverter, phase - voltage divider 115, 1 16, 117

I Ion, def. of - Positive 22, 87

I -f alignment 180, 233 Ionosphere, absorption 13, 414

" Amplifier 208 Ionosphere, Sporadic -E layer 414

" Noise limiter 225 Ionospheric cycle 414

" Pass -bond 217 Ionospheric fading 414

Rejection notch 220 Ionospheric layers - propagation 412, 413

" Shape factor 217 Ionospheric reflection 415

" Tuned circuit 216 IR drop 24

Ignition noise 523 Iron vane meter 733

Images -image interference- ratio....209, 211, 381 Isotropic radiator 406

Impedance 46, 47

" Antenna 404 J

" Bridge 737 Jitter 188

" Complex 51 Johnson 0 -feed system 442

" Coupling 56, 113 'J" Operator 47

" Match, audio 127 Joule, def. of 30, 37

" Reflected 56, 63 Junction, transistor 93

" Resonant 54

" Screen circuit 287 K

" Surge 420

" Tank circuit 260 K, dielectric constant 32

" Transformation 63 Key clicks 392

" Transistor 97 Keyer, "9T0" electronic 597

" Transmission line 417 Keying - blocked grid 394

" Triangle 48 Cathode circuit 393

Incident voltage, transmission line 742 " Differential 396

Indicator, Antennascope 748 Frequency shift 322

Indicator, Selsyn 510 Screen grid 399

Indicator - standing wave - twin- lamp..741, 745 Transmitter 395

Induced current 42 Kilocycle 42

Inductance 37 Kinescope tube 84

" Capacitor 355 Kirchhoff's Laws 28

" Cathode lead 153 Kylstron - reflex 81, 82

" Critical 686

" Lead - Magnetic - Mutual 81, 37 L

" Parallel - Series 38 Lamb noise limiter 226

" Resistor 353 Layout, chassis 726

" Screen lead 255 Lazy -H antenna 464

Induction, def. of 42 L/C ratio 56

Inductive reactance 46 Lead inductance 81

Inductive coupling 269 Leakage, capacitor 34

Inductive tuning 609 Leakage reactance 63

Inductor, iron core 38 Left -hand rule 35

Inductor, time constant 40 Length, antenna 401

Infinite impedance detector 222 L- section filter 64

801

www.americanradiohistory.com

Licenses, amateur 12 Mechanical filter, use of 571

Limiter, audio 226 Megacycle 42

Limiter, diode 185 Megohm 24

Limiter, F -M 320 Memory circuit 204

Limiter, full -wave 228 Mercury vapor rectifier 694

Limiter, noise 225 Meteor bursts 416

Limiter, series diode 185 Meter, d'Arsonval 731

Limiter, INS 228 Meter, iron vane 733

Limiting circuit 185, 186, 202 Meter, multi -range 732

Limiting diode 204 Meter, rectifier 734

Line filter 227 Meter shunt 731

Line regulation 384 Meter, thermocouple 734, 736

Line, Slotted 740 Mho, def. of 74

Linear amplifier 284 Mica dielectric 32

" Class B 129 Microfarad 30

" Kilowatt 649 Microhenry 37

" Tuning 285 Micro -ohm 23

" 2KW, P.E.P. 654 Middle C 134

Linear matching transformer 442 Miller effect 107, 112, 218

Linearity, distortion products 613 Miller feedback - oscillator 199, 245

Linearity tracer 185 Miller integrator 199

Link coupling - antenna 270, 448 Milliammeter 731

Lissajous figures 176 Millihenry, def. of 37

Litz, wire 55 Mixer - circuits 210

L- network design 263 " Diode 80

Load line 74 " Noise 210

Load line, r.f. amplifier 166 " Products, SSB - spurious 337, 339

Load line, transistor 99 " SSB 336

Load resistance amplifier 119 " Stage 209

Load, r.f. dummy 736 " Transistor 100

loaded -O 57 " Triode 211

Loftin -White amplifier 118 " Tube 79

Long -wire antenna 457 Mobile, Antenna coupling 516, 520

Loop feedback 192 " Control circuit 521

Loran, band 12 " Dynamotor 522

Loudness control, audio 140 Equipment construction - design -...520, 511

Loudspeaker - def. of - response 139, 140 Filter circuits 514

Low- frequency parasitics 362 Noise limiter 512

Low -pass TVI filter 372 Noise sources 525

Power supply 515,517,697

M Mode of resonance 230

Magic eye tube 88, 225 Modulation 237

Magnetic - Air gap 38 " Amplitude 280, 669

" Circuit - field - flux 35 Automatic control 670

" Eddy current 38 Bandwidth 281

" Hysteresis loss 38 Cathode 295

" Induction 37 Class B 292

" Left hand rule 35 Constant efficiency 284

" Permeability - reluctance - saturation....36 Distortion 305

Magnetism - residual 35, 37 Frequency 308

Magnetomotive force (M. M. F.) 36 Grid 250, 286

Magnetron 83 Heising 292

Magnitude, scalar 48 Index, F -M 310

Majority carrier 93 Pattern, oscilloscope 178

Marconi antenna 408, 428 Percentage 282

Mast, A -frame 444 Phase 311,315

Matching stub, antenna 441, 499 Plate 249, 291, 293

Matching systems antenna 438 Screen 286

Matching transformer 425 Suppressor 290

Mathematics, radio 752 Transformer 293

Maximum usable frequency 13, 413 Variable efficiency 283

Maxwell, def. of 36 Velocity 81

M- derived filter 64 Modulator - Adjustment 676

Measurements - Antenna 740 " Balanced - S.S.B. 328, 334, 339

Bridge 738 " Beam deflection 350

Circuit constants 737 " Bias -shift 305

Current - Voltage 731, 733 Cathode follower 288

Frequency 739 Class B 125

Parallel wire line 745 Construction 672

Power 731,735 Crosby 590

Transmission line 740, 744 Diode 328

Mechanical filter 220 Matching - impedance match 125, 127

802

www.americanradiohistory.com

" Reactance tube

" Semi -conductor

" SSB

" Tetrode

" Zero bias

" 200 -watt

Molecule

Monitor oscilloscope

Morse code

Mu factor (µ)

Multee antenna

Multi -band antenna

Multiplication, electronic

Multiplier, frequency

Multiplier resistor

Multi -range meters

Multivibrator - circuits

Multivibrator, free running

Multi -wire doublet

Music, def. of

Music systems - scale

Mutual conductance

Mutual coupling

Mutual coupling, antennas

Mutual inductance

Mycalex, dielectric

N

Narrow -band F -M

NBS bridge -T oscillator

Negative feedback loop

Negative peak clipping

Network, R -L integrator

Networks, L and Pi

Neutralization - Amplifier

Capacitance

" Grid

" Grounded -grid

Procedure

R -F

Shunt

Test

Transistor

Neutralizing procedure

NI (ampere turns)

Noise - Factor

" Check, mobile

Generator, silicon

Limiters - Mobile

Mixer

Regulator - sources

Suppression

Thermal agitation

Voltage

Wow and flutter

Nomenclature, color code

Nomenclature, component

Nonconductor

Nondirectional VHF antenna

Non -linear distortion

Non-linear function

Non -resonant transmission line

Nonsinusoidal wave

N -P -N transistor

"Nuvistor" tube, use

Octave, def. of

Oersted, def. of

Ohm - Ohm's Law

Ohmmeter, low range

Ohm's Law (a.c.)

o

312 Ohm's Law (complex quantities) 49

104 Ohm's law for magnetic circuit 36

336 Ohm's Law (Resonant Circuit) 54

669 Ohms - per -volt 732

683 Omega, def. of 44

697 One -shot multivibrator 189

21 On -off circuits 195

179 Open wire line 417

15 Operating desk, constuction of 724

78 Operational amplifier 199

436 Optimum working frequency 414

432, 510 Orbital electron 21

198 Oscillation, parasitic 277, 361, 608, 615

256 Oscillator - Beat 222

732 " Blocking 102, 189, 190

732 " Bridge -T 192

102, 188 Circuits 248

189 Clapp - Colpitts 239

425, 439 Code practice 19

134 Crystal 242

138, 135 Dynatron 240

73 E.c.o. 239

216 Franklin 241

471 Free running 173

37 Harmonic 245

32 Hartley 238

Keying 247

Miller 245

311 NBS bridge -T 192

192 Overtone 247

141 Phase shift 191

670 Pierce 245

199 RC 191

263 Relaxation 102

250 T.p.t.g. 239

107 Transistor 101

251 Transistron 241

615 V.F.O. 242

277 Wein- bridge 191

274 Oscilloscope 170

252 " Circuit 174

276 " Linearity tracer 183

100 " Lissajous figures 176

253 62

Modulation pattern -phase patterns.178, 177

1. Monitor 179, 181, 750

152 Sideband measurements 182

524 Trapezoidal pattern -wave pattern..178, 180

749 Output, peak power 124

225, 512 Overloading, TV receiver 367

210 Overtone crystals 247

524 Overtone, music 135

225, 523 Oxide filament 67, 69

188

151 P

135 Paper dielectric 32

527 Parallel circuit - resistance 25

526 " Diode limiter 186

22 " Feed 271

478 " Resonance 54

135 " Wire line measurements 745

202 Parasitic antenna, Design chart 494

417 " Antenna, VHF 484

58 " Beam design 490

92 " Check for 364

547 " Coupling 360

Element, antenna 493

Oscillation 277, 361, 608

135 Resonance 360

36 Pass band, I -F 217

23, 24 Pass band, mechanical filter 221

733 Pass band, SSB filters 332

45 Patterns, antenna 400, 456

803

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PE -103 dynamotor

Peak amplifier current

Peak current (a.c.)

Peak envelope power, SSB

Peak limiter

Peak noise limiter

Peak power output

Peaked wave

Pentode amplifier

Pentode tube

Period, sound

Permeability

Phantom signal

Phase - angle (a.c.)

" Angle - difference

" Distortion

" Inverter

Modulation

Shift, clipping

Shift, feedback

Shift, oscillator

Phasing, antenna

Phasing generator, SSB

Phasing networks, audio

Phonograph reproduction

Photoelectric emission

Pickups

Pickup, spurious

Pierce oscillator

Pi- network amplifier

Pi- network chart - design

Pi- network coupling

Pilot carrier, SSB

Pitch, sound

Plate current flow

Plate current, static

Plate detector

Plate efficiency amplifier

Plate modulation

Plate resistance

P -N -P transistor

Point contact transistor

Polar notation

Polarization, antenna - VHF

Polyphase rectifier

Potential difference

Potential, electrode

Potential energy

Power amplifier design

Power amplifier triode

Power, feed through

Power, gain

Power gain antenna

Power gain SSB

Power measurement

Power interlock

Power -line filter

Power, resistive

Power supplies

" A.C. -D.C.

Bridge

Design

Dual voltage

Mobile

Oscilloscope

Regulated

Screen

Three -phase

Transistorized

Voltage doubler

1. Voltage multiplier

Voltage quadrupler

177,

109,

311,

265,

249, 73,

400,

731,

716,

522

1 25

45

324

185

225

124 59

120

77

134 36

378 46

178

135

115

315

299

193

191

462

333

334

136 67

137

380

245

609

263

449

323

134

257

166

222

159

291 74

91 92 48

475

693 22

106 30

602

118

612 94

401

324

735

391

225 29

684

694

716

713

718

697

173

712

267

517

703

695

" 300 volt

" 1500 volt

" 2500 volt

Power system, primary

Power transformer

Power transistor

Powerstat auto- transformer

Preamplifier, hi -fi

Pre -emphasis, recording

Preselector

Primary transformer

Probe, R.F. (v.t.v.m.)

Product detector

Propagation

Propagation, sporadic -E

Pulsating alternating current

Pulse- repetition frequency

Push -pull amplifier

Push -pull transformer

Push -pull tripler - doubler

Push -to -talk circuit

Q

Q amplifier tank

Q antenna

Q coils

Q, def. of

Q loaded

Q multiplier

Q tank circuit

Q transformer

Q tuned circuit

Quad antenna

Quadrant, sine

Quantity, complex

Quench oscillator

R

Radian, def. of

Radiation, angle of 407,

Radiation, def. of

Radiation, harmonic

Radiation pattern, distortion

Radiation resistance

Radiator, isotropic

Radiator cross- section, VHF

Radio frequency, def. of

Radio mathematics

Radio propagation

Radio teletype

Ramp function

Ratio detector

Ratio, image

RC amplifier

RC audio network

RC differentialor - integrator

RC oscillator - circuits

RC time constant

RC transient

Reactance, antenna

Reactance, capacitive - inductive - net

Reactance, leakage

Reactance, resonance

Reactance tube modulator

Read -out

Receiver, alignment

Receiver, superheterodyne

Receiver, transistor

Receivers, High -frequency

" Broadcast transistorized

" DX operator

" Mobile transceiver, 10 m.

716

717

718

383

709 99

386

138

137

212

61

133

235, 348

409 -416

414 45

103, 190

604, 121

113

258

522

158

405

356 55 57

234

259

425

214

467 43 49

207

43

456, 474

399

369

474

400, 491

406

475 42

752

409

322

203

319

211

109

139 59

191 60 38

404 46 63 47

312

201

180, 233

208

103, 529

526

529

564

555

804

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" Transceiver, 10 -15 m. 539 " Oscilloscope pattern 181

" Transistorized b.c. 529 " Parasitic 360

Receivers, UHF 229 " Speaker 140

Reception, frequency modulation 317 Resonant cavity 230

Reception, mobile 511 Resonant circuit 53, 54

Reception, SSB 347 Resonant transmission line 420

Recording, constant amplitude, velocity ....136, 137 Response, audio 112, 136

Recording, crossover point 136 Return trace 172

Recording, high fidelity 136 R -F alignment 234

Recording, pre- emphasis - RIAA curve 137 R -F amplifier 211, 249

Rectification, audio 379 " Class AB' 166

Rectification, stray 379 Class B 168

Rectified A.C. 45 Construction 608

Rectifier, bridge 690 Grounded -grid 61 2

" Full -wave 689 Grounded screen 611, 612

" Half -wave 689 Inductive tuning 609

" Hash 694 Instability - neutralization 607, 608

" Mercury vapor - polyphase 693,

" Selenium 694

695

Oscillation

1. Pi- network 608

609

" Silicon 696 Push -pull 604

" Type meter 734 Self -neutralization 608

" Vacuum 693 Tetrode 606

" Voltage doubler 695 R -F chokes 270, 357

" Voltage quadrupler 695 R -F dummy loads 736

" V.t.v.m. 133 R -F feedback 272

Reflected impedance 57, 63 R -F ground 358

Reflected voltage, transmission line 743 R -F shielding 358

Reflection, ionospheric 414 Rhombic antenna 460, 482

Ref lectometer, coaxial 742 Rhumbatron cavity 230

Reflectometer, SWR meter 742 RIAA equalizer curve 137

Regeneration 206 Ribbon TV line 474

Regulated power supply 71 2 "Ribbon" (TV) transmission line 418

Regulation, power line 384 Ring diode modulator 329

Regulation, power supply 715 Ripple factor, filter 688

Regulation, voltage 686 Ripple voltage 686

Regulator noise 524 RL circuit 40

Regulator tube 686 R -L integrator network 199

Regulator tube (VR) 709 RL transient 38

Regulator, voltage 687 RLC circuits 48

Rejection notch, I -F 220 Root- mean -square (a.c.) 45

Rel, def. of 36 Rotary beam antenna 490

Relaxation oscillator 102, 188 Rotator, antenna 508

Reluctance, def. of 36 Ruggedized tube 88

Remote cut -off tube

Residual magnetism 78 37 S

Resistance 23 S- curve, linear amplifier 166

" Capacitance filter 687 Safety bleeder 391

Dynamic 95 Safety precautions 389

1. Gain 94 Saturation current 72

Ground 406 Saturation, magnetic 36

Input 107, 214 Sawtooth wave 59, 172

Internal - parallel - series 25 Scalar notation 48

Load 119 Scale, musical 135

Plate 73, 74 Scatter, ionospheric 414

Radiation 400 Scatter signals 415

Resistive power 29 Screen, CRT 87

Resistivity, table of 23 Screen grid keying 395

Resistor, bleeder 27, 709 Screen grid tube 77

Resistor, characteristics of 352 Screen lead inductance 255

Resistor, color code 527 Screen modulation 286

Resistor, equalizing 34 Screen supply 267

Resistor, inductance of 353 Secondary emission 71, 77

Resistor multiplier 732 Secondary transformer 61

Resistor, non -inductive, use of 661 Selective fading, SSB 327

Resistor, typical 24 Selectivity, arithmetical 209

Resonance 47 Selectivity chart 338

" Antenna 400 Selectivity control, I -F 219

Current 56 Selectivity, mobile reception 513

Curve - parallel - series 53, 54 Selectivity, resonant circuit 54

Filter 688 Selectivity, tuned circuits 339

Impedance -Q 54 Selenium rectifier 695

Mode 230 Self inductance 37

805

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Self neutralization amplifier 608

Selsyn indicator 510

Semi -conductor 90, 21

Semi -conductor, heat sink 104

Semi- conductor rectifier 695

Semi -conductor, zener 105

Series -cathode modulator 295

Series circuit 24

Series -derived filter 64

Series -diode limiter 185

Series feed 271

Series -feed amplifier 604

Series -parallel circuit 25

Series resistance - resonance 25, 53

Shape factor, I -F 217

Shielding, R -F 358

Shot effect 153

Shunt -derived filter 64

Shunt loading, a.v.c. 224

Shunt, meter 731

Shunt neutralization 252

Shunt -regulated bias supply 712

Sidebands, def. of 280

Signal, error 193

Signal, phantom 378

Signal -to- distortion ratio - SSB 153, 340

Signal -to -noise ratio 152

Silicon crystal noise generator 749

Silicon rectifier 696

Sine wave 42, 43

Single -ended amplifier 603

Single sideband (S513) 283, 323, 325

" Envelopes 326

" Filter 324

" Jr. exciter 342

" Measurements 150

" Reception 347

" Transmission 323

Single signal reception 220

Single swing oscillator 189, 190

Single -wire antenna tuner 452

Single -wire feed, antenna 437

Single -wire feeder 426

Six- shooter antenna 468

Skeleton VHF antenna 477

Skin effect 55

Skip distance 414

Sky wave 410

Sleeve antenna 476

Slide -wire bridge 738

Slope detection 208

Slotted transmission line 740

S -meter circuits 224

Soldering techniques 727

Sound in air 134

Space charge 71, 78

Space wave 410

Speaker response 140

Speaker tweeter 139

Specific resistance 23

Speech amplifier construction 677

Speech clipper, circuitry 678

Speech clipper filter 302

Speech clipping 124, 298, 670

Speech filter, high level 303

Speech waveforms 283, 292, 299

Splatter suppressor 302, 669

Sporadic -E propagation 414

Spotter, frequency 740

Spurious emission - spurious pickup 379, 380

Spurious products, mixer 339

Square wave - square wave test 58, 61, 62

SSB, monitor oscilloscope 750

Stacked antenna - stacked dipole 494, 464

Standard frequency 739

Standard frequency, WWV 739

Standard pitch 135

Standing wave 399, 419

Standing wave indicator 741

Static, corona 525

Static plate current 166

Static, wheel 524

Steering diode 103

Step -by -step counter 190

Sterba antenna 465

Storage time, diode 103

Storm, ionospheric 415

Stray capacitance 216

"Strip line" amplifier 595

Stub match, antenna 499

Summation amplifier 200

Summation voltage 197

Summing amplifier 201

Sunspot cycle 414, 415

Superheterodyne receiver 208

Super- regenerative detector 207

Suppression, audio 304

Suppression circuits, parasitics 362

Suppressor 77

Suppressor modulation 290

Suppressor, splatter 300

Surface wave 410

Surge impedance 420

Susceptance, def. of 52

Sweep frequency 172

Switching, transistor action 704

SWR bridge 454, 746

SWR meter, antennascope 747

SWR meter, balanced line 745

SWR meter, bridge 741

SWR meter, reflectometer 742

SWR meter, "twin lamp" 745

Symmetry, amplifier 605

Synchronization, generator 173

Synchronizing voltage 172

Synchronous detection, DSB 349

T

Tank capacitance 259

Tank circuit - efficiency 55, 57, 58

Tank circuit impedance - loading 260, 262

Tank circuit Q 156

Technician class amateur license 12

Teletype, radio 322

Television interference 367

Temperature coefficient 32

Ten -A, SSB exciter 343

Terman- Woodyard amplifier 296

Termination, transmission line 421

Termination, VHF rhombic 483

Test equipment 731

Tetrode, grounded grid use 614

Tetrode modulator 669

Tetrode, zero -bias 683

Tetrode neutralization 254

Tetrode r -f amplifier 606

Tetrode tube 77

Thermal agitation noise 188

Thermionic emission 67

Thermocouple meter 734, 736

Three -phase power supply 517

Threshold voltage 696

Thyratron tube 87

Time -base generator 171

806

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Time constant 38, 60 Transmitter, High- Frequency 576

Time sequence keying 396 " Deluxe 200 -watt 581

T -match antenna 497 " 813 grounded -grid amplifier 627

TNS limiter 228 " 350 -Watt grounded -grid 617

Tools, radio 721 Kilowatt 4 -400A amplifier 649

T.P.T.G. oscillator 239 KW -2 grounded -grid amplifier 634

Trace - cathode ray 86, 172 "Strip- line" amplifier 595

Tracking capacitor 215 Transistorized, 6 m. 577

Transceiver 528 "Tri- band" amplifier 622

Transceivers, High Frequency 526 3 -1000Z amplifier 661

" 10 -15 m. 539 4CX1000A amplifier 654

" 10 m. mobile 555 4 -400A amplifier 643

Transconductance 74, 79 Trapezoidal pattern, oscilloscope 178

Transducer - pickup 137, 138 Trap -type antenna 510

Transformation, impedance 63 Travelling wave tube 84

Transformation ratio 62 Travis discriminator 318

Transformer - ampere -turns 61, 62 Triode amplifier 110

" Antenna match 498 Triode mixer 211

" Auto 63 Triode power amplifier 118

" Coupling 113 Triode tube 72

" I -F 209 Tripler, push -pull 258

" Linear matching 442 Triplex antenna 471

" Matching - impedance match 425, 63 Tropospheric propagation 411

" Modulator 293 T- section filter 64

" Power 709 Tube, vacuum 67

Transformerless power supply 694 Tube, VHF design 232

Transient circuit (a.c.) 59 Tubular transmission line 418

Transient distortion 135 Tuned circuit, coaxial 230

Transient, RC, RL - transient wave 38, 58 Tuned circuit, I -F 216

Transistor 90, 92 Tuned circuit, r.f. amplifier 153

" Bias 96 Tuned circuit Q 214

Code oscillator 19 Tuned circuits, selectivity 339

Complementary circuit 97 Tuner, antenna 452

Drift 93 Tuning, bandspread 215

Equivalent circuit 95 Tuning indicators 224

Heat sink 104, 707 Tuning, inductive 609

Impedance 97 Turnstile antenna 476

Junction 91 TVI filters 369, 374

Mixer

Multivibrator

Oscillator

100

102

101

TVI -proof enclosures - openings

TVI suppression

TVI -type filter

724, 725

371 65

Point -contact 95 Tweeter speaker 139

Power 99 "Twin lamp" SWR meter 745

" Receiver 529 Two -band Marconi antenna 434

" Switching action 704

" 6 m, transmitter 577 u

Transistorized broadcast receiver 529 Ultra- linear amplifier 142

Transistorized mobile supply 703 Unidirectional antenna 500

Transistorized modulator 104 Uni -potential cathode 70

Transit time 81 Unity coupling 269

Transitron oscillator 241 Unloaded Q 57

Transmission line - antennascope

" Balanced, SWR meter 747

745 V

Characteristics 418 Vacuum capacitor 356

Chart 418 Vacuum tube 67

Circuits 229 " Amplification 73, 74

Coaxial 419 Beam power 78

Discontinuities 421 Cathode ray 84

Impedance 417 Classes - classification 107, 87

Incident voltage

Measurements 742

740, 744 Conductance

tl Constant- current curve 73

157

Reflected voltage 743 Diode 71

Resonant - non- resonant 420, 417 Equivalent noise resistance 153

Slotted 740 Foreign 89

Termination 421 Gas 87

" VHF 474 Input loading 152

Transmitter - Control circuits

" Design 387

352 Klystron

is Load line 81 74

" Ground 390 Magic eye 88

" Keying 391 Magnetron 83

" Low Power 576 Mixer 79

" Oscilloscope, monitor of 179 Operation 75

807

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" Parameters

" Pentode

" Plate resistance

" Polarity reversal

" Remote cutoff

Vacuum, Shot effect

" Space charge

" Tetro.Je

" Thyrtron

" Travelling wave

" Triode

" Upper frequency

" Variable mu (A)

" VHF

" Voltage regulator

" Voltmeter

V- Antenna

Variable -mu (11) tube

Variable reluctance pickup

Variac autotronsformer

Vector, sine -wave

Vehicular noise suppression

Velocity modulation

Vertical antenna

Vertical directivity

VHF amplifier

" Antennas

" Antenna polarization

" Antenna relay

" Bands, characteristics

" Corner reflector antenna

" Definition of

73,

130,

106 77 74 76 78

153 89 77 87 84

72

230

154

80 87

734

458

154

137

386 43

523

81

426

407

212

473

475

474 14

481

473

" Discone antenna -ground -plane antenna -.477

" Helical antenna -Horn antenna -- -.479, 482

" Multi- element antenna 484

" Nondirectional antenna

" Parasitics

" Radiation angle - radiation pattern

" Rhombic antenna

" Screen antenna

" Sleeve antenna

" "Strip -line" amplifier

" Transmission line

" Turnstile antenna

" Wavelength table

Vibrator, split -reed

Video amplifier

Volt, def. of

Voltage - Amplifier

" Avalanche

" Break

" Breakdown

" Decay

" Divider

" Divider, phase inverter

" Doubler supply

" Drop, summation

Gradient

Incident

Injection

Initial condition

Instantaneous

Inverse

Measurement

Multiplier power supply

Noise

Output

Quadrupler supply

Reflected

Regulation

Regulator tube 87,

Resonant

478

362

474

482

487

476

595

474

476

475

698

112, 128

22

110

697

203 32 40

26, 52

117

695 28 39

742

211

203 44

693

731, 733

695

151

685

695

742

686, 687

686, 709 54

" Ripple 686

" Summation 197

" Synchronizing 172

" Threshold 696

Voltmeter 732

Voltmeter, A -C V -T 734

Voltmeter, Diode 734

Voltmeter, vacuum tube 130, 734

Volt- ohmmeters 732

Volume compression 670

w

Wagner ground 739

Watt, def. of 29

Wave - Carrier 205

" Ground 410

" Harmonic - nonsinusoidal 58

" Pattern, oscilloscope 180

" Peaked - sawtooth 59

" Sky - space 410

" Square - transient 58

" Surface 410

Waveform 59, 175

Waveform, speech 283, 299, 670

Wavelength, def. of 401

Wavelength table, VHF 475

Wave- shaping circuits 185

Weston cell 24

Wheatstone bridge 738

Wheel static 524

Wien- bridge oscillator 191

Williamson amplifier 141

Wire, litz 55

Wiring hints 527

Wiring technique, audio 142

Workshop practice - layout 720, 729

W8JK antenna 469

WWV Transmissions 739

x

X -array antenna 465

Y

Y, (admittance, def. of) 52

Yogi, adjustment 505

Yagi antenna - Feed systems 494

Yogi antenna, VHF 484

Yogi, constructions 502

Yoke match 494

z

Zener diode 105

Zeppelin antenna 423

Zero -bias modulator 683

Zero -bias tetrode 683

808

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CONVERT SURPLUS RADIO GEAR

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-3 volumes give complete conversion data,

including instructions, photos and diagrams

SURPLUS RADIO CONVERSION MANUALS - 3 Volumes - $3.00 ea. (foreign $3.50)

Contain data which has become standard for the most commonly available surplus items. Each conversion shown

yields a practical piece of equipment . . . proved by testing.

VOLUME I - ARC -5; BC -221, 224, 312. 342, 348, 412, 453/455, 457/459, 624/625, 645, 696, 946B, 1068A/

1161A; PE -103A; SCR -211, 268/271, 274N, 522. 542. TBY; Electronic Surplus Index; Cross Index of A/N V.T.

and Commercial Tubes.

VOLUME II - AIC; AM -26; APS -13; ARB (Schematic only); ARC -5; ART -13; ATC; AVT -112A; BC -I91, 357, 375.

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T -23 /ARC -5; Selenium- Rectifier Power Units; Simplified Coil- Winding Data; Surplus Beam Rotating Mechanisms.

VOLUME III - APN -1; ARC -5; ART -13; BC -191, 312. 342, 348, 375, 442, 453, 455, 456 to 459, 603, 624,

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MD7 /ARC -5; RM -52, 53: R -9 /APN -4; R-28 /ARC -5; RT -19 /ARC -4; RT -159; SCR -274N series; SCR -508. 522,

528, 538; T -15 to T -23 /ARC -5; URC -4; WE- 701 -A. Schematics only: APA -10; APT -2; APT -5; ARR -2; ASB -5;

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THE SURPLUS HANDBOOK (Receivers and Transceivers) - $3.00 (foreign $3.50)

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ARC -5 (V.H.F.); ARN -5; ARR -2; ASB -7; BC -222, 312, 314, 342, 344, 348. 603, 611, 624. 652, 654, 659. 669,

683, 728, 745, 764. 779, 794, 923, 1000, 1004, 1066, 1206, 1306, 1335; BC -AR -231; CRC -7; DAK -3; GF -11;

Mark Il; MN-26; RAK -5; RAL -5; RAX -1; SCR -522; Super Pro; TBY; TCS; Resistor and Capacitor Color Codes; Cross

Index of A/N V.T. and Commercial Tubes.

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WORLD'S RADIO TUBES

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Characteristics of all existing radio tubes

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authoritative tube book. All types classified

numerically and alphabetically. To find a

given tube is only a matter of seconds! Also

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Characteristics of all TV picture and cathode

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tubes. Invaluable to technicians and

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Radio Handbook 16 1962

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