RT7291A, RT7291B Datasheet. Www.s Manuals.com. R02 Richtek

User Manual: Marking of electronic components, SMD Codes 3906, 393I, 3958NL, 399*, 39=**, 39=***, 39A, 39A*, 39B*, 39C*, 39D*, 39E*, 39F*, 39G*, 39H*, 39I*, 39J*, 39K*, 39L*, 39M*, 39N*, 39O*, 39P*, 39Q*, 39R*, 39S*, 39T, 39T*, 39U*, 39V*, 39Y. Datasheets 1.5SMC39AT3, APL5309-15A, APL5309-15B, APL5309-16A, APL5309-16B, APL5309-17A, APL5309-17B, APL5309-18A, APL5309-18B, APL5309-19A, APL5309-19B, APL5309-20A, APL5309-20B, APL5309-21A, APL5309-21B, APL5309-22A, APL5309-22B, APL5309-23A, APL5309-23B, APL5309-24A, APL5309-2

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RT7291A/B
6A, 23V, 500kHz, ACOTTM Synchronous Buck Converter
with LDO for System 5V
General Description

Features

The RT7291A/B is a synchronous Buck converter with
Advanced Constant On-Time (ACOTTM) mode control. The
main control loop of RT7291A/B uses an ACOTTM mode
control which provides a very fast transient response with
no external compensators. The RT7291A/B operates from
5V to 23V input voltage, provides a 5V LDO and a 300kHz
CLK to drive an external charge pump. OCP, UVP and
OVP are included in the RT7291A/B. This IC also provides
a 1.5ms internal soft-start function and an open-drain power
good indicator.

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
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

Applications










Laptop Computers
Tablet PCs
Networking Systems
Servers
Personal Video Recorders
Flat Panel Television and Monitors
Distributed Power Systems



5V to 23V Input Voltage Range
Up to 98% Duty for 2S Battery Application
PWM Frequency Fixed 500kHz
ACOTTM Mode Performs Fast Transient Response
Integrated MOSFETs
 31mΩ
Ω of High-Side MOSFET
 20mΩ
Ω of Low-Side MOSFET
Support Output MLCC Stable
Internal Soft-Start (1.5ms typ)
Built-in OVP/UVP/OCP
Power Good Indicator
Fixed 300kHz VCLK to Support Charge Pump
Individual EN for PWM and LDO
Thermal Shutdown

Simplified Application Circuit
D1

VOUT

VCP
C1

CIN
VLDO

VOUT

CLDO

CB L
VOUT

SW
COUT

VOUT
PGND

RBYP

VBYP

RPGOOD
PGOOD
VCC

ENLDO AGND

September 2015

BOOT

RT7291A/B
EN
LDO

CBYP

DS7291A/B-02

CLK

VIN

VCC
CVCC

is a registered trademark of Richtek Technology Corporation.

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1

RT7291A/B
Ordering Information

Pin Configurations

RT7291A/B

AGND
EN
ENLDO
VCC
BOOT

(TOP VIEW)

Package Type
QUF : UQFN-16L 3x3 (FC) (U-Type)
Lead Plating System
G : Green (Halogen Free and Pb Free)

14 13 12 11 10

Output Voltage
A : 5V
B : 5.1V

Note :

VIN

PGND

15
16

Richtek products are :

RoHS compliant and compatible with the current requirements of IPC/JEDEC J-STD-020.



VBYP
PGOOD
CLK
LDO
VOUT



Suitable for use in SnPb or Pb-free soldering processes.

UQFN-16L 3x3 (FC)

Marking Information
RT7291AGQUF

RT7291BGQUF
39= : Product Code

39=YM
DNN

YMDNN : Date Code

4M= : Product Code

4M=YM
DNN

YMDNN : Date Code

Functional Pin Description
Pin No.

Pin Name

Pin Function

VIN

Power Input Connect to High-Side MOSFET Drain.

PGND

VBYP

Power Ground.
Switch Over Source Voltage for VCC. A low pass filter should be connected to
AGND, if VBYP is applied. If VBYP is not used, then connect to AGND. Do not
connect to VCC pin.

PGOOD

Open-Drain Power Good Indicator Output.

CLK

300kHz Clock Output to Drive the External Charge Pump.

LDO

5V Linear Regulator Output. Decouple with a minimum 4.7F ceramic capacitor.

VOUT

8, 9, 15, 16

Output Voltage Sense Input. An internal discharging circuit is connected to this
pin.
Switch Node.

BOOT

Bootstrap Supply for High-Side Gate Driver. A capacitor is needed to drive the
power switch's gate above the supply voltage. It is connected between the SW
and BOOT pins to form a floating supply across the power switch driver.

VCC

5V Linear Regulator Output for Internal Control Circuit. A capacitor (typical 2.2F)
should be connected to AGND. VCC can only supply internal circuits. Do not
connect to external loads.

ENLDO

Enable Control Input for Linear Regulator. This pin is internally pulled up to high by
10A.

Enable Control Input. Do not leave this pin floating.

AGND

Analog Ground.

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is a registered trademark of Richtek Technology Corporation.

DS7291A/B-02

September 2015

RT7291A/B
Function Block Diagram
VBYP

VCC

VIN
POR &
Reference

Soft-Start
VOUT
VREF
VFB

BSTREG

VCC
Switch-Over

+
+
-

BOOT
VIN

On-Time
One shot

Gate
Control
Logic

Min off Time

SW
VCC

EN
PGND

VOUT
SW
VOC

120% x VREF
-

OCP
PGOOD

OVP

Fault
Logic

90% x VREF

POK

AGND

60% x VREF

UVP

VOUT
CLK

CLK
Generator

VOUT

VCC
VCC
Regulator

VIN

LDO
Control

LDO
Switch-Over

ENLDO LDO

Operation
Overall

OCP

The RT7291A/B is a synchronous step-down converter
with advanced constant on-time control mode. Using the
ACOTTM control mode can reduce the output capacitance
and provide fast transient response. It can minimize the
component size without additional external compensation
network.

The inductor valley current is monitored via the internal
switches in cycle-by-cycle. Once the output voltage drops
below UV threshold, the device enters latch mode.
Power Good
After soft-start is finished, the power good function will be
activated. The PGOOD pin is an open-drain output.

Internal VCC Regulator
The regulator provides 5V power to supply the internal
control circuit. Connecting a 2.2μF ceramic capacitor for
decoupling and stability is required.

CLK Generator
Provide a 300kHz clock to drive external charge pump.
VCC Switch-Over

Soft-Start
In order to prevent the converter output voltage from
overshooting during the startup period, the soft-start
function is necessary. The soft-start time is internal setting
and the duration is around 1.5ms

DS7291A/B-02

September 2015

The internal regulator output will switch over to VBYP if
VBYP level is higher than 4.6V.
LDO
Built-in 5V, 100mA LDO with 1% accuracy. The LDO output
will switch over to VOUT once PGOOD goes high.
is a registered trademark of Richtek Technology Corporation.

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3

RT7291A/B
Absolute Maximum Ratings















(Note 1)

Supply Input Voltage, VIN ---------------------------------------------------------------------------------- −0.3V to 27V
Switch Voltage, SW ----------------------------------------------------------------------------------------- −0.3V to (V IN + 0.3V)
<30ns ----------------------------------------------------------------------------------------------------------- −5V to 28V
BOOT Switch Voltage --------------------------------------------------------------------------------------- (VSW − 0.3V) to (VSW + 6V)
EN, ENLDO Pin Voltages ---------------------------------------------------------------------------------- −0.3V to 27V
Other I/O Pin Voltages -------------------------------------------------------------------------------------- −0.3V to 6V
Power Dissipation, PD @ TA = 25°C
UQFN-16L 3x3 (FC) ------------------------------------------------------------------------------------------ 1.4W
Package Thermal Resistance (Note 2)
UQFN-16L 3x3 (FC), θJA ------------------------------------------------------------------------------------ 70°C/W
UQFN-16L 3x3 (FC), θJC ------------------------------------------------------------------------------------ 15°C/W
Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------ 260°C
Junction Temperature ---------------------------------------------------------------------------------------- 150°C
Storage Temperature Range ------------------------------------------------------------------------------- −65°C to 150°C
ESD Susceptibility (Note 3)
HBM (Human Body Model) --------------------------------------------------------------------------------- 2kV
MM (Machine Model) ---------------------------------------------------------------------------------------- 200V

Recommended Operating Conditions




(Note 4)

Supply Input Voltage, VIN ---------------------------------------------------------------------------------- 5V to 23V
Junction Temperature Range ------------------------------------------------------------------------------- −40°C to 125°C
Ambient Temperature Range ------------------------------------------------------------------------------- −40°C to 85°C

Electrical Characteristics
(VIN = 12V, TA = 25°C, unless otherwise specified)

Parameter

Symbol

Test Conditions

Min

Typ

Max

Unit

Supply Current
Shutdown Current

VEN = VENLDO = 0V

2.5

A

Quiescent Current

VEN = 2V, VENLDO = 2V, No Switching

100

130

A

Standby Current

VEN = 0V, VENLDO = 2V, LDO Load
Current = 0A

A

RDS(ON)_H VBOOT – VSW = 5V

RDS(ON)_L

7.6

11.4

450

500

550

kHz

200

115

120

125

s

Switch On-Resistance
Switch On-Resistance

m

Current Limit
Current Limit

IOC

Valley current of low-side switch

Switching Frequency and Minimum Off Timer
Switching Frequency

f SW

Minimum Off-Time

TOFF

Protections
OVP Trip Threshold

VOVP

OVP Propagation Delay

TOVPDLY

With respect to output voltage

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DS7291A/B-02

September 2015

RT7291A/B
Parameter

Symbol

UVP Trip Threshold

VUVP

UVP Propagation Delay

TUVPDLY

Test Conditions

Min

Typ

Max

Unit

s

RT7291A

4.95

5.05

RT7291B

5.049

5.1

5.151

1.5

RT7291A

1.25

1.35

1.45

RT7291B

1.3

1.4

1.5

200

VEN = 2V

VEN = 0V

With respect to output voltage

Reference and Soft-Start
Output Voltage Valley

VOUT

Soft-Start Time

TSS

From EN high to PGOOD high

V
ms

Enable and UVLO
EN Input High Voltage

VENH

EN Hysteresis

VENHYS

EN Input Current

IEN

VCC UVLO Rising

VCCUVLO

4.2

VCC UVLO Hysteresis

VCCHYS

400

RT7291A

5.05

RT7291B

5.151

0.1

0.2

300

RT7291A

4.95

5.05

RT7291B

5.049

5.1

5.151

EN = GND,
LDO Load Current = 5mA

EN = GND,
LDO Load Current = 100mA

RT7291A

4.805

5.295

RT7291B

4.905

5.1

5.395

RT7291A

4.45

4.6

4.75

RT7291B

4.542

4.692

4.842

200



mV
A

CLK Output
CLK Output
Voltage

High-Level

VCLKH

IVCLK = 10mA

Low-Level

VCLKL

IVCLK = 10mA

CLK Frequency

f CLK

kHz

LDO Regulator
LDO Regulator

VLDO

LDO Load Regulation

Switch On-Resistance

RSW



VCC Regulator
VCC Regulator

VVCC

VCC Switch Over Threshold to
VBYP

VBYP Rising
Edge

VCC Switch Over Hysteresis

VBYP Falling Edge

Switch Over On-Resistance

DS7291A/B-02

September 2015

is a registered trademark of Richtek Technology Corporation.

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5

RT7291A/B
Parameter

Symbol

Test Conditions

Min

Typ

Max

Unit

Power Good Indicator
PGOOD Threshold From Lower

VOUT Rising

PGOOD Low Hysteresis

VOUT Falling

10

0.5

PGOOD Low to High Delay

TPGDLY

PGOOD Sink Current Capability VPGSINK

Sink 4mA

0.4

PGOOD Leakage Current

VPGOOD = 5V

100

135

150

°C

IPGLEAK

Thermal Shutdown
Thermal Shutdown Threshold

TSD

Thermal Shutdown Hysteresis

Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in
the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may
affect device reliability.
Note 2. θJA is measured at TA = 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7.
Note 3. Devices are ESD sensitive. Handling precaution is recommended.
Note 4. The device is not guaranteed to function outside its operating conditions.

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is a registered trademark of Richtek Technology Corporation.

DS7291A/B-02

September 2015

RT7291A/B
Typical Application Circuit
D1

C1
100nF
/ 50V

C2
100nF
/ 50V

C3
100nF
/ 50V

VOUT

VIN
5.2V to 23V

VLDO
5V

VCP

1
CIN
10µF
/25V x 2

5
CLK

VIN

C5
100nF
/ 50V

BOOT

RB
2.2
10 (Optional)

RT7291A/B
13
6

CLDO
4.7µF /10V

EN
LDO

8, 9, 15, 16

CB
0.1µF / 50V
L
1.5µH

VOUT 7
2
PGND

RBYP
5.1
VOUT

C4
100nF
/ 50V

3
CBYP
2.2µF /
10V

VBYP

12 ENLDO

AGND

COUT
22µF /
6.3V x 4

VOUT
5V/6A

PGOOD 4
VCC 11

RPGOOD
100k
VCC
CVCC
2.2µF /
10V

Figure 1. Typical Application Circuit with Pure MLCC Solution

DS7291A/B-02

September 2015

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RT7291A/B
Typical Operating Characteristics
Efficiency vs. Load Current
100

Efficiency vs. Load Current
100

VIN = 7.4V, EN = 2V, ENLDO = floating

VIN = 12V, EN = 2V, ENLDO = floating
95

Efficiency (%)

90
85
80

85
75
80
0.001

0.01

0.1

70
0.001

0.01

Load Current (A)

Switching Frequency vs. Load Current
550

Switching Frequency (kHz)1

VIN = 19V, EN = 2V, ENLDO = floating

Efficiency (%)

Load Current (A)

Efficiency vs. Load Current
100

0.1

90
85
80
75

VIN = 7.4V, EN = 2V, ENLDO = floating

500
450
400
350
300
250
200
150
100
50

70
0.001

0.01

0.1

0
0.001

0.01

Load Current (A)

Switching Frequency vs. Load Current
VIN = 12V, EN = 2V, ENLDO = floating

500
450
400
350
300
250
200
150
100
50
0
0.001

500

VIN = 19V, EN = 2V, ENLDO = floating

450
400
350
300
250
200
150
100
50

0.01

0.1

Load Current (A)

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Switching Frequency vs. Load Current
550

Switching Frequency (kHz)1

550

0.1

Load Current (A)

0
0.001

0.01

0.1

Load Current (A)

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DS7291A/B-02

September 2015

RT7291A/B
Standby Current vs. Input Voltage
80

Standby Current (µA)

Quiescent Current (µA) 1

Quiescent Current vs. Input Voltage
100

70
60
50
40
30
20

60
50
40
30
20
10

EN = 2V, ENLDO = floating, No Switching

EN = 0V, ENLDO = 2V, ILDO = 0A
0

0
5

Input Voltage (V)

Shutdown Current vs. Input Voltage

Output Voltage vs. Load Current

5.25

5.20

5.15

Output Voltage (V)

Shutdown Current (µA)1

7
6
5
4
3

5.10
5.05
5.00
4.95
4.90
4.85

2
1

EN = ENLDO = 0V

0
5

4.80
4.75
0.001

VIN = 12V, EN = 2V, ENLDO = floating
0.01

0.1

Input Voltage (V)

Load Current (A)

LDO Output Voltage vs. Input Voltage

Start Up Through EN

5.25

LDO Output Voltage (V)

5.20

VOUT
(5V/Div)

5.15
5.10
5.05

V CC
(5V/Div)
PGOOD
(5V/Div)

5.00
4.95
4.90

ILDO = 0mA
ILDO = 50mA
ILDO = 100mA

4.85
4.80

EN
(5V/Div)

VIN = 12V, ENLDO = GND, No Load

4.75
4

Time (500μs/Div)

Input Voltage (V)

DS7291A/B-02

September 2015

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RT7291A/B
Start Up Through ENLDO

VLDO
(5V/Div)

VOUT
(5V/Div)

V CC
(5V/Div)

VCP
(5V/Div)
ENLDO
(5V/Div)

Power Off Through EN

VIN = 12V, EN = GND, No Load

PGOOD
(5V/Div)
EN
(5V/Div)

Time (500μs/Div)

Power Off Through ENLDO

Load Transient Response

VLDO
(5V/Div)

VOUT
(30mV/Div)

V CC
(5V/Div)
VCP
(5V/Div)

SW
(20V/Div)

ENLDO
(5V/Div)

VIN = 12V, ENLDO = GND, No Load

VIN = 12V, EN = GND, No Load

IL
(5A/Div)

VIN = 12V, EN = ENLDO = High

Time (500μs/Div)

Time (50μs/Div)

UVP

OVP

VIN = 12V, EN = ENLDO = High

VOUT
(5V/Div)

PGOOD
(5V/Div)

SW
(10V/Div)

VIN
(10V/Div)
IL
(5A/Div)
Time (50μs/Div)

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VIN = 12V, VOUT = 6V, EN = ENLDO = High

Time (50μs/Div)

is a registered trademark of Richtek Technology Corporation.

DS7291A/B-02

September 2015

RT7291A/B
Application Information
The RT7291A/B is high-performance 500kHz 6A step-down
regulators with internal power switches and synchronous
rectifiers. It features an Advanced Constant On-Time
(ACOT TM) control architecture that provides stable
operation for ceramic output capacitors without
complicated external compensation, among other benefits.
The input voltage range is from 5V to 23V, and the output
voltage is fixed 5V.
The proprietary ACOT TM control scheme improves
conventional constant on-time architectures, achieving
nearly constant switching frequency over line, load, and
output voltage ranges. Since there is no internal clock,
response to transients is nearly instantaneous and inductor
current can ramp quickly to maintain output regulation
without large bulk output capacitance.
ACOTTM Control Architecture
In order to achieve good stability with low-ESR ceramic
capacitors, ACOTTM uses a virtual inductor current ramp
generated inside the IC. This internal ramp signal replaces
the ESR ramp normally provided by the output capacitor's
ESR. The ramp signal and other internal compensations
are optimized for low-ESR ceramic output capacitors.
Making the on-time proportional to VOUT and inversely
proportional to VIN is not sufficient to achieve good
constant-frequency behavior for several reasons. First,
voltage drops across the MOSFET switches and inductor
cause the effective input voltage to be less than the
measured input voltage and the effective output voltage to
be greater than the measured output voltage as sensing
input and output voltage. When the load changes, the
switch voltage drops change causing a switching
frequency variation with load current. Also, at light loads
if the inductor current goes negative, the switch deadtime between the synchronous rectifier turn-off and the
high-side switch turn-on allows the switching node to rise
to the input voltage. This increases the effective on-time
and causes the switching frequency to drop noticeably.
One way to reduce these effects is to measure the actual
switching frequency and compare it to the desired range.
This has the added benefit eliminating the need to sense

DS7291A/B-02

September 2015

the actual output voltage, potentially saving one pin
connection. The ACOTTM uses this method, measuring
the actual switching frequency and modifying the on-time
with a feedback loop to keep the average switching
frequency in the desired range.
ACOTTM One-shot Operation
The RT7291A/B control algorithm is simple to understand.
The feedback voltage, with the virtual inductor current ramp
added, is compared to the reference voltage. When the
combined signal is less than the reference, the on-time
one-shot is triggered, as long as the minimum off-time
one-shot is clear and the measured inductor current
(through the synchronous rectifier) is below the current
limit. The on-time one-shot turns on the high-side switch
and the inductor current ramps up linearly. After the ontime, the high-side switch is turned off and the synchronous
rectifier is turned on and the inductor current ramps down
linearly. At the same time, the minimum off-time one-shot
is triggered to prevent another immediate on-time during
the noisy switching time and allow the feedback voltage
and current sense signals to settle. The minimum off-time
is kept short (200ns typical) so that rapidly-repeated ontimes can raise the inductor current quickly when needed.
Diode Emulation Mode
In diode emulation mode, the RT7291A/B automatically
reduces switching frequency at light load conditions to
maintain high efficiency. This reduction of frequency is
achieved smoothly. As the output current decreases from
heavy load conditions, the inductor current is also reduced,
and eventually comes to the point that its current valley
touches zero, which is the boundary between continuous
conduction and discontinuous conduction modes. To
emulate the behavior of diodes, the low-side MOSFET
allows only partial negative current to flow when the
inductor free wheeling current becomes negative. As the
load current is further decreased, it takes longer and longer
time to discharge the output capacitor to the level that
requires the next “ON” cycle. In reverse, when the output
current increases from light load to heavy load, the
switching frequency increases to the preset value as the

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RT7291A/B
inductor current reaches the continuous conduction. The
transition load point to the light load operation is shown in
Figure 2 and can be calculated as follows :
IL
Slope = (VIN - VOUT) / L
IPEAK

ILOAD = IPEAK / 2

When VOUT is powered on and PGOOD is pulled high,
an internal 3Ω P-MOSFET switch connects VOUT to the
LDO pin while the internal linear regulator is simultaneously
turned off.
The RT7291A/B also includes a 5V linear regulator (VCC).
The VCC regulator steps down input voltage to supply
both internal circuitry and gate drivers. Do not connect
the VCC pin to external loads. When PGOOD is pulled
high and BYP pin voltage is above 4.6V, an internal 3Ω
P-MOSFET switch connects VCC to the BYP pin while
the VCC linear regulator is simultaneously turned off.

tON

Figure 2. Boundary Condition of CCM/DEM
(V  VOUT )
ILOAD  IN
 tON
2L
where tON is the on-time.

The switching waveforms may appear noisy and
asynchronous when light load causes diode emulation
operation. This is normal and results in high efficiency.
Trade offs in DEM noise vs. light load efficiency is made
by varying the inductor value. Generally, low inductor values
produce a broader efficiency vs. load curve, while higher
values result in higher full load efficiency (assuming that
the coil resistance remains fixed) and less output voltage
ripple. Penalties for using higher inductor values include
larger physical size and degraded load transient response
(especially at low input voltage levels).
During discontinuous switching, the on-time is immediately
increased to add “hysteresis” to discourage the IC from
switching back to continuous switching unless the load
increases substantially. The IC returns to continuous
switching as soon as an on-time is generated before the
inductor current reaches zero. The on-time is reduced back
to the length needed for 500kHz switching and encouraging
the circuit to remain in continuous conduction, preventing
repetitive mode transitions between continuous switching
and discontinuous switching.
Linear Regulators (LDO & VCC)
The RT7291A/B includes a 5V linear regulator (LDO). The
LDO regulator can supply up to 100mA for external loads.
Bypass LDO with a minimum 4.7μF ceramic capacitor.

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Current Limit
The RT7291A/B current limit is a cycle-by-cycle “valley”
type, measuring the inductor current through the
synchronous rectifier during the off-time while the inductor
current ramps down. The current is determined by
measuring the voltage between Source and Drain of the
synchronous rectifier, adding temperature compensation
for greater accuracy. If the current exceeds the current
limit, the on-time one-shot is inhibited until the inductor
current ramps down below the current limit. Thus, only
when the inductor current is well below the current limit,
another on-time is permitted. If the output current exceeds
the available inductor current (controlled by the current
limit mechanism), the output voltage will drop. If it drops
below the output under-voltage protection level (see next
section), the IC will stop switching to avoid excessive
heat.
The RT7291A/B also features a negative current limit to
protect the IC against sinking excessive current and
possibly damage. If the voltage across the synchronous
rectifier indicates the negative current is too high, the
synchronous rectifier turns off.
Output Over-Voltage Protection and Under-Voltage
Protection
The RT7291A/B features an output Over-Voltage Protection
(OVP). If the output voltage rises above the regulation
level, the IC stops switching and is latched off. The
RT7291A/B also features an output Under-Voltage
Protection (UVP). If the output voltage drops below the
UVP trip threshold for longer than 2μs (typical), the UVP

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RT7291A/B
is triggered, and the IC will shut down. The IC stops
switching and is latched off. To restart operation, toggle
EN or power the IC off and then on again.
Input Under-Voltage Lockout
In addition to the enable function, the RT7291A/B features
an Under-Voltage Lockout (UVLO) function that monitors
the input voltage. To prevent operation without fullyenhanced internal MOSFET switches, this function inhibits
switching when input voltage drops below the UVLO-falling
threshold. The IC resumes switching when input voltage
exceeds the UVLO-rising threshold.
Over-Temperature Protection
The RT7291A/B features an Over-Temperature Protection
(OTP) circuitry to prevent overheating due to excessive
power dissipation. The OTP shuts down switching
operation when the junction temperature exceeds 150°C.
Once the junction temperature cools down by
approximately 25°C the IC resumes normal operation with
a complete soft-start. For continuous operation, provide
adequate cooling so that the junction temperature does
not exceed 150°C. Note that the VCC and LDO regulator
remains on as the OTP is triggered.
Enable and Disable
The enable input (EN) has a logic-low level of 1.15V. When
VEN is below this level, the IC enters shutdown mode and
supply current drops to less than 5μA (typical). When
VEN exceeds its logic-high level of 1.35V, the IC is fully
operational. The logics of EN and ENLDO to control the
VOUT, CLK, LDO and VCC are stated in Table 1.
Table 1. EN/ENLDO Control Logics

ENLDO

VOUT/CLK

LDO

VCC

OFF

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Soft-Start
The RT7291A/B provides an internal soft-start function to
prevent large inrush current and output voltage overshoot
when the converter starts up. The soft-start (SS)
automatically begins once the chip is enabled. During softstart, it clamps the ramp of internal reference voltage which
is compared with FB signal. The typical soft-start duration
is 1.5ms.
Power Off
When VEN is pulled to GND or lower than the logic-low
level of 1.15V, there is an internal discharging resistor to
discharge the residual charge inside the output capacitors.
Besides, the value of discharging resistor is about twenty
ohms.
Power Good Output (PGOOD)
The power good output is an open-drain output that requires
a pull-up resistor. When the output voltage is 20% (typical)
below its set voltage, PGOOD will be pulled low. It is held
low until the output voltage returns to 90% of its set voltage
once more. During soft-start, PGOOD is actively held low
and only allowed to be pulled high after soft-start is over
and the output reaches 90% of its set voltage. There is a
2μs delay built into PGOOD circuitry to prevent false
transition.
External Bootstrap Capacitor (CBOOT)
Connect a 0.22μF low ESR ceramic capacitor between
the BOOT and SW pins. This bootstrap capacitor provides
the gate driver supply voltage for the high-side N-MOSFET
switch.
The internal power MOSFET switch gate driver is
optimized to turn the switch on fast enough for low power
loss and good efficiency, and slow enough to reduce EMI.
Switch turn-on is when most EMI occurs since VSW rises
rapidly. During switch turn-off, SW is discharged relatively
slowly by the inductor current during the dead-time
between high-side and low-side switch on-times. In some
cases it is desirable to reduce EMI further, at the expense
of some additional power dissipation. The switch turn-on
can be slowed by placing a small (<10Ω) resistance
between BOOT and the external bootstrap capacitor. This
will slow the high-side switch turn-on and VSW's rise.
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RT7291A/B
Inductor Selection

Input Capacitor Selection

Selecting an inductor involves specifying its inductance
and also its required peak current. The exact inductor value
is generally flexible and is ultimately chosen to obtain the
best mix of cost, physical size, and circuit efficiency.
Lower inductor values benefit from reduced size and cost
and they can improve the circuit's transient response.
However, they increase the inductor ripple current and
output voltage ripple and reduce the efficiency due to the
resulting higher peak currents. Conversely, higher inductor
values increase efficiency, but the inductor will either be
physically larger or have higher resistance since more
turns of wire are required and transient response will be
slower since more time is required to change current (up
or down) in the inductor. A good compromise between
size, efficiency, and transient response is to use a ripple
current (ΔIL) about 20-50% of the desired full output load
current. Calculate the approximate inductor value by
selecting the input and output voltages, the switching
frequency (fSW), the maximum output current (IOUT(MAX))
and estimating a ΔIL as some percentage of that current.

High quality ceramic input decoupling capacitor, such as
X5R or X7R, with values greater than 20μF are
recommended for the input capacitor. The X5R and X7R
ceramic capacitors are usually selected for power regulator
capacitors because the dielectric material has less
capacitance variation and more temperature stability.
Voltage rating and current rating are the key parameters
when selecting an input capacitor. Generally, selecting an
input capacitor with voltage rating 1.5 times greater than
the maximum input voltage is a conservatively safe design.
The input capacitor is used to supply the input RMS
current, which can be approximately calculated using the
following equation :

V
 (VIN  VOUT )
L  OUT
VIN  fSW  IL
Once an inductor value is chosen, the ripple current (ΔIL)
is calculated to determine the required peak inductor
current.

V
 (VIN  VOUT )
I
IL  OUT
and IL(PEAK)  IOUT(MAX)  L
VIN  fSW  L
2
To guarantee the required output current, the inductor
needs a saturation current rating and a thermal rating that
exceeds IL(PEAK). These are minimum requirements. To
maintain control of inductor current in overload and shortcircuit conditions, some applications may desire current
ratings up to the current limit value. However, the IC's
output under-voltage shutdown feature make this
unnecessary for most applications.
For best efficiency, choose an inductor with a low DC
resistance that meets the cost and size requirements.
For low inductor core losses some type of ferrite core is
usually best and a shielded core type, although possibly
larger or more expensive, it will probably give fewer EMI
and other noise problems.

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14

VOUT 
V
I 2 
 (1  OUT )  IOUT 2  L 
VIN
VIN
12 

The next step is to select a proper capacitor for RMS
IRMS 

current rating. One good design uses more than one
capacitor with low Equivalent Series Resistance (ESR) in
parallel to form a capacitor bank. The input capacitance
value determines the input ripple voltage of the regulator.
The input voltage ripple can be approximately calculated
using the following equation :
VIN 

IOUT  VIN
V
 (1  OUT )
CIN  fSW  VOUT
VIN

The typical operating circuit is recommended to use two
10μF low ESR ceramic capacitors on the input.
Output Capacitor Selection
The RT7291A/B is optimized for ceramic output capacitors
and best performance will be obtained by using them. The
total output capacitance value is usually determined by
the desired output voltage ripple level and transient response
requirements for sag (undershoot on positive load steps)
and soar (overshoot on negative load steps).
Output ripple at the switching frequency is caused by the
inductor current ripple and its effect on the output
capacitor's ESR and stored charge. These two ripple
components are called ESR ripple and capacitive ripple.
Since ceramic capacitors have extremely low ESR and
relatively little capacitance, both components are similar
in amplitude and both should be considered if ripple is
critical.
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RT7291A/B
VRIPPLE  VRIPPLE(ESR)  VRIPPLE(C)
VRIPPLE(ESR)  IL  RESR
VRIPPLE(C) 

IL
8  COUT  fSW

In addition to voltage ripple at the switching frequency,
the output capacitor and its ESR also affect the voltage
sag (undershoot) and soar (overshoot) when the load steps
up and down abruptly. The ACOT transient response is
very quick and output transients are usually small.
However, the combination of small ceramic output
capacitors (with little capacitance), low output voltages
(with little stored charge in the output capacitors), and
low duty cycle applications (which require high inductance
to get reasonable ripple currents with high input voltages)
increases the size of voltage variations in response to
very quick load changes. Typically, load changes occur
slowly with respect to the IC's 500kHz switching frequency.
However, some modern digital loads can exhibit nearly
instantaneous load changes and the following section
shows how to calculate the worst-case voltage swings in
response to very fast load steps.
The amplitude of the ESR step up or down is a function of
the load step and the ESR of the output capacitor :
VESR_STEP  IOUT  RESR
The amplitude of the capacitive sag is a function of the
load step, the output capacitor value, the inductor value,
the input-to-output voltage differential, and the maximum
duty cycle. The maximum duty cycle during a fast transient
is a function of the on-time and the minimum off-time since
the ACOTTM control scheme will ramp the current using
on-times spaced apart with minimum off-times, which is
as fast as allowed. Calculate the approximate on-time
(neglecting parasitics) and maximum duty cycle for a given
input and output voltage as :
VOUT
t ON
t ON 
and DMAX 
VIN  fSW
tON  tOFF(MIN)
The actual on-time will be slightly longer as the IC
compensates for voltage drops in the circuit, but we can
neglect both of these since the on-time increases
compensations for the voltage losses. Calculate the output
voltage sag as :
VSAG 

L  (IOUT )2
2  COUT  ( VIN(MIN)  DMAX  VOUT )

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September 2015

The amplitude of the capacitive soar is a function of the
load step, the output capacitor value, the inductor value
and the output voltage :
VSOAR 

L  ( IOUT )2
2  COUT  VOUT

Most applications never experience instantaneous full load
steps and the RT7291A/B's high switching frequency and
fast transient response can easily control voltage regulation
at all times. Therefore, sag and soar are seldom an issue
except in very low-voltage CPU core or DDR memory
supply applications, particularly for devices with high clock
frequencies and quick changes into and out of sleep
modes. In such applications, simply increasing the amount
of ceramic output capacitor (sag and soar are directly
proportional to capacitance) or adding extra bulk
capacitance can easily eliminate any excessive voltage
transients.
In any application with large quick transients, it should
calculate soar and sag to make sure that over-voltage
protection and under-voltage protection will not be triggered.
Thermal Considerations
For continuous operation, do not exceed absolute
maximum junction temperature. The maximum power
dissipation depends on the thermal resistance of the IC
package, PCB layout, rate of surrounding airflow, and
difference between junction and ambient temperature. The
maximum power dissipation can be calculated by the
following formula :
PD(MAX) = (TJ(MAX) − TA) / θJA
where TJ(MAX) is the maximum junction temperature, TA is
the ambient temperature, and θJA is the junction to ambient
thermal resistance.
For recommended operating condition specifications, the
maximum junction temperature is 125°C. The junction to
ambient thermal resistance, θJA, is layout dependent. For
UQFN-16L 3x3 (FC) package, the thermal resistance, θJA,
is 70°C/W on a standard JEDEC 51-7 four-layer thermal
test board. The maximum power dissipation at TA = 25°C
can be calculated by the following formula :
P D(MAX) = (125°C − 25°C) / (70°C/W) = 1.4W for
UQFN-16L 3x3 (FC) package
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RT7291A/B

Maximum Power Dissipation (W)1

The maximum power dissipation depends on the operating
ambient temperature for fixed T J(MAX) and thermal
resistance, θJA. The derating curve in Figure 3 allows the
designer to see the effect of rising ambient temperature
on the maximum power dissipation.

Layout Considerations
Layout is very important in high frequency switching
converter design. The PCB can radiate excessive noise
and contribute to converter instability with improper layout.
Certain points must be considered before starting a layout
using the RT7291A/B.

2.0

Four-Layer PCB



Make traces of the main current paths as short and wide
as possible.



Put the input capacitor as close as possible to the device
pins (VIN and PGND).



SW node encounters high frequency voltage swings so
it should be kept in a small area. Keep sensitive
components away from the SW node to prevent stray.



The PGND pin should be connected to a strong ground
plane for heat sinking and noise protection.



Avoid using vias in the power path connections that have
switched currents (from CIN to PGND and CIN to VIN)
and the switching node (SW).

1.6

1.2

0.8

0.4

0.0
0

100

125

Ambient Temperature (°C)

Figure 3. Derating Curve of Maximum Power Dissipation

GND
VIN

VIN

ENLDO

VCC

BOOT

The input capacitor must
be placed as close to the
IC as possible.

AGND

CVCC

1
4

1
3

1
2

1
1

1
0

L
15

CIN

PGND

VOUT

VBYP

PGOOD

CLK

LDO

VOUT

GND

The output capacitor must
be placed near the IC

CBOOT

VOUT

COUT

SW should be connected to
inductor by wide and short
trace.
Keep sensitive components
away from this trace.

CLDO

Impedance between PGND and AGND should be as small as possible for unified ground voltage.

Figure 4. Layout Guide

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RT7291A/B
Outline Dimension

Symbol

Dimensions In Millimeters

Dimensions In Inches

Min.

Max.

Min.

Max.

0.500

0.600

0.020

0.024

0.000

0.050

0.000

0.002

0.100

0.175

0.004

0.007

2.900

3.100

0.114

0.122

2.900

3.100

0.114

0.122

0.150

0.250

0.006

0.010

0.100

0.200

0.004

0.008

0.350

0.450

0.014

0.018

0.750

0.850

0.030

0.033

0.550

0.650

0.022

0.026

0.400

0.016

0.975

0.038

1.335

0.053

1.675

0.066

1.935

0.076

0.975

0.038

1.675

0.066

U-Type 16L QFN 3x3 (FC) Package

DS7291A/B-02

September 2015

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RT7291A/B

Richtek Technology Corporation
14F, No. 8, Tai Yuen 1st Street, Chupei City
Hsinchu, Taiwan, R.O.C.
Tel: (8863)5526789
Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should
obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot
assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be
accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third
parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.

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