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User Manual: Marking of electronic components, SMD Codes 3906, 393I, 3958NL, 399*, 39=**, 39=***, 39A, 39A*, 39B*, 39C*, 39D*, 39E*, 39F*, 39G*, 39H*, 39I*, 39J*, 39K*, 39L*, 39M*, 39N*, 39O*, 39P*, 39Q*, 39R*, 39S*, 39T, 39T*, 39U*, 39V*, 39Y. Datasheets 1.5SMC39AT3, APL5309-15A, APL5309-15B, APL5309-16A, APL5309-16B, APL5309-17A, APL5309-17B, APL5309-18A, APL5309-18B, APL5309-19A, APL5309-19B, APL5309-20A, APL5309-20B, APL5309-21A, APL5309-21B, APL5309-22A, APL5309-22B, APL5309-23A, APL5309-23B, APL5309-24A, APL5309-2

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RT7291A/B

DS7291A/B-02 September 2015 www.richtek.com

6A, 23V, 500kHz, ACOTTM Synchronous Buck Converter

with LDO for System 5V

General Description

The RT7291A/B is a synchronous Buck converter with

Advanced Constant On-Time (ACOTTM) mode control. The

main control loop of RT7291A/B uses an ACOTTM mode

control which provides a very fast transient response with

no external compensators. The RT7291A/B operates from

5V to 23V input voltage, provides a 5V LDO and a 300kHz

CLK to drive an external charge pump. OCP, UVP and

OVP are included in the RT7291A/B. This IC also provides

a 1.5ms internal soft-start function and an open-drain power

good indicator.

Simplified Application Circuit

Features

-5V to 23V Input Voltage Range

-Up to 98% Duty for 2S Battery Application

-PWM Frequency Fixed 500kHz

-ACOTTM Mode Performs Fast Transient Response

-Integrated MOSFETs



 31mΩΩ

ΩΩ

Ω of High-Side MOSFET



 20mΩΩ

ΩΩ

Ω of Low-Side MOSFET

-Support Output MLCC Stable

-Internal Soft-Start (1.5ms typ)

-Built-in OVP/UVP/OCP

-Power Good Indicator

-Fixed 300kHz VCLK to Support Charge Pump

-Individual EN for PWM and LDO

-Thermal Shutdown

Applications

-Laptop Computers

-Tablet PCs

-Networking Systems

-Servers

-Personal Video Recorders

-Flat Panel Television and Monitors

-Distributed Power Systems

RT7291A/B

VIN BOOT

ENLDO

VBYP

LDO

CLK

VOUT

PGND

AGND

VOUT

PGOOD

VCC

VCP

VOUT

VIN

VOUT

VLDO

VCC

C1 C2

D2 D3

C3 C4

CIN

CBL

COUT

RPGOOD

CVCC

CLDO

CBYP

RBYP

RT7291A/B

DS7291A/B-02 September 2015www.richtek.com

Functional Pin Description

Pin No. Pin Name Pin Function

1 VIN Power Input Connect to High-Side MOSFET Drain.

2 PGND Power Ground.

3 VBYP

Switch Over Source Voltage for VCC. A low pass filter should be connected to

AGND, if VBYP is applied. If VBYP is not used, then connect to AGND. Do not

connect to VCC pin.

4 PGOOD Open-Drain Power Good Indicator Output.

5 CLK 300kHz Clock Output to Drive the External Charge Pump.

6 LDO 5V Linear Regulator Output. Decouple with a minimum 4.7F ceramic capacitor.

7 VOUT

Output Voltage Sense Input. An internal discharging circuit is connected to this

pin.

8, 9, 15, 16 SW Switch Node.

10 BOOT Bootstrap Supply for High-Side Gate Driver. A capacitor is needed to drive the

power switch's gate above the supply voltage. It is connected between the SW

and BOOT pins to form a floating supply across the power switch driver.

11 VCC 5V Linear Regulator Output for Internal Control Circuit. A capacitor (typical 2.2F)

should be connected to AGND. VCC can only supply internal circuits. Do not

connect to external loads.

12 ENLDO

Enable Control Input for Linear Regulator. This pin is internally pulled up to high by

10A.

13 EN Enable Control Input. Do not leave this pin floating.

14 AGND Analog Ground.

Pin Configurations

(TOP VIEW)

UQFN-16L 3x3 (FC)

Ordering Information

Note :

Richtek products are :

 RoHS compliant and compatible with the current require-

ments of IPC/JEDEC J-STD-020.

 Suitable for use in SnPb or Pb-free soldering processes.

PGND

VIN

PGOOD

LDO

VCC

BOOT

VBYP

ENLDO

CLK

VOUT

AGND

3 4 5 6 7

14 13 12 11 10

16 8

Marking Information

RT7291AGQUF RT7291BGQUF

39=YM

DNN 4M=YM

DNN

39= : Product Code

YMDNN : Date Code

4M= : Product Code

YMDNN : Date Code

Package Type

QUF : UQFN-16L 3x3 (FC) (U-Type)

RT7291A/B

Lead Plating System

G : Green (Halogen Free and Pb Free)

Output Voltage

A : 5V

B : 5.1V

RT7291A/B

DS7291A/B-02 September 2015 www.richtek.com

Function Block Diagram

Operation

Overall

The RT7291A/B is a synchronous step-down converter

with advanced constant on-time control mode. Using the

ACOTTM control mode can reduce the output capacitance

and provide fast transient response. It can minimize the

component size without additional external compensation

network.

Internal VCC Regulator

The regulator provides 5V power to supply the internal

control circuit. Connecting a 2.2μF ceramic capacitor for

decoupling and stability is required.

Soft-Start

In order to prevent the converter output voltage from

overshooting during the startup period, the soft-start

function is necessary. The soft-start time is internal setting

and the duration is around 1.5ms

OCP

The inductor valley current is monitored via the internal

switches in cycle-by-cycle. Once the output voltage drops

below UV threshold, the device enters latch mode.

Power Good

After soft-start is finished, the power good function will be

activated. The PGOOD pin is an open-drain output.

CLK Generator

Provide a 300kHz clock to drive external charge pump.

VCC Switch-Over

The internal regulator output will switch over to VBYP if

VBYP level is higher than 4.6V.

LDO

Built-in 5V, 100mA LDO with 1% accuracy. The LDO output

will switch over to VOUT once PGOOD goes high.

POR &

Reference

VOUT

VCC

On-Time

One shot

VIN LDO

VBYP

Gate

Control

Logic

Fault

Logic

CLK

PGND

VIN

BSTREG BOOT

PGOOD

AGND

Min off Time

Soft-Start

VOUT

OCP

UVP

60% x VREF

POK

OVP

90% x VREF

120% x VREF

VREF

VFB

LDO

Switch-Over

LDO

Control

VCC

Regulator

VCC VOUT

CLK

Generator

VOUT

ENLDO

VCC

Switch-Over

VIN

VOC

VCC

RT7291A/B

DS7291A/B-02 September 2015www.richtek.com

Parameter Symbol Test Conditions Min Typ Max Unit

Supply Current

Shutdown Current V

EN = VENLDO = 0V -- 2.5 5 A

Quiescent Current V

EN = 2V, VENLDO = 2V, No Switching -- 100 130 A

Standby Current VEN = 0V, VENLDO = 2V, LDO Load

Current = 0A -- 35 45 A

Switch On-Resistance

Switch On-Resistance RDS(ON)_H V

BOOT – VSW = 5V -- 31 --

m

RDS(ON)_L -- 20 --

Current Limit

Current Limit IOC Valley current of low-side switch 7.6 -- 11.4 A

Switching Frequency and Minimum Off Timer

Switching Frequency fSW 450 500 550 kHz

Minimum Off-Time TOFF -- 200 -- ns

Protections

OVP Trip Threshold VOVP With respect to output voltage 115 120 125 %

OVP Propagation Delay TOVPDLY -- 5 -- s

Electrical Characteristics

(VIN = 12V, TA = 25°C, unless otherwise specified)

Absolute Maximum Ratings (Note 1)

-Supply Input Voltage, VIN ---------------------------------------------------------------------------------- −0.3V to 27V

-Switch Voltage, SW ----------------------------------------------------------------------------------------- −0.3V to (VIN + 0.3V)

<30ns ----------------------------------------------------------------------------------------------------------- −5V to 28V

-BOOT Switch Voltage ---------------------------------------------------------------------------------------(VSW − 0.3V) to (VSW + 6V)

-EN, ENLDO Pin Voltages ---------------------------------------------------------------------------------- −0.3V to 27V

-Other I/O Pin Voltages -------------------------------------------------------------------------------------- −0.3V to 6V

-Power Dissipation, PD @ TA = 25°C

UQFN-16L 3x3 (FC) ------------------------------------------------------------------------------------------1.4W

-Package Thermal Resistance (Note 2)

UQFN-16L 3x3 (FC), θJA ------------------------------------------------------------------------------------ 70°C/W

UQFN-16L 3x3 (FC), θJC ------------------------------------------------------------------------------------ 15°C/W

-Lead Temperature (Soldering, 10 sec.)------------------------------------------------------------------260°C

-Junction Temperature ----------------------------------------------------------------------------------------150°C

-Storage Temperature Range ------------------------------------------------------------------------------- −65°C to 150°C

-ESD Susceptibility (Note 3)

HBM (Human Body Model)--------------------------------------------------------------------------------- 2kV

MM (Machine Model) ---------------------------------------------------------------------------------------- 200V

Recommended Operating Conditions (Note 4)

-Supply Input Voltage, VIN ----------------------------------------------------------------------------------5V to 23V

-Junction Temperature Range------------------------------------------------------------------------------- −40°C to 125°C

-Ambient Temperature Range ------------------------------------------------------------------------------- −40°C to 85°C

RT7291A/B

DS7291A/B-02 September 2015 www.richtek.com

Parameter Symbol Test Conditions Min Typ Max Unit

UVP Trip Threshold VUVP With respect to output voltage 55 60 65 %

UVP Propagation Delay TUVPDLY -- 5 -- s

Reference and Soft-Start

Output Voltage Valley VOUT RT7291A 4.95 5 5.05

RT7291B 5.049 5.1 5.151

Soft-Start Time TSS From EN high to PGOOD high 1 1.5 2 ms

Enable and UVLO

EN Input High Voltage VENH RT7291A 1.25 1.35 1.45

RT7291B 1.3 1.4 1.5

EN Hysteresis VENHYS -- 200 -- mV

EN Input Current IEN VEN = 2V -- 1 -- A

VEN = 0V -- 0 --

VCC UVLO Rising VCCUVLO -- 4.2 -- V

VCC UVLO Hysteresis VCCHYS -- 400 -- mV

CLK Output

Voltage

High-Level VCLKH I

VCLK = 10mA RT7291A -- -- 5.05

RT7291B -- -- 5.151

Low-Level VCLKL I

VCLK = 10mA 0 0.1 0.2

CLK Frequency fCLK -- 300 -- kHz

LDO Regulator

LDO Regulator VLDO RT7291A 4.95 5 5.05

RT7291B 5.049 5.1 5.151

LDO Load Regulation

EN = GND,

LDO Load Current = 5mA -- 1 --

EN = GND,

LDO Load Current = 100mA -- 5 --

Switch On-Resistance RSW -- 3 5 

VCC Regulator

VCC Regulator VVCC RT7291A 4.805 5 5.295

RT7291B 4.905 5.1 5.395

VCC Switch Over Threshold to

VBYP VBYP Rising

Edge

RT7291A 4.45 4.6 4.75 V

RT7291B 4.542 4.692 4.842

VCC Switch Over Hysteresis VBYP Falling Edge -- 200 -- mV

Switch Over On-Resistance -- 3 5 

RT7291A/B

DS7291A/B-02 September 2015www.richtek.com

Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are

stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in

the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may

affect device reliability.

Note 2. θJA is measured at TA = 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7.

Note 3. Devices are ESD sensitive. Handling precaution is recommended.

Note 4. The device is not guaranteed to function outside its operating conditions.

Parameter Symbol Test Conditions Min Typ Max Unit

Power Good Indicator

PGOOD Threshold From Lower VOUT Rising 85 90 95 %

PGOOD Low Hysteresis VOUT Falling -- 10 -- %

PGOOD Low to High Delay TPGDLY -- 0.5 -- ms

PGOOD Sink Current Capability VPGSINK Sink 4mA -- -- 0.4 V

PGOOD Leakage Current IPGLEAK V

PGOOD = 5V -- -- 100 nA

Thermal Shutdown

Thermal Shutdown Threshold TSD 135 150 -- °C

Thermal Shutdown Hysteresis -- 25 -- °C

RT7291A/B

DS7291A/B-02 September 2015 www.richtek.com

Typical Application Circuit

Figure 1. Typical Application Circuit with Pure MLCC Solution

RT7291A/B

VIN BOOT

ENLDO

VBYP

LDO

10µF

/25V x 2

CLK

VOUT

PGND

AGND

(Optional)

0.1µF / 50V

1.5µH

22µF /

6.3V x 4

VOUT

5V/6A

PGOOD

VCC

2.2µF /

10V

100k

VCP

VOUT

100nF

/ 50V

VIN

5.2V to 23V

4.7µF /10V

VOUT

VLDO

110

8, 9, 15, 16

11 VCC

C1 C2

D2 D3

C3 C4

CIN

COUT

RPGOOD

CVCC

CLDO

2.2

CBYP

2.2µF /

10V

RBYP

5.1

100nF

/ 50V 100nF

/ 50V

RT7291A/B

DS7291A/B-02 September 2015www.richtek.com

Efficiency vs. Load Current

100

0.001 0.01 0.1 1 10

Load Current (A)

Efficiency (%)

Efficiency vs. Load Current

100

0.001 0.01 0.1 1 10

Load Current (A)

Efficiency (%)

Efficiency vs. Load Current

100

0.001 0.01 0.1 1 10

Load Current (A)

Efficiency (%)

Switching Frequency vs. Load Current

100

150

200

250

300

350

400

450

500

550

0.001 0.01 0.1 1 10

Load Current (A)

Switching Frequency (kHz) 1

Switching Frequency vs. Load Current

100

150

200

250

300

350

400

450

500

550

0.001 0.01 0.1 1 10

Load Current (A)

Switching Frequency (kHz) 1

Switching Frequency vs. Load Current

100

150

200

250

300

350

400

450

500

550

0.001 0.01 0.1 1 10

Load Current (A)

Switching Frequency (kHz) 1

Typical Operating Characteristics

VIN = 7.4V, EN = 2V, ENLDO = floating VIN = 12V, EN = 2V, ENLDO = floating

VIN = 19V, EN = 2V, ENLDO = floating

VIN = 12V, EN = 2V, ENLDO = floating VIN = 19V, EN = 2V, ENLDO = floating

VIN = 7.4V, EN = 2V, ENLDO = floating

RT7291A/B

DS7291A/B-02 September 2015 www.richtek.com

Quiescent Current vs. Input Voltage

100

5 7 9 11131517192123

Input Voltage (V)

Quiescent Current (µA) 1

Standby Current vs. Input Voltage

5 7 9 11131517192123

Input Voltage (V)

Standby Current (µA

)

Shutdown Current vs. Input Voltage

5 7 9 11131517192123

Input Voltage (V)

Shutdown Current (µA) 1

Output Voltage vs. Load Current

4.75

4.80

4.85

4.90

4.95

5.00

5.05

5.10

5.15

5.20

5.25

0.001 0.01 0.1 1 10

Load Current (A)

Output Voltage (V)

Time (500μs/Div)

Start Up Through EN

(5V/Div)

VOUT

(5V/Div)

VCC

(5V/Div)

PGOOD

(5V/Div)

VIN = 12V, ENLDO = GND, No Load

VIN = 12V, EN = 2V, ENLDO = floating

EN = ENLDO = 0V

EN = 0V, ENLDO = 2V, ILDO = 0AEN = 2V, ENLDO = floating, No Switching

LDO Output Voltage vs. Input Voltage

4.75

4.80

4.85

4.90

4.95

5.00

5.05

5.10

5.15

5.20

5.25

4 6 8 1012141618202224

Input Voltage (V)

LDO Output Voltage (V)

ILDO = 0mA

ILDO = 50mA

ILDO = 100mA

RT7291A/B

DS7291A/B-02 September 2015www.richtek.com

Time (500μs/Div)

Start Up Through ENLDO

ENLDO

(5V/Div)

VLDO

(5V/Div)

VCC

(5V/Div)

VIN = 12V, EN = GND, No Load

VCP

(5V/Div)

Time (500μs/Div)

Power Off Through EN

VIN = 12V, ENLDO = GND, No Load

(5V/Div)

VOUT

(5V/Div)

VCC

(5V/Div)

PGOOD

(5V/Div)

Time (500μs/Div)

Power Off Through ENLDO

VIN = 12V, EN = GND, No Load

ENLDO

(5V/Div)

VLDO

(5V/Div)

VCP

(5V/Div)

VCC

(5V/Div)

Time (50μs/Div)

Load Transient Response

(20V/Div)

VOUT

(30mV/Div)

VIN = 12V, EN = ENLDO = High

(5A/Div)

Time (50μs/Div)

UVP

(10V/Div)

VOUT

(5V/Div)

VIN = 12V, EN = ENLDO = High

PGOOD

(5V/Div)

Time (50μs/Div)

OVP

PGOOD

(5V/Div)

VOUT

(5V/Div)

VIN

(10V/Div)

(5A/Div) VIN = 12V, VOUT = 6V, EN = ENLDO = High

RT7291A/B

DS7291A/B-02 September 2015 www.richtek.com

Application Information

The RT7291A/B is high-performance 500kHz 6A step-down

regulators with internal power switches and synchronous

rectifiers. It features an Advanced Constant On-Time

(ACOTTM) control architecture that provides stable

operation for ceramic output capacitors without

complicated external compensation, among other benefits.

The input voltage range is from 5V to 23V, and the output

voltage is fixed 5V.

The proprietary ACOTTM control scheme improves

conventional constant on-time architectures, achieving

nearly constant switching frequency over line, load, and

output voltage ranges. Since there is no internal clock,

response to transients is nearly instantaneous and inductor

current can ramp quickly to maintain output regulation

without large bulk output capacitance.

ACOTTM Control Architecture

In order to achieve good stability with low-ESR ceramic

capacitors, ACOTTM uses a virtual inductor current ramp

generated inside the IC. This internal ramp signal replaces

the ESR ramp normally provided by the output capacitor's

ESR. The ramp signal and other internal compensations

are optimized for low-ESR ceramic output capacitors.

Making the on-time proportional to VOUT and inversely

proportional to VIN is not sufficient to achieve good

constant-frequency behavior for several reasons. First,

voltage drops across the MOSFET switches and inductor

cause the effective input voltage to be less than the

measured input voltage and the effective output voltage to

be greater than the measured output voltage as sensing

input and output voltage. When the load changes, the

switch voltage drops change causing a switching

frequency variation with load current. Also, at light loads

if the inductor current goes negative, the switch dead-

time between the synchronous rectifier turn-off and the

high-side switch turn-on allows the switching node to rise

to the input voltage. This increases the effective on-time

and causes the switching frequency to drop noticeably.

One way to reduce these effects is to measure the actual

switching frequency and compare it to the desired range.

This has the added benefit eliminating the need to sense

the actual output voltage, potentially saving one pin

connection. The ACOTTM uses this method, measuring

the actual switching frequency and modifying the on-time

with a feedback loop to keep the average switching

frequency in the desired range.

ACOTTM One-shot Operation

The RT7291A/B control algorithm is simple to understand.

The feedback voltage, with the virtual inductor current ramp

added, is compared to the reference voltage. When the

combined signal is less than the reference, the on-time

one-shot is triggered, as long as the minimum off-time

one-shot is clear and the measured inductor current

(through the synchronous rectifier) is below the current

limit. The on-time one-shot turns on the high-side switch

and the inductor current ramps up linearly. After the on-

time, the high-side switch is turned off and the synchronous

rectifier is turned on and the inductor current ramps down

linearly. At the same time, the minimum off-time one-shot

is triggered to prevent another immediate on-time during

the noisy switching time and allow the feedback voltage

and current sense signals to settle. The minimum off-time

is kept short (200ns typical) so that rapidly-repeated on-

times can raise the inductor current quickly when needed.

Diode Emulation Mode

In diode emulation mode, the RT7291A/B automatically

reduces switching frequency at light load conditions to

maintain high efficiency. This reduction of frequency is

achieved smoothly. As the output current decreases from

heavy load conditions, the inductor current is also reduced,

and eventually comes to the point that its current valley

touches zero, which is the boundary between continuous

conduction and discontinuous conduction modes. To

emulate the behavior of diodes, the low-side MOSFET

allows only partial negative current to flow when the

inductor free wheeling current becomes negative. As the

load current is further decreased, it takes longer and longer

time to discharge the output capacitor to the level that

requires the next “ON” cycle. In reverse, when the output

current increases from light load to heavy load, the

switching frequency increases to the preset value as the

RT7291A/B

DS7291A/B-02 September 2015www.richtek.com

inductor current reaches the continuous conduction. The

transition load point to the light load operation is shown in

Figure 2 and can be calculated as follows :

Figure 2. Boundary Condition of CCM/DEM

Slope = (VIN - VOUT) / L

IPEAK

ILOAD = IPEAK / 2

tON

IN OUT

LOAD ON

(V V )





where tON is the on-time.

The switching waveforms may appear noisy and

asynchronous when light load causes diode emulation

operation. This is normal and results in high efficiency.

Trade offs in DEM noise vs. light load efficiency is made

by varying the inductor value. Generally, low inductor values

produce a broader efficiency vs. load curve, while higher

values result in higher full load efficiency (assuming that

the coil resistance remains fixed) and less output voltage

ripple. Penalties for using higher inductor values include

larger physical size and degraded load transient response

(especially at low input voltage levels).

During discontinuous switching, the on-time is immediately

increased to add “hysteresis” to discourage the IC from

switching back to continuous switching unless the load

increases substantially. The IC returns to continuous

switching as soon as an on-time is generated before the

inductor current reaches zero. The on-time is reduced back

to the length needed for 500kHz switching and encouraging

the circuit to remain in continuous conduction, preventing

repetitive mode transitions between continuous switching

and discontinuous switching.

Linear Regulators (LDO & VCC)

The RT7291A/B includes a 5V linear regulator (LDO). The

LDO regulator can supply up to 100mA for external loads.

Bypass LDO with a minimum 4.7μF ceramic capacitor.

When VOUT is powered on and PGOOD is pulled high,

an internal 3Ω P-MOSFET switch connects VOUT to the

LDO pin while the internal linear regulator is simultaneously

turned off.

The RT7291A/B also includes a 5V linear regulator (VCC).

The VCC regulator steps down input voltage to supply

both internal circuitry and gate drivers. Do not connect

the VCC pin to external loads. When PGOOD is pulled

high and BYP pin voltage is above 4.6V, an internal 3Ω

P-MOSFET switch connects VCC to the BYP pin while

the VCC linear regulator is simultaneously turned off.

Current Limit

The RT7291A/B current limit is a cycle-by-cycle “valley”

type, measuring the inductor current through the

synchronous rectifier during the off-time while the inductor

current ramps down. The current is determined by

measuring the voltage between Source and Drain of the

synchronous rectifier, adding temperature compensation

for greater accuracy. If the current exceeds the current

limit, the on-time one-shot is inhibited until the inductor

current ramps down below the current limit. Thus, only

when the inductor current is well below the current limit,

another on-time is permitted. If the output current exceeds

the available inductor current (controlled by the current

limit mechanism), the output voltage will drop. If it drops

below the output under-voltage protection level (see next

section), the IC will stop switching to avoid excessive

heat.

The RT7291A/B also features a negative current limit to

protect the IC against sinking excessive current and

possibly damage. If the voltage across the synchronous

rectifier indicates the negative current is too high, the

synchronous rectifier turns off.

Output Over-Voltage Protection and Under-Voltage

Protection

The RT7291A/B features an output Over-Voltage Protection

(OVP). If the output voltage rises above the regulation

level, the IC stops switching and is latched off. The

RT7291A/B also features an output Under-Voltage

Protection (UVP). If the output voltage drops below the

UVP trip threshold for longer than 2μs (typical), the UVP

RT7291A/B

DS7291A/B-02 September 2015 www.richtek.com

Soft-Start

The RT7291A/B provides an internal soft-start function to

prevent large inrush current and output voltage overshoot

when the converter starts up. The soft-start (SS)

automatically begins once the chip is enabled. During soft-

start, it clamps the ramp of internal reference voltage which

is compared with FB signal. The typical soft-start duration

is 1.5ms.

Power Off

When VEN is pulled to GND or lower than the logic-low

level of 1.15V, there is an internal discharging resistor to

discharge the residual charge inside the output capacitors.

Besides, the value of discharging resistor is about twenty

ohms.

Power Good Output (PGOOD)

The power good output is an open-drain output that requires

a pull-up resistor. When the output voltage is 20% (typical)

below its set voltage, PGOOD will be pulled low. It is held

low until the output voltage returns to 90% of its set voltage

once more. During soft-start, PGOOD is actively held low

and only allowed to be pulled high after soft-start is over

and the output reaches 90% of its set voltage. There is a

2μs delay built into PGOOD circuitry to prevent false

transition.

External Bootstrap Capacitor (CBOOT)

Connect a 0.22μF low ESR ceramic capacitor between

the BOOT and SW pins. This bootstrap capacitor provides

the gate driver supply voltage for the high-side N-MOSFET

switch.

The internal power MOSFET switch gate driver is

optimized to turn the switch on fast enough for low power

loss and good efficiency, and slow enough to reduce EMI.

Switch turn-on is when most EMI occurs since VSW rises

rapidly. During switch turn-off, SW is discharged relatively

slowly by the inductor current during the dead-time

between high-side and low-side switch on-times. In some

cases it is desirable to reduce EMI further, at the expense

of some additional power dissipation. The switch turn-on

can be slowed by placing a small (<10Ω) resistance

between BOOT and the external bootstrap capacitor. This

will slow the high-side switch turn-on and VSW's rise.

is triggered, and the IC will shut down. The IC stops

switching and is latched off. To restart operation, toggle

EN or power the IC off and then on again.

Input Under-Voltage Lockout

In addition to the enable function, the RT7291A/B features

an Under-Voltage Lockout (UVLO) function that monitors

the input voltage. To prevent operation without fully-

enhanced internal MOSFET switches, this function inhibits

switching when input voltage drops below the UVLO-falling

threshold. The IC resumes switching when input voltage

exceeds the UVLO-rising threshold.

Over-Temperature Protection

The RT7291A/B features an Over-Temperature Protection

(OTP) circuitry to prevent overheating due to excessive

power dissipation. The OTP shuts down switching

operation when the junction temperature exceeds 150°C.

Once the junction temperature cools down by

approximately 25°C the IC resumes normal operation with

a complete soft-start. For continuous operation, provide

adequate cooling so that the junction temperature does

not exceed 150°C. Note that the VCC and LDO regulator

remains on as the OTP is triggered.

Enable and Disable

The enable input (EN) has a logic-low level of 1.15V. When

VEN is below this level, the IC enters shutdown mode and

supply current drops to less than 5μA (typical). When

VEN exceeds its logic-high level of 1.35V, the IC is fully

operational. The logics of EN and ENLDO to control the

VOUT, CLK, LDO and VCC are stated in Table 1.

Table 1. EN/ENLDO Control Logics

EN ENLDO VOUT/CLK LDO VCC

1 1 ON ON ON

1 0 ON ON ON

0 1 OFF ON ON

0 0 OFF OFF OFF

RT7291A/B

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Inductor Selection

Selecting an inductor involves specifying its inductance

and also its required peak current. The exact inductor value

is generally flexible and is ultimately chosen to obtain the

best mix of cost, physical size, and circuit efficiency.

Lower inductor values benefit from reduced size and cost

and they can improve the circuit's transient response.

However, they increase the inductor ripple current and

output voltage ripple and reduce the efficiency due to the

resulting higher peak currents. Conversely, higher inductor

values increase efficiency, but the inductor will either be

physically larger or have higher resistance since more

turns of wire are required and transient response will be

slower since more time is required to change current (up

or down) in the inductor. A good compromise between

size, efficiency, and transient response is to use a ripple

current (ΔIL) about 20-50% of the desired full output load

current. Calculate the approximate inductor value by

selecting the input and output voltages, the switching

frequency (fSW), the maximum output current (IOUT(MAX))

and estimating a ΔIL as some percentage of that current.

OUT IN OUT

IN SW L

V(VV)

LVf I





Once an inductor value is chosen, the ripple current (ΔIL)

is calculated to determine the required peak inductor

current.

OUT IN OUT L

L L(PEAK) OUT(MAX)

IN SW

V(VV) I

I and I I

Vf L 2

 

  



To guarantee the required output current, the inductor

needs a saturation current rating and a thermal rating that

exceeds IL(PEAK). These are minimum requirements. To

maintain control of inductor current in overload and short-

circuit conditions, some applications may desire current

ratings up to the current limit value. However, the IC's

output under-voltage shutdown feature make this

unnecessary for most applications.

For best efficiency, choose an inductor with a low DC

resistance that meets the cost and size requirements.

For low inductor core losses some type of ferrite core is

usually best and a shielded core type, although possibly

larger or more expensive, it will probably give fewer EMI

and other noise problems.

OUT OUT L

RMS OUT

IN IN

VV I

I(1)I

VV 12











The next step is to select a proper capacitor for RMS

current rating. One good design uses more than one

capacitor with low Equivalent Series Resistance (ESR) in

parallel to form a capacitor bank. The input capacitance

value determines the input ripple voltage of the regulator.

The input voltage ripple can be approximately calculated

using the following equation :

OUT IN OUT

IN IN SW OUT IN

IV V

V(1)

Cf V V



 



The typical operating circuit is recommended to use two

10μF low ESR ceramic capacitors on the input.

Output Capacitor Selection

The RT7291A/B is optimized for ceramic output capacitors

and best performance will be obtained by using them. The

total output capacitance value is usually determined by

the desired output voltage ripple level and transient response

requirements for sag (undershoot on positive load steps)

and soar (overshoot on negative load steps).

Output ripple at the switching frequency is caused by the

inductor current ripple and its effect on the output

capacitor's ESR and stored charge. These two ripple

components are called ESR ripple and capacitive ripple.

Since ceramic capacitors have extremely low ESR and

relatively little capacitance, both components are similar

in amplitude and both should be considered if ripple is

critical.

Input Capacitor Selection

High quality ceramic input decoupling capacitor, such as

X5R or X7R, with values greater than 20μF are

recommended for the input capacitor. The X5R and X7R

ceramic capacitors are usually selected for power regulator

capacitors because the dielectric material has less

capacitance variation and more temperature stability.

Voltage rating and current rating are the key parameters

when selecting an input capacitor. Generally, selecting an

input capacitor with voltage rating 1.5 times greater than

the maximum input voltage is a conservatively safe design.

The input capacitor is used to supply the input RMS

current, which can be approximately calculated using the

following equation :

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RIPPLE RIPPLE(ESR) RIPPLE(C)

RIPPLE(ESR) L ESR

RIPPLE(C) OUT SW

VV V

VIR

V8C f



 





In addition to voltage ripple at the switching frequency,

the output capacitor and its ESR also affect the voltage

sag (undershoot) and soar (overshoot) when the load steps

up and down abruptly. The ACOT transient response is

very quick and output transients are usually small.

However, the combination of small ceramic output

capacitors (with little capacitance), low output voltages

(with little stored charge in the output capacitors), and

low duty cycle applications (which require high inductance

to get reasonable ripple currents with high input voltages)

increases the size of voltage variations in response to

very quick load changes. Typically, load changes occur

slowly with respect to the IC's 500kHz switching frequency.

However, some modern digital loads can exhibit nearly

instantaneous load changes and the following section

shows how to calculate the worst-case voltage swings in

response to very fast load steps.

The amplitude of the ESR step up or down is a function of

the load step and the ESR of the output capacitor :

The amplitude of the capacitive sag is a function of the

load step, the output capacitor value, the inductor value,

the input-to-output voltage differential, and the maximum

duty cycle. The maximum duty cycle during a fast transient

is a function of the on-time and the minimum off-time since

the ACOTTM control scheme will ramp the current using

on-times spaced apart with minimum off-times, which is

as fast as allowed. Calculate the approximate on-time

(neglecting parasitics) and maximum duty cycle for a given

input and output voltage as :

ESR_STEP OUT ESR

VIR 

OUT ON

ON MAX

IN SW ON OFF(MIN)

t and D

Vf t t





The actual on-time will be slightly longer as the IC

compensates for voltage drops in the circuit, but we can

neglect both of these since the on-time increases

compensations for the voltage losses. Calculate the output

voltage sag as :

()

OUT

SAG OUT IN(MIN) MAX OUT

V2C V D V



 

The amplitude of the capacitive soar is a function of the

load step, the output capacitor value, the inductor value

and the output voltage :

()

OUT

SOAR OUT OUT

V2C V





Most applications never experience instantaneous full load

steps and the RT7291A/B's high switching frequency and

fast transient response can easily control voltage regulation

at all times. Therefore, sag and soar are seldom an issue

except in very low-voltage CPU core or DDR memory

supply applications, particularly for devices with high clock

frequencies and quick changes into and out of sleep

modes. In such applications, simply increasing the amount

of ceramic output capacitor (sag and soar are directly

proportional to capacitance) or adding extra bulk

capacitance can easily eliminate any excessive voltage

transients.

In any application with large quick transients, it should

calculate soar and sag to make sure that over-voltage

protection and under-voltage protection will not be triggered.

Thermal Considerations

For continuous operation, do not exceed absolute

maximum junction temperature. The maximum power

dissipation depends on the thermal resistance of the IC

package, PCB layout, rate of surrounding airflow, and

difference between junction and ambient temperature. The

maximum power dissipation can be calculated by the

following formula :

PD(MAX) = (TJ(MAX) − TA) / θJA

where TJ(MAX) is the maximum junction temperature, TA is

the ambient temperature, and θJA is the junction to ambient

thermal resistance.

For recommended operating condition specifications, the

maximum junction temperature is 125°C. The junction to

ambient thermal resistance, θJA, is layout dependent. For

UQFN-16L 3x3 (FC) package, the thermal resistance, θJA,

is 70°C/W on a standard JEDEC 51-7 four-layer thermal

test board. The maximum power dissipation at TA = 25°C

can be calculated by the following formula :

PD(MAX) = (125°C − 25°C) / (70°C/W) = 1.4W for

UQFN-16L 3x3 (FC) package

RT7291A/B

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Layout Considerations

Layout is very important in high frequency switching

converter design. The PCB can radiate excessive noise

and contribute to converter instability with improper layout.

Certain points must be considered before starting a layout

using the RT7291A/B.

Make traces of the main current paths as short and wide

as possible.

Put the input capacitor as close as possible to the device

pins (VIN and PGND).

SW node encounters high frequency voltage swings so

it should be kept in a small area. Keep sensitive

components away from the SW node to prevent stray.

The PGND pin should be connected to a strong ground

plane for heat sinking and noise protection.

Avoid using vias in the power path connections that have

switched currents (from CIN to PGND and CIN to VIN)

and the switching node (SW).

Figure 3. Derating Curve of Maximum Power Dissipation

0.0

0.4

0.8

1.2

1.6

2.0

0 25 50 75 100 125

Ambient Temperature (°C)

Maximum Power Dissipation (W) 1

Four-Layer PCB

The maximum power dissipation depends on the operating

ambient temperature for fixed TJ(MAX) and thermal

resistance, θJA. The derating curve in Figure 3 allows the

designer to see the effect of rising ambient temperature

on the maximum power dissipation.

Figure 4. Layout Guide

The output capacitor must

be placed near the IC

The input capacitor must

be placed as close to the

IC as possible.

SW should be connected to

inductor by wide and short

trace.

Keep sensitive components

away from this trace.

VOUT

COUT

GND

AGND

ENLDO

VCC

BOOT

34567

VIN

PGND

VBYP

PGOOD

CLK

LDO

VOUT

CIN

VOUT

GND

CBOOT

CLDO

CVCC

VIN

Impedance between PGND and AGND should be as small as possible for unified ground voltage.

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Outline Dimension

U-Type 16L QFN 3x3 (FC) Package

Min. Max. Min. Max.

A 0.500 0.600 0.020 0.024

A1 0.000 0.050 0.000 0.002

A3 0.100 0.175 0.004 0.007

D 2.900 3.100 0.114 0.122

E 2.900 3.100 0.114 0.122

b 0.150 0.250 0.006 0.010

b1 0.100 0.200 0.004 0.008

L 0.350 0.450 0.014 0.018

L1 0.750 0.850 0.030 0.033

L2 0.550 0.650 0.022 0.026

0.975 0.038

1.675 0.066

1.935 0.076

0.975 0.038

1.335 0.053

Symbol Dimensions In Millimeters Dimensions In Inches

0.400 0.016

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Richtek Technology Corporation

14F, No. 8, Tai Yuen 1st Street, Chupei City

Hsinchu, Taiwan, R.O.C.

Tel: (8863)5526789

Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should

obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot

assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be

accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third

parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.

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