Fairchild Semiconductor An 7502 Users Manual Power MOSFET Switching Waveforms

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Power MOSFET Switching Waveforms:
A New Insight
Application Note

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The examination of power MOSFET voltage and current
waveforms during switching transitions reveals that the
device characterization now practiced by industry is inadequate. In this Note, device waveforms are explained by considering the interaction of a vertical JFET driven in cascode
from a lateral MOSFET in combination with the interelectrode capacitances. Particular attention is given to the
drain-voltage waveform and its dual-slope nature. The
three terminal capacitances now published by the industry
are shown to be valid only for zero drain current. For cases
where the gate drive is a voltage step generator with internal fixed resistance, the drain voltage characteristics are
inferred from the gate current drive behavior and compared
to observed waveforms. The nature of the “asymmetric
switching times” is explained.
A waveform family is proposed as a more descriptive and
accurate method of characterization. This new format is a
plot of drain voltage and gate voltage versus normalized
time. A family of curves is presented for a constant load
resistance with VDS varied. Gate drive during switching
transitions is a constant current with voltage compliance
limits of 0 and 10 volts. Time is normalized by the value of
gate driving current. The normalization shows excellent
agreement with data over five orders of magnitude, and is
bounded on one extreme by gate propagation effects and
on the other by transition time self-heating (typically tens of
nanoseconds to hundreds of microseconds).

Device Models
The keystone of an understanding of power MOSFET
switching performance is the realization that the active
device is bimodal and must be described using a model that
accounts for the dual nature. Buried in today’s power MOSFET devices is the equivalent of a depletion layer JFET that
contributes significantly to switching speed. Figure 1 is a
cross-sectional view of a typical power MOSFET, with MOSFET/JFET symbols superimposed on the structure.
Figure 2 is obtained by taking the lateral MOS and vertical
JFET from this conception and adding all the possible nodeto-node capacitances. Computed values of the six capacitances for a typical device structure suggest that device
behavior may be adequately modeled using only three
capacitors in the manner of Figure 3. This is the model to be
employed for analysis and study.

Pageode
Useutines

October 1999

AN-7502

SOURCE METAL
POLY GATE GLASS GATE OXIDE
n+ SOURCE
p BODY

MOS

JFET

p+

0
10 VOLTS
DEPLETION EDGE

n-

40 VOLTS
n+ DRAIN

FIGURE 1. CROSS-SECTION VIEW OF MOSFET SHOWING
EQUIVALENT MOS TRANSISTOR AND JFET
C6
C3

C2

DRAIN

GATE
C4

C5

C1

SOURCE

FIGURE 2.

MOS TRANSISTOR WITH CASCODE-CONNECTED
JFET AND ALL CAPACITORS
Cx
DRAIN

GATE

CDS
CGS

SOURCE

FIGURE 3.

FIGURE 2 SIMPLIFIED

Gate Drive: Constant Voltage or
Constant Current
Before moving on to the study of the equivalent circuit states
of the model, a gate-drive forcing function which is easy to
represent, relates to reality, and best illustrates device
behavior must be chosen. The choice may be immediately
narrowed to two:
(1) An instantaneous step voltage with internal resistance R,
Figure 5.
(2) An instantaneous step current with infinite internal resistance, Figure 6.

©2002 Fairchild Semiconductor Corporation

Application Note 7502 Rev. A1

Application Note 7502
i(t)

IG

DRAIN
VOLTAGE

VG(SAT)

-VG =

v(t)

-IG

2

3

4

STATES

5

6

TURN ON
IGt
v(t) =
C
i(t) = IG, 0 < t < T

IG = CONSTANT

T

T

IG

t

TURN OFF

VT
VDK

-I t
v(t) = 2VG G
C

VD(SAT)

-IG

i(t) = IG, T < t < 2T
TIME

FIGURE 4.

FIGURE 6.

STEP CURRENT FORCING FUNCTION

IDEALIZED POWER MOSFET WAVEFORMS
RO

Equivalent Circuit

v(t)

-VG

The lumped-parameter model of Figure 3, with the cascodeconnected JFET, can now be reduced to the linear equivalent circuit of Figure 7, and the six device states investigated
from full off to full on.

v(t)

VG

i(t)

C
t
-IPK = VG/RO

GATE
VGS

i(t)

TURN ON
v(t) = VG (1 - e) -t/ROC
i(t) = VG e -t/ROC

VX

gMJ VX

VD

gM VG

CGS

RL

CDS

IPK = VG/RO

RO

FIGURE 5.

IG

t

G

CX

DRAIN

RO
TURN OFF
v(t) = VG e -t/ROC
i(t) = - V e -t/ROC

SOURCE

STEP-VOLTAGE FORCING FUNCTION

Power MOSFET devices are highly capacitive in nature;
hence, simple capacitor responses to the forcing functions
offer a good vehicle for comparison. The advantageous
choice is immediately obvious: Figure 6. Voltage/time
responses dominated by capacitance are straight lines
(when constant current is used). The slope of these lines is
proportional to current and inversely proportional to capacitance. Analytically, then, constant current is most convenient. It is quite another matter, however, to build a
bidirectional current drive that is accurate across the many
decades of both current and time required to establish
experimental verification.

Six States
To completely characterize power MOSFET switching waveforms, the six states that a device assumes, Figure 6, must
be addressed:
STATE

t

i(t)

VOLTAGE

C
1

IGT
C

v(t)

GATE VOLTAGE

VGS

VDD

MOS

JFET

Turn-on 1

Off

Off

Turn-on 2

Active

Active

Turn-on 3

Active

Saturated†

Turn-off 4

Saturated

Saturated†

Turn-off 5

Active

Saturated

Turn-off 6

Active

Active

†The

term saturated is taken to mean a constant low-voltage drain-source
condition.

©2002 Fairchild Semiconductor Corporation

LEGEND
VGS - Gate Voltage

CDS

- Drain Source Capacitance

VX

- JFET Driving Voltage

gM

- MOSFET Transconductance

VD

- Drain Voltage

gMJ

- JFET Transconductance

CGS - Gate Source
Capacitance

RL

- Drain Load Resistance

CX

IG

- Constant Current Amplitude

- MOSFET Feedback
Capacitance
FIGURE 7.

POWER MOSFET EQUIVALENT CIRCUIT

State 1: MOS Off, JFET Off
In a power-MOSFET device, no drain current will flow until
the device’s gate threshold voltage, Vgs(TH), is reached. During this time, the gate’s current drive is only charging the
gate source capacitance. More accurately, IG is charging
CISS (CISS = CGS + CGD, CDS shorted), the capacitance
designation published by the industry.
The current generators, gMVG and gMJVX are open circuits
for zero drain current, and RL is presumed to be so low as to
represent a short circuit (generally true for practical applications). This is academic however since CGS is very much
larger that CX. The time to reach threshold, then, is simply:

T1 =

CISS
Vgs(TH)
IG

Application Note 7502 Rev. A1

Application Note 7502
State 2: MOS ActIve, JFET ActIve
This state graphically illustrates the dramatic influence that
the JFET has on the power MOSFET drain-voltage waveform. Instead of having to discharge C x from VDD to ground,
the lateral MOSFET need only swing VX to ground, a much
smaller voltage thanks to the grounded gate JFET. Since the
interaction of RL with the device capacitances has a secondorder effect on the drain voltage, the equivalent circuit of Figure 7 predicts a drain voltage change of:
dVG/dt = gMRLlG/[CGS + CX(1 + gM/gMJ)]
In all but the smallest power-MOSFET devices, Cx is several
thousand picofarads and gM/gMJ is of the order of 3:1.
Power-MOSFET devices exhibit a high dVD/dt switching rate
because of the cascode-connected JFET, not because
CRSS (CRSS = CGD) is a small value, as zero-drain-current
data sheet capacitance values might lead one to believe. If
CRSS were, in actuality, small, long drain voltage tails would
not exist. The tail response is a direct result of JFET saturation. In order to delineate the transition from state 2 to state
3, a drain voltage at which the transition occurs must be
defined. VDK is the knee voltage at which linear extrapolations of drain-voltage slopes intersect. The time duration of
state 2 is:

Experimental Verification
The four switching states just analyzed indicate that for a
given device, all four switching state times are inversely proportional to the magnitude of the gate drive current. Figure 8
illustrates the switching performance of a typical power
MOSFET across three decades of gate drive current and
time. In each case the data slope is almost a perfect -1.

A New Device Characterization
Figure 8 could not be a reasonable device data sheet presentation because it does not give the designer any information on a typical value for CX, nor does it convey how VDK,
gM, gM/gMJ, and VG(sat) vary with drain current. What would
be of enormous value to the designer is a plot of VD(t), VG(t)
for selected values of VDD and ID within device ratings.
A reasonable characterization would be as follows:
1. The x axis would be normalized in terms of gate current drive.
2. The y axis would be normalized in terms of percent maximum rated
BVDSS (0 to 100%).
3. RL = BVDSS/ID(max) would define the drain load resistance.
4. Four plots of VD(t), VG(t) at 100%, 75%, 50%, and 25% BVDSS(max)
would be shown.
10

t2(t6) = (VDD - VDK)[CGS + CX(1 + gM/gMJ)]/gMRLIG

RFM15N15
VDD = 75V
ID = 7.5A
RO = ∞ Ω
VG = 10V

State 3: MOS Active, JFET Saturated

dVD/dt = gMRLIG/[CGS + CX(1 + gMRL)]
This is the Miller effect so often referred to in older texts that
describe the behavior of grounded-cathode vacuum-tube
amplifier circuits. Allowing for the fact that 1 + gMRL is
approximately equal to gMRL and CX(1 + gMRL) is very
much larger than CGS, the expression for drain-voltage tail
time is:

(t) - MICROSECONDS

When the JFET saturates, the gMJVX current generator
becomes a short circuit and the equivalent circuit predicts:
1

0.1
DATA THEORY

t3(t5) = (VDK - VD(SAT))Cx/lG

tD(OFF)
tR
tF
tD(ON)

State 4: MOS Saturated, JFET Saturated (Turn-Off)
In this state, in addition to gMJVX being shorted, the gMVG current generator is shorted, and IG is occupied with charging CX
and CGS, in parallel, from the peak value of V G to VG(SAT). The
time required for this is:
t4 = (VG - VG(SAT))(CGS + Cx)/IG
Since a value for CGS may be measured independently of
switching time, the method described is the simplest way of
determining CX.

0.01
1

FIGURE 8.

10
100
(IG) - MILLIAMPERES

1000

CONSTANT GATE CURRENT SWITCHING TIME

Figure 9 is such a plot for the RFM15N15 power MOSFET.
With such a plot, a designer can estimate device switching
performance under any resistive gate/drain conditions.

On turn-off, the state time equations are equally applicable,
but in reverse order (states 5 and 6); see the idealized waveform of Figure 4.

©2002 Fairchild Semiconductor Corporation

Application Note 7502 Rev. A1

Application Note 7502
sis for resistive step voltage inputs, which is complex
because the gate current is no longer constrained to be constant, but is a function of device gate-voltage response, is
covered in Appendix A. (A second, shorter appendix, B, has
been added to illustrate the estimation of RO for some practical gate drive circuits.) Table 1 summarizes the common
switching equations, and indicates the appropriate 1G to be
used in each state for relating step voltage drives to the characterization curves.

100

% RATES VDSS

75

50
RFM15N15
IT = 1mA
VG = 10 VOLTS
RL = VDSS/ID(RMS)

25

Experimental Verification
Since the switching equations for step currents and voltages
differ only by gate-current magnitudes for the same device
type, one would expect a plot of switching time versus 1/RO
to be of the same form as those obtained for a step current
drive. This is exactly the case, as Figure 10 is merely a variation of Figure 8. Using the relationships of Table 1, the
observed differences between Figures 7 and 9 can be pinpointed. The two sets of experimental curves confirm that,
on the basis of the short-circuit drive current VG/RO equalling the constant IG, tD(on), tR, tD(off), and tF will all be
longer, as predicted by the ratios of the gate drive currents of
Table 1. Notice also that tR, tF switching symmetry is disrupted by the use of a step voltage with source resistance
RO. For states 2 and 6 the time ratio is:

0
20IT/IG

40IT/IG
60IT/IG
TIME - microseconds

80IT/IG

FIGURE 9. NORMALIZED RFM15N15 SWITCHING WAVEFORMS FOR CANSTANT GATE-CURRENT DRIVE.

Step-Voltage Gate Drive
The majority of power MOSFET applications employ a step
gate-voltage input with a finite source resistance RO. Often
RO for turn-on is not the same as RO for turn-off. How can
switching times for these situations be estimated using the
switching characterization curves just described? The analy-

TABLE 1.
CONSTANT CURRENT
t=
T
U
R
N
O
N

COMMON SWITCHING EQUATIONS
STATE 1: MOS OFF, JFET OFF

CISS VGS(TH)

CONSTANT VOLTAGE
[1]
t = RO CISS In

IG
IG = IT

STATE 2: ACTIVE, ACTIVE

[1 - VGS(TH)/VG]

IG = (VG - VGS(TH))/RO

[VDD - VDK] [CGS + Cx (1 + gM/gMJ)]

t=

gMRLIG
IG = IT

STATE 3: ACTIVE, SATURATED

IG = (VG - VG(SAT))/RO

(VDK - VD(SAT))CX

t=

IG
IG = IT
t=
T
U
R
N
O
F
F

STATE 4: SATURATED, SATURATED
(CGS + CX)(VG - VG(SAT))

IG = -VG/RO
t = RO(CGS + CX) In (VG/VG(SAT))

IG
IG = IT

STATE 5: ACTIVE, SATURATED
t=

IG = (VG - VG(SAT))/RO

(VDK - VD(SAT))CX
IG

IG = IT

STATE 6: ACTIVE, ACTIVE
t=

IG = (VG - VG(SAT))/RO

[VDD - VDK] [CGS + CX (1 + gM/gMJ)]
gMRLIG

©2002 Fairchild Semiconductor Corporation

Application Note 7502 Rev. A1

Application Note 7502
Experimental Verification

State 1: MOS Off, JFET Off

Since the switching equations for step currents and voltages
differ only by gate-current magnitudes for the same device
type, one would expect a plot of switching time versus 1/RO
to be of the same form as those obtained for a step current
drive. This is exactly the case, as Figure 10 is merely a variation of Figure 8. Using the relationships of Table 1, the
observed differences between Figures 7 and 9 can be pinpointed. The two sets of experimental curves confirm that,
on the basis of the short-circuit drive current VG/RO equalling the constant IG, tD(on), tR, tD(off), and tF will all be
longer, as predicted by the ratios of the gate drive currents of
Table 1. Notice also that tR, tF switching symmetry is disrupted by the use of a step voltage with source resistance
RO. For states 2 and 6 the time ratio is:

This time can be estimated without recourse to the curves

tTURN-ON
tTURN-OFF

VG(SAT)
VG - VGS(TH)

=

t=

100(1200 x 10-12) ln [1/(1 - 4/10)]

t=

61 ns

State 2 & 6: MOS Active, JFET Active
IG =

tTURN-OFF

9
60

=

150ns

IG =

=

467ns

(10 - 7)/100 = 30mA
(curve divisions) x IT µs

t=

30

=

14
30

State 4: MOS Saturated, JFET Saturated
CGS + Cx

Utilization of available maximum gate drive voltage and current can be optimized for fastest power MOSFET switching
speed through the use of constant-current gate drive at the
expense of increased gate-drive circuit complexity.

=

(gate voltage slope)(test current)

=

(1.5 x 10-6s/5 volts)(10mA)

=

3000pF

t=

100(3000 x 10-12) ln [10/6.6]

t=

125ns

State 5: MOS Active, JFET Saturated
IG =

10
RFM15N15
VDD = 75V
ID = 7.5A
VG = 10V

6.6/100 = 66mA
(curve divisions) x IT µs

t=

66

=

8
66

=

121ns

Figure 11 shows RFM15N15 waveforms using the conditions
specified in the example.

1

75
VD

0.1
DATA THEORY
tD(OFF)
tR
tF
tD(ON)
0.01
10-4

10-3

1/RO

VGS

DRAIN VOLTAGE - VOLTS

(t) - MICROSECONDS

60

=

State 3: MOS Active, JFET Saturated

VG(SAT)
VG - VG(SAT)

=

(curve divisions) x IT µs

t=

For states 3 and 5 the time ratio is:
tTURN-ON

(10 - 4)/100 = 60mA

10-2

RFM15N15
VDD = 75 VOLTS
RL = 10 OHMS
VG = 10 VOLTS
RO = 100 OHMS

10-1
0

FIGURE 10.

0

CONSTANT GATE VOLTAGE SWITCHING TIME

Using the Characterization Curve,
Figure 9

FIGURE 11.

To estimate the switching times for an RFM15N15 power
MOSFET under the conditions VG = 10V, VDD = 75V, RO =
100 ohms, and RL = 10 ohms, precedes as follows:

STATE

1

©2002 Fairchild Semiconductor Corporation

1.5
TIME - MICROSECONDS

3

STEP GATE VOLTAGE INPUT TO AN RFM15N15
CALCULATED
TIME

MEASURED
TIME

RATIO

(tC, ns)

(tM, ns)

(tC/tM)

61

60

1.02

2+3

617

670

0.92

4

125

137

0.91

5+6

271

375

0.72

Application Note 7502 Rev. A1

Application Note 7502
For peak gate voltages other than 10 volts, and load resistances other than BVDSS/ID(MAX), the equations of Table 1
may be used in conjunction with slope estimates from the
characterization curves for CX and CGS + CX(1 + gM/gMJ) at
the appropriate drain-current level.

states must be examined using the same device equivalent
circuit as was used for the constant-gate-current case, but
with the forcing function replaced wIth a step voltage with
internal resistance RO, Figure A-1.
GATE

Characterization-Curve Limits
The switching-time range over which the characterization can be
applied is very impressive. For gate currents of the order of
microamperes, device dissipation is the limiting factor. For gate
currents of the order of amperes, the device response will be
slowed by gate propagation delay. This delay, of course,
degrades the linear switching relationship to gate current. However, as Figure 12 graphically shows, the characterization is valid
across five decades of gate current and switching time, allowing
all but a very few switching applications to be described by the
characterization curves of Figure 9.
104
RFM15N15
tD(OFF)
tR
tF
tD(ON)

TIME(t) - MICROSECONDS

103

CX

RO VGS

102

VX

gMJ VX

VD
DRAIN

VG

gM VG

CGS

RL

CDS

SOURCE

LEGEND
VGS

- Gate Voltage

VX

- JFET Driving Voltage gM

- MOSFETTransconductance

VD

- Drain Voltage

gMJ

- JFET Transconductance

CGS

- Gate Source
Capacitance

RL

- Drain Load Resistance

CX

- MOSFET Feedback
Capacitance

IG

- Constant Current Amplitude

FIGURE A-1.

CDS

- Drain Source Capacitance

POWER MOSFET EQUIVALENT CIRCUIT

State 1: Mos Off, JFET Off
101

As before, both current generators are open circuits, reducing
the equivalent circuit to simply charging CISS through RO.

100

t=

ROCISSIn(1/(1 - VGS(TH)/VG)]

State 2: Mos Active, JFET Active
10-1

10-2

100

101
102
103
104
105
GATE CURRENT (IG) - MICROAMPERES

FIGURE 12.

106

FIVE DECADES OF LINEAR RESPONSE

Before proceeding, it is wise to examine an actual device
response and make use of available simplifications. Figure A-2
shows iG(t) and iD(t) for a typical power MOSFET driven by a
step gate voltage. For truly resistive switching, realize that these
waveforms are only mirror images of their voltage counterparts
vG(t) and vD(t). Using Figure A-2, applicable gate currents for
each of the device states may be listed.
IPK1

Conclusions

Appendix A - Analysis for Resistive Step
Voltage Inputs

IPK2

iD(t)
IPK3

CURRENT

The viability of the proposed characterization curves using constant current has been demonstrated and the limits of application defined. The existence of a vertical JFET in a power
MOSFET makes data-sheet capacitances of little use for estimating switching times. The classical method of defining
switching time by 10% and 90% is a poor representation for
power MOSFETs because of the dual-slope nature of the drain
waveforms. Switching influences are masked because the 10%
level is controlled by one mechanism and the 90% level by
another. Device comparisons based on the classical switching
definition can be very misleading.

iG(t)

IPK6
IPK4

IPK5

TIME

FIGURE A-2. iG(t) AND iD(t) FOR A TYPICAL POWER MOSFET
DRIVEN BY A STEP GATE VOLTAGE

Step Voltage Gate Drive
To obtain the necessary relationships, six device switching

©2002 Fairchild Semiconductor Corporation

Application Note 7502 Rev. A1

Application Note 7502
Turn-On

State 5: Mos Active, JFET Saturated
The JFET current generator VxgmJ, is operative.

State 1: MOS Off, JFET Off
IPK1 = VG/RO

[VDK - VD[SAT])CX

t=

State 2: MOS Active, JFET Active
IPK2 = (VG - VGS(TH))/RO

IPK5

IPK5 = VG(SAT)/RO

State 3: MOS Active, JFET Saturated

State 6: Mos Active, JFET Active

IPK3 = (VG - VG(SAT))/RO

The Miller effect is now reduced by the activation of VGgMJ,
and the equivalent circuit predicts:

Turn-Off
State 4: MOS Saturated, JFET Saturated

[VDD - VDK][CGS + CX(1 + gM/gMJ)]

t=

IPK4 = VG/RO
State 5: MOS Active, JFET Saturated

gMRL IPAK6

IPAK6 = VG(SAT)/RO

IPK5 = VG(SAT)/RO

Appendix B - Estimating RO for Some
Typical Gate-Drive Circuits

State 6: MOS Active, JFET Active
IPK6 = VG(SAT)/RO
The equivalent circuit of Figure A-1 predicts that:

Case 1: Typical Pulse-Generator Drive, Figure B-1

dVD/dt = (-gMRL(VG - VGS(TH))e-t/T1) /T1

VDD

where T1 = ROCGS + (1 + gM/gMJ)ROCX
Note that gMRL(VG - VGS(TH)) is usually an order of magnitude
greater than VDD, indicating that the drain voltage is discharging toward a very large negative value. The device operation,
then, is on the early, almost linear, portion of the exponential,
where e-t/T1 approximates unity. The drain current of Figure A2, and hence the drain voltage, does indeed exhibit a linear
decrease with time.
Thus, for state 2:

RGEN

TYPICAL PULSE-GENERATOR DRIVE CIRCUIT

Turn-On and Turn-Off
RO = RGENRGS/(RGEN + RGS)

gMRL IPK2

For the typical case where R GEN = 50Ω, and a coaxial-cable
termination of 50 ohms, RO = 25Ω and VG = VGEN/2.

where IPK2 = (VG - VGS(TH))/RO
State 3: Mos Active, JFET Saturated

Case 2: Voltage-Follower Gate Drive, Figure B-2

Because of the Miller effect, the gate voltage and, hence, the
gate current, is almost constant during the tail time. The
equivalent circuit then predicts:
dVD
dt

=

gMRLlG
CGS + (1 + gMRL)CX

=

+
RL
VDD

lG
CX
RS

lG = IPK3 = (VG - VG(SAT))/RO
and

t=

VG

RGS

FIGURE B-1.

[VDD - VDK][CGS + CX(1 + gM/gMJ)]

t=

RL

VGEN

(VDK - VD[SAT])Cx
IPK3

State 4: Mos Saturated, JFET Saturated (Turn-off)
Both equivalent-circuit generators are short circuits, and the
gate drive is discharging CX in parallel with CGS through RO.
t = RO(CGS + CX) ln[VG/VG(SAT)]
IPK4 = VG/RO

FIGURE B-2. VOLTAGE-FOLLOWER GATE-DRIVE CIRCUIT

Turn-On
RO is approximately equal to 1/gM for RS very much
greater than 1/gM.
gm = transconductance of driving MOSFET transistor.
Turn Off
RO = RS

©2002 Fairchild Semiconductor Corporation

Application Note 7502 Rev. A1

Application Note 7502
Case 3 :Common-Source Gate Drive, Figure B-3
+

RL

RD

VDD

10V
0V

FIGURE B-3. COMMON-SOURCE GATE-DRIVE CIRCUIT

Turn-On
RO = RD
(drain-to-ground capacitance of driving device adds to
CGS of driven MOSFET.)
Turn Off
RO = rDS(ON) of driving MOSFET when
RD is very much greater than RDS(ON)

©2002 Fairchild Semiconductor Corporation

Application Note 7502 Rev. A1

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SuperSOT™-6
SuperSOT™-8
SyncFET™
TinyLogic™
TruTranslation™

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FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER
NOTICE TO ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD
DOES NOT ASSUME ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT
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FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT
DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF FAIRCHILD SEMICONDUCTOR CORPORATION.
As used herein:
2. A critical component is any component of a life
1. Life support devices or systems are devices or
support device or system whose failure to perform can
systems which, (a) are intended for surgical implant into
be reasonably expected to cause the failure of the life
the body, or (b) support or sustain life, or (c) whose
support device or system, or to affect its safety or
failure to perform when properly used in accordance
with instructions for use provided in the labeling, can be
effectiveness.
reasonably expected to result in significant injury to the
user.
PRODUCT STATUS DEFINITIONS
Definition of Terms
Datasheet Identification

Product Status

Definition

Advance Information

Formative or
In Design

This datasheet contains the design specifications for
product development. Specifications may change in
any manner without notice.

Preliminary

First Production

This datasheet contains preliminary data, and
supplementary data will be published at a later date.
Fairchild Semiconductor reserves the right to make
changes at any time without notice in order to improve
design.

No Identification Needed

Full Production

This datasheet contains final specifications. Fairchild
Semiconductor reserves the right to make changes at
any time without notice in order to improve design.

Obsolete

Not In Production

This datasheet contains specifications on a product
that has been discontinued by Fairchild semiconductor.
The datasheet is printed for reference information only.

Rev. H5



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Title                           : AN-7502 Power MOSFET Switching Waveforms: A New Insight

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